Broadband dual-polarized radar antenna system based on multilayer metasurface

Abstract

The application discloses a wide-band dual-polarization radar antenna system based on a multilayer metasurface and relates to the technical field of radar antennas, which comprises a modulation module, a receiving module and a decoupling module.The modulation module converts a multilayer spatial electromagnetic response parameter set into a high-frequency digital pulse sequence and applies the high-frequency digital pulse sequence to a multilayer adjustable metasurface, introduces a periodic time-varying non-reciprocal electromagnetic control on a time axis, and outputs dual-polarization microwave radiation field and reflection field signals.The receiving module digitizes the dual-polarization microwave radiation field and reflection field signals and outputs dual-channel transient baseband signals.The decoupling module combines a current linear frequency modulation slope of the radar, carries out time-frequency coherent demodulation and dispersion phase compensation on the dual-channel transient baseband signals, and carries out component stripping along orthogonal polarization axes, and outputs completely decoupled high-dynamic dual-polarization radar target data.The application suppresses metasurface wide-band dispersion defects and transient electromagnetic lag crosstalk, and obtains high-dynamic dual-polarization radar target data with extremely high polarization isolation and complete decoupling.

Description

Technical Field

[0001] This invention relates to the field of radar antenna technology, and in particular to a broadband dual-polarization radar antenna system based on a multilayer metasurface. Background Technology

[0002] As next-generation radar systems evolve towards high-resolution detection, wideband and dual-polarization synergistic modulation have become core support. Traditional polarization antennas rely on heterogeneous feed networks or mechanical conversion structures, resulting in volume redundancy and bandwidth limitations imposed by the material's natural dispersion. In recent years, information metasurfaces, as two-dimensional artificial electromagnetic metamaterials, have provided a new approach to reshaping ultrathin, wideband dual-polarization antenna systems by leveraging their digital programming control capabilities over the amplitude, phase, and polarization state of spatial electromagnetic waves. Currently, dynamic electromagnetic modulation technology based on multi-layer cascaded metasurface evolution, through the introduction of active microwave components such as diodes, can establish reconfigurable electromagnetic boundary conditions on spatially discrete grid points, thereby achieving decoupled radiation of horizontal and vertical polarized waves within specific frequency bands and significantly improving the spatial multiplexing efficiency of radar apertures. This cascaded modulation not only achieves dynamic beam shaping but also significantly reduces the system profile height, aligning with the trends of low cost and integration.

[0003] However, existing technologies using multilayer metasurfaces for broadband dual-polarization modulation are limited by the physical limits of the unit mass factor. This results in nonlinear polarization phase dispersion at the bandwidth edge, and time-varying control signals induce microscopic electromagnetic transient hysteresis and crosstalk. For example, in conventional radar systems, when the metasurface undergoes high-speed periodic switching for signal modulation, the physical hysteresis in the storage and release of electromagnetic energy by each layer of hardware prevents the electromagnetic wave from immediately reaching a stable state at the moment of switching, resulting in severe signal tailing on the time axis. This tailing directly causes severe signal adhesion and crosstalk between the originally perpendicular horizontal and vertical polarization channels in space. This dynamic electromagnetic pollution caused by high-speed time-varying hardware switching makes it impossible for backend algorithms to accurately remove and separate clean orthogonal polarization components when faced with echoes containing transient hysteresis spurious signals, thus preventing the acquisition of fully decoupled high-dynamic polarization target data. Summary of the Invention

[0004] In view of the aforementioned existing problems, the present invention is proposed.

[0005] Therefore, this invention provides a broadband dual-polarization radar antenna system based on multilayer metasurfaces to solve the problem that the back-end cannot accurately separate orthogonal polarization components and obtain completely decoupled target data due to crosstalk caused by polarization phase dispersion and transient electromagnetic hysteresis.

[0006] To solve the above-mentioned technical problems, the present invention provides the following technical solution: This invention provides a broadband dual-polarization radar antenna system based on a multilayer metasurface, comprising: The sensing module captures radar operating parameters and interlayer state parameters of the multilayer tunable metasurface in real time, and outputs a set of operating parameters and boundary conditions. The solution module, based on the set of working parameters and boundary conditions, reconstructs the spatial electromagnetic boundary parameters of each layer of metasurface, counteracts the polarization phase dispersion generated at the broadband edge, and generates a multi-layer spatial electromagnetic response parameter set. The modulation module converts the multi-layer spatial electromagnetic response parameter set into a high-frequency digital pulse sequence and applies it to the multi-layer tunable metasurface. It introduces periodic time-varying non-reciprocal electromagnetic control on the time axis and outputs dual-polarization microwave radiation field and reflection field signals. The receiving module digitizes the dual-polarization microwave radiation field and reflection field signals and outputs a dual-channel transient baseband signal. The decoupling module, combined with the current linear frequency modulation slope of the radar, performs time-frequency coherent demodulation and dispersion phase compensation on the dual-channel transient baseband signal, and performs component stripping along the orthogonal polarization axis to output fully decoupled high dynamic dual-polarization radar target data.

[0007] Preferably, the method for setting the output working parameters and boundary conditions includes: Real-time capture of physical operating status commands issued by the radar radio frequency front end, and extraction of the radar's current center frequency, operating bandwidth, and linear frequency modulated pulse parameters as the target polarization mode; The transient feedback power of each layer of the multilayer tunable metasurface and the initial isolation between channels are acquired simultaneously. The radar's current center frequency, operating bandwidth, linear frequency modulated pulse parameters, transient feedback power of each metasurface layer, and initial isolation between channels are time-synchronously sampled and digitally converted, and then packaged and combined into a set of operating parameters and boundary conditions.

[0008] Preferably, the method for reconstructing the spatial electromagnetic boundary parameters of each metasurface layer includes: By connecting the radar’s current center frequency to the pre-stored electromagnetic parameter table, the basic electromagnetic characteristic parameters of the multilayer tunable metasurface are obtained. The basic electromagnetic characteristic parameters are then calibrated using transient feedback power and initial isolation between channels to lock the surface electromagnetic boundary response benchmark under the current hardware physical state. The surface electromagnetic boundary response reference is discretized along the working bandwidth and mapped to form a spatial electromagnetic boundary control matrix, thus obtaining the reconstructed spatial electromagnetic boundary parameters.

[0009] Preferably, the method of counteracting the polarization phase dispersion generated at the wideband edge includes: Extract the nonlinear phase deviation value of the reconstructed spatial electromagnetic boundary parameters at the edge frequency of the working bandwidth, lock the magnitude of polarization phase dispersion and the tilt angle of polarization axis caused by the wide bandwidth edge, and generate an inverse counteracting phase value with the opposite sign to the nonlinear phase deviation value. The reverse offset phase value is directly superimposed and compensated into the reconstructed spatial electromagnetic boundary parameters to reverse offset the polarization phase dispersion generated at the broadband edge. The spatial electromagnetic boundary parameters after superimposed reverse offset phase compensation are allocated to the corresponding physical array elements of each metasurface according to the physical spatial hierarchy of the multilayer tunable metasurface, and converted into step bias level amplitudes that directly drive the physical control terminals of the corresponding active control devices, thereby generating a multilayer spatial electromagnetic response parameter set.

[0010] Preferably, the method of converting the multilayer spatial electromagnetic response parameter set into a high-frequency digital pulse sequence and applying it to the multilayer tunable metasurface includes: Parallel high-frequency sawtooth waves are generated using a reference clock signal. The amplitude of each step bias level in the multi-layer spatial electromagnetic response parameter set is compared with the amplitude of the high-frequency sawtooth waves. Based on the comparison result, a level transition is triggered, which is then converted into a high-frequency digital pulse sequence. The high-frequency digital pulse sequence is amplified to convert it into a dynamic bias voltage waveform, and then applied directly to the physical control terminals of the corresponding active control devices in the multilayer tunable metasurface through physical wire paths.

[0011] Preferably, the method of introducing periodic time-varying non-reciprocal electromagnetic control on the time axis includes: By periodically driving the physical control terminals of each active control device using a dynamic bias voltage waveform, a periodic time-varying change in the physical surface impedance of the multilayer tunable metasurface is generated, thereby destroying the Lorentz reciprocal characteristics. By utilizing the periodically time-varying physical surface impedance, the cross-polarization parasitic component in the broadband microwave signal passing through a multilayer tunable metasurface is modulated to outside the preset receiving frequency band. The modulated broadband microwave signal is controlled to perform spatial radiation and echo capture on a multilayer tunable metasurface, and outputs a dual-polarized microwave radiation field and reflection field signal.

[0012] Preferably, the method of outputting the dual-channel transient baseband signal includes: A dual-polarized antenna array is used to capture dual-polarized microwave radiation and reflection signals from space, and the dual-polarized microwave radiation and reflection signals are separated into two spatially orthogonal radio frequency signals through an orthogonal coupling path; The two spatially orthogonal RF signals are respectively amplified by low noise linear power, and then mixed, filtered and down-converted using a shared local oscillator signal to convert them into two orthogonal analog baseband signals. Two orthogonal analog baseband signals are synchronously converted and digitized under the control of the same sampling clock, and then assembled in parallel to output dual-channel transient baseband signals.

[0013] Preferably, the method for performing time-frequency coherent demodulation and dispersion phase compensation on the dual-channel transient baseband signal includes: Based on the current linear frequency modulation slope of the radar, a time-frequency coherent reference signal is generated that follows the frequency jump. The time-frequency coherent reference signal and the dual-channel transient baseband signal are multiplied by complex frequency modulation and stripped to cancel the linear frequency modulation component in the dual-channel transient baseband signal and convert it into an orthogonal dual-channel beat frequency signal. The orthogonal dual-channel beat frequency signal is sent into the all-pass filter path. The physical time delay compensation amount, which is opposite to the nonlinear polarization phase dispersion phase generated at the edge of the broadband, is used to re-normalize and flatten the orthogonal dual-channel beat frequency signal to obtain the corrected dual-channel beat frequency signal.

[0014] Preferably, the method for outputting fully decoupled high-dynamic dual-polarization radar target data includes: The corrected dual-channel beat frequency signal is split into mutually independent horizontal polarization beat frequency characteristic components and vertical polarization beat frequency characteristic components. The vertical polarization beat frequency characteristic component is physically flipped and injected into the horizontal polarization beat frequency characteristic component by using the phase inversion drive path, thereby canceling the residual cross-polarization crosstalk components between the horizontal polarization channel and the vertical polarization channel, and obtaining the component stripped horizontal polarization beat frequency characteristic component and vertical polarization beat frequency characteristic component. The horizontally polarized beat frequency characteristic components and the vertically polarized beat frequency characteristic components after component separation are time-series recombined to output fully decoupled high-dynamic dual-polarization radar target data.

[0015] Preferably, the method for generating the reverse offset phase value with the opposite sign to the nonlinear phase deviation value includes: The nonlinear phase deviation value is decomposed into an ionospheric dispersion-type deviation component and a Lorentz-type deviation component; By adjusting the equivalent medium physical thickness between the layers in the multilayer tunable metasurface, group delay compensation is introduced to counteract the ionospheric dispersion bias component. By adjusting the imaginary part of the impedance of each layer in the multilayer tunable metasurface, phase jump compensation is introduced to counteract the Lorentz-type deviation component; the group delay compensation and phase jump compensation are vector-superimposed to synthesize the reverse counteracting phase value.

[0016] The beneficial effects of this invention are as follows: By jointly controlling the interlayer physical thickness and the imaginary part of the surface impedance of the multilayer metasurface at the transmitting end to synthesize the reverse counter-phase value, and by using dynamic voltage drive to destroy the Lorentz reciprocal characteristics, and by performing time-frequency coherent renormalization and cross-channel crosstalk adaptive cancellation on the dual-channel echo signal at the receiving end, waveform pre-distortion of the physical layer of the radiating front end and multi-dimensional decoupling of the signal layer of the receiving back end are realized. This acts on the data closed loop of the entire radar antenna transmission and reception process, and is used to perform all-round coherent purification and spurious interception of radar electromagnetic signals in space, time and polarization domain. It suppresses the broadband dispersion defects of the metasurface and transient electromagnetic hysteresis crosstalk, and obtains high dynamic dual-polarization radar target data with extremely high polarization isolation and complete decoupling. Attached Figure Description

[0017] To more clearly illustrate the technical solutions of the embodiments of the present invention, the drawings used in the following description of the embodiments will be briefly introduced. Obviously, the drawings described below are only some embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0018] Figure 1 This is a schematic diagram of the broadband dual-polarization radar antenna system based on multilayer metasurfaces in this invention.

[0019] Figure 2 This is a flowchart of the process for generating the dynamic bias voltage waveform in this invention.

[0020] Figure 3 This is a flowchart for outputting completely decoupled target data in this invention.

[0021] Figure 4 This is a flowchart for generating the reverse offset phase value in this invention. Detailed Implementation

[0022] To make the above-mentioned objects, features and advantages of the present invention more apparent and understandable, the specific embodiments of the present invention will be described in detail below with reference to the accompanying drawings.

[0023] Many specific details are set forth in the following description in order to provide a full understanding of the invention. However, the invention may also be practiced in other ways different from those described herein, and those skilled in the art can make similar extensions without departing from the spirit of the invention. Therefore, the invention is not limited to the specific embodiments disclosed below.

[0024] Secondly, the term "one embodiment" or "embodiment" as used herein refers to a specific feature, structure, or characteristic that may be included in at least one implementation of the present invention. The phrase "in one embodiment" appearing in different places in this specification does not necessarily refer to the same embodiment, nor is it a single or selective embodiment that is mutually exclusive with other embodiments.

[0025] Reference Figure 1 , Figure 2 , Figure 3 and Figure 4 This is one embodiment of the present invention, which provides a broadband dual-polarization radar antenna system based on a multilayer metasurface, comprising the following steps: The methods for outputting the set of working parameters and boundary conditions include: The system captures physical operating status commands issued by the radar RF front-end in real time. These commands are binary encoded data frames periodically sent by the radar RF front-end to the multi-layer adjustable metasurface controller via a digital control bus. The digital control bus is exemplarily a serial peripheral interface bus. The data frames carry the radar's current center frequency, operating bandwidth, and linear frequency modulation pulse parameters corresponding to the target polarization mode in different field positions according to a preset communication protocol. The preset communication protocol exemplarily defines the first to fourth bytes of the data frame as the frame header identification field, the fifth to eighth bytes as the center frequency field, the ninth to twelfth bytes as the operating bandwidth field, the thirteenth to twentieth bytes as the linear frequency modulation pulse parameter field, and the twentieth to twentieth bytes as the checksum field.

[0026] After the data frame is converted from serial to parallel, the binary encoded data corresponding to the center frequency field, working bandwidth field and linear frequency modulation pulse parameter field are extracted and converted into decimal values ​​according to the start address and field length of each field defined in the preset communication protocol. The linear frequency modulation pulse parameters include, for example, the linear frequency modulation start frequency, the linear frequency modulation end frequency, the linear frequency modulation slope and the pulse repetition period. The target polarization mode is, for example, a dual polarization mode in which horizontal polarization and vertical polarization are transmitted simultaneously.

[0027] The transient feedback power of each layer of the multilayer tunable metasurface and the initial isolation between channels are acquired synchronously. Specifically, the multilayer tunable metasurface is composed of multiple metasurfaces stacked sequentially along the microwave propagation direction. A miniature electromagnetic probe is placed in the gap between two adjacent metasurfaces. The induced voltage signal output by the miniature electromagnetic probe is amplified by a logarithmic amplifier and converted into a power level value to obtain the transient feedback power of each metasurface. Directional couplers are set at the input and output ports of the multilayer tunable metasurface. The initial isolation between channels is obtained by measuring the crosstalk power difference between the horizontally polarized and vertically polarized signals at the input and output ports, exemplarily expressed in decibels.

[0028] The radar's current center frequency, operating bandwidth, linear frequency modulation pulse parameters, transient feedback power of each metasurface layer, and initial isolation between channels are sampled and digitized synchronously. Specifically, the global clock inside the multi-layer adjustable metasurface controller is used as the synchronization reference clock. At the rising edge of each sampling clock, the register values ​​corresponding to the above parameters are simultaneously latched, and the analog-to-digital conversion of each power level value and crosstalk power difference is simultaneously triggered. All data obtained after time synchronization sampling and digitization conversion are encapsulated and combined according to a preset data frame format. The preset data frame format is exemplarily defined by starting with a frame header identifier, sequentially arranging each parameter field, and ending with a checksum field. The encapsulated and combined data frame is output as a set of working parameters and boundary conditions.

[0029] Methods for reconstructing the spatial electromagnetic boundary parameters of each metasurface layer include: The radar's current center frequency is matched with the pre-stored electromagnetic parameter table. The pre-stored electromagnetic parameter table is a three-dimensional lookup table that is stored in the non-volatile memory inside the multi-layer tunable metasurface controller. The pre-stored electromagnetic parameter table uses the center frequency, transient feedback power, and initial isolation between channels as the three index dimensions, and uses the real part of the surface impedance, the imaginary part of the surface impedance, the equivalent permittivity, and the equivalent permeability of each metasurface in the multi-layer tunable metasurface as the stored data. The pre-stored electromagnetic parameter table is constructed as follows: before the multilayer tunable metasurface leaves the factory, the surface impedance and equivalent electromagnetic parameters of each layer of the metasurface are measured layer by layer under different center frequencies, different feed powers and different initial isolation conditions using a vector network analyzer, and the measurement results are written into a lookup table according to a three-dimensional index. The basic electromagnetic property parameters of the multilayer tunable metasurface are obtained as follows: the current center frequency of the radar in the set of operating parameters and boundary conditions is used as the first dimension index, the average value of the transient feedback power of each metasurface layer is used as the second dimension index, and the initial isolation between channels is used as the third dimension index. The grid node that is closest to the three index values ​​is found in the pre-stored electromagnetic parameter table, and the real part of the surface impedance, the imaginary part of the surface impedance, the equivalent permittivity and the equivalent permeability of each metasurface layer corresponding to the grid node are read as the basic electromagnetic property parameters. The transient feedback power of each metasurface layer is compared layer by layer with the reference power value corresponding to that metasurface layer in the pre-stored electromagnetic parameter table, and the percentage power deviation between the two is calculated. Multiply the power deviation percentage by the real and imaginary parts of the surface impedance of the metasurface layer in the basic electromagnetic characteristic parameters to obtain the correction amount of the surface impedance. Simultaneously, the initial isolation degree between channels acquired in real time is compared with the corresponding reference isolation degree value in the pre-stored electromagnetic parameter table, the isolation degree deviation value is calculated, the isolation degree deviation value is divided by 20 and then the result is converted into the change of coupling coefficient by a power of 10, and the change of coupling coefficient is multiplied by the equivalent dielectric constant between two adjacent metasurfaces in the basic electromagnetic characteristic parameters for coupling correction. The modified real part of surface impedance, imaginary part of surface impedance, equivalent permittivity and equivalent permeability are used as the surface electromagnetic boundary response benchmark under the current hardware physical state. The surface electromagnetic boundary response benchmark reflects the true electromagnetic response characteristics of each layer of metasurface under the current center frequency, current feed power and current isolation.

[0030] The surface electromagnetic boundary response reference is mapped to discrete frequency points along the working bandwidth. The process of discrete frequency point mapping is as follows: the current center frequency of the radar in the set of working parameters and boundary conditions is used as the reference center frequency point, the working bandwidth is used as the frequency mapping range, and the discrete frequency points are divided within the frequency range of the working bandwidth at fixed frequency intervals. The fixed frequency interval is exemplarily taken as one-twentieth of the working bandwidth. For each discrete frequency point, the real part of the surface impedance, the imaginary part of the surface impedance, the equivalent permittivity and the equivalent permeability of each layer of metasurface corresponding to that frequency point in the surface electromagnetic boundary response reference are arranged in the order of the physical spatial hierarchy of each layer of metasurface along the microwave propagation direction to form the spatial electromagnetic boundary vector corresponding to that discrete frequency point. Arrange the spatial electromagnetic boundary vectors corresponding to all discrete frequency points within the working bandwidth into a two-dimensional matrix in order of frequency from low to high. The row direction of the two-dimensional matrix corresponds to the spatial layer dimension of each metasurface layer, and the column direction corresponds to the frequency dimension of the working bandwidth. The two-dimensional matrix formed by the arrangement is the spatial electromagnetic boundary control matrix. Each matrix element of the spatial electromagnetic boundary control matrix represents the electromagnetic boundary control parameter of a specific metasurface layer at a specific frequency point, thus obtaining the reconstructed spatial electromagnetic boundary parameters.

[0031] Modern radars, when using multi-layer metasurfaces for dual polarization modulation in a broadband configuration, inevitably experience severe nonlinear polarization phase dispersion at the edge of their operating bandwidth due to the physical limitations of the unit's quality factor. Furthermore, the hardware's high-speed time-varying switching processes induce microscopic transient lags in the release of electromagnetic energy, leading to cross-channel polarization signal adhesion and severe crosstalk, making it impossible for backend algorithms to accurately extract clean orthogonal polarization components. Therefore, this invention addresses this issue by introducing dispersion physical offsetting, three-dimensional spatial hierarchical mapping, and time-varying non-reciprocal out-of-band shift modulation at the transmission front end. This achieves three-dimensional interception and adaptive purification of cross-polarization parasitic signals at both the physical and frequency domains. The specific steps are as follows: Methods for counteracting polarization phase dispersion generated at the edges of a wide bandwidth include: The nonlinear phase deviation value of the reconstructed spatial electromagnetic boundary parameters at the edge frequency of the working bandwidth is extracted by locating the lower and upper edge frequency points of the working bandwidth along the frequency dimension in the spatial electromagnetic boundary control matrix, extracting the imaginary part of the surface impedance of each metasurface layer and converting it into the actual phase response value through arctangent operation, and calculating the difference between the actual phase response value and the ideal phase value at the corresponding frequency point on the ideal linear phase response curve to obtain the nonlinear phase deviation value. The magnitude of polarization phase dispersion and the tilt angle of the polarization axis are locked. The locking method is as follows: the magnitude of polarization phase dispersion is obtained by calculating the difference between the nonlinear phase deviation values ​​between the horizontal and vertical polarization channels at the lower and upper edge frequency points. The polarization phase dispersion magnitude is divided by the angle between the orthogonal polarization axes of the two channels to obtain the tilt angle of the polarization axis. The sign of the nonlinear phase deviation value is reversed while keeping its amplitude unchanged, that is, the nonlinear phase deviation value is multiplied by negative 1 to obtain the reverse offset phase value. The reverse offset phase value and the nonlinear phase deviation value are positive and negative mirror images of each other about the zero phase axis on the phase plane.

[0032] The reverse offset phase value is directly superimposed and compensated into the reconstructed spatial electromagnetic boundary parameters. The superposition compensation method is as follows: in the spatial electromagnetic boundary control matrix, the imaginary part of the surface impedance of each layer of metasurface corresponding to the lower edge frequency point and the upper edge frequency point is added to the reverse offset phase value corresponding to that frequency point. The summed imaginary part of the surface impedance is the imaginary part of the surface impedance after reverse offset compensation. The actual phase response generated by the compensated imaginary part of the surface impedance at the broadband edge frequency point is close to the corresponding ideal phase value on the ideal linear phase response curve. In this way, the polarization phase dispersion generated at the broadband edge is reverse offset, so that the phase consistency deviation between the horizontal polarization channel and the vertical polarization channel in the entire working bandwidth is compressed to the preset allowable range.

[0033] The electromagnetic boundary control parameters of each layer of metasurface corresponding to each row in the spatial electromagnetic boundary control matrix are mapped row by row to the three-dimensional spatial coordinates of the corresponding metasurface physical array element according to the correspondence between the row index and the physical arrangement order of the metasurface physical array element along the microwave propagation direction. The electromagnetic boundary control parameters allocated to each metasurface physical element are converted into step bias level amplitudes that directly drive the physical control terminals of the corresponding active control devices. The active control devices are varactor diodes integrated on each layer of metasurface physical elements. The correspondence curve between the reverse bias voltage and junction capacitance of the varactor diodes is obtained in advance through offline calibration. The imaginary part of the surface impedance allocated to each metasurface physical element is mapped to the desired junction capacitance target value. Then, according to the correspondence curve, the junction capacitance target value is reverse-searched and mapped to the corresponding reverse bias voltage amplitude, which is the step bias level amplitude. The step bias level amplitudes corresponding to all metasurface physical array elements are arranged in physical spatial hierarchy order to generate a multi-layer spatial electromagnetic response parameter set.

[0034] Methods for converting multilayer spatial electromagnetic response parameter sets into high-frequency digital pulse sequences and applying them to multilayer tunable metasurfaces include: Parallel high-frequency sawtooth waves are generated using a reference clock signal. The reference clock signal is generated by a high-frequency FPGA clock source inside the multilayer adjustable metasurface controller. For example, the output frequency of the high-frequency FPGA clock source is 10 times the repetition frequency of the radar linear frequency modulation pulse. The parallel high-frequency sawtooth wave generation method is as follows: multiple parallel digital sawtooth wave generators are set inside the high-frequency FPGA, the number of which is equal to the number of step bias level amplitudes in the multilayer spatial electromagnetic response parameter set. Each digital sawtooth wave generator uses the reference clock signal as the counting clock to accumulate at a fixed step size. The high-order output of the accumulation register is connected to the digital-to-analog converter to obtain the sawtooth wave voltage waveform. The sawtooth wave voltage waveform rises linearly from the lowest level to the highest level and then automatically resets. The amplitude of each step bias level in the multi-layer spatial electromagnetic response parameter set is compared with the amplitude of the high-frequency sawtooth wave. When the instantaneous voltage amplitude of the high-frequency sawtooth wave is greater than or equal to the amplitude of the step bias level, a high level is output; otherwise, a low level is output. Each channel outputs a high-frequency digital pulse signal whose pulse width changes linearly with the amplitude of the step bias level. All channels output in parallel to form a high-frequency digital pulse sequence.

[0035] Each channel signal of the high-frequency digital pulse sequence is connected to a multi-channel high-speed power amplifier circuit for voltage amplitude amplification. For example, the input logic level of 3.3 volts is amplified to the driving level of 0 to 30 volts required by the varactor diode. After low-pass filtering and smoothing, it is converted into a dynamic bias voltage waveform. The dynamic bias voltage waveform is directly applied to the physical control terminals of the corresponding active control devices in the multilayer tunable metasurface through physical wire paths. Each dynamic bias voltage waveform is connected to the cathode of the varactor diode through an independent physical wire path, and the anode of the varactor diode is grounded. The dynamic bias voltage waveform directly controls the reverse bias voltage of the varactor diode, adjusting the junction capacitance of the varactor diode in real time and changing the surface impedance of the corresponding metasurface physical element. For example, when the reverse bias voltage output by the dynamic bias voltage waveform is 15 volts, the junction capacitance of the varactor diode is 0.5 picofarads, making the imaginary part of the surface impedance of the corresponding metasurface physical element inductive and the equivalent inductance 2 nanohenries; when the reverse bias voltage is reduced to 3 volts, the junction capacitance increases to 2 picofarads, making the imaginary part of the surface impedance of the corresponding metasurface physical element capacitive and the equivalent capacitance 1.5 picofarads.

[0036] Methods for introducing periodic time-varying non-reciprocating electromagnetic control on the time axis include: The physical control terminals of each active control device are periodically driven by a dynamic bias voltage waveform. The dynamic bias voltage waveform is a square wave signal with an adjustable duty cycle that is continuously output with the reference clock signal of the high-frequency FPGA clock source as the beat. Each square wave signal cycle contains a high level and a low level. The high level corresponds to the reverse bias voltage of the varactor diode as the first voltage value, and the low level corresponds to the reverse bias voltage as the second voltage value. The amplitude difference between the first voltage value and the second voltage value causes the junction capacitance of the varactor diode to switch back and forth between the two capacitance values. For example, the first voltage value is 15V, the second voltage value is 3V, the square wave frequency is 10MHz, and the junction capacitance of the varactor diode changes periodically between 0.5pF and 2pF at a rate of 10MHz. At the rising and falling edges of each square wave signal, the junction capacitance of the varactor diode undergoes a transient jump. This transient jump in junction capacitance is coupled to the equivalent circuit of the metasurface physical array element through the varactor diode, causing a synchronous transient jump in surface impedance on each layer of the metasurface physical array element of the multilayer tunable metasurface. The jump period of the surface impedance is equal to the square wave period of the dynamic bias voltage waveform. Because the surface impedance jumps repeat at a fixed period on the time axis, the physical surface impedance of the multilayer tunable metasurface produces periodic time-varying abrupt changes. These periodic time-varying abrupt changes cause the time-varying impedance boundary conditions experienced by the microwave signal in the forward and reverse propagation paths to become asymmetrical. The surface impedance distribution at a certain moment in the forward propagation path is different from the surface impedance distribution at the same moment in the reverse propagation path, thereby destroying the Lorentz reciprocal characteristics of electromagnetic waves.

[0037] By utilizing the periodically time-varying physical surface impedance, the cross-polarization parasitic component in a broadband microwave signal passing through a multilayer tunable metasurface is modulated outside a preset receiving frequency band. The modulation method is as follows: the broadband microwave signal contains the main polarization component of the target polarization mode and the cross-polarization parasitic component of the non-target polarization direction. When the broadband microwave signal passes through the multilayer tunable metasurface, the periodically time-varying physical surface impedance is equivalent to superimposing a time-varying modulation function on the microwave signal transmission path. ;in, It is a time-varying modulation function, with an example value of or The periodic square wave function, It is a sign function, taking the value when the internal independent variable is positive. When it is negative, take , It is the modulation frequency. It is a time variable, with the unit being seconds.

[0038] The modulation frequency of the time-varying modulation function is equal to the square wave frequency of the dynamic bias voltage waveform. The time-varying modulation function is a product-type modulation transfer function that periodically switches the reflection phase and transmission phase of the surface impedance at the square wave frequency on the time axis. The complex amplitude of the cross-polarization parasitic component is multiplied with the time-varying modulation function in the time domain. After being processed by a time-varying modulation function, the cross-polarized parasitic component generates upper and lower sideband spectrum shift components on its original spectrum, with the modulation frequency as the interval. By using a preset modulation frequency (the preset modulation frequency is exemplarily set to be more than twice the operating bandwidth) that is greater than the receiving bandwidth of the radar receiver, the upper and lower sideband frequencies of the spectrum-shifted cross-polarized parasitic component fall outside the preset receiving frequency band. For example, if the radar's current center frequency is 10 GHz, the operating bandwidth is 1 GHz, the preset receiving frequency band is 9.5 GHz to 10.5 GHz, and the modulation frequency is set to 2 GHz, then the original spectrum of the cross-polarized parasitic component is shifted to below 8.5 GHz and above 11.5 GHz, far away from the preset receiving frequency band, and cannot be captured by the subsequent receiving link.

[0039] The modulated broadband microwave signal is controlled to perform spatial radiation and echo capture on a multi-layer tunable metasurface. The control method is as follows: after the broadband microwave signal emitted by the radar radio frequency front end passes through the multi-layer tunable metasurface, the main polarization component maintains its original spectral characteristics within a preset receiving frequency band (exemplarily the radar's current center frequency ± half of the working bandwidth, i.e., 9.5GHz to 10.5GHz) and radiates into space along the target polarization direction to form a dual-polarized microwave radiation field signal. After the dual-polarized microwave radiation field signal is transmitted to the target object in space, it generates a reflected echo. The reflected echo passes through the multi-layer tunable metasurface again and is captured by the radar receiving antenna to form a reflected field signal. The cross-polarized parasitic component is modulated outside the receiving frequency band and is filtered out by the bandpass filter at the front end of the radar receiver, so it no longer causes crosstalk to the main polarization component. Thus, the dual-polarized microwave radiation field and reflected field signals are output. The horizontal polarization channel and the vertical polarization channel in the dual-polarized microwave radiation field and reflected field signals achieve polarization isolation enhanced by time-domain modulation. For example, after time-domain modulation enhancement, the isolation between the horizontal polarization channel and the vertical polarization channel is increased from 25dB before modulation to 45dB after modulation.

[0040] By employing a design that combines "front-end hardware decoupling, spatial hierarchy mapping, and non-reciprocal time-domain modulation," the inherent bandwidth edge phase distortion of multilayer metasurface structures is fundamentally reversed, achieving extremely high phase consistency across wideband dual channels. Furthermore, the traditional Lorentz reciprocity limitation is broken, successfully and adaptively shifting previously extremely difficult-to-handle cross-channel parasitic crosstalk sources outside the radar receiving bandwidth. This closed-loop design, combining pre-stage physical predistortion with spatiotemporal joint coding, constructs a rigid polarization physical protection network at the microwave radiation front end, improving the polarization purity of the antenna radiation field, eliminating dynamic electromagnetic pollution, and enhancing inter-channel polarization isolation.

[0041] The methods for outputting dual-channel transient baseband signals include: The dual-polarized antenna array is used to capture dual-polarized microwave radiation and reflection signals from space. The dual-polarized antenna array consists of a set of antenna elements polarized along the horizontal polarization direction and the vertical polarization direction, respectively. The horizontally polarized antenna element only generates induced current for the component of the electric field vector in the space electromagnetic wave that vibrates along the horizontal direction, and the vertically polarized antenna element only generates induced current for the component of the electric field vector that vibrates along the vertical direction. The dual-polarized microwave radiation field and reflection field signals are separated into two spatially orthogonal radio frequency signals through an orthogonal coupling path. The orthogonal coupling path consists of two mutually isolated radio frequency transmission lines located at the output end of the dual-polarized antenna array. The induced current output by the horizontally polarized antenna unit enters the horizontally polarized receiving link through the first radio frequency transmission line. The induced current output by the vertical polarization antenna unit enters the vertical polarization receiving link through the second radio frequency transmission line. An isolation barrier or shielding layer is set between the two radio frequency transmission lines. For example, the isolation between the two radio frequency transmission lines is greater than 40dB. The two spatially orthogonal radio frequency signals carry horizontal polarization component information and vertical polarization component information, respectively.

[0042] The radio frequency signal in the horizontal polarization receiving link is sent to the first low-noise amplifier circuit, and the radio frequency signal in the vertical polarization receiving link is sent to the second low-noise amplifier circuit. While amplifying the power of radio frequency signals, extremely low additional noise is introduced. For example, the noise figure of the low-noise amplifier is less than 1.5dB and the gain is 25dB. A single-frequency continuous sine wave is generated by a local oscillator signal source as a shared local oscillator signal. The frequency of the shared local oscillator signal is equal to the current center frequency of the radar. The shared local oscillator signal is divided into two local oscillator signal components with the same frequency and phase by a power divider, and then sent to the local oscillator input terminals of the first mixer circuit and the second mixer circuit, respectively. The first mixer circuit mixes and multiplies the amplified radio frequency signal in the horizontal polarization link with the first local oscillator signal component to generate a sum frequency component and a difference frequency component. After the sum frequency component is filtered out by a low-pass filter, the difference frequency component is retained to obtain the analog baseband signal corresponding to the horizontal polarization link. The second mixer circuit mixes the amplified RF signal in the vertical polarization link with the second local oscillator signal component in the same way and performs low-pass filtering to obtain the analog baseband signal corresponding to the vertical polarization link. This is then converted into two orthogonal analog baseband signals, the frequency range of which is exemplarily from 0Hz to half of the operating bandwidth.

[0043] Two orthogonal analog baseband signals are synchronously converted and digitized under the control of the same sampling clock. A sampling clock signal is generated by a common sampling clock source. For example, the frequency of the sampling clock signal is 2.5 times the working bandwidth. The sampling clock signal is simultaneously sent to the first analog-to-digital conversion channel and the second analog-to-digital conversion channel. The analog baseband signals of the horizontal polarization link and the vertical polarization link are sampled and quantized at equal intervals on the same clock edge. The horizontally polarized baseband digital signal and the vertically polarized baseband digital signal at the same sampling time are combined into a dual-channel sampling data pair. For example, the high 16 bits store the horizontally polarized channel sampling value and the low 16 bits store the vertically polarized channel sampling value. The dual-channel sampling data pairs at all sampling times are output sequentially in chronological order to form a dual-channel transient baseband signal.

[0044] Traditional dual-polarization radars often struggle to effectively eliminate dynamic adhesion and phase distortion of cross-channel polarization signals when facing edge phase dispersion introduced by metasurfaces and transient spurious signals generated by high-speed switching. This is often due to a lack of accurate physical correspondence between time delays and backend algorithms, making it difficult to focus the echo pulses. Therefore, this invention constructs a time-frequency coherent reference signal based on the current frequency modulation slope for frequency modulation stripping, and utilizes the full-pass filter re-normalization delay difference combined with phase inversion drive injection to counteract crosstalk across channels. The specific steps are as follows: Methods for time-frequency coherent demodulation and dispersion phase compensation of dual-channel transient baseband signals include: Based on the current linear frequency modulation slope of the radar, a time-frequency coherent reference signal that follows the frequency jump is generated. The linear frequency modulation slope is extracted from the linear frequency modulation pulse parameters in the set of operating parameters and boundary conditions. For example, the linear frequency modulation slope is equal to the difference between the linear frequency modulation termination frequency and the linear frequency modulation start frequency divided by the pulse repetition period. The method for generating the time-frequency coherent reference signal that follows the frequency jump is as follows: using the sampling timestamp sequence of the dual-channel transient baseband signal as the time axis, at each sampling timestamp, the linear frequency modulation slope is multiplied by the time offset of that sampling timestamp relative to the start time of the pulse repetition period to obtain the instantaneous frequency modulation frequency at that sampling timestamp. The complex sine signal corresponding to the instantaneous frequency modulation is generated using the instantaneous frequency modulation as the angular frequency. The real part of the complex sine signal is the cosine component, and the imaginary part is the sine component. The complex sine signals at all sampling timestamps are arranged in chronological order to form a time-frequency coherent reference signal.

[0045] The time-frequency coherent reference signal and the dual-channel transient baseband signal are separated by complex multiplication and frequency modulation. The method of complex multiplication and frequency modulation separation is as follows: the horizontal polarization baseband digital signal and the vertical polarization baseband digital signal in the dual-channel transient baseband signal are each a complex baseband signal. The complex baseband signal takes the sampling timestamp as the independent variable and the complex sample value as the dependent variable. For each sampling timestamp, the complex sampled value of the horizontal polarization baseband digital signal and the complex conjugate value of the time-frequency coherent reference signal at that sampling timestamp are multiplied by a complex number. The result of the complex multiplication is the beat frequency signal after stripping the linear frequency modulation of the horizontal polarization channel. Simultaneously, the complex sampled value of the vertical polarization baseband digital signal and the complex conjugate value of the time-frequency coherent reference signal at the sampling timestamp are multiplied by a complex number to obtain the beat frequency signal after stripping the linear frequency modulation of the vertical polarization channel. Complex multiplication cancels out the time-varying frequency component introduced by linear frequency modulation in the dual-channel transient baseband signal with the conjugate frequency component in the time-frequency coherent reference signal. The remaining frequency component after cancellation contains only the constant beat frequency caused by the distance delay of the target object and the residual phase error caused by the dispersion effect of the multilayer tunable metasurface. Thus, the dual-channel transient baseband signal is transformed into an orthogonal dual-channel beat frequency signal, which includes a horizontally polarized beat frequency signal and a vertically polarized beat frequency signal.

[0046] The quadrature dual-channel beat frequency signal is sent into the all-pass filter path. The all-pass filter path is a set of phase correction filter paths whose amplitude frequency response remains flat within the working frequency band and introduces different physical time delays only to different frequency components of the signal. The polarization phase dispersion magnitude between the horizontal and vertical polarization channels at broadband edge frequencies is extracted from the multi-layer spatial electromagnetic response parameter set. The polarization phase dispersion magnitude is then converted into the corresponding physical time delay compensation amount. The conversion method is as follows: divide the polarization phase dispersion magnitude by 2π and then divide by the instantaneous frequency modulation frequency corresponding to that frequency point to obtain the physical time delay compensation amount in seconds; or equivalently, divide the polarization phase dispersion magnitude directly by the linear frequency modulation slope, with the unit of the linear frequency modulation slope being Hertz per second, and the result is the physical time delay compensation amount in seconds. In the all-pass filter path, half of the positive physical delay compensation is applied to the horizontally polarized beat signal, and half of the negative physical delay compensation is applied to the vertically polarized beat signal. This ensures that the time delay difference between the horizontally and vertically polarized beat signals at the output of the all-pass filter path equals the negative physical delay compensation, thereby canceling the dispersion delay difference between the two channels caused by broadband edge polarization phase dispersion. After time delay reshaping and leveling, the phase difference between the horizontally and vertically polarized beat signals at the same sampling timestamp is restored to the theoretical value corresponding to the actual scattering characteristics of the target object, and the corrected dual-channel beat signal is output.

[0047] Methods for outputting fully decoupled high-dynamic dual-polarization radar target data include: The corrected dual-channel beat frequency signal is split into two independent horizontal polarization beat frequency characteristic components and vertical polarization beat frequency characteristic components. The splitting method is as follows: the corrected dual-channel beat frequency signal contains two independent complex signal sequences, namely the horizontal polarization corrected beat frequency signal and the vertical polarization corrected beat frequency signal. The complex sampled value sequence of the horizontal polarization corrected beat frequency signal is directly extracted as the horizontal polarization beat frequency characteristic component, and the complex sampled value sequence of the vertical polarization corrected beat frequency signal is directly extracted as the vertical polarization beat frequency characteristic component.

[0048] The vertical polarization beat frequency characteristic component is physically flipped and injected into the horizontal polarization beat frequency characteristic component by using the phase inversion drive path to cancel the cross-polarization crosstalk component remaining between the horizontal polarization channel and the vertical polarization channel. The cancellation method is: the phase inversion drive path is a digital signal processing path that multiplies the complex sampled value of the vertical polarization beat frequency characteristic component by negative 1 and then superimposes it onto the horizontal polarization beat frequency characteristic component. For each sampling timestamp, the complex sampled value of the vertical polarization beat frequency characteristic component at that time is multiplied by a cross-coupling coefficient and then inverted to obtain the crosstalk cancellation term. The cross-coupling coefficient is the residual cross-polarization crosstalk intensity between the horizontal polarization channel and the vertical polarization channel. The cross-coupling coefficient is obtained by offline calibration measurement, and its value is a dimensionless complex number between 0.01 and 0.05. The crosstalk cancellation term is added to the complex sampled value of the horizontal polarization beat frequency characteristic component at the sampling timestamp. The result of the addition is the horizontal polarization beat frequency characteristic component after removing cross-polarization crosstalk. Similarly, the complex sampled value of the horizontal polarization beat frequency characteristic component at that moment is multiplied by the same cross-coupling coefficient and then inverted. The inverted result is then added to the complex sampled value of the vertical polarization beat frequency characteristic component at that sampling timestamp to obtain the vertical polarization beat frequency characteristic component after removing cross-polarization crosstalk. This yields the horizontal polarization beat frequency characteristic component and the vertical polarization beat frequency characteristic component after component removal. After component removal, the cross-polarization crosstalk residual between the two channels is further compressed. For example, after component removal, the cross-polarization isolation is further improved from 45dB to over 55dB.

[0049] The real and imaginary parts of the horizontally polarized beat frequency characteristic components after component stripping are arranged in the order of sampling timestamps to form the first set of time-domain data sequences, and the real and imaginary parts of the vertically polarized beat frequency characteristic components after component stripping are arranged in the order of sampling timestamps to form the second set of time-domain data sequences. The first and second sets of time-domain data sequences are arranged into a frame structure according to the standard data output format of dual-polarization radar. For example, the current sampling timestamp, center frequency, and working bandwidth are added to the frame header of each frame. The frame body alternately arranges the horizontal polarization real part data, the horizontal polarization imaginary part data, the vertical polarization real part data, and the vertical polarization imaginary part data, ending with a frame tail check code. The time-reassembled data frame sequence is used as the fully decoupled high-dynamic dual-polarization radar target data output.

[0050] This receiver design, which combines time-frequency coherent renormalization with cross-channel phase inversion self-cancellation, not only transforms the residual phase deviation caused by broadband edge dispersion into deterministic physical time delay compensation, achieving perfect coherent focusing of the echo pulse, but also precisely offsets spatial depolarization crosstalk through phase inversion injection in the digital signal path. This digital purification scheme establishes a high-precision polarization isolation barrier at the back end, successfully enhancing polarization isolation to an extremely high level, achieving the effect of outputting highly dynamic, fully decoupled dual-polarization radar target data.

[0051] Methods for generating a counter-current phase value with the opposite sign to the nonlinear phase deviation value include: The nonlinear phase deviation value is decomposed into an ionospheric dispersion type deviation component and a Lorentz type deviation component. The decomposition method is as follows: the nonlinear phase deviation value is a nonlinear function of frequency, and in the frequency domain it presents a deviation curve that deviates from the ideal linear phase curve. The nonlinear phase deviation value is fitted with a polynomial along the frequency axis, and the first power term of the frequency in the polynomial is extracted as the ionospheric dispersion deviation component. The ionospheric dispersion deviation component is linearly related to the frequency and is a group delay error that increases or decreases monotonically with the frequency. This component is equivalent to the dispersion effect introduced by the change of electron density when the electromagnetic wave passes through the ionosphere. The higher-order terms of the polynomial, excluding the first-order terms, are extracted as Lorentz-type deviation components. These Lorentz-type deviation components exhibit a resonant shape with local peaks or valleys near specific frequencies. This component is equivalent to the phase jump error caused by the Lorentz resonant response of the metasurface resonant unit near the resonant frequency.

[0052] The equivalent physical thickness of the medium between the metasurfaces in a multilayer tunable metasurface is determined by the dielectric constant of the interlayer medium material and the physical spacing between the layers. The equivalent physical thickness of the medium is equal to the physical spacing between the layers multiplied by the square root of the relative dielectric constant of the interlayer medium material. Dividing the ionospheric dispersion bias component by the angular frequency change corresponding to the broadband edge frequency point yields the required group delay time compensation. For example, the ionospheric dispersion bias component is 0.5 radians, the upper edge frequency point angular frequency is 2π×10.5GHz, the lower edge frequency point angular frequency is 2π×9.5GHz, the angular frequency change is 2π×1GHz, and the group delay time is equal to 0.5 divided by 2π×1GHz, which is approximately 80 picoseconds. Based on the linear relationship between the group delay time and the equivalent physical thickness of the medium, the required adjustment amount for the equivalent physical thickness of the medium between layers is obtained by multiplying the group delay time by the propagation speed of electromagnetic waves in vacuum and then dividing by the square root of the equivalent dielectric constant of the interlayer medium. For example, the propagation speed of electromagnetic waves in vacuum is... Meters per second, the interlayer dielectric material is quartz with a relative permittivity of 4, and the equivalent dielectric physical thickness adjustment is equal to 80 picoseconds multiplied by 1. Meters per second divided by 2 equals approximately 12 micrometers; By adjusting the physical spacing between each metasurface layer using a microelectromechanical actuator, the physical spacing between the layers can be increased or decreased by half of the corresponding adjustment amount, while keeping the interlayer dielectric material unchanged. For example, the initial physical spacing between each metasurface layer is 100 micrometers, and the adjusted physical spacing becomes 112 micrometers or 88 micrometers, causing a corresponding change in the physical thickness of the equivalent dielectric. This introduces group delay compensation with the opposite sign to the ionospheric dispersion deviation component in the transmission path of electromagnetic waves through the multilayer tunable metasurface, and the two cancel each other out in the transmission path.

[0053] The adjustment amount for the imaginary part of the impedance is obtained by multiplying the phase jump deviation value of the Lorentz-type bias component at the broadband edge frequency by a pre-calibrated impedance imaginary part-phase sensitivity coefficient. For example, the phase jump deviation value of the Lorentz-type bias component at the upper edge frequency of 10.5 GHz is 0.3 radians. By performing a small-signal frequency sweep analysis of the metasurface layer near the resonant frequency using electromagnetic simulation software, the impedance imaginary part-phase sensitivity coefficient is obtained as 20 ohms per radian. Therefore, the required adjustment amount for the impedance imaginary part is 0.3 radians multiplied by 20 ohms per radian, which equals 6 ohms. By changing the reverse bias voltage of the varactor diode on the metasurface, the junction capacitance of the varactor diode is adjusted, thereby changing the imaginary part of the impedance of the corresponding metasurface physical element. For example, when the reverse bias voltage of the varactor diode is adjusted from the current 10V to 8V, the junction capacitance increases from 1.2pF to 1.8pF, and the imaginary part of the impedance of the corresponding metasurface physical element decreases from 50 ohms to 44 ohms, resulting in an adjustment of 6 ohms. This introduces a phase jump compensation with the opposite sign to the Lorentz-type deviation component on the reflection path of electromagnetic waves on the metasurface. The phase jump compensation is -0.3 radians, so that the two cancel each other out on the transmission path. The group delay compensation of 80 picoseconds multiplied by the angular frequency corresponding to the edge frequency of 10.5 GHz (2π × 10.5 GHz) is converted into the equivalent phase compensation amount. The equivalent phase compensation amount is equal to 80 times 10 to the power of negative 12 seconds multiplied by 2π times 10.5 times 10 to the power of 9 radians per second, which is approximately equal to 5.28 radians. The equivalent phase compensation of 5.28 radians and the phase jump compensation of -0.3 radians are combined into a complex vector in the electromagnetic wave complex amplitude space. The combination method is as follows: on the unit complex plane, the equivalent phase compensation of 5.28 radians is used as the argument of the first complex amplitude vector, and the phase jump compensation of -0.3 radians is used as the argument of the second complex amplitude vector. The first complex amplitude vector and the second complex amplitude vector are multiplied by a complex number. This is equivalent to algebraically superimposing the equivalent phase compensation and the phase jump compensation on the phase plane. The total argument of the resulting composite complex amplitude vector is calculated. The total argument is the sum of 5.28 radians and -0.3 radians, which equals 4.98 radians. The total argument of 4.98 radians is the phase angle of the composite reverse offset phase value. The reverse offset phase value is completely inversely matched with the original nonlinear phase deviation value in phase.

[0054] In summary, this invention achieves waveform predistortion at the radiating front-end physical layer and multidimensional decoupling of the receiving back-end signal layer by jointly controlling the interlayer physical thickness and the imaginary part of the surface impedance of the multilayer metasurface at the transmitting end, and by using dynamic voltage drive to destroy the Lorentz reciprocal characteristics, and by performing time-frequency coherent renormalization and adaptive crosstalk cancellation on the dual-channel echo signal at the receiving end. This results in a data closed loop that acts on the entire radar antenna transmission and reception process, enabling comprehensive coherent purification and spurious interception of radar electromagnetic signals in the spatial, temporal, and polarization domains. It suppresses the broadband dispersion defects and transient electromagnetic hysteresis crosstalk of the metasurface, and acquires highly dynamic dual-polarization radar target data with extremely high polarization isolation and complete decoupling.

[0055] It should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and are not intended to limit it. Although the present invention has been described in detail with reference to preferred embodiments, those skilled in the art should understand that modifications or equivalent substitutions can be made to the technical solutions of the present invention without departing from the spirit and scope of the technical solutions of the present invention, and all such modifications or substitutions should be covered within the scope of the claims of the present invention.

Claims

1. A broadband dual-polarization radar antenna system based on a multilayer metasurface, characterized in that, include: The sensing module captures radar operating parameters and interlayer state parameters of the multi-layer tunable metasurface in real time, and outputs a set of operating parameters and boundary conditions, specifically including: Real-time capture of physical operating status commands issued by the radar radio frequency front end, and extraction of the radar's current center frequency, operating bandwidth, and linear frequency modulated pulse parameters as the target polarization mode; The transient feedback power of each layer of the multilayer tunable metasurface and the initial isolation between channels are acquired simultaneously. The radar's current center frequency, operating bandwidth, linear frequency modulated pulse parameters, transient feedback power of each metasurface layer, and initial isolation between channels are time-synchronized sampled and digitally converted, and then packaged and combined into a set of operating parameters and boundary conditions. The solution module, based on the set of operating parameters and boundary conditions, reconstructs the spatial electromagnetic boundary parameters of each metasurface layer, counteracts the polarization phase dispersion generated at the broadband edges, and generates a multi-layer spatial electromagnetic response parameter set; the method of reconstructing the spatial electromagnetic boundary parameters of each metasurface layer includes: By connecting the radar’s current center frequency to the pre-stored electromagnetic parameter table, the basic electromagnetic characteristic parameters of the multilayer tunable metasurface are obtained. The basic electromagnetic characteristic parameters are then calibrated using transient feedback power and initial isolation between channels to lock the surface electromagnetic boundary response benchmark under the current hardware physical state. The surface electromagnetic boundary response reference is discretized along the working bandwidth and arranged to form a spatial electromagnetic boundary control matrix, thus obtaining the reconstructed spatial electromagnetic boundary parameters. The methods for counteracting polarization phase dispersion generated at the wideband edge include: Extract the nonlinear phase deviation value of the reconstructed spatial electromagnetic boundary parameters at the edge frequency of the working bandwidth, lock the magnitude of polarization phase dispersion and the tilt angle of polarization axis caused by the wide bandwidth edge, and generate an inverse counteracting phase value with the opposite sign to the nonlinear phase deviation value. The reverse offset phase value is directly superimposed and compensated into the reconstructed spatial electromagnetic boundary parameters to reverse offset the polarization phase dispersion generated at the broadband edge. The spatial electromagnetic boundary parameters after superimposed reverse counter-phase compensation are allocated to the corresponding physical array elements of each metasurface according to the physical spatial hierarchy of the multi-layer tunable metasurface, and converted into the step bias level amplitude that directly drives the physical control terminal of the corresponding active control device to generate a multi-layer spatial electromagnetic response parameter set. The modulation module converts the multi-layer spatial electromagnetic response parameter set into a high-frequency digital pulse sequence and applies it to the multi-layer tunable metasurface. It introduces periodic time-varying non-reciprocal electromagnetic control on the time axis and outputs dual-polarization microwave radiation field and reflection field signals. The receiving module digitizes the dual-polarization microwave radiation field and reflection field signals and outputs a dual-channel transient baseband signal. The decoupling module, combined with the current linear frequency modulation slope of the radar, performs time-frequency coherent demodulation and dispersion phase compensation on the dual-channel transient baseband signal, and performs component stripping along the orthogonal polarization axis to output fully decoupled high dynamic dual-polarization radar target data.

2. The broadband dual-polarization radar antenna system based on multilayer metasurfaces as described in claim 1, characterized in that, The method of converting a multilayer spatial electromagnetic response parameter set into a high-frequency digital pulse sequence and applying it to a multilayer tunable metasurface includes: Parallel high-frequency sawtooth waves are generated using a reference clock signal. The amplitude of each step bias level in the multi-layer spatial electromagnetic response parameter set is compared with the amplitude of the high-frequency sawtooth waves. Based on the comparison result, a level transition is triggered, which is then converted into a high-frequency digital pulse sequence. The high-frequency digital pulse sequence is amplified to convert it into a dynamic bias voltage waveform, and then applied directly to the physical control terminals of the corresponding active control devices in the multilayer tunable metasurface through physical wire paths.

3. The broadband dual-polarization radar antenna system based on multilayer metasurfaces as described in claim 1, characterized in that, The method of introducing periodic time-varying non-reciprocal electromagnetic control on the time axis includes: By periodically driving the physical control terminals of each active control device using a dynamic bias voltage waveform, a periodic time-varying change in the physical surface impedance of the multilayer tunable metasurface is generated, thereby destroying the Lorentz reciprocal characteristics. By utilizing the periodically time-varying physical surface impedance, the cross-polarization parasitic component in the broadband microwave signal passing through a multilayer tunable metasurface is modulated to outside the preset receiving frequency band. The modulated broadband microwave signal is controlled to perform spatial radiation and echo capture on a multilayer tunable metasurface, and outputs a dual-polarized microwave radiation field and reflection field signal.

4. The broadband dual-polarization radar antenna system based on multilayer metasurfaces as described in claim 1, characterized in that, The methods for outputting dual-channel transient baseband signals include: A dual-polarized antenna array is used to capture dual-polarized microwave radiation and reflection signals from space, and the dual-polarized microwave radiation and reflection signals are separated into two spatially orthogonal radio frequency signals through an orthogonal coupling path; The two spatially orthogonal RF signals are respectively amplified by low noise linear power, and then mixed, filtered and down-converted using a shared local oscillator signal to convert them into two orthogonal analog baseband signals. Two orthogonal analog baseband signals are synchronously converted and digitized under the control of the same sampling clock, and then assembled in parallel to output dual-channel transient baseband signals.

5. The broadband dual-polarization radar antenna system based on multilayer metasurfaces as described in claim 1, characterized in that, The methods for performing time-frequency coherent demodulation and dispersion phase compensation on dual-channel transient baseband signals include: Based on the current linear frequency modulation slope of the radar, a time-frequency coherent reference signal is generated that follows the frequency jump. The time-frequency coherent reference signal and the dual-channel transient baseband signal are multiplied by complex frequency modulation and stripped to cancel the linear frequency modulation component in the dual-channel transient baseband signal and convert it into an orthogonal dual-channel beat frequency signal. The orthogonal dual-channel beat frequency signal is sent into the all-pass filter path. The physical time delay compensation amount, which is opposite to the nonlinear polarization phase dispersion phase generated at the edge of the broadband, is used to re-normalize and flatten the orthogonal dual-channel beat frequency signal to obtain the corrected dual-channel beat frequency signal.

6. The broadband dual-polarization radar antenna system based on multilayer metasurfaces as described in claim 1, characterized in that, The methods for outputting fully decoupled high-dynamic dual-polarization radar target data include: The corrected dual-channel beat frequency signal is split into mutually independent horizontal polarization beat frequency characteristic components and vertical polarization beat frequency characteristic components. The vertical polarization beat frequency characteristic component is physically flipped and injected into the horizontal polarization beat frequency characteristic component by using the phase inversion drive path, thereby canceling the residual cross-polarization crosstalk components between the horizontal polarization channel and the vertical polarization channel, and obtaining the component stripped horizontal polarization beat frequency characteristic component and vertical polarization beat frequency characteristic component. The horizontally polarized beat frequency characteristic components and the vertically polarized beat frequency characteristic components after component separation are time-series recombined to output fully decoupled high-dynamic dual-polarization radar target data.

7. The broadband dual-polarization radar antenna system based on multilayer metasurfaces as described in claim 1, characterized in that, The methods for generating the reverse offset phase value with the opposite sign to the nonlinear phase deviation value include: The nonlinear phase deviation value is decomposed into an ionospheric dispersion-type deviation component and a Lorentz-type deviation component; By adjusting the equivalent medium physical thickness between the layers in the multilayer tunable metasurface, group delay compensation is introduced to counteract the ionospheric dispersion bias component. By adjusting the imaginary part of the impedance of each layer in the multilayer tunable metasurface, phase jump compensation is introduced to counteract the Lorentz-type deviation component; the group delay compensation and phase jump compensation are vector-superimposed to synthesize the reverse counteracting phase value.

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