Input phase-shift tuning method for dual-load quadrature dual-channel inductive power transfer system

Abstract

The application relates to the technical field of electric energy transmission, and particularly discloses an input phase-shifting tuning method for a dual-load orthogonal dual-energy-channel inductive electric energy transmission system. The method aims at the problem that the existing tuning technology is out of tune and the transmission performance is reduced when the position of a receiving end is shifted. The output voltage and output current of inverters of two energy channels are collected to obtain a voltage-current phase difference, and a smaller phase difference is taken as a feedback quantity and compared with a resonance setting value to determine whether the system is out of tune. When the system is out of tune, a main energy transmission channel is determined and current system parameters are obtained, an expected input voltage phase difference is calculated based on the target that the input reactive power of the main energy transmission channel is zero, and the PWM driving signal of the inverter of the second energy channel is adjusted according to the phase difference, so that the main energy transmission channel works in a resonance state. The method does not need to increase a variable inductor, an auxiliary coil or an additional power conversion circuit, and can realize tuning by adjusting the input voltage phase difference.

Description

Technical Field

[0001] This invention relates to the field of power transmission technology, and in particular to an input phase-shifting tuning method for a dual-load orthogonal dual-channel inductive power transmission system. Background Technology

[0002] Wireless power transfer (WPT) technology is widely used in consumer electronics, industrial robots, and medical devices due to its advantages such as flexible power supply and high reliability. Among them, inductive power transfer (IPT) technology, as an important branch of WPT, has received continuous attention from academia and industry due to its characteristics of not requiring physical connections and strong environmental adaptability. With the continuous expansion of application scenarios, the need for IPT systems to simultaneously power multiple terminal loads such as battery management systems and foreign object detection modules is becoming increasingly urgent, placing higher demands on the spatial adaptability and power supply freedom of the system.

[0003] Quadrature Double Channels (QDC) IPT systems, by constructing two spatially orthogonal power transmission channels and utilizing the synergistic effect of magnetic coupling between the channels, can effectively expand the power supply area and improve the flexibility of power supply to multiple loads, becoming one of the effective solutions to the aforementioned needs. However, in actual operation, the receiving mechanism often cannot maintain a perfectly aligned state with the transmitting mechanism. The positional deviation of the receiving end causes the magnetic coupling parameters such as coil self-inductance and mutual inductance to drift, causing the system to deviate from the preset resonant operating point. After the system is detuned, the inverter output reactive power increases, and the transmission efficiency decreases significantly, severely restricting the engineering application of QDC-IPT systems.

[0004] To address the detuning problem caused by receiver offset, existing tuning methods can be mainly categorized into the following three types:

[0005] The first category is the mechanical position adjustment method. This method uses auxiliary sensing devices (such as cameras or electromagnetic positioning) to obtain the relative position of the transceiver mechanism and drives the actuator to adjust the coil alignment state to restore system resonance. Although the principle of this type of solution is intuitive, it requires additional configuration of position detection and mechanical adjustment devices, resulting in slow system response speed, limited control accuracy, and mechanical wear and reliability issues, which greatly limits its practical application.

[0006] The second category is system parameter adjustment, which can be further subdivided into two approaches. One involves using variable inductors or capacitors to achieve parameter adjustment. For example, adjusting the control current of the dual-E core changes the equivalent inductance, or using a switch-controlled capacitor (SCC) to adjust the equivalent compensation capacitor. This method requires the control switch to be strictly synchronized with the system operating frequency, but in practice, frequency deviations are difficult to completely eliminate, affecting tuning accuracy. The second approach is frequency conversion tuning, which changes the system operating frequency to make the input impedance purely resistive. However, under wide-range offset conditions, the resonant frequency curve may exhibit bifurcation, increasing the complexity and difficulty of frequency tracking and identification. Furthermore, the frequency conversion method is not suitable for medium- to high-power applications requiring high frequency stability.

[0007] The third category is coil structure optimization methods. For example, designing adaptive repeater coils, improving the shape of the transmitting coil to concentrate the magnetic field, or adding auxiliary coils to inject reactive power to achieve dynamic tuning. Although these methods can suppress the effects of offset to some extent, they often require complex coil designs or additional auxiliary channels, resulting in high system costs and implementation complexity. Furthermore, some methods are only effective for offsets in specific directions, and their ability to resist offset is limited. Summary of the Invention

[0008] This invention provides an input phase-shifting tuning method for a dual-load orthogonal dual-channel inductive power transmission system. The technical problem it solves is that existing tuning techniques in QDC-IPT systems generally have shortcomings in response speed, tuning accuracy, system cost, or offset adaptability, making it difficult to ensure both transmission performance and ease of implementation while maintaining the resonant state.

[0009] To solve the above technical problems, this invention provides an input phase-shifting tuning method for a dual-load orthogonal dual-channel inductive power transfer system. The dual-load orthogonal dual-channel inductive power transfer system includes a DC power supply, a first channel, and a second channel. The first channel includes a first full-bridge inverter connected in parallel with the DC power supply, and a first primary-side compensation network, a first transmitting coil, a first receiving coil, a first secondary-side compensation network, a first rectifier filter, and a first load cascaded with the first full-bridge inverter. The second channel includes a second full-bridge inverter connected in parallel with the DC power supply, and a second primary-side compensation network, a second transmitting coil, a second receiving coil, a second secondary-side compensation network, a second rectifier filter, and a second load cascaded with the second full-bridge inverter. The phase difference between the output voltages of the first full-bridge inverter and the second full-bridge inverter is... ;

[0010] The input phase-shifting tuning method includes:

[0011] The inverter output voltage and output current of the first and second energy channels are collected to obtain the corresponding voltage and current phase difference. , ;

[0012] According to the voltage and current phase difference , Determine if the system is detuned. If detuned, identify the main energy transfer channel and calculate the desired input voltage phase difference based on the system parameters under the current offset state. ;

[0013] Based on the calculated desired input voltage phase difference The PWM drive signal of the second energy channel is adjusted to change the input voltage phase of the second energy channel inverter, so that the main energy transmission channel operates in a resonant state.

[0014] Further, the step based on the voltage-current phase difference , To determine if a system is detuned, the following steps are taken:

[0015] The voltage and current phase difference , The smaller one is used as the feedback quantity. The feedback quantity With the preset resonance state setting value If a comparison is made, If so, then the system is determined to satisfy the resonance condition; if If so, the system is determined to be in a detuned state.

[0016] Furthermore, determining the main energy transmission channel includes:

[0017] Compare the phase difference of piezocurrents and ,like Then the first energy channel is determined to be the main energy transmission channel; if If so, then the second energy channel is determined to be the main energy transmission channel.

[0018] Furthermore, the step of calculating the desired input voltage phase difference based on the system parameters under the current offset state... include:

[0019] Obtain system parameters, including: the effective value of the inverter output voltage of the two channels. , Change in self-inductance of transmitting coil , Change in self-inductance of the receiving coil , Directly opposite to the coupled mutual inductance , Cross-coupled mutual inductance , Equivalent load resistance , intermediate variables , and the system operating angular frequency Wherein, subscript 1 represents the first channel, subscript 2 represents the second channel, subscript t represents the transmitting coil, and subscript r represents the receiving coil; the intermediate variable , The expression is: ;

[0020] When the main energy transmission channel is the first energy channel, the desired input voltage phase difference is calculated according to the following formula. :

[0021] ,

[0022] in, , , For custom parameters, the expression is:

[0023] ,

[0024] When the main energy transmission channel is the second energy channel, the desired input voltage phase difference is calculated according to the following formula. :

[0025] ,

[0026] in, For custom parameters, the expression is:

[0027] ;

[0028] The coordinate system between the coupling mechanisms is defined as follows: With the geometric center of the transmitting coupling mechanism as the origin O, mutually perpendicular X and Y axes are established in the plane containing the transmitting coupling mechanism, and the direction perpendicular to this plane is defined as the Z axis. The coordinate system of the plane offset of the receiving coupling mechanism relative to the transmitting coupling mechanism is defined as follows: The regions where x and y are both positive and negative are in the first and third quadrants, respectively; the regions where x and y are both negative and positive are in the second quadrant; and the regions where x and y are both positive and negative are in the fourth quadrant.

[0029] Furthermore, adjusting the PWM drive signal of the second energy channel to change the input voltage phase of the second energy channel inverter, so that the main energy transfer channel operates in a resonant state, includes:

[0030] Based on the calculated desired input voltage phase difference The PWM drive signal of the second energy channel is adjusted to change the input voltage phase of the second energy channel inverter, so that the voltage and current phase difference of the main energy transfer channel is... Reduced to less than or equal to .

[0031] Furthermore, the adjustment of the PWM drive signal of the second channel to change the input voltage phase of the second channel inverter specifically includes:

[0032] Compare the current control cycle , absolute value of phase difference absolute value of the phase difference with the previous control cycle ;like Then the scanning direction flag value is maintained. Otherwise, change the scan direction flag value. Multiply To change the scanning direction;

[0033] judge If it is within the preset range, then it is determined based on the scan direction marker. The desired input voltage phase difference is iteratively updated using a step size. Otherwise, the receiving device is deemed to be outside the location range, and the process ends.

[0034] Furthermore, the preset interval is .

[0035] Furthermore, after adjusting the PWM drive signal of the second channel, the process also includes:

[0036] Sampling the first channel load of the dual-load orthogonal dual-channel inductive power transfer system voltage at both ends With current Calculate the actual received power ;

[0037] like If so, the current control parameters will be maintained, where The received power of the previous control cycle; if This triggers a recalculation and outputs a new desired phase difference. .

[0038] Furthermore, the first primary-side compensation network and the second primary-side compensation network adopt LCC-type compensation networks, and the first secondary-side compensation network and the second secondary-side compensation network adopt S-type compensation networks.

[0039] The present invention provides an input phase-shifting tuning method for a dual-load quadrature dual-channel inductive power transfer system. The system includes a first channel and a second channel, and the phase difference between the inverter output voltages of the two channels is [missing information]. By regulating To achieve resonant operation of the main energy transfer channel, the method first acquires the inverter output voltage and output current of the two channels to obtain the voltage and current phase difference, and determines whether the system is detuned. When detuned, the main energy transfer channel is determined. Then, the system parameters under the current offset state are obtained and substituted into the analytical formula established based on the analysis of the circulating current coupling path and power interaction mechanism to calculate the optimal value that makes the reactive power input of the main energy transfer channel zero. The value, finally based on this The PWM drive signal of the second energy channel inverter is adjusted to change its input voltage phase, so that the main energy transmission channel operates in a resonant state. This method achieves dynamic tuning of the main energy transmission channel when the receiver position shifts by adjusting the input voltage phase difference in real time. It does not require the addition of variable inductors / capacitors, auxiliary coils, or additional power conversion circuits, effectively improving the system's transmission power and efficiency. Attached Figure Description

[0040] Figure 1 This is a circuit diagram of the dual-load orthogonal dual-channel inductive power transmission system provided in an embodiment of the present invention.

[0041] Figure 2 This is the fundamental equivalent circuit diagram of the dual-load orthogonal dual-channel inductive power transmission system provided in this embodiment of the invention;

[0042] Figure 3 This is a schematic diagram of the coupled circulating current path in the dual-load orthogonal dual-channel inductive power transmission system provided in an embodiment of the present invention;

[0043] Figure 4 This is an equivalent circuit diagram with mapped impedance in a dual-load orthogonal dual-channel induced power transmission system provided in an embodiment of the present invention.

[0044] Figure 5 This is a structural block diagram of the input phase-shifting tuning control system provided in an embodiment of the present invention;

[0045] Figure 6 This is a control flowchart of the input phase-shifting tuning control method provided in the embodiments of the present invention;

[0046] Figure 7 This is a distribution diagram of the quality factor in the XOY plane provided by an embodiment of the present invention;

[0047] Figure 8 This is the envelope diagram of each system parameter during the tuning process when the receiving end is located in the first quadrant, as provided in the embodiment of the present invention.

[0048] Figure 9 This is a transient waveform diagram of the system before and after tuning when the receiving end is located in the first quadrant, as provided in an embodiment of the present invention.

[0049] Figure 10 This is the envelope diagram of each system parameter during the tuning process when the receiving end is located in the second quadrant, as provided in the embodiment of the present invention.

[0050] Figure 11 This is a transient waveform diagram of the system before and after tuning when the receiving end is located in the second quadrant, as provided in an embodiment of the present invention.

[0051] Figure 12 This is a comparison diagram of the system transmission characteristics provided in the embodiments of the present invention before and after tuning in the XOY plane;

[0052] Figure 13 This is a diagram showing the change in the phase difference of the main energy transfer channel before and after tuning under XOY plane offset, provided in an embodiment of the present invention.

[0053] Figure 14 This is an experimental waveform diagram of the system before and after tuning when the receiving mechanism provided in this embodiment of the invention is offset to the (100mm, 100mm) position;

[0054] Figure 15 This is an experimental waveform diagram of the system before and after tuning when the receiving mechanism provided in the embodiment of the present invention is offset to the position of (-100mm, 100mm);

[0055] Figure 16 This is a comparison chart of the system transmission performance before and after tuning, provided in an embodiment of the present invention. Detailed Implementation

[0056] The embodiments of the present invention are described in detail below with reference to the accompanying drawings. The embodiments are given for illustrative purposes only and should not be construed as limiting the present invention. The accompanying drawings are for reference and illustration only and do not constitute a limitation on the scope of patent protection of the present invention, because many changes can be made to the present invention without departing from the spirit and scope of the present invention.

[0057] Figure 1 This is the circuit diagram of a dual-load QDC-IPT system. (Example:) Figure 1 As shown, the dual-load QDC-IPT system mainly consists of a DC power supply ( ), and DC power supply ( The first energy channel (referred to as channel one) and the second energy channel (referred to as channel two) are connected in parallel. The first energy channel includes the DC power supply ( The first full-bridge inverter is connected in parallel with the first full-bridge inverter, and the first primary-side compensation network (LCC type) cascaded with the first full-bridge inverter, the first coupling mechanism (including the first transmitting coil and the first receiving coil), the first secondary-side compensation network (S type), the first rectifier filter, and the first load, and the second channel includes and is connected to the DC power supply ( The second full-bridge inverter is connected in parallel with the second full-bridge inverter, and the second primary-side compensation network (LCC type), the second coupling mechanism (including the second transmitting coil and the second receiving coil), the second secondary-side compensation network (S type), the second rectifier filter and the second load are cascaded with the second full-bridge inverter. Here, "first" corresponds to the first channel and "second" corresponds to the second channel.

[0058] Between the two energy channels , These are the output currents of the first and second full-bridge inverters, respectively (the corresponding fundamental currents are expressed as follows). , ), , They are respectively The current on the transmitting and receiving coils of the channel (the corresponding fundamental current is expressed as) , ), , , It constitutes The LCC compensation network of the channel, for Series compensation capacitor of the energy channel, , They are respectively The inductance of the transmitting and receiving coils. , For the mutual inductance of the direct coupling between the transmitting and receiving coils, , For the cross-coupling mutual inductance between the transmitting and receiving coils, for The filter capacitor in the rectifier filter of the power supply, for The load resistance of the channel, subscript These correspond to the first and second energy channels, respectively. The subscript t corresponds to transmission, and the subscript r corresponds to reception.

[0059] This system converts DC power into two high-frequency AC voltages with equal amplitude and independently adjustable phase difference by controlling the drive signals of two inverters. , .Right now:

[0060] (1)

[0061] in, This represents the AC input voltage of the two channels (i.e., the inverter output voltage). , Phase difference between , , They represent , Valid value.

[0062] The input voltage is changed by adjusting the PWM drive signal of the second energy channel in real time. The phase of the input voltage is adjusted, thereby controlling the phase difference between the two channels. Ultimately, this achieves the main energy transfer channel operating in a resonant state. The main energy transfer channel refers to the channel where power transmission is dominant; additionally, the phase difference... The distribution of the strong magnetic coupling region was also adjusted.

[0063] Figure 2 This is the fundamental equivalent circuit of a dual-load QDC-IPT system. Figure 2 middle and The equivalent resistance at the input terminals of the two-channel rectifier is expressed as:

[0064] (2)

[0065] Figure 2 middle, for The self-inductance of the transmitting coil at the correct alignment position, express The change in self-inductance of the transmitting coil relative to the aligned state; for The self-inductance of the receiving coil when it is in the correct alignment position. Then it means The change in self-inductance of the receiving coil relative to the aligned state.

[0066] When the QDC mechanism is in the correct alignment position, according to Figure 2 Write Kirchhoff's voltage law (KVL) equations:

[0067] (3)

[0068] in, This indicates that the coils of the coupling mechanism are mutually inducted. The induced voltage generated in the single-channel transmission circuit, This indicates that the coils of the coupling mechanism are mutually inducted. The induced voltage generated in the single-channel transmission circuit, This indicates that the coils of the coupling mechanism are mutually inducted. The induced voltage generated in the receiving circuit of a single energy channel, This indicates that the coils of the coupling mechanism are mutually inducted. The induced voltage generated in the receiving circuit of a single energy channel, This indicates that the coils of the coupling mechanism are mutually inducted. The induced voltage generated in the two-channel emitter circuit, This indicates that the coils of the coupling mechanism are mutually inducted. The induced voltage generated in the two-channel emitter circuit, This indicates that the coils of the coupling mechanism are mutually inducted. The induced voltage generated in the two-channel receiving circuit, This indicates that the coils of the coupling mechanism are mutually inducted. The induced voltage generated in the two-channel receiving circuit.

[0069] The custom parameter Z in equation (3) ii (i=1,2,3,4,5,6) are represented as follows:

[0070] (4)

[0071] symbol and The reactances of the resonant inductor and capacitor are respectively:

[0072] (5)

[0073] in, This represents the system's operating angular frequency.

[0074] The parameter configuration conditions for the resonant capacitor in the LCC-S topology are as follows:

[0075] (6)

[0076] Cross-inductance between the transmitter and receiver and Make the voltage source and Two inductive coupling paths are formed between them, such as Figure 3 Paths ① and ② are shown in the diagram. Two excitation currents. and An additional voltage is induced in the transmitting coil through the coupling path, thereby generating a coupled circulating current in the compensation network of each energy channel.

[0077] The current of each branch can be further derived from equations (4) to (6), and its expression is shown in equation (7).

[0078] (7)

[0079] Combining the symmetry of the magnetic coupling mechanism with the system excitation condition (1), it can be seen that, Represents current through Figure 3 Path in In the coil The induced voltage generated in the middle, Represents current through Figure 3 Path in In the coil The induced voltage in exist Current is generated in the branch where the voltage source is located , exist Current is generated in the branch where the voltage source is located Its expression can be derived as follows:

[0080] (8)

[0081] in, , They are respectively .

[0082] From equation (8), the input and output currents of the two channels can be obtained:

[0083] (9)

[0084] The expression is:

[0085] (10)

[0086] In equation (9), and They are respectively the First Path Sutra and The energy transfer current component, and These are respectively the second path of power. and The component is expressed as:

[0087] (11)

[0088] From equation (9), it can be seen that the receiving coil The induced current in the middle is from and The two components together form the signal; similarly, the receiving coil... The induced current is then generated by and It consists of two parts. Its expression is:

[0089] (12)

[0090] Overall, equation (9) shows that the system's input and output currents have symmetrical characteristics in their structure. Combined with equations (11) and (12), it can be seen that the influence of the circulating current components in path ① and path ② on the energy transfer current of the two energy channels depends on the phase difference α. and As can be seen from the reactive component, the circulating current introduces additional reactive power into the system. Under the condition of constant output power, the reactive component generated by the circulating current not only increases the input capacity requirement of the QDC-IPT system, but also weakens the overall transmission performance.

[0091] The interaction coupling between the two energy channels can be characterized as follows: Figure 4 The equivalent mapped impedance of the transmitting coil shown is the RC series branch, which is:

[0092] (13)

[0093] Among them, Z t1_1 Z t1_2 Z t2_1 and Z t2_2 Z represents the equivalent mapped impedance of the two energy channels via the circulating current path at the corresponding transmitting coil side; t1_r1 Z t1_r2 Z t2_r1 and Z t2_r2 These are the total impedances of the receiving circuit passing through... , and , Converted to transmitting coil The mapped impedance on the side.

[0094] The equivalent mapped impedance of the transmitting coil branch characterizes the power of the circulating current in paths ① and ②. The mapped impedance corresponding to each circulating current can be derived as follows:

[0095] (14)

[0096] Based on the superposition relationship of the circulating current components in path ① and path ②, the total mapped impedance of the two circulating currents referred to the corresponding transmitting coil side can be expressed as the superposition of the real and imaginary mapped impedances in equation (14).

[0097] In addition to the mapped impedance formed by the circulating current referred to the transmitting coil side, the total impedance of the receiving circuit is also mapped to the transmitting coil through mutual inductance, thus forming a corresponding equivalent impedance, the expression of which is:

[0098] (15)

[0099] It is evident that the effect of the two coupled circulating currents is equivalent to introducing a capacitive mapped impedance component into the transmitting coil branch. This capacitive component, together with the change in self-inductance, affects the total impedance of the transmitting coil branch, ultimately influencing the resonant state of the system under the preset parameter configuration method.

[0100] Based on the currents of each loop given in equation (7), the power injected into the system by the two inverter equivalent voltage sources can be further calculated, and its expression is:

[0101] (16)

[0102] in, This refers to the power injected into the system by the two inverter equivalent voltage sources. The active power transferred for the input voltage component. The active power transferred for the input voltage component. The circulating active power caused by energy-consuming components. The active power of the complementary circulating current caused by the coupling mechanism. The circulating reactive power caused by energy-consuming components, The complementary circulating reactive power caused by the coupling mechanism, voltage source The resulting loss of function voltage source The resulting loss of function This refers to the reactive power caused by the transmitting coil.

[0103] As can be seen from equation (16), the power injected by the inverter equivalent power supply of the two channels contains a non-zero imaginary part, which will cause the system output voltage and current to not operate in phase, and the system cannot operate in resonance.

[0104] Combining equations (14) and (7), the circulating power can be derived as follows:

[0105] (17)

[0106] in, This refers to the circulating power caused by energy-consuming components; The complementary circulating power caused by the coupling mechanism is expressed as:

[0107] (18)

[0108] In equations (16) and (18), represents the active power transferred by the input voltage component. and the active power transferred to the circulating component. As in equation (19):

[0109] (19)

[0110] Compared to the directly aligned position, the receiving coil in the offset position causes a change in the self-inductance of the transmitting and receiving coils. This change in self-inductance will induce reactive power in the system. From equation (7), the generated reactive power is:

[0111] (20)

[0112] Among them, the variable inductance reactive power caused by the input voltage component And the circulating reactive power caused by the circulating component , is represented as:

[0113] .(twenty one)

[0114] As shown in equation (16), the non-zero imaginary part of the power injected by the inverter equivalent power supply in the two channels can be attributed to the combined effect of the change in coil self-inductance and the coupled circulating current. Furthermore, from equations (18) and (21), it can be seen that the reactive power of the two channels in this dual-load QDC-IPT system is not only related to the mutual inductance between the transmitting and receiving coils, but also depends on the phase difference of the input voltage. .

[0115] When the system experiences receiver offset, the equivalent load resistance and Generally, equation (22) is satisfied.

[0116] ,(twenty two)

[0117] Then there is , .

[0118] As can be seen from equation (7), the first channel input current The imaginary part is not only related to and , Related, and also affected by the phase difference of the input voltage Function; Second energy channel input current The same applies to the imaginary part.

[0119] Based on the above analysis of reactive power, the system reactive power can be mainly classified into circulating reactive power and variable inductance reactive power. Among them, circulating reactive power is affected by the phase difference of the input voltage. Therefore, it can be regulated by adjusting By changing its value, active compensation for variable reactive power can be achieved, ultimately making the total reactive power input of the corresponding energy channel zero.

[0120] When the first energy channel is the main energy transmission channel, it can be derived from equation (16) When it is 0, the corresponding As shown in equation (23):

[0121] ,(twenty three)

[0122] Among them, custom parameters , , express:

[0123] .(twenty four)

[0124] Analysis of equation (24) shows that the parameter , , The sign is determined by both direct mutual inductance and cross mutual inductance.

[0125] To determine the quadrant of the receiving coil, the geometric center of the transmitting end QDC coupling mechanism is taken as the origin O, and the plane containing the transmitting end coupling mechanism is taken as the XOY plane. X-axis and Y-axis are established along the symmetry axes of the two sets of orthogonal transmitting coils, respectively. The offset coordinates of the geometric center of the receiving end coupling mechanism relative to the origin O in the XOY plane are denoted as (x, y). When x > 0 and y > 0, the receiving coil is located in the first quadrant; when x < 0 and y > 0, the receiving coil is located in the second quadrant; when x < 0 and y < 0, the receiving coil is located in the third quadrant; and when x > 0 and y < 0, the receiving coil is located in the fourth quadrant.

[0126] Based on the magnetic field characteristics of the dual-load QDC mechanism, two typical distribution states can be summarized. The first state is when the receiving coil is in the first and third quadrants, because the cross-inductance and the opposing mutual inductance in this region have the same polarity. , and All values ​​are positive. At this point, the phase difference of the input voltage is adjusted. To a specific value This allows the main energy transfer channel to operate in a resonant state. The second state corresponds to the case where the receiving coil is located in the second or fourth quadrant. Cross-inductance and direct-acting inductance have opposite polarities within the region, leading to... , and The signs are all represented as negative values. Similarly, by... Adjust to another specific value This ensures that the main energy transfer channel remains in a resonant operating state.

[0127] When the second energy channel is the main energy transmission channel, it can be derived from equation (16) When it is 0, the corresponding As shown in equation (25):

[0128] (25)

[0129] Among them, custom parameters express:

[0130] (26)

[0131] Similarly, when the receiving coil is in the first or third quadrant, or the second or fourth quadrant, the phase difference of the input voltage can be... The corresponding adjustment is as follows: or This ensures the main energy transmission channel operates resonantly. Therefore, this principle effectively enables the interactive tuning of the dual-channel system when a misalignment occurs at the receiving end of the coupling mechanism.

[0132] In a dual-load QDC-IPT system, based on the input phase-shift tuning principle, the two energy channels can work collaboratively through interactive tuning; however, the criteria for determining the main energy transfer channel need to be clearly defined. Therefore, a quality factor is selected. As a criterion for switching between dual-channel operation modes:

[0133] (27)

[0134] The core basis lies in the fact that the quality factor is directly related to the system's reactive power and transmission efficiency, and can sensitively reflect the energy efficiency status of the channel. Equation (27) shows that, under the initial orthogonal excitation conditions, the quality factor of each channel is... It is a functional relationship between the self-inductance and mutual inductance of its coil, where, . No. The phase difference between the inverter output voltage and the inverter output current is denoted as . The phase difference This is the phase angle of the input impedance of the channel. Due to the quality factor... with the absolute value of the phase difference They have a monotonic correspondence; therefore, the phase difference between the two energy channels can be detected and compared. and Indirectly determine the quality factor and The size relationship. When the following conditions are met. At that time, there is a corresponding The first energy channel is designated as the main energy transmission channel; when the conditions are met... At that time, there is a corresponding The second energy channel is designated as the primary energy transfer channel. Therefore, the quality factor... The phase difference between the inverter output voltage and current serves as a theoretical criterion for selecting the main energy transfer channel. As a measurable and quantifiable parameter in the actual control process, it is detected and compared in real time. and It can identify the main energy transfer channel and dynamically switch between dual-channel operating modes. The physical meaning of this criterion is that the quality factor directly characterizes the relative strength of reactive circulating current and active power transmission within the system. By real-time monitoring and comparison of the quality factor values ​​of the two channels, the system can quickly identify the main energy transfer channel and dynamically switch operating modes to adjust... The value is used to tune the main energy transmission channel, thereby maintaining the efficient and stable operation of the system even if the receiving end of the coupling mechanism is offset.

[0135] To achieve input phase-shift interactive tuning of a dual-load QDC-IPT system under receiver offset conditions, a control system was designed to achieve resonance based on the phase difference of the input voltage, such as... Figure 5 As shown, it includes a transmitter control circuit and a receiver control circuit. This control system determines the main energy transfer channel by comparing the phase difference between the input voltage and current of the two energy channels, and precisely adjusts the phase of the input voltage of the second energy channel to enable the main energy transfer channel to operate in resonance.

[0136] To achieve the system's control objectives, this embodiment constructs the following closed-loop control system: First, at the transmitting ends of the two energy transfer channels, voltage sensors and current sensors are used to collect the inverter's output voltage, respectively. , With output current , Subsequently, the input signal is converted into a square wave by the zero-crossing detection module and input to the phase detection module, which outputs the voltage and current phase difference of the two channels respectively. and By comparison and The energy channel with the smaller phase difference is determined as the main energy transfer channel, and its phase difference is taken as the feedback quantity. . This feedback quantity With resonance state setting value Subtraction yields the phase error signal. This error signal is then processed by the controller via a linear scan to obtain the desired phase difference. Ultimately, the system based on... Generate a corresponding PWM drive signal to change the phase of the second channel input voltage. This puts the main energy transfer channel in a resonant working state.

[0137] After the main energy transfer channel achieves resonance, the system measures the load using voltage and current sensors. voltage at both ends With current And calculate the actual received power. .like Higher than the power value before tuning If the current control parameters are not adjusted, the current control parameters will be maintained without introducing feedback adjustment; otherwise, a feedback signal will be input to the controller to recalculate and output the desired phase difference. This drives the system to adjust to a new resonant state, thereby ensuring that the transmitted power tends to be optimal.

[0138] Figure 6 The control flow of input phase-shift tuning is described in detail. First, the system parameters are initialized: desired values... (=3°), Current control cycle value , and the previous control cycle value , and load The initial received power (all are 0), and the initial phase difference value is set. At the same time, define the number of iterations. With scan direction marker .

[0139] After the system process begins, the feedback volume will be... and expected value If a comparison is made, If the system meets the resonance condition, the control process enters the monitoring state; otherwise, it is determined that the system does not. The system is then determined to be in a detuned state, which triggers the tuning mechanism.

[0140] During the tuning process, the first comparison is... absolute value and absolute value ,like A smaller value indicates that the current linear scan direction is correct and the target is being gradually approached. Conversely, it will Multiply by -1 to get the negative value, thus changing the scanning direction. Then, the phase difference must be ensured. Within the range of (0°, 180°), iterative updates are performed with a preset step size of 1°. The updated... The value was used to adjust the PWM drive signal of the second-channel inverter in real time, and then the phase difference was remeasured. Then, proceed to the next cycle of judgment and loop.

[0141] When the main energy transfer channel resonates, the system further verifies its transmission performance: If If so, maintain the current control parameters; Then, a feedback signal is input to the controller, triggering it to recalculate and output a new value. The value is set to ensure that the transmission power tends to be optimal while the system remains in a resonant state.

[0142] Finally, safety boundaries were set for the entire process; if during iterative calculations... If the value exceeds the range, the receiver is determined to be in an untunable position, and the process will automatically terminate.

[0143] A system simulation circuit model was built using Matlab / Simulink, and the main parameters of the model are shown in Table 1. The dynamic offset process of the receiver was simulated using the Ansys finite element simulation platform. The specific parameter values ​​of the QDC mechanism are shown in Table 2. During the offset process, the alignment position was set as the starting point, 25mm was used as the step size, and the offset range was set to ±50% on the X-axis and ±50% on the Y-axis (X=±150mm, Y=±150mm).

[0144] Table 1. Main parameter values ​​of the simulation circuit

[0145]

[0146] Table 2 Dimensional parameters of magnetic coupling mechanism

[0147]

[0148] Figure 7 The distribution of the quality factor of the QDC mechanism in the XOY plane is presented. There are 169 offset points in the plane, where ▲ and ● represent the location of the receiver center point; ▲ indicates that the first channel has a smaller quality factor at that location, and ● indicates that the second channel has a smaller quality factor at that location. Figure 7 As can be seen, the positions of ▲ are distributed in the second and fourth quadrants of the XOY plane, while ● is mainly distributed in the first and third quadrants. This indicates that the transmission performance of the first channel is superior in the second and fourth quadrants, and the QDC-IPT system operates in mode where the first channel is the primary transmission channel. Similarly, in the first and third quadrants of the XOY plane, the second channel operates as the primary transmission channel. This demonstrates the feasibility of using the quality factor as a criterion for switching between dual-channel operating modes.

[0149] The simulation model was used to verify five positions: (0mm, 0mm), (100mm, 100mm), (150mm, 150mm), (-100mm, 100mm), and (-150mm, 150mm). The parameters for each position are shown in Table 3.

[0150] Table 3 Self-inductance and mutual inductance parameters of the coupling mechanism

[0151]

[0152] When the center point of the receiver shifts to the first quadrant region, the system switches to the working mode of using the second energy channel as the main energy transmission channel. Figure 8 The parameter envelope diagram of the main energy transfer channel of the QDC-IPT system during the tuning process is given. Figure 8 (a) Figure 8 (b) shows the voltage and current envelope diagrams of the main energy transfer channel, respectively. The envelope diagram. Figure 8 In the diagram, time T1 is when the receiver starts from position 0. # To position 1 # Perform the offset; at time T2, adjust the input phase shift angle. Time T3 is when the receiver starts from position 1. # To position 2 # Offset; at time T4, the input phase shift angle is adjusted. .

[0153] from Figure 8 It can be seen that:

[0154] During the time interval 0 to T1, the inverter output of the main energy channel is stable and the phase difference θ2 is 5.32°, indicating that the main energy channel is operating in resonance. At t=T1, the receiver outputs to position 1... # During the offset operation, the main channel output current decreases significantly and When the angle is increased to 21.09°, it is in a detuned state; at t=T2, the input phase shift angle is adjusted. The output current of both channels gradually increases and stabilizes, and the phase difference of the main channel is... The angle then drops from 21.09° to 7.12°, reaching a resonant state; at t=T3, the receiver shifts back to position 2. # The output current of the main channel decreased again and was lower than 1. # The position is smaller, and the phase difference increases to 37.52°, putting the system in a detuned state. When t=T4, a new input phase shift angle is obtained by rescanning. At this point, the main channel output current increases significantly and the phase difference... It is reduced to 8.56° and operates in a resonant state.

[0155] Figure 9 Detailed waveform diagrams of the voltage and current of the two channels are presented before and after each moment when the device shifts to the first quadrant region. Figure 9 (a) is and waveform diagram, Figure 9 (b) is and waveform diagram, Figure 9 (c) is and waveform diagram, Figure 9 (d) is and The waveform diagram.

[0156] Depend on Figure 9 As can be seen in (a) and (c), before time T1, and , and Being in phase indicates that the two channels are operating in resonance; after time T1 and before time T2, the receiver is offset to position 1. # The voltage and current waveforms of the two channels begin to show a phase difference, and the system is in a detuned state; after time T2, adjust the phase angle of the input voltage of the second channel. =130°, utilizing the inter-channel coupling effect to introduce circulating reactive power to compensate for the reactive component generated by changes in coil parameters, thereby offsetting the reactive power of the main channel. At this time, the voltage With current Phase difference The angle decreased to approximately 7°, indicating that the main energy channel is approaching its resonant operating state again. Simultaneously, due to... , As can be seen from the waveform, changing the input phase angle can suppress higher harmonics and effectively reduce the degree of inverter output current distortion.

[0157] joint Figure 9 As can be seen in (b) and (d), the input voltage and current on the two rectifier sides, i.e. and , and The fact that the two loads remained in phase throughout the tuning process demonstrates that both loads could stably acquire active power before and after tuning. Furthermore, the significant increase in the input voltage and current of both rectified circuits after tuning indicates the effectiveness of the input phase-shift tuning method in enhancing transmission power. Similarly, the waveform changes before and after times T3 and T4 effectively verify that continuous tuning of the system can be achieved when the receiver position changes multiple times. When the receiver center point shifts to the second quadrant, the system switches to the operating mode where the first energy channel serves as the main power transmission channel. Figure 10 The output voltage, current, and phase difference of the main channel are displayed. Dynamic changes, Figure 10 (a) is the voltage and current envelope diagram of the main energy channel (first energy channel). Figure 10 (b) represents the phase difference. The envelope diagram.

[0158] from Figure 10 It can be seen that:

[0159] At t=T1, the receiver sends a signal to position 3. # The offset is then performed. At this time, the main channel output current drops significantly and... Increased to 32.37°, it is in a detuned state; at t=T2, adjust the input phase shift angle. The main channel output current gradually increases and stabilizes, and the main channel phase difference... The angle then drops from 32.37° to 7.05°, reaching a resonant state; at t=T3, the receiver shifts again to position 4. # The output current of the main channel begins to decrease, and at the same time, the phase difference of the main channel increases to 41.93°, causing the system to detune again; at t=T4, a new scan is performed to obtain a new input phase shift angle. At this point, the main channel output current increases significantly and the phase difference... It decreases to 8.57° and returns to the resonant state.

[0160] Figure 11 Detailed waveforms of the two energy channels' voltage and current are presented before and after each moment when the device shifts to the second quadrant region. Figure 11 (a) is and waveform diagram, Figure 11 (b) is and waveform diagram, Figure 11 (c) is and waveform diagram, Figure 11 (d) is and The waveform diagram.

[0161] like Figure 11 As shown in (a) and (c), at time T1, the receiver is offset to position 3. # At time T2, a phase difference begins to appear in the voltage and current waveforms of the two channels, and the system is in a detuned state; at time T2, the phase angle of the input voltage of the second channel is adjusted. =14°, main channel voltage and current Phase difference When the angle is reduced to around 7°, the main energy channel is in a resonant state. For example... Figure 11 As shown in (b) and (d), the input voltage and current of the two rectified circuits are also significantly improved after tuning.

[0162] Figure 12 This is a comparison of the system's transmission characteristics before and after tuning in the XOY plane. Figure 12 (a) and (b) show the results before and after tuning. , Figure 12 (c) and (d) represent the state before and after tuning. .from Figure 12 It can be seen that when the receiver is offset in the four quadrants, the untuned load pickup power decreases with increasing offset and exhibits a symmetrical change. After the tuning method is applied, the pickup power in the four quadrants is significantly improved, and the system transmission performance is effectively improved.

[0163] Figure 13This represents the change in the phase difference θ of the main energy channel before and after tuning within a ±150mm offset range along the X and Y axes at the receiver. When the receiver is offset to any of the four quadrants, the phase difference θ before tuning increases with increasing offset distance, exceeding 15° in all cases. The largest θ value is found at the diagonal of each quadrant, reaching approximately 43°, at which point the system is detuned. After input phase shift tuning, the phase difference across the entire XOY plane... The values ​​were all controlled at approximately 7°.

[0164] This demonstrates that when a shift occurs at the receiving end, the proposed input phase-shifting tuning method can not only improve the output power but also ensure that the main channel operates in a resonant state.

[0165] Based on the parameters listed in Tables 1 and 2, a prototype of the dual-load QDC-IPT system was built. During the experiment, the receiving mechanism was offset within ±150mm along the X and Y axes. Three offset positions—(0, 0), (100, 100) (mm), and (-100, 100) (mm)—were selected for experimental verification. The equivalent load used in the experiment was 12Ω.

[0166] Figure 14 The experimental waveforms before and after tuning of the two channels and the phase difference of the output voltage and current of the two channels when the receiving mechanism shifts from (0, 0) to position (100, 100) are given. Figure 14 In the middle, (a) and (c) show the experimental waveforms and phase difference of the inverter output before tuning at positions (0, 0) and (100, 100), respectively. Figure 14 In the middle, (b) and (d) are the experimental waveforms of the rectified input voltage and current before tuning of the two energy channels at positions (0, 0) and (100, 100), respectively. Figure 14 In the diagram, (e) and (f) are the experimental waveforms of the inverter output and rectifier input voltage and current before tuning of the two channels at (100, 100), respectively.

[0167] Figure 14 (a) shows that the phase difference θ of the main energy channel is 5.14° when in the correct alignment position. Figure 14 As can be seen in (c), as the receiving mechanism shifts to the (100, 100) position, the phase difference θ of the main energy channel increases to 22.35°. And compared to... Figure 14 From (b) and (c), it can be seen that the rectified input voltage and current decrease with the offset. Comparison Figure 14 From (c), (d), (e), and (f), it can be seen that after the main power channel is tuned, the inverter output phase difference θ is reduced to about 6° and the rectified input voltage and current are significantly improved, indicating that the system transmission power is effectively improved.

[0168] Figure 15Experimental waveforms of the two energy channels before and after tuning when the receiving mechanism is offset to position (-100, 100). Figure 15 In the middle (b) and (d), the experimental waveforms of the rectified input voltage and current before tuning of the two energy channels at positions (0, 0) and (-100, 100) are respectively. Figure 14 In the middle (e) and (f), the experimental waveforms of the inverter output and rectifier input voltage and current before tuning of the two channels at (-100, 100) are respectively. Figure 15 (a) shows that the phase difference θ of the main energy channel is 5.03° at the aligned position. (Combined) Figure 15 As can be seen in (c), as the receiving mechanism shifts to the (-100, 100) position, the phase difference θ of the main energy channel increases to 31.09°. And compared to... Figure 15 From (b) and (c), it can be seen that the rectified input voltage and current decrease with the offset. Comparison Figure 15 As can be seen from (c), (d), (e), and (f), after adjusting the input phase shift angle α=14°, the inverter output phase difference θ of the main channel is reduced to about 6° after the tuning effect, and the rectified input voltage and current are also significantly improved.

[0169] The consistency between the simulation and experimental results verifies the correctness and effectiveness of the input phase-shift-based interactive tuning method. Furthermore, the phase difference of the main power channel inverter output still exceeds 3° after tuning, making the resonant network slightly inductive, thus creating conditions for achieving ZVS in the transmitter power devices.

[0170] Figure 16 A comparison of the system's transmission performance before and after interactive tuning at different offset positions is presented. Figure 16 It can be seen that after tuning, both the system's transmission power and efficiency are significantly improved. The average load pickup power is doubled, and the average system transmission efficiency is improved by 15%. The highest transmission efficiency is 94.87% at position (-100, 100). At position (150, 150), the tuning effect is optimal, with the transmission efficiency increasing from 72.40% to 93.03%, an increase of 20.63%. The load pickup power corresponding to the main channel increases from 138.8W to 330.6W, an increase of 1.38 times. This proves that the input phase-shifting tuning method proposed in this embodiment can ensure that the main channel operates in a resonant state while achieving a significant improvement in system transmission performance even when the receiving mechanism is offset.

[0171] Table 4 shows the differences between the tuning method proposed in this embodiment and existing methods in terms of operating frequency, transmission performance, system cost, implementation complexity, and anti-offset performance.

[0172] Table 4 Comparison with existing tuning methods

[0173]

[0174] As shown in Table 4, the frequency conversion tuning method requires real-time adjustment of the system's operating frequency, making it unsuitable for medium- to high-power applications. The method in this embodiment achieves tuning at a fixed frequency of 85 kHz by adjusting the phase difference between the two energy channels, which is beneficial for achieving soft switching of the converter and stable operation of the compensation network. The experimental prototype built in this embodiment can output 1450 W with an efficiency of 94.87%, only lower than the highest of 3300 W, indicating that the proposed method can balance system transmission performance during tuning.

[0175] Furthermore, regarding system cost and implementation complexity, the variable inductor / capacitor tuning method requires additional passive components, power switches, and drive circuits. The coil parameter self-tuning and auxiliary coil tuning methods typically require designing suitable coupling mechanisms for specific applications, or adding auxiliary coils and power conversion channels. Therefore, all three methods increase system cost, design difficulty, and control complexity. In contrast, the method presented in this paper utilizes the inherent cross-coupling relationship and power interaction mechanism of the orthogonal dual-channel system, eliminating the need for variable inductors / capacitors, auxiliary coils, and additional power conversion circuits. It only requires sampling the phase difference between the inverter output voltage and current of the two channels, thus resulting in relatively lower system cost and implementation complexity.

[0176] Regarding offset resistance, the self-tuning method for coil parameters has a relatively limited offset resistance range, while other methods exhibit stronger offset resistance. The prototype constructed in this embodiment maintains resonant operation of the main energy transmission channel within a ±50% offset range by adjusting the input voltage phase of the two energy channels, while simultaneously improving the detuning degree and transmission performance of the secondary energy transmission channel. Furthermore, at the 50% maximum offset position, the proposed method improves transmission efficiency by 20.63% compared to the untuned state, while increasing pickup power to 1.38 times that before tuning.

[0177] Comprehensive comparison shows that, under fixed frequency operation conditions, the tuning method proposed in this embodiment can balance high transmission power and efficiency, low system cost and implementation complexity, while also having high anti-migration capability.

[0178] In summary, this invention proposes an input phase-shifting tuning method for a dual-load orthogonal dual-channel inductive power transfer system. It reveals the interaction mechanism between the two channels in the dual-channel IPT system. The proposed tuning method allows the main power transfer channel to operate in a resonant state even with a horizontal offset of ±50% in the magnetic coupling mechanism, while simultaneously improving the transmission performance of the secondary power transfer channel. The conclusions are as follows:

[0179] 1) This method only uses the phase difference between the inverter output voltage and current of the two channels as the resonant state characterization quantity, without the need to collect other complex parameters; the tuning of the main channel is achieved by adjusting the phase of the drive signal of the second channel inverter, which simplifies the sampling control circuit.

[0180] 2) Within the offset range, the voltage and current waveform distortion of the two channels is significantly reduced during phase-shifting tuning, the high-order harmonic components of the system are effectively suppressed, the harmonic loss is reduced, and the additional stress of high-order harmonics on power devices is alleviated.

[0181] 3) As the offset range increases, this method exhibits superior tuning performance. Experiments show that, under the condition of a maximum design offset of 50%, the system transmission efficiency is improved by 20.63%, and the pickup power is increased by 1.38 times.

[0182] 4) While ensuring the main energy transfer channel achieves resonant operation and efficient energy transfer, the power distribution characteristics of the secondary energy transfer channel are indirectly optimized through the inter-channel interaction mechanism. The detuning degree of the secondary energy channel is significantly reduced and the pickup power is effectively improved.

[0183] Furthermore, to improve the system's adaptability, advanced control algorithms need to be introduced under complex operating conditions. These algorithms shorten the tuning response time and enhance the system's robustness, laying the foundation for the engineering application of dual-channel IPT systems.

[0184] The above embodiments are preferred embodiments of the present invention, but the embodiments of the present invention are not limited to the above embodiments. Any changes, modifications, substitutions, combinations, or simplifications made without departing from the spirit and principle of the present invention shall be considered equivalent substitutions and shall be included within the protection scope of the present invention.

Claims

1. An input phase-shifting tuning method for a dual-load orthogonal dual-channel inductive power transfer system, characterized in that: The dual-load orthogonal dual-channel inductive power transfer system includes a DC power supply, a first channel, and a second channel. The first channel includes a first full-bridge inverter connected in parallel with the DC power supply, and a first primary-side compensation network, a first transmitting coil, a first receiving coil, a first secondary-side compensation network, a first rectifier filter, and a first load cascaded with the first full-bridge inverter. The second channel includes a second full-bridge inverter connected in parallel with the DC power supply, and a second primary-side compensation network, a second transmitting coil, a second receiving coil, a second secondary-side compensation network, a second rectifier filter, and a second load cascaded with the second full-bridge inverter. The phase difference between the output voltages of the first full-bridge inverter and the second full-bridge inverter is [missing information]. ; The input phase-shifting tuning method includes: The inverter output voltage and output current of the first and second energy channels are collected to obtain the corresponding voltage and current phase difference. , ; According to the voltage and current phase difference , Determine if the system is detuned. If detuned, identify the main energy transfer channel and calculate the desired input voltage phase difference based on the system parameters under the current offset state. ; Based on the calculated desired input voltage phase difference The PWM drive signal of the second energy channel is adjusted to change the input voltage phase of the second energy channel inverter, so that the main energy transmission channel operates in a resonant state.

2. The input phase-shifting tuning method for the dual-load orthogonal dual-channel inductive power transfer system according to claim 1, characterized in that, The voltage and current phase difference , To determine if a system is detuned, the following steps are taken: The voltage and current phase difference , The smaller one is used as the feedback quantity. The feedback quantity With the preset resonance state setting value If a comparison is made, If so, then the system is determined to satisfy the resonance condition; if If so, the system is determined to be in a detuned state.

3. The input phase-shifting tuning method for the dual-load orthogonal dual-channel inductive power transfer system according to claim 2, characterized in that, The determination of the main energy transmission channel includes: Compare the phase difference of piezocurrents and ,like Then the first energy channel is determined to be the main energy transmission channel; if If so, then the second energy channel is determined to be the main energy transmission channel.

4. The input phase-shifting tuning method for the dual-load orthogonal dual-channel inductive power transfer system according to claim 2, characterized in that, The desired input voltage phase difference is calculated based on the system parameters under the current offset state. include: Obtain system parameters, including: the effective value of the inverter output voltage of the two channels. , Change in self-inductance of transmitting coil , Change in self-inductance of the receiving coil , Directly opposite to the coupled mutual inductance , Cross-coupled mutual inductance , Equivalent load resistance , intermediate variables , and the system operating angular frequency Wherein, subscript 1 represents the first channel, subscript 2 represents the second channel, subscript t represents the transmitting coil, and subscript r represents the receiving coil; the intermediate variable , The expression is: ; When the main energy transmission channel is the first energy channel, the desired input voltage phase difference is calculated according to the following formula. : ， in, , , For custom parameters, the expression is: ， When the main energy transmission channel is the second energy channel, the desired input voltage phase difference is calculated according to the following formula. : ， in, For custom parameters, the expression is: ； The coordinate system between the coupling mechanisms is defined as follows: with the geometric center of the transmitting coupling mechanism as the origin O, mutually perpendicular X-axis and Y-axis are established in the plane where the transmitting coupling mechanism is located, and the direction perpendicular to this plane is defined as the Z-axis. The coordinate system of the plane offset of the receiving coupling mechanism relative to the transmitting coupling mechanism is defined as follows: The regions where x and y are both positive and negative are in the first and third quadrants, respectively; the regions where x and y are both negative and positive are in the second quadrant; and the regions where x and y are both positive and negative are in the fourth quadrant.

5. The input phase-shifting tuning method for the dual-load orthogonal dual-channel inductive power transfer system according to claim 4, characterized in that: The step of adjusting the PWM drive signal of the second energy channel to change the input voltage phase of the second energy channel inverter, so that the main energy transfer channel operates in a resonant state, includes: Based on the calculated desired input voltage phase difference The PWM drive signal of the second energy channel is adjusted to change the input voltage phase of the second energy channel inverter, so that the voltage and current phase difference of the main energy transfer channel is... Reduced to less than or equal to .

6. The input phase-shifting tuning method for the dual-load orthogonal dual-channel inductive power transfer system according to claim 5, characterized in that, The adjustment of the PWM drive signal of the second channel to change the input voltage phase of the second channel inverter specifically includes: Compare the current control cycle , absolute value of phase difference absolute value of the phase difference with the previous control cycle ;like Then the scanning direction flag value is maintained. Otherwise, change the scan direction flag value. Multiply To change the scanning direction; judge If it is within the preset range, then it is determined based on the scan direction marker. The desired input voltage phase difference is iteratively updated using a step size. Otherwise, the receiving device is deemed to be outside the location range, and the process ends.

7. The input phase-shifting tuning method for the dual-load orthogonal dual-channel inductive power transfer system according to claim 6, characterized in that: The preset interval is: .

8. The input phase-shifting tuning method for the dual-load orthogonal dual-channel inductive power transfer system according to claim 1, characterized in that, After adjusting the PWM drive signal of the second channel, the following is also included: Sampling the first channel load of the dual-load orthogonal dual-channel inductive power transfer system voltage at both ends With current Calculate the actual received power ; like If so, the current control parameters will be maintained, where The received power of the previous control cycle; if This triggers a recalculation and outputs a new desired phase difference. .

9. The input phase-shifting tuning method for the dual-load orthogonal dual-channel inductive power transfer system according to claim 1, characterized in that: The first primary-side compensation network and the second primary-side compensation network adopt LCC type compensation network, and the first secondary-side compensation network and the second secondary-side compensation network adopt S type compensation network.

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