Energy efficiency optimization collaborative control method, system and device of bidirectional converter and medium

Abstract

The application provides an energy efficiency optimization collaborative control method, system, device and medium of a bidirectional converter, and relates to the technical field of power electronic conversion. The method is applied to a single-stage high-frequency isolation main circuit without an internal intermediate direct-current voltage stabilizing large capacitor, and comprises the following steps: acquiring real-time operation condition data of the converter system; constructing a global loss model containing high-frequency magnetic element loss and power switch tube device loss; solving control variables of the global loss model within a safety boundary drawn by a multi-dimensional nonlinear constraint model constituted by instantaneous power, a soft switching interval and frequency limiting, to obtain an optimal switching frequency and an optimal phase shift angle that minimize system predicted total loss as a feedforward reference; and introducing the feedforward reference into a closed-loop control architecture to generate a driving signal by coupling with a real-time feedback regulation amount. The application can realize bidirectional power smooth transmission and global comprehensive energy efficiency improvement of the single-stage bidirectional converter under all operation conditions.

Description

Technical Field

[0001] This application relates to the field of power electronic conversion technology, and in particular to a method, system, device and medium for energy efficiency optimization and coordinated control of a bidirectional converter. Background Technology

[0002] Currently, methods for achieving single optimization objectives such as soft switching and minimizing current stress in single-stage variable frequency isolated bidirectional converters are relatively mature. However, research on optimizing global energy efficiency is still limited. The difficulty in achieving this global energy efficiency optimization lies in the difficulty of solving for the system's global losses (especially high-frequency magnetic component losses). This is because the single-stage topology eliminates the large intermediate capacitor, resulting in direct high-frequency AC-DC coupling. On the one hand, this causes the high-frequency transformer to operate under complex excitations such as wide frequency conversion and asymmetric square waves for extended periods, leading to extremely complex non-sinusoidal magnetic field distributions that make traditional empirical formulas inapplicable. On the other hand, the optimization process easily disrupts the converter's original zero-voltage soft-switching conditions and power transmission balance. These conditions make it difficult for single-stage variable frequency isolated bidirectional converters to achieve global energy efficiency optimization under complex and variable operating conditions. Summary of the Invention

[0003] This application provides a method, system, device, and medium for energy efficiency optimization and collaborative control of a bidirectional converter, in order to solve the technical bottleneck of existing single-stage converters that are difficult to achieve global energy efficiency optimization under complex operating conditions; by constructing a global loss model that includes high-precision transformer magnetic component losses and switching device losses, the global comprehensive energy efficiency optimization of the single-stage converter under all operating conditions is realized.

[0004] In a first aspect, this application provides an energy efficiency optimization and coordinated control method for a bidirectional converter, applied to a single-stage high-frequency isolated main circuit that does not have an internal intermediate DC voltage regulator capacitor. The method includes: Acquire real-time operating condition data of the converter system, including at least the grid voltage and grid current on the AC side and the bidirectional power demand command on the DC side. A system comprehensive energy efficiency evaluation framework for single-stage topologies is constructed. The system comprehensive energy efficiency evaluation framework includes a global loss model with the goal of minimizing the total predicted loss of the system and a multidimensional nonlinear constraint model for limiting the operating boundary of the converter. The global loss model includes the device losses of the main circuit power switching transistors and the magnetic component losses of the high-frequency transformer and series inductor. Based on the operating condition data, within the dynamic feasible solution set defined by the multidimensional nonlinear constraint model, the parameters of the global loss model are optimized to obtain the optimal switching frequency and optimal phase shift angle that minimize the total predicted loss of the system under the current operating condition, and these are used as the feedforward reference. The feedforward reference is introduced into the closed-loop control architecture of the converter system, coupled with the real-time closed-loop feedback adjustment to generate a comprehensive control command, which is then parsed into a drive signal that drives the power switching devices in the single-stage high-frequency isolated main circuit.

[0005] In one possible design, the specific steps for calculating the loss of the magnetic component in the global loss model include: The piecewise equivalent frequency method is used to analyze the non-sinusoidal and high-frequency transient magnetic field parameters of asymmetric duty cycle and each conduction and turn-off time period under frequency conversion conditions. Extracting the time-domain waveform of the magnetic flux density of a high-frequency transformer within a complete high-frequency switching cycle. B ( t A fundamental calculus equation for the loss density of a transformer core at any transient moment is constructed. The transient loss density is calculated using this fundamental calculus equation, which is: In the formula, ΔB This represents the peak-to-peak value of the magnetic flux density within the current complete hysteresis loop. dB(t) / dt The instantaneous rate of change of magnetic flux density; α , β The dynamic correction coefficient is extracted from the physical properties of the magnetic core material; k 1 represents the waveform characteristic coefficient that is strongly correlated with the nonlinear magnetization law of the material; For the equivalent sinusoidal alternating magnetic field, the waveform characteristic coefficients are derived using the following formula. k 1: In the formula, C is the basic Steinmetz constant of the physical core material under standard operating conditions; It is the phase angle of the sinusoidal alternating magnetic field; The transient loss density is integrated over a complete switching cycle, and equivalent frequency mapping is performed using the piecewise equivalent frequency method. A modified Steinmetz formula, which incorporates material coefficients identified from a database, is then introduced to derive the total dynamic magnetic loss. P total The dynamic magnetic loss P total The expression is: In the formula, D This represents the phase shift angle between the primary and secondary sides of the high-frequency main circuit. f s This refers to the system's real-time switching frequency. B m The peak values ​​of magnetic flux density under square wave excitation with different duty cycles are given.

[0006] In one possible design, the specific steps for calculating the device losses of the main circuit power switch in the global loss model include: Establish transient and steady-state time-domain fitting models applicable to the voltage and current of power switches within adjacent switching cycles; Based on the transient and steady-state time-domain fitting model, the switching loss of the power switch in the transient process of turn-on and turn-off and the conduction loss in the steady-state process of conduction are solved respectively, and the device loss is obtained by summing the solved switching loss and conduction loss.

[0007] In one possible design, based on the transient and steady-state time-domain fitting model, the switching losses of the power switch during the turn-on and turn-off transient processes and the conduction losses during the conduction steady-state process are solved respectively. The device losses are obtained by summing the solved switching losses and conduction losses, including: The dynamic switching loss of a high-frequency switch is mapped as a time-varying function modulated by the real-time pulsating voltage and the time-varying high-frequency switching frequency. P switch The analytical model is: In the formula, P on.ac For the switching loss on the AC side, P on.dc This refers to the turn-off loss on the DC side. P off.ac For the switching loss on the AC side, P off.dc This refers to the turn-off loss on the DC side. Calculate the turn-on energy of AC-side power devices E on.ac The calculation formula is: In the formula, N ac This refers to the number of power devices connected in parallel on the AC side. i L ( t 0) represents the initial current during the switching cycle of the power device; Q oss.ac Output capacitor during the turn-on process C oss.ac The cumulative amount of charge; I c.ac The minimum current required to release this portion of the charge; v ac ( t () represents the instantaneous value of the AC side voltage at the moment the switching transistor is turned on; LThe inductance value of the series inductor in the converter; t To measure the turn-on time of the AC switching transistor; t 0 represents the number of switching transistors in the calculated switching cycle. S 1. The opening time.

[0008] Calculate the turn-on loss of power devices during the turn-on process P on.loss The calculation formula is: In the formula, E on.dc ( t )for; E on.dc.1 ( t )for; E on.dc.2 ( t )for; f s ( t )for; f The power grid frequency; Calculate the turn-off energy of AC-side power devices E off.ac The calculation formula is: In the formula, E on.ac ( t () is a function of the turn-on energy of the AC-side power device as a function of time; E on.dc.1 ( t () is a function of the turn-on energy of the first totem pole power device on the DC side as a function of time; E on.dc.2 ( t () is a function of the turn-on energy of the second totem pole power device on the DC side as a function of time; f s ( t () represents the switching frequency of the switching transistor; f The power grid frequency; Calculate the turn-off loss of AC-side power devices P off.ac The calculation formula is: The formula for calculating the turn-off energy of DC-side power devices is as follows: In the formula, t off.dc This refers to the turn-off time of the DC-side power devices. i L ( t2) is the current at the moment of turn-off. v dc ( t () represents the instantaneous value of the DC-side voltage. E off.dc.1 Power is turned off for the first totem-pole power device on the DC side; E off.dc.2 Power is cut off for the second totem pole power device on the DC side. i L ( t 1) The magnitude of the inductor current at the moment when the switching transistor S4 is turned on during the calculated switching cycle; Calculate the DC-side turn-off loss P off.dc The calculation formula is: In the formula, E off.dc.1 ( t () is a function of the turn-off energy of the first totem-pole power device on the DC side as a function of time. E off.dc.2 ( t () is a function of the turn-off energy of the second totem pole power device on the DC side as a function of time; Calculate conduction loss P cond The calculation formula is: In the formula, R ds.on It is the equivalent resistance between the drain and source when a single power device is turned on; N The number of power devices connected in parallel on the DC or AC side; I rms The total root-mean-square current is calculated using the following formula: In the formula, v dc ( t () represents the instantaneous value of the DC-side voltage; n For transformer turns ratio; D 1. D 2 represents the phase shift angle within the bridge of the entire bridge H1 and the phase shift angle between the bridges of the entire bridge H1 and the entire bridge H2, respectively. Dynamic switching losses P switch With conduction loss P cond Adding them together gives the total loss of the switching devices. P IGBT .

[0009] In one possible design, the multidimensional nonlinear constraint model includes an instantaneous power constraint model, a soft-switching interval constraint model, and a frequency limiting model; the method for parameter optimization of the global loss model includes: A comprehensive optimization function is constructed by introducing Lagrange multipliers for solving the problem. L(D, f s ,λ,u) The expression is: In the formula, P loss (D, f s ) The total system loss predicted by the global loss model; λ and u j These are the multiplier coefficients; h ( D, f s The equation constraint mapped by the instantaneous power constraint model is used to ensure the balance of bidirectional power transmission. g j ( D, f s The inequality constraint mapping the soft-switching interval constraint model to the frequency limiting model is used to ensure zero-voltage turn-on under all operating conditions and limit the frequency range. D This represents the phase shift angle between the primary and secondary sides of the high-frequency main circuit. f s This refers to the system's real-time switching frequency.

[0010] In one possible design, the instantaneous power constraint model maps to equality constraints. h ( D, f s The instantaneous power balance equation is satisfied, and the instantaneous power balance equation is: In the formula, L The inductance value of the series inductor; P ( D, f s () is the instantaneous power transferred by the converter; P out Set the output power for the converter; t For a moment; N For transformer turns ratio; D 1. D 2 represents the phase shift angle within the bridge of the entire bridge H1 and the phase shift angle between the bridges of the entire bridge H1 and the entire bridge H2, respectively. u in This is the input voltage of the full-bridge H1; uout This is the output voltage of the full-bridge H1; The inequality constraints mapping the soft-switching interval constraint model to the frequency limiting model g j ( D, f s The zero-voltage turn-on condition is satisfied, which is characterized by the following set of inequalities: In the formula, i L ( t 1) The magnitude of the inductor current at the moment when the switching transistor S4 is turned on during the calculated switching cycle; i L ( t 3) The magnitude of the inductor current at the moment when the switching transistor S5 is turned on during the calculated switching cycle; v dc This is the DC side voltage at the moment of turn-off; C oss,ac For AC-side switch junction capacitance; C oss,dc This refers to the junction capacitance of the DC-side switching transistor. L It is a transformer connected in series with an inductor.

[0011] In one possible design, the specific steps of introducing the feedforward reference into the closed-loop control architecture of the converter system, coupling it with the real-time closed-loop feedback adjustment to generate a comprehensive control command, and parsing it into a drive signal to drive the power switching devices in the single-stage high-frequency isolated main circuit include: Set the command reference value of the grid current based on the acquired bidirectional power demand command on the DC side; The deviation between the real-time feedback value of the grid current and the command reference value is calculated by the error regulator to generate a closed-loop dynamic compensation amount. The feedforward reference quantity is coupled with the closed-loop dynamic compensation quantity to generate a comprehensive control command that includes the compensated target switching frequency and target phase shift angle. The target switching frequency and target phase shift angle in the integrated control command are mapped to high-frequency modulation signals that control the primary and secondary full-bridge circuits in the single-stage high-frequency isolation main circuit. The zero-crossing signal of the grid voltage is extracted synchronously to generate a low-frequency drive signal to control the synchronous inversion bridge in the main circuit, so that the synchronous inversion bridge reverses its polarity at the zero-crossing point of the grid voltage, thereby realizing the bidirectional folding and unfolding conversion between pulsating DC power containing twice the power frequency component and AC power of the same frequency and phase.

[0012] Secondly, this application provides an energy efficiency optimization and coordinated control system for a bidirectional converter, the system comprising: Both DC and AC side interfaces support bidirectional energy flow. A single-stage high-frequency isolation main circuit is connected between the DC side interface and the AC side interface. The single-stage high-frequency isolation main circuit does not contain an intermediate DC voltage regulator capacitor. The single-stage high-frequency isolation main circuit includes, in sequence, a primary full-bridge, a series inductor, a high-frequency transformer, a secondary full-bridge, a high-frequency filter capacitor for filtering out switching frequency ripple, a synchronous inverting bridge, and an LR filter circuit. The controller is communicatively connected to the single-stage high-frequency isolated main circuit and is configured to execute the energy efficiency optimization and coordinated control method for the bidirectional converter as described in the first aspect and various possible designs of the first aspect.

[0013] Thirdly, embodiments of this application provide an electronic device, including: at least one processor and a memory; the memory stores computer-executable instructions; the at least one processor executes the computer-executable instructions stored in the memory, causing the at least one processor to execute the energy efficiency optimization and coordinated control method for the bidirectional converter as described in the first aspect and various possible designs of the first aspect.

[0014] Fourthly, embodiments of this application provide a computer-readable storage medium storing computer-executable instructions. When a processor executes the computer-executable instructions, it implements the energy efficiency optimization and coordinated control method for the bidirectional converter described in the first aspect and various possible designs of the first aspect.

[0015] Fifthly, embodiments of this application provide a computer program product, including a computer program that, when executed by a processor, implements the energy efficiency optimization and coordinated control method for bidirectional converters as described in the first aspect and various possible designs of the first aspect.

[0016] The energy efficiency optimization and coordinated control method, system, equipment, and medium for bidirectional converters provided in this application have at least the following beneficial effects: This application breaks through the limitation of existing single-stage variable frequency isolated bidirectional converters that can only achieve local or single-objective optimization, and proposes a global energy efficiency optimization framework for complex and variable operating conditions.

[0017] First, this application constructs a global loss model that includes the losses of high-precision transformer magnetic components and switching devices, especially addressing the complex magnetic losses of high-frequency transformers that are generally difficult to quantify in previous energy efficiency studies.

[0018] Secondly, at the optimization execution level, this application uses the constructed high-precision global loss model as the optimization target, and combines instantaneous power transmission balance, the full-condition zero-voltage soft switching (ZVS) range and frequency regulation range as multi-dimensional nonlinear constraints to optimize control parameters, thus completely avoiding the destruction of the converter's safe operation boundary and soft switching conditions during the optimization process.

[0019] Finally, this application introduces the optimal switching frequency and phase shift angle trajectory of the optimized output into the converter closed-loop control, successfully overcoming the control coupling problem caused by the lack of a large intermediate capacitor and the direct high-frequency coupling between AC and DC in a single-stage topology. Based on the above technical solution, this application not only ensures smooth and stable bidirectional power transmission of the system but also realizes efficient operation of the single-stage bidirectional converter under all operating conditions.

[0020] Additional aspects and advantages of this application will be set forth in part in the description which follows, and in part will be obvious from the description, or may be learned by practice of this application. Attached Figure Description

[0021] The accompanying drawings, which are incorporated in and form part of this specification, illustrate embodiments consistent with this application and, together with the description, serve to explain the principles of this application.

[0022] Figure 1 This application provides a schematic diagram of the main circuit topology of a single-stage variable frequency isolated bidirectional converter. Figure 2 A flowchart of an energy efficiency optimization and collaborative control method for a bidirectional converter provided in this application embodiment; Figure 3 This is a schematic diagram of the working waveform and the synthesis process of a symmetrical three-level square wave of a single-stage variable frequency isolated converter provided in the embodiments of this application; wherein, (a), the switching cycle T s Internal primary bridge output voltage u p Secondary side converted voltage nu s and series inductance or equivalent leakage inductance current i Lk (b) Typical operating waveforms; u in S(t) , u in S(tD 1 T) as well as nu out S(tD 2 T) The process of constructing a symmetrical three-level square wave; (c) Equivalent model; Figure 4 The schematic diagram of the asymmetric hysteresis loop prediction and abnormal loss magnetic field calculation based on the piecewise equivalent frequency method provided in the embodiments of this application; (a) Voltage under the original asymmetric square wave excitation u With magnetic flux density B The time-domain evolution curve, in duty cycle D Under the influence of magnetic flux density B In 0 to (1 -D ) T and (1) -D ) T to T The linear changes exhibit different slopes in the two time intervals; (b) and (c) indicate that the duty cycle will change. D Under the influence of magnetic flux density B In 0 to (1 -D ) T and (1) -D ) T to T The two segments within the two time intervals are respectively equivalent to sub-waveforms after local symmetrical excitation; Figure 5 This is a schematic diagram of the function Tr(t) used in the embodiments of this application; Figure 6 The open-loop trajectory diagram of the control variable for optimization output under multidimensional constraints provided in the embodiments of this application; (a) Optimal switching frequency per unit value f s *In power frequency phase ωt per-unit value of alternating current I ac *The open-loop trajectory surface within the three-dimensional operating plane; (b)Optimal phase-shift control quantity D 1 and D 2 The open-loop trajectory surface; Figure 7 A block diagram of a frequency-shifting coordinated control strategy coupled with global energy efficiency optimization and current closed-loop feedback provided for embodiments of this application; wherein, (a) current closed-loop control block diagram; (b) modulation and drive generation block diagram; Figure 8 This is a structural diagram of the energy efficiency optimization and collaborative control system for a bidirectional converter provided in an embodiment of this application.

[0023] The accompanying drawings illustrate specific embodiments of this application, which will be described in more detail below. These drawings and descriptions are not intended to limit the scope of the concept in any way, but rather to illustrate the concepts of this application to those skilled in the art through reference to specific embodiments. Detailed Implementation

[0024] Exemplary embodiments will now be described in detail, examples of which are illustrated in the accompanying drawings. When the following description relates to the drawings, unless otherwise indicated, the same numbers in different drawings represent the same or similar elements. The embodiments described in the following exemplary embodiments do not represent all embodiments consistent with this application. Rather, they are merely examples of systems and methods consistent with some aspects of this application as detailed in the appended claims.

[0025] The collection, storage, use, processing, transmission, provision, and disclosure of relevant data and information in the technical solution of this application all comply with the provisions of relevant laws and regulations and do not violate public order and good morals.

[0026] It should be noted that in the embodiments of this application, certain software, components, models and other existing solutions in the industry may be mentioned. These should be regarded as exemplary and are only intended to illustrate the feasibility of implementing the technical solution of this application. However, they do not mean that the applicant has used or necessarily used the solution.

[0027] The technical solution of this application and how the technical solution of this application solves the above-mentioned technical problems are described in detail below with specific embodiments. These specific embodiments can be combined with each other, and the same or similar concepts or processes may not be described again in some embodiments. The embodiments of this application will be described below with reference to the accompanying drawings.

[0028] This application provides an energy efficiency optimization and coordinated control method for a bidirectional converter. To facilitate understanding of the physical execution environment of this control method, the hardware main circuit topology upon which this embodiment is based is first described. Please refer to... Figure 1 The converter system used in this embodiment mainly includes a DC-side interface 101 for connecting a DC voltage source. U dc It may include a DC-side supporting capacitor. C dc and input-side inductor L dc AC side interface 102 is used to connect AC side voltage. u ac and AC side current i ac A single-stage high-frequency isolation main circuit 103 is disposed between the DC side interface 101 and the AC side interface 102. When the single-stage high-frequency isolation main circuit 103 is running, the DC side input current... i inAfter modulation by the primary-side full-bridge 1031 containing switching transistors S1-S4, high-frequency energy transfer is achieved through series inductor 1032 and high-frequency transformer 1033. The output voltage is then formed under the action of the secondary-side full-bridge 1034 containing switching transistors S5-S8 and high-frequency filter capacitor 1035. u out and output current i out Subsequently, the synchronous inversion bridge 1036 and LR The filter circuit 1037 completes the pulsating DC to AC side voltage conversion. u ac AC side current i ac The bidirectional folding and unfolding transformation. Figure 1 In u in , u out These represent the equivalent voltages on both sides of the high-frequency isolation stage. i out Indicates the secondary output current. i Lk This represents the current flowing through the series inductor.

[0029] To further explain the working mechanism of the above-mentioned single-stage high-frequency isolation main circuit, please refer to [link / reference needed]. Figure 3 . Figure 3 Figure (a) gives a switching cycle. T s Internal primary bridge output voltage u p Secondary side converted voltage nu s and series inductance or equivalent leakage inductance current i Lk The typical operating waveform, among which, t 0 to t 6 This is the boundary moment between adjacent working modes. T hs For half a switching cycle, D 1 Ths and D 2 Ths These represent the corresponding time delays determined by the phase-shift control quantity. In the figure, , , and These represent the conduction states of the corresponding bridge arm switching devices at different time periods.

[0030] In each operating mode, the primary-side output voltage u p Secondary-side voltage nus The difference between them acts on the series inductance or equivalent leakage inductance, thus determining the current. i Lk The slope changes. When u p Greater than nu s hour, i Lk Linear increase; when u p Less than nu s hour, i Lk Linear decrease; when u p and nu s When they are close to or equal, i Lk The slope of the change decreases. Therefore, the change in phase shift directly alters the duration of each mode and the shape of the current waveform.

[0031] Figure 3 (b) further illustrates the basis u in S(t) , u in S(tD 1 T) as well as nu out S(tD 2 T) The process of constructing a symmetrical three-level square wave. Figure 3 (c) shows its equivalent model, in which L t For equivalent transmission inductance, i Lt This corresponds to the current. Through this symmetrical three-level square wave synthesis process, the phase shift control quantity of the primary and secondary sides can be obtained. D 1. D 2. A clear correspondence is established with the voltage and current evolution process on the equivalent transmission inductance, providing waveform basis for subsequent root mean square current calculation, switching loss modeling, and instantaneous power transmission constraint solution.

[0032] Based on the above hardware topology environment, such as Figure 2 As shown, the energy efficiency optimization and collaborative control method for bidirectional converters in this application embodiment specifically includes the following steps S10-S40.

[0033] First, step S10 is executed to acquire real-time operating condition data of the converter system. This data includes at least the AC-side grid voltage and current, as well as the bidirectional power demand command from the DC side. During actual converter operation, the system controller uses high-precision sensors configured on the AC side to collect real-time data on the instantaneous amplitude and phase of the AC-side grid voltage and the feedback status of the grid current. Simultaneously, it receives bidirectional power demand commands from the DC side via the communication bus or energy management system.

[0034] After acquiring real-time operating condition data, this embodiment further executes step S20 to construct a system comprehensive energy efficiency evaluation framework for single-stage topologies, taking into account the physical characteristics of direct high-frequency AC-DC coupling in the single-stage main circuit. The system comprehensive energy efficiency evaluation framework includes a global loss model with the goal of minimizing the total predicted loss of the system and a multi-dimensional nonlinear constraint model for limiting the operating boundary of the converter. The global loss model includes the device losses of the main circuit power switching transistors and the magnetic component losses of the high-frequency transformer and series inductor.

[0035] This embodiment first extracts the high-frequency transformer within a complete high-frequency switching cycle. T s Time-domain waveform of magnetic flux density within B ( t For this non-sinusoidal waveform, this embodiment introduces an improved generalized Steinmetz physical model to construct the fundamental calculus equation for the loss density of the transformer core at any transient moment: In the formula, ΔB This represents the peak-to-peak value of the magnetic flux density within the current complete hysteresis loop. dB(t) / dt The instantaneous rate of change of magnetic flux density; α , β These are dynamic correction coefficients extracted from the specific physical properties of magnetic core materials; k 1 represents the waveform characteristic coefficient that is strongly correlated with the nonlinear magnetization law of the material. For an equivalent sinusoidal alternating magnetic field, this waveform characteristic coefficient... k 1 It can be obtained through rigorous derivation of the following formula: in, C This is the basic Steinmetz constant for the material under standard operating conditions; It is the phase angle of the sinusoidal alternating magnetic field.

[0036] After establishing the aforementioned transient loss differential equation, the piecewise equivalent frequency method is used to slice the asymmetric waveform unique to the single-stage topology into time-domain segments according to the various conduction and turn-off modes of the main circuit. This is affected by the phase shift angle between the primary and secondary sides. DDirect control of the magnetic flux density allows the magnetization and demagnetization processes within a single switching cycle to be divided into segments of varying durations. D*T s and (1-D)*T s The asymmetric linear segment. For specific implementation details, please refer to the relevant documentation. Figure 4 ,in, Figure 4 In the middle (a), the voltage under the original asymmetric square wave excitation is represented. u With magnetic flux density B The time-domain evolution curve, in duty cycle D Under the influence of magnetic flux density B In 0 to (1 -D ) T and (1) -D ) T to T The linear changes exhibit different slopes in the two time intervals; Figure 4 In diagrams (b) and (c), the aforementioned two segments are respectively equated to sub-waveforms after local symmetrical excitation, and the segmented equivalent frequencies are obtained accordingly. The duration is (1... -D ) T The equivalent frequency corresponding to the rising segment of the magnetic flux The equivalent frequency corresponding to the flux descent segment of duration DT , and The parsing expression is as follows: In the formula, The switching period of the original asymmetric square wave. This represents the duty cycle corresponding to the phase shift angle of the primary and secondary sides. Through the above equivalent frequency mapping, the magnetic loss calculation under asymmetric excitation can be decomposed into a loss calculation problem under two symmetric excitations, providing a time-domain mapping basis for the separate calculation of multiple loss components.

[0037] Figure 4 The complete calculation process for intermediate parameter identification follows the principle of loss separation: first, sinusoidal dynamic hysteresis loop data is measured at high frequencies to obtain magnetic flux density. B With total dynamic magnetic field strength H dyn The correspondence was then established; subsequently, the eddy current magnetic field was calculated using classical eddy current field theory. H cl The magnetic field of hysteresis loss was calculated using the Preisach model. H h The abnormal loss magnetic field is then obtained by separating it through the subtraction relationship of the magnetic field components. H ex The corresponding loss separation formula is: Based on the separation results at multiple frequencies, the abnormal loss statistical parameters corresponding to the rising and falling segments can be identified separately. These parameters characterize the nonlinear statistical features of the abnormal loss magnetic field as a function of the magnetic flux density under different flux change slopes. Building upon the parameter identification, for high-frequency square wave excitation with arbitrary duty cycles, the abnormal loss magnetic field throughout the entire switching cycle can be calculated using piecewise analytical expressions. Its piecewise calculation formula is: In the formula, The conductivity of the magnetic core material of a high-frequency transformer; is the geometric correction coefficient corresponding to the magnetic core lamination structure, used to characterize the geometric constraint effect of the magnetic core lamination structure on magnetic field loss; This represents the effective magnetic cross-sectional area of ​​the magnetic core. , These are the statistical characteristic coefficients of abnormal losses and the amplitude correction coefficients corresponding to the rising magnetic flux interval, respectively. , These are the statistical characteristic coefficients of abnormal loss and the amplitude correction coefficients corresponding to the magnetic flux decrease interval, respectively. Both sets of parameters were identified from high-frequency hysteresis loop experimental data. This represents the instantaneous flux density change rate within the corresponding time-domain interval during the switching cycle. This piecewise formula is used to characterize the dynamic evolution of the abnormal loss magnetic field in different slope intervals under asymmetric flux waveforms.

[0038] Figure 4 Dynamic hysteresis prediction is used to realize time-domain reconstruction of the total magnetic field under asymmetric square wave excitation: for input high-frequency square wave excitation with arbitrary duty cycle The hysteresis loss magnetic field was calculated using the Preisach model. The eddy current magnetic field was calculated using classical eddy current theory. The abnormal loss magnetic field is calculated using the above piecewise formula. The total dynamic magnetic field is finally obtained through the superposition of components. The corresponding superposition formula is: .

[0039] It should be noted that, This provides an instantaneous analytical calculation formula for the abnormal loss magnetic field, with specific expressions given for two time-domain segments within a single switching cycle. It serves as the solution formula for calculating the value of the abnormal loss magnetic field at any given moment. In dynamic hysteresis prediction, the time-domain component of the abnormal loss magnetic field under square wave excitation with arbitrary duty cycle is called the total dynamic magnetic field. Its instantaneous value throughout the entire period is obtained by substituting the above piecewise formula into the magnetic flux density change rate of the corresponding time period.

[0040] In this embodiment, the actual instantaneous magnetic flux density change rate within each linear time period is analyzed and substituted into the above-mentioned basic calculus equation to calculate the local loss of each slice mode.

[0041] Finally, the local losses of each mode are calculated over a complete switching cycle. T s Time-weighted integration and superposition aggregation are performed internally. Through rigorous mathematical integral derivation and equivalent frequency mapping, the total dynamic magnetic loss of this single-stage topology under complex asymmetric phase-shifting excitation is determined. P total The final, precise analytical physical model is derived as follows: In the final analytical model described above, D Characterizes the phase shift angle between the primary and secondary sides of the high-frequency main circuit; f s =1 / T s This refers to the system's real-time switching frequency. B m The peak magnetic flux density is the response under asymmetric square wave excitation with different duty cycles.

[0042] In the single-stage capacitorless topology of this application, the direct AC / DC coupling physical characteristic causes the terminal voltage of the power switch to exhibit significant power frequency ripple characteristics. Simultaneously, the system operates under a frequency conversion and phase-shift coordinated control mode, resulting in dynamic changes in both the switching frequency and on-state current of the devices. To accurately quantify this highly dynamic dissipation process, this embodiment analyzes the physical switching characteristics of the switching module and establishes transient and steady-state time-domain fitting models applicable to the voltage and current evolution within adjacent switching cycles. Specifically, regarding the dynamic switching losses of high-frequency switches P switch In this embodiment, the switching on and off processes on the AC and DC sides are mapped to being influenced by real-time pulsating voltage and time-varying high-frequency switching frequency. f s (t) The time-varying function of joint modulation. In a complete AC power grid frequency cycle (its frequency is... f The period is 1 / f Within the aforementioned transient overlap region between on and off states, the energy dissipation is reconstructed using time-domain integration. The analytical model for this dynamic switching loss is shown in the following equation: In the formula, P on.ac For the switching loss on the AC side, P on.dc This refers to the turn-off loss on the DC side. Poff.ac For the switching loss on the AC side, P off.dc This refers to the turn-off loss on the DC side.

[0043] Switching losses are generated by the interaction of voltage and current. The energy required for a single turn-on and a single turn-off is determined by the voltage and current waveforms during the switching process.

[0044] The turn-on energy of the AC-side power devices is as follows: In the formula, N ac This refers to the number of power devices connected in parallel on the AC side. i L ( t 0) represents the initial current during the switching cycle of the power device; Q oss.ac Output capacitor during the turn-on process C oss.ac The cumulative amount of charge; I c.ac The minimum current required to release this portion of the charge; v ac ( t () represents the instantaneous value of the AC side voltage at the moment the switching transistor is turned on; L The inductance value of the series inductor in the converter; t To measure the turn-on time of the AC switching transistor; t 0 represents the number of switching transistors in the calculated switching cycle. S 1. The opening time.

[0045] The turn-on energy of DC-side power devices is similar to that of AC-side devices, but the drain-source voltage and initial current during turn-on need to be determined based on the turn-on time of the power device. In summary, the turn-on loss of the power device during the turn-on process can be obtained. P on.loss As shown below: In the formula, E on.ac ( t () is a function of the turn-on energy of the AC-side power device as a function of time; E on.dc.1 ( t () is a function of the turn-on energy of the first totem pole power device on the DC side as a function of time; E on.dc.2 ( t () is a function of the turn-on energy of the second totem pole power device on the DC side as a function of time; f s ( t () represents the switching frequency of the switching transistor;f The power grid frequency; Turn-off energy of AC-side power devices E off.ac As shown below: In the formula, t off.ac For the turn-off time of the AC-side power devices, i L ( t 3) is the current at the moment of turn-off. v ac ( t () represents the instantaneous value of the AC side voltage at the moment the switching transistor is turned off; Therefore, the turn-off loss of the AC-side power device can be obtained as follows: Since the power devices on the DC side are affected by the internal phase shift angle, the turn-off energy of the power devices on the DC side needs to be discussed based on the actual situation.

[0046] First, the turn-off energy of the DC-side power devices is as follows: in, t off.dc This refers to the turn-off time of the DC-side power devices. i L ( t 2) is the current at the moment of turn-off. v dc ( t () represents the instantaneous value of the DC-side voltage. E off.dc.1 Power is turned off for the first totem-pole power device on the DC side; E off.dc.2 Power is cut off for the second totem pole power device on the DC side. i L ( t 1) The magnitude of the inductor current at the moment when the switching transistor S4 is turned on during the calculated switching cycle; Therefore, the turn-off loss on the DC side P off.dc for: In the formula, E off.dc.1 ( t () is a function of the turn-off energy of the first totem-pole power device on the DC side as a function of time. E off.dc.2 ( t () is a function of the turn-off energy of the second totem pole power device on the DC side as a function of time; The conduction loss can be calculated using the following formula: in, I rms This represents the total root-mean-square current; R ds.on It is the equivalent resistance between the drain and source when a single power device is turned on; N This refers to the number of power devices connected in parallel on the DC or AC side.

[0047] The root mean square current is shown in the following equation: in, v ac ( t () represents the instantaneous value of the AC side voltage; v dc ( t () represents the instantaneous value of the DC-side voltage; n For transformer turns ratio; D 1. D 2. D 3 and D 4 represents the phase-shift control quantity of DAB; f This refers to the power grid frequency.

[0048] Total losses of switching devices P IGBT for: After completing the above-mentioned construction, which includes high-frequency magnetic component losses... P total Total losses of switching devices P IGBT After obtaining a high-precision global loss model, the predicted total loss of the system can be expressed as: P loss =P total +P IGBT .

[0049] In order to minimize the predicted total loss under complex operating conditions, this embodiment further performs step S30, which optimizes the parameters of the global loss model within the dynamic feasible solution set defined by the multidimensional nonlinear constraint model based on the operating condition data, and obtains the optimal switching frequency and optimal phase shift angle that minimize the predicted total loss of the system under the current operating conditions, and uses them as the feedforward reference.

[0050] Specifically, because the single-stage high-frequency isolation main circuit eliminates the intermediate large DC capacitor, adjusting its control variables can easily disrupt the converter's original zero-voltage soft-switching condition and power transmission balance. Therefore, when solving for the control variables, it is necessary to construct multi-dimensional nonlinear constraints to define the safe operating boundaries of the converter. These multi-dimensional nonlinear constraints specifically include: Instantaneous power constraint: Used to ensure accurate bidirectional power transfer and real-time balance, forming an equality constraint, as shown in the instantaneous power balance equation below: In the formula, L The inductance value of the series inductor; P ( D, f s () is the instantaneous power transferred by the converter; P out Set the output power for the converter; t For a moment; N For transformer turns ratio; D 1. D 2 represents the phase shift angle within the bridge of the entire bridge H1 and the phase shift angle between the bridges of the entire bridge H1 and the entire bridge H2, respectively. u in This is the input voltage of the full-bridge H1; u out This is the output voltage of the full-bridge H1; In this embodiment, Tr ( t Functions such as Figure 5 As shown. Figure 5 The x-axis represents time. t The ordinate represents the function value. Tr(t) and mark 0、T、 2 T、 3 T and 4 T Waiting for the crucial moment. Tr ( t The function is a periodic piecewise trigonometric function: in 0≤t≤T Within the interval, Tr(t) It increases linearly from negative values, and in t=T It reached a peak value of 0.25. T ;exist T≤t≤ 2 T Within the interval, Tr(t) From 0.25 T Linear decrease, and in t=2T The value reached its lowest point - 0.25 T ; in 2 T≤t≤ 3 T and 3 T≤t≤4T The above pattern of change repeats within the interval, therefore Tr(t)It can be regarded as 2 T It is a periodically extended symmetric trigonometric function. This function is used to characterize the time-varying equivalent modulation trajectory in instantaneous power constraints, so as to establish the analytical relationship between output power, phase shift, and switching frequency.

[0051] Soft-switching range constraint: Used to ensure zero-voltage turn-on (ZVS) of the target power switch under all operating conditions, avoiding the huge losses and device stress caused by hard switching, as shown below: In the formula, i L ( t 1) The magnitude of the inductor current at the moment when the switching transistor S4 is turned on during the calculated switching cycle; i L ( t 3) The magnitude of the inductor current at the moment when the switching transistor S5 is turned on during the calculated switching cycle; v ac ( t () represents the instantaneous value of the AC side voltage; v dc This is the DC side voltage at the moment of turn-off; C oss,ac For AC-side switch junction capacitance; C oss,dc This refers to the junction capacitance of the DC-side switching transistor. L It is a transformer connected in series with an inductor.

[0052] Frequency limiting: Used to limit the switching frequency within the safe operating range (5kHz~20kHz) of hardware magnetic components and switching devices.

[0053] Within the dynamically feasible solution set defined by the aforementioned multidimensional nonlinear constraint model, this embodiment introduces the Lagrange multiplier method to construct a comprehensive optimization function. L(D,f s ,λ,u) Solve for the control variables: In the formula, P loss (D,f s ) The total system loss predicted by the global loss model; λ and u j These are the multiplier coefficients; h(D,f s ) The equality constraints mapped to the instantaneous power constraint model are used to ensure accurate bidirectional power transfer balance. g j(D, f s ) The inequality constraint mapping the soft-switching range constraint model to the frequency limiting model is used to ensure zero-voltage turn-on of the target power switch and limit the frequency range.

[0054] By solving for the extreme values ​​of the above-mentioned comprehensive optimization function, the system can obtain the optimal control variable trajectory (i.e., the optimal switching frequency) that minimizes the total predicted loss of the system under the current real-time operating conditions. f s With the optimal phase shift angle D (Sequence), and use it as the feedforward reference for the system. See also Figure 6 , Figure 6 (a) represents the optimal switching frequency per unit value. f s *In power frequency phase ωt per-unit value of alternating current I ac *The open-loop running trajectory surface within the three-dimensional working plane; Figure 6 (b) represents the optimal phase-shift control quantity corresponding to the above operating condition. D 1 and D 2 The open-loop trajectory surface. Figure 6 In (a), it is shown that in different ωt and I ac *Optimal switching frequency under operating conditions f s *It is not a constant, but changes continuously under multidimensional constraints; Figure 6 Figure (b) shows that the optimal phase shift is... D 1 and D 2 There is a synergistic adjustment relationship between them, and their values ​​are dynamically adjusted according to changes in operating conditions. Figure 6 The two subgraphs together illustrate that the control variable trajectory obtained by this application can maintain a continuous and achievable open-loop output within the feasible region jointly defined by the soft-switching interval, power balance constraint, and frequency limiting constraint, thereby providing a stable feedforward reference for subsequent closed-loop control.

[0055] After obtaining the feedforward reference, this embodiment executes step S40, which introduces the feedforward reference into the closed-loop control architecture of the converter system, couples it with the real-time closed-loop feedback adjustment to generate a comprehensive control command, and parses it into a drive signal to drive the power switching devices in the single-stage high-frequency isolation main circuit.

[0056] Please refer to the following: Figure 7The diagram shows a variable frequency phase-shift coordinated control strategy that couples global energy efficiency solution with current closed-loop feedback. During dynamic operation, the controller compares the actual feedback value of the grid current with the command reference value and generates a closed-loop dynamic compensation amount through an error regulator (such as a PI controller). This compensation amount is then compared with the optimal phase shift angle obtained in step S30. D and optimal switching frequency f s Feedforward coupling is performed, and the final mapping is a high-frequency PWM modulation signal that controls the primary and secondary full-bridge circuits.

[0057] Specifically, in Figure 7 In (a), the AC side current i ac After processing by the second-order generalized integrator SOGI, the following is obtained α , β 0 component, and the power frequency phase output of the phase-locked loop (PLL). Under the influence of Park transformation αβ0 → dq0 Obtain the d-axis current id and q-axis current i q . i d , i q After comparing with the reference value, the error signal i is obtained. d.err and i q.err Then through PI ( i d.err ) and PI ( i q.err After adjustment, input the complex number synthesis module C, which is composed of its real part. R e and the virtual part I m Generate amplitude respectively I ac and phase quantity φ b .

[0058] exist Figure 7 In (b), the DC side voltage U dc Amplitude I ac Phase quantity φ b and AC side voltage u ac As input to the modulation optimization module; wherein, the AC side voltage u ac The power frequency phase is extracted using PLL and Fourier modules. ω t and voltage fundamental amplitude U ac The above quantities are input together into the "Frequency Conversion Phase Shift Modulation and Optimization" module to solve for the optimal switching frequency. f s and optimal phase shift D The desired result f s and D Further input to the DAB drive signal generation unit generates... V 1a , V 1b , V 2a , V 2b , V 3a , V 3b , V 4a and V 4b High-frequency drive signals are used to drive high-frequency power devices in single-stage isolated converters. Simultaneously, the signal... u SR via symbolic function module sgn(u SR ) After subsequent logical transformations, it generates V 5a(6b) and V 6a(5b) Low-frequency drive signals are used to control the synchronous inverting bridge to complete polarity reversal near the zero-crossing point of the power frequency.

[0059] This application also provides an energy efficiency optimization and coordinated control system for a bidirectional converter, such as... Figure 8 As shown, the energy efficiency optimization and collaborative control system of the bidirectional converter includes: The DC side interface 101 and the AC side interface 102 support bidirectional energy flow; A single-stage high-frequency isolation main circuit 103 is connected between the DC side interface 101 and the AC side interface 102. The single-stage high-frequency isolation main circuit 103 does not contain an intermediate DC voltage regulator capacitor. The single-stage high-frequency isolation main circuit includes, in sequence, a primary-side full bridge 1031, a series inductor 1032, a high-frequency transformer 1033, a secondary-side full bridge 1034, a high-frequency filter capacitor 1035 for filtering out switching frequency ripple, a synchronous inverting bridge 1036, and an LR filter circuit 1037. The controller 104 is communicatively connected to the single-stage high-frequency isolated main circuit 103, and the controller 104 is configured to execute the energy efficiency optimization and collaborative control method of the bidirectional converter in the above embodiment.

[0060] This application provides an electronic device. The electronic device may include a processor and a memory, wherein the processor and the memory can communicate; exemplarily, the processor and the memory communicate via a communication bus.

[0061] The processor executes computer execution instructions stored in memory, causing the processor to perform the scheme in the above embodiments. The processor can be a general-purpose processor, including a central processing unit (CPU), a network processor (NP), etc.; it can also be a digital signal processor (DSP), an application-specific integrated circuit (ASIC), a field-programmable gate array (FPGA), or other programmable logic devices, discrete gate or transistor logic devices, or discrete hardware components.

[0062] The communication bus can be a peripheral component interconnect (PCI) bus or an extended industry standard architecture (EISA) bus, etc. The system bus can be divided into address bus, data bus, control bus, etc. Transceivers are used to enable communication between the database access system and other computers (e.g., clients, read-write libraries, and read-only libraries). Memory may include random access memory (RAM) and may also include non-volatile memory.

[0063] The electronic device provided in this application embodiment can be the terminal device described in the above embodiments.

[0064] This application also provides a computer-readable storage medium storing computer instructions. When the computer instructions are executed on a computer, the computer performs the technical solution of the energy efficiency optimization and coordinated control method for the bidirectional converter described above.

[0065] This application also provides a computer program product, which includes a computer program stored in a computer-readable storage medium. At least one processor can read the computer program from the computer-readable storage medium. When the at least one processor executes the computer program, it can implement the technical solution of the energy efficiency optimization and coordinated control method of the bidirectional converter in the above embodiments.

[0066] In the several embodiments provided in this application, it should be understood that the disclosed devices and methods can be implemented in other ways. For example, the device embodiments described above are merely illustrative; for instance, the division of modules is only a logical functional division, and in actual implementation, there may be other division methods. For example, multiple modules may be combined or integrated into another system, or some features may be ignored or not executed. Furthermore, the coupling or direct coupling or communication connection shown or discussed may be through some interfaces; the indirect coupling or communication connection between systems or modules may be electrical, mechanical, or other forms.

[0067] The modules described as separate components may or may not be physically separate. The components shown as modules may or may not be physical units; that is, they may be located in one place or distributed across multiple network units. Some or all of the modules can be selected to implement the solution of this embodiment according to actual needs.

[0068] Furthermore, the functional modules in the various embodiments of this application can be integrated into one processing unit, or each module can exist physically separately, or two or more modules can be integrated into one unit. The unit composed of the above modules can be implemented in hardware or in the form of hardware plus software functional units.

[0069] The integrated modules described above, implemented as software functional modules, can be stored in a computer-readable storage medium. These software functional modules, stored in a storage medium, include several instructions to cause a computer device (which may be a personal computer, server, or network device, etc.) or processor to execute some steps of the methods of the various embodiments of this application.

[0070] It should be understood that the aforementioned processor can be a Central Processing Unit (CPU), or other general-purpose processors, digital signal processors (DSPs), application-specific integrated circuits (ASICs), etc. A general-purpose processor can be a microprocessor or any conventional processor. The steps of the method disclosed in this invention can be directly manifested as execution by a hardware processor, or execution by a combination of hardware and software modules within the processor.

[0071] The memory may include high-speed RAM, and may also include non-volatile storage (NVM), such as at least one disk storage device, and may also be a USB flash drive, external hard drive, read-only memory, disk or optical disc, etc.

[0072] Buses can be Industry Standard Architecture (ISA) buses, Peripheral Component Interconnect (PCI) buses, or Extended Industry Standard Architecture (EISA) buses, etc. Buses can be categorized into address buses, data buses, control buses, etc.

[0073] The aforementioned storage medium can be implemented by any type of volatile or non-volatile storage device or a combination thereof, such as static random access memory (SRAM), electrically erasable programmable read-only memory (EEPROM), erasable programmable read-only memory (EPROM), programmable read-only memory (PROM), read-only memory (ROM), magnetic storage, flash memory, magnetic disk, or optical disk. The storage medium can be any available medium that can be accessed by a general-purpose or special-purpose computer.

[0074] An exemplary storage medium is coupled to a processor, enabling the processor to read information from and write information to the storage medium. Alternatively, the storage medium can be an integral part of the processor. The processor and storage medium can reside in an Application Specific Integrated Circuit (ASIC). Alternatively, the processor and storage medium can exist as discrete components in an electronic control unit or main control device.

[0075] Those skilled in the art will understand that all or part of the steps of the above-described method embodiments can be implemented by hardware related to program instructions. The aforementioned program can be stored in a computer-readable storage medium. When executed, the program performs the steps of the above-described method embodiments; and the aforementioned storage medium includes various media capable of storing program code, such as ROM, RAM, magnetic disks, or optical disks.

[0076] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of this application, and are not intended to limit them. Although this application has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some or all of the technical features therein. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the scope of the technical solutions of the embodiments of this application.

Claims

1. A method for optimizing the energy efficiency of a bidirectional converter through coordinated control, applied to a single-stage high-frequency isolated main circuit that does not have an internal intermediate DC voltage regulator capacitor, characterized in that... The method includes: Acquire real-time operating condition data of the converter system, including at least the grid voltage and grid current on the AC side and the bidirectional power demand command on the DC side. A system comprehensive energy efficiency evaluation framework for single-stage topologies is constructed. The system comprehensive energy efficiency evaluation framework includes a global loss model with the goal of minimizing the total predicted loss of the system and a multidimensional nonlinear constraint model for limiting the operating boundary of the converter. The global loss model includes the device losses of the main circuit power switching transistors and the magnetic component losses of the high-frequency transformer and series inductor. Based on the operating condition data, within the dynamic feasible solution set defined by the multidimensional nonlinear constraint model, the parameters of the global loss model are optimized to obtain the optimal switching frequency and optimal phase shift angle that minimize the total predicted loss of the system under the current operating condition, and these are used as the feedforward reference. The feedforward reference is introduced into the closed-loop control architecture of the converter system, coupled with the real-time closed-loop feedback adjustment to generate a comprehensive control command, which is then parsed into a drive signal that drives the power switching devices in the single-stage high-frequency isolated main circuit.

2. The method according to claim 1, characterized in that, In the global loss model, the specific steps for calculating the loss of the magnetic component include: The piecewise equivalent frequency method is used to analyze the non-sinusoidal and high-frequency transient magnetic field parameters of asymmetric duty cycle and each conduction and turn-off time period under frequency conversion conditions. Extracting the time-domain waveform of the magnetic flux density of a high-frequency transformer within a complete high-frequency switching cycle. B ( t A fundamental calculus equation for the loss density of a transformer core at any transient moment is constructed. The transient loss density is calculated using this fundamental calculus equation, which is: In the formula, ΔB This represents the peak-to-peak value of the magnetic flux density within the current complete hysteresis loop. dB(t) / dt The instantaneous rate of change of magnetic flux density; α , β The dynamic correction coefficient is extracted from the physical properties of the magnetic core material; k 1 represents the waveform characteristic coefficient that is strongly correlated with the nonlinear magnetization law of the material; For the equivalent sinusoidal alternating magnetic field, the waveform characteristic coefficients are derived using the following formula. k 1: In the formula, C is the basic Steinmetz constant of the physical core material under standard operating conditions; It is the phase angle of the sinusoidal alternating magnetic field; The transient loss density is integrated over a complete switching cycle, and equivalent frequency mapping is performed using the piecewise equivalent frequency method. A modified Steinmetz formula, which incorporates material coefficients identified from a database, is then introduced to derive the total dynamic magnetic loss. P total The dynamic magnetic loss P total The expression is: In the formula, D This represents the phase shift angle between the primary and secondary sides of the high-frequency main circuit. f s This refers to the system's real-time switching frequency. B m The peak values ​​of magnetic flux density under square wave excitation with different duty cycles are given.

3. The method according to claim 1, characterized in that, In the global loss model, the specific steps for calculating the device losses of the main circuit power switch include: Establish transient and steady-state time-domain fitting models applicable to the voltage and current of power switches within adjacent switching cycles; Based on the transient and steady-state time-domain fitting model, the switching loss of the power switch in the transient process of turn-on and turn-off and the conduction loss in the steady-state process of conduction are solved respectively, and the device loss is obtained by summing the solved switching loss and conduction loss.

4. The method according to claim 3, characterized in that, Based on the transient and steady-state time-domain fitting model, the switching losses of the power switch during the turn-on and turn-off transient processes and the conduction losses during the conduction steady-state process are solved respectively. The device losses are obtained by summing the solved switching losses and conduction losses, including: The dynamic switching loss of a high-frequency switch is mapped as a time-varying function modulated by the real-time pulsating voltage and the time-varying high-frequency switching frequency. P switch The analytical model is: In the formula, P on.ac For the switching losses on the AC side, P on.dc This refers to the turn-off loss on the DC side. P off.ac For the switching losses on the AC side, P off.dc This refers to the turn-off loss on the DC side. Calculate the turn-on energy of AC-side power devices E on.ac The calculation formula is: In the formula, N ac This refers to the number of power devices connected in parallel on the AC side. i L ( t 0) represents the initial current during the switching cycle of the power device; Q oss.ac Output capacitor during the turn-on process C oss.ac The cumulative amount of charge; I c.ac The minimum current required to release this portion of the charge; v ac ( t () represents the instantaneous value of the AC side voltage at the moment the switching transistor is turned on; L The inductance value of the series inductor in the converter; t To measure the turn-on time of the AC switching transistor; t 0 represents the number of switching transistors in the calculated switching cycle. S 1. The opening time; Calculate the turn-on loss of power devices during the turn-on process P on.loss The calculation formula is: In the formula, E on.ac ( t () is a function of the turn-on energy of the AC-side power device as a function of time; E on.dc.1 ( t () is a function of the turn-on energy of the first totem pole power device on the DC side as a function of time; E on.dc.2 ( t () is a function of the turn-on energy of the second totem pole power device on the DC side as a function of time; f s ( t () represents the switching frequency of the switching transistor; f The power grid frequency; Calculate the turn-off energy of AC-side power devices E off.ac The calculation formula is: In the formula, t off.ac For the turn-off time of the AC-side power devices, i L ( t 3) is the current at the moment of turn-off. v ac ( t () represents the instantaneous value of the AC side voltage at the moment the switching transistor is turned off; Calculate the turn-off loss of AC-side power devices P off.ac The calculation formula is: The formula for calculating the turn-off energy of DC-side power devices is as follows: In the formula, t off.dc This refers to the turn-off time of the DC-side power devices. i L ( t 2) is the current at the moment of turn-off. v dc ( t () represents the instantaneous value of the DC-side voltage. E off.dc.1 Power is turned off for the first totem-pole power device on the DC side; E off.dc.2 Power is cut off for the second totem pole power device on the DC side. i L ( t 1) The magnitude of the inductor current at the moment when the switching transistor S4 is turned on during the calculated switching cycle; Calculate the DC-side turn-off loss P off.dc The calculation formula is: In the formula, E off.dc.1 ( t () is a function of the turn-off energy of the first totem-pole power device on the DC side as a function of time. E off.dc.2 ( t () is a function of the turn-off energy of the second totem pole power device on the DC side as a function of time; Calculate conduction loss P cond The calculation formula is: In the formula, R ds.on It is the equivalent resistance between the drain and source when a single power device is turned on; N The number of power devices connected in parallel on the DC or AC side; I rms The total root-mean-square current is calculated using the following formula: In the formula, v dc ( t () represents the instantaneous value of the DC-side voltage; n For transformer turns ratio; D 1. D 2 represents the phase shift angle within the bridge of the entire bridge H1 and the phase shift angle between the bridges of the entire bridge H1 and the entire bridge H2, respectively. Dynamic switching losses P switch With conduction loss P cond Adding them together gives the total loss of the switching devices. P IGBT .

5. The method according to claim 1, characterized in that, The multidimensional nonlinear constraint model includes an instantaneous power constraint model, a soft-switching interval constraint model, and a frequency limiting model; the methods for parameter optimization of the global loss model include: A comprehensive optimization function is constructed by introducing Lagrange multipliers for solving the problem. L(D, f s ,λ,u) The expression is: In the formula, P loss (D, f s ) The total system loss predicted by the global loss model; λ and u j These are the multiplier coefficients; h ( D, f s The equation constraint mapped by the instantaneous power constraint model is used to ensure the balance of bidirectional power transmission. g j ( D, f s The inequality constraint mapping the soft-switching interval constraint model to the frequency limiting model is used to ensure zero-voltage turn-on under all operating conditions and limit the frequency range. D This represents the phase shift angle between the primary and secondary sides of the high-frequency main circuit. f s This refers to the system's real-time switching frequency.

6. The method according to claim 5, characterized in that, The equality constraints mapped by the instantaneous power constraint model h ( D, f s The instantaneous power balance equation is satisfied, and the instantaneous power balance equation is: In the formula, L The inductance value of the series inductor; P ( D, f s () is the instantaneous power transferred by the converter; P out Set the output power for the converter; t For a moment; N For transformer turns ratio; D 1. D 2 represents the phase shift angle within the bridge of the entire bridge H1 and the phase shift angle between the bridges of the entire bridge H1 and the entire bridge H2, respectively. u in This is the input voltage of the full-bridge H1; u out This is the output voltage of the full-bridge H1; The inequality constraints mapping the soft-switching interval constraint model to the frequency limiting model g j ( D, f s The zero-voltage turn-on condition is satisfied, which is characterized by the following set of inequalities: In the formula, i L ( t 1) The magnitude of the inductor current at the moment when the switching transistor S4 is turned on during the calculated switching cycle; i L ( t 3) The magnitude of the inductor current at the moment when the switching transistor S5 is turned on during the calculated switching cycle; v ac ( t () represents the instantaneous value of the AC side voltage; v dc This is the DC side voltage at the moment of turn-off; C oss,ac For AC-side switch junction capacitance; C oss,dc This refers to the junction capacitance of the DC-side switching transistor. L It is a transformer connected in series with an inductor.

7. The method according to claim 1, characterized in that, The specific steps of introducing the feedforward reference into the closed-loop control architecture of the converter system, coupling it with the real-time closed-loop feedback adjustment to generate a comprehensive control command, and parsing it into a drive signal to drive the power switching devices in the single-stage high-frequency isolation main circuit include: Set the command reference value of the grid current based on the acquired bidirectional power demand command on the DC side; The deviation between the real-time feedback value of the grid current and the command reference value is calculated by the error regulator to generate a closed-loop dynamic compensation amount. The feedforward reference quantity is coupled with the closed-loop dynamic compensation quantity to generate a comprehensive control command that includes the compensated target switching frequency and target phase shift angle. The target switching frequency and target phase shift angle in the integrated control command are mapped to high-frequency modulation signals that control the primary and secondary full-bridge circuits in the single-stage high-frequency isolation main circuit. The zero-crossing signal of the grid voltage is extracted synchronously to generate a low-frequency drive signal to control the synchronous inversion bridge in the main circuit, so that the synchronous inversion bridge reverses its polarity at the zero-crossing point of the grid voltage, thereby realizing the bidirectional folding and unfolding conversion between pulsating DC power containing twice the power frequency component and AC power of the same frequency and phase.

8. An energy efficiency optimization and collaborative control system for a bidirectional converter, characterized in that, The system includes: Both DC and AC side interfaces support bidirectional energy flow. A single-stage high-frequency isolation main circuit is connected between the DC side interface and the AC side interface. The single-stage high-frequency isolation main circuit does not contain an intermediate DC voltage regulator capacitor. The single-stage high-frequency isolation main circuit includes, in sequence, a primary full-bridge, a series inductor, a high-frequency transformer, a secondary full-bridge, a high-frequency filter capacitor for filtering out switching frequency ripple, a synchronous inverting bridge, and an LR filter circuit. The controller is communicatively connected to the single-stage high-frequency isolated main circuit, and the controller is configured to perform the energy efficiency optimization and collaborative control method for the bidirectional converter as described in any one of claims 1-7.

9. An electronic device, characterized in that, include: A processor, and a memory communicatively connected to the processor; The memory stores computer-executed instructions; The processor executes computer execution instructions stored in the memory to implement the energy efficiency optimization and collaborative control method for the bidirectional converter as described in any one of claims 1-7.

10. A computer-readable storage medium, characterized in that, The computer-readable storage medium stores computer-executable instructions, which, when executed by a processor, are used to implement the energy efficiency optimization and coordinated control method for a bidirectional converter as described in any one of claims 1-7.

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