Loop-free adaptive inertia network configuration control method and system

Abstract

The application discloses a non-locked loop adaptive inertia network construction control method and system, comprising obtaining current three-phase voltage and current and current instantaneous power according to an instantaneous power calculation strategy; obtaining current electric angular frequency and determining current inertia according to a double-dimension adaptive inertia determination strategy and current frequency change rate; determining current voltage phase in combination with a power direct self-synchronization mechanism; inputting the current three-phase voltage and current into a non-current inner loop direct voltage control model to obtain current adjusted two-phase voltage and process current three-phase modulation wave; and generating a current pulse width modulation signal through space vector pulse width modulation. The embodiment of the application can adopt a non-locked loop direct power synchronization mechanism, directly control the output voltage phase based on instantaneous active power deviation, cancel the locked loop from the architecture level, and does not need to measure the grid phase through the locked loop, thereby avoiding the problems of weak grid instability and fixed virtual inertia that cannot be dynamically adjusted.

Description

Technical Field

[0001] This application relates to the field of intelligent control engineering technology for energy storage power stations, and in particular to a phase-locked loop-free adaptive inertia network control method and system. Background Technology

[0002] Currently, with the deepening of the construction of new power systems, the penetration rate of new energy sources such as wind power and photovoltaics continues to increase, and the power system's electronic characteristics are becoming increasingly prominent. The installed capacity of traditional synchronous generators is declining, the total inertia of the system is decreasing, and frequency stability issues are becoming increasingly prominent. Grid-forming converters, by simulating the external characteristics of synchronous generators through control algorithms, can actively provide voltage and inertia support to the grid, and have become a key core technology for solving system stability problems under high-proportion new energy access.

[0003] Currently, there are three main technical solutions for grid-based control. The first is traditional virtual synchronous generator (VSG) control, which detects the grid phase through a phase-locked loop (PLL) and simulates the characteristics of a synchronous generator based on the rotor motion equation. The second is droop control, which achieves power distribution based on the droop characteristics of Pf and QU. It has a simple structure but lacks inertia simulation capabilities. The third is an improved virtual synchronous generator scheme, which adds various improved components to the VSG, but the virtual inertia parameters remain fixed and cannot be dynamically adjusted.

[0004] The above-mentioned existing technology has the following technical drawbacks: 1) There is a problem of phase-locked loop instability in weak power grids. Traditional VSG control relies heavily on phase-locked loops for synchronization. Under weak power grid conditions (SCR<3, where SCR stands for Short Circuit Ratio), phase-locked loops are prone to phase detection errors, which can lead to power oscillations and ultimately cause system instability and grid disconnection. This is the biggest bottleneck in the application of current grid construction technology. 2) There is a problem of fixed and inflexible virtual inertia. The existing solution has fixed virtual inertia parameters, which cannot be dynamically adjusted according to the power grid conditions. If the inertia is set too large, the system response will be slow. If it is set too small, the inertia support will be insufficient. It is impossible to simultaneously take into account dynamic performance and steady-state performance. 3) There is a problem of the current inner loop limiting the response speed. In the traditional dual-loop control architecture, the existence of the current inner loop limits the control bandwidth, and the inertia response time is usually on the order of 80-150ms, which cannot meet the requirements of fast frequency support. Summary of the Invention

[0005] This application provides a phase-locked loop-free adaptive inertia grid-building control method and system, aiming to solve the problems in the prior art where grid-building converters use phase-locked loops to measure the grid phase during grid-building control, resulting in instability in weak grids and fixed virtual inertia that cannot be dynamically adjusted.

[0006] In a first aspect, embodiments of this application provide a phase-locked loop-free adaptive inertia network control method, which includes: In response to the grid control command generated according to the preset control cycle, the current three-phase voltage and current three-phase current are obtained, and the corresponding current instantaneous power is obtained according to the preset instantaneous power calculation strategy; The current electrical angular frequency corresponding to the current virtual synchronizer is obtained, and the current inertia is determined according to the preset two-dimensional adaptive inertia determination strategy, the current electrical angular frequency and the corresponding current frequency change rate; the two-dimensional adaptive inertia determination strategy is used to determine the current inertia based on the first inertia adjustment value determined by the difference between the current electrical angular frequency and the preset rated instantaneous frequency and the second inertia adjustment value determined by the current electrical angular frequency, and the constraint condition corresponding to the two-dimensional adaptive inertia determination strategy is that the inertia value range is [1s, 20s]; The current voltage phase is determined based on the current instantaneous power, the current inertia, and the preset power direct self-synchronization mechanism. The current three-phase voltage and the current three-phase current are transformed by Parker transformation based on the current voltage phase angle in the current voltage phase and then input to a preset direct voltage control model without current inner loop to obtain the current adjusted two-phase voltage; the direct voltage control model without current inner loop includes a voltage control loop but does not include a current inner loop; The current two-phase voltages after adjustment are subjected to inverse Parker transform and inverse Clarke transform in sequence to obtain the current three-phase modulated wave; The current three-phase modulated wave is generated into a current pulse width modulation signal through space vector pulse width modulation, and then input to the connected insulated gate bipolar transistor driver board; wherein, the insulated gate bipolar transistor driver board is located in the energy storage converter connected to the energy storage power station control system.

[0007] Secondly, embodiments of this application also provide a phase-locked loop-free adaptive inertia network construction control system, which includes: The sampling unit is used to respond to the grid control command generated according to the preset control cycle, obtain the current three-phase voltage and current three-phase current, and obtain the corresponding current instantaneous power according to the preset instantaneous power calculation strategy; The inertia acquisition unit is used to acquire the current electrical angular frequency corresponding to the current virtual synchronizer, and determine the current inertia according to the preset two-dimensional adaptive inertia determination strategy, the current electrical angular frequency and the corresponding current frequency change rate. The voltage phase acquisition unit is used to determine the current voltage phase based on the current instantaneous power, the current inertia, and a preset power direct self-synchronization mechanism. The voltage phase acquisition unit is used to input the current three-phase voltage and the current three-phase current into a preset current-free inner loop direct voltage control model after performing Parker transformation based on the current voltage phase angle in the current voltage phase, so as to obtain the current adjusted two-phase voltage. The three-phase modulation wave acquisition unit is used to sequentially perform inverse Parker transform and inverse Clarke transform on the current adjusted two-phase voltage to obtain the current three-phase modulation wave. The modulation signal generation unit is used to generate a current pulse width modulation signal from the current three-phase modulation wave through space vector pulse width modulation, and input it to the connected insulated gate bipolar transistor driver board; wherein the insulated gate bipolar transistor driver board is located in the energy storage converter connected to the energy storage power station control system.

[0008] Thirdly, embodiments of the present invention also provide a computer device, which includes a memory and a processor, wherein the memory stores a computer program, and the processor executes the computer program to implement the method described in the first aspect above.

[0009] Fourthly, embodiments of the present invention also provide a computer-readable storage medium storing a computer program, the computer program including program instructions that, when executed by a processor, can implement the method described in the first aspect above.

[0010] This application provides a phase-locked loop-free adaptive inertia network construction control method and system. The method includes: responding to a network construction control command generated according to a preset control cycle; acquiring the current three-phase voltage and current three-phase current, and acquiring the corresponding current instantaneous power according to a preset instantaneous power calculation strategy; acquiring the current electrical angular frequency corresponding to the current virtual synchronizer, and determining the current inertia according to a preset two-dimensional adaptive inertia determination strategy, the current electrical angular frequency, and the corresponding current frequency change rate; determining the current voltage phase according to the current instantaneous power, the current inertia, and a preset power direct self-synchronization mechanism; and... The current three-phase voltage and current are input to a preset current-free inner-loop direct voltage control model after undergoing Parker transformation based on the current voltage phase angle in the current voltage phase, to obtain the current regulated two-phase voltage. The regulated two-phase voltage is then subjected to inverse Parker transformation and inverse Clarke transformation sequentially to obtain the current three-phase modulated wave. This current three-phase modulated wave is then sequentially processed through space vector pulse width modulation to generate a current pulse width modulation signal, which is input to the connected insulated-gate bipolar transistor (IGBT) driver board. The IGBT driver board is located within the energy storage converter connected to the energy storage power station control system. This embodiment of the application employs a phase-locked loop (PLL)-free direct power synchronization mechanism, directly controlling the output voltage phase based on instantaneous active power deviation. This eliminates the need for a PLL at the architectural level and also avoids the problems of PLL instability in weak grids and the inability to dynamically adjust fixed virtual inertia. Attached Figure Description

[0011] To more clearly illustrate the technical solutions of the embodiments of this application, the drawings used in the description of the embodiments will be briefly introduced below. Obviously, the drawings described below are some embodiments of this application. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0012] Figure 1 A schematic diagram illustrating an application scenario of the phase-locked loop-free adaptive inertia network control method provided in this application embodiment; Figure 2 A schematic flowchart illustrating the phase-locked loop-free adaptive inertia network control method provided in this application embodiment; Figure 3 This is a schematic diagram of the first sub-process of the phase-locked loop-free adaptive inertia network control method provided in the embodiments of this application; Figure 4 This is a schematic diagram of the second sub-process of the phase-locked loop-free adaptive inertia network control method provided in the embodiments of this application; Figure 5 This is a schematic diagram of the third sub-process of the phase-locked loop-free adaptive inertia network control method provided in the embodiments of this application; Figure 6This is a schematic diagram of the fourth sub-process of the phase-locked loop-free adaptive inertia network control method provided in the embodiments of this application; Figure 7 This is a schematic diagram of the fifth sub-process of the phase-locked loop-free adaptive inertia network control method provided in the embodiments of this application; Figure 8 This is a schematic diagram of the sixth sub-process of the phase-locked loop-free adaptive inertia network control method provided in the embodiments of this application; Figure 9 A schematic block diagram of the phase-locked loop-free adaptive inertia network control system provided in an embodiment of the present invention; Figure 10 A schematic block diagram of a computer device provided for an embodiment of the present invention. Detailed Implementation

[0013] The technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of this application, not all embodiments. Based on the embodiments of this application, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of this application.

[0014] It should be understood that, when used in this specification and the appended claims, the terms "comprising" and "including" indicate the presence of the described features, integrals, steps, operations, elements and / or components, but do not exclude the presence or addition of one or more other features, integrals, steps, operations, elements, components and / or collections thereof.

[0015] It should also be understood that the terminology used in this specification is for the purpose of describing particular embodiments only and is not intended to limit the scope of the application. As used in this specification and the appended claims, the singular forms “a,” “an,” and “the” are intended to include the plural forms unless the context clearly indicates otherwise.

[0016] It should also be further understood that the term “and / or” as used in this application specification and the appended claims means any combination of one or more of the associated listed items and all possible combinations, and includes such combinations.

[0017] Please also refer to Figures 1-2 ,in Figure 1 A schematic diagram illustrating an application scenario of the phase-locked loop-free adaptive inertia network control method provided in this application embodiment; Figure 2 This is a schematic flowchart illustrating the phase-locked loop-free adaptive inertia network control method provided in an embodiment of this application. Figure 1As shown, the phase-locked loop-free adaptive inertia network control method is applied to the energy storage power station control system 10. The energy storage power station control system 10 is deployed in the energy storage power station and is connected to the power grid 20 (i.e., the mains power grid), and also connected to the energy storage converter group 30, which includes multiple energy storage converters. Figure 2 As shown, the phase-locked loop-free adaptive inertia network control method includes steps S110 to S160.

[0018] S110: In response to the grid control command generated according to the preset control cycle, obtain the current three-phase voltage and current three-phase current, and obtain the corresponding current instantaneous power according to the preset instantaneous power calculation strategy.

[0019] In this embodiment, the power storage station control system 10 can be pre-set with a preset control cycle (e.g., 50 μs; however, the value can be flexibly adjusted according to actual needs and is not limited to the listed values). Whenever a grid connection control command is detected, the current three-phase voltage and current are collected from the power grid via the current sampling module and voltage sampling module connected to the power storage station control system. Then, after a series of transformations, data processing, and calculations using the instantaneous power calculation strategy, the current instantaneous power is obtained.

[0020] In one embodiment, such as Figure 3 As shown, step S110 includes: S111. The current three-phase voltage and the current three-phase current are both subjected to first-order recursive low-pass filtering and sampling correction processing to obtain the current corrected three-phase voltage and the current corrected three-phase current. S112. Obtain the Clarke transformation matrix in the instantaneous power calculation strategy, and transform the current corrected three-phase voltage and the current corrected three-phase current through the Clarke transformation matrix to obtain the first transformed two-phase voltage and the first transformed two-phase current in the αβ domain; wherein, the αβ domain corresponds to the two-phase stationary coordinate system; S113. Obtain the Parker transformation matrix in the instantaneous power calculation strategy, and transform the first transformed two-phase voltage and the first transformed two-phase current through the Parker transformation matrix to obtain the second transformed two-phase voltage and the second transformed two-phase current in the dq domain; wherein, the dq domain corresponds to an orthogonal rotating coordinate system that rotates synchronously with the grid voltage vector; S114. The two-phase voltage and the two-phase current after the second transformation are both used as input parameters of the three-phase instantaneous active power acquisition model and the three-phase instantaneous reactive power acquisition model in the instantaneous power calculation strategy to obtain the current instantaneous active power and the current instantaneous reactive power, and form the current instantaneous power.

[0021] In this embodiment, when the current three-phase voltage and the current three-phase current are both processed by first-order recursive low-pass filtering and sampling correction, specifically, first-order recursive low-pass filtering (i.e., first-order IIR low-pass filtering, IIR stands for Infinite Impulse Response) is used to filter out high-frequency noise in the current three-phase voltage and the current three-phase current. Then, DC bias correction (subtracting the sampling zero-point drift from both the filtered output three-phase voltage and three-phase current, for example, subtracting the sampling zero-point drift voltage from the filtered output three-phase voltage and subtracting the sampling zero-point drift current from the filtered output three-phase current) or phase synchronization correction (introducing phase offset into both the filtered output three-phase voltage and three-phase current for correction, so as to synchronize with the ADC sampling time of the three-phase voltage and the ADC sampling time of the three-phase current) is performed to obtain the current corrected three-phase voltage and the current corrected three-phase current.

[0022] Then, the current corrected three-phase voltages and currents are transformed using the Clarke transform matrix, converting the voltages or currents of phases a, b, and c into α and β components in a two-phase stationary coordinate system in the αβ domain. If the three-phase voltages in the current corrected three-phase voltages are denoted as ua, ub, and uc, and the current corrected three-phase currents as ia, ib, and ic, respectively, the Clarke transform matrix is ​​represented by T. Clarke Let the two-phase voltages after the first transformation be denoted as [uα;uβ], and the two-phase currents after the first transformation be denoted as [iα;iβ], then [uα;uβ] = (2 / 3) * T Clarke *[ua;ub;uc],[iα;iβ]=(2 / 3)*T Clarke *[ia; ib; ic].

[0023] Subsequently, both the first transformed two-phase voltages and the first transformed two-phase currents are transformed using the Parker transformation matrix. This transforms the α and β components in the two-phase stationary coordinate system of the αβ domain into a dq coordinate system rotating at an angular frequency using a virtual synchronizing machine. The dq domain in the dq coordinate system corresponds to an orthogonal rotating coordinate system that rotates synchronously with the grid voltage vector. If the second transformed two-phase voltages are denoted as ud and uq, and the second transformed two-phase currents as id and iq, the Parker transformation matrix is ​​represented by T. park This means that [ud;uq]=T park *[uα;uβ]=[uαcosθ+ uβsinθ;-uαsinθ+ uβcosθ], [id;iq]=T park *[iα;iβ]= [iαcosθ+ iβsinθ;-iαsinθ+ iβcosθ], where the Parker transformation matrix T park In this context, θ represents the voltage phase angle output by the virtual synchronous machine.

[0024] Finally, since the calculation formula for the three-phase instantaneous active power acquisition model in the instantaneous power calculation strategy is P=(ud*id+uq*iq)*3 / 2, and the calculation formula for the three-phase instantaneous reactive power acquisition model is Q=(uq*id-ud*iq)*3 / 2, the two-phase voltages and two-phase currents after the second transformation are used as input parameters for the three-phase instantaneous active power acquisition model and the three-phase instantaneous reactive power acquisition model in the instantaneous power calculation strategy to obtain the current instantaneous active power P and the current instantaneous reactive power Q. The current instantaneous power is composed of the above two powers, and the obtained current instantaneous power is used to determine the current inertia and the current voltage phase in the subsequent process.

[0025] S120. Obtain the current electrical angular frequency corresponding to the current virtual synchronizer, and determine the current inertia according to the preset two-dimensional adaptive inertia determination strategy, the current electrical angular frequency and the corresponding current frequency change rate.

[0026] In this embodiment, the acquisition of the current instantaneous power previously incorporated the angular frequency of the virtual synchronous machine (i.e., the current electrical angular frequency ω corresponding to the current virtual synchronous machine, which is directly given by the active-frequency control loop without relying on the phase-locked loop to measure the grid phase). Now, it is also necessary to combine the current rate of change of frequency (ROCOF) and the calculation model corresponding to the two-dimensional adaptive inertia determination strategy to jointly determine the current inertia. The two-dimensional adaptive inertia determination strategy is used to determine the current inertia based on a first inertia adjustment value determined by the difference between the current electrical angular frequency and a preset rated instantaneous frequency, and a second inertia adjustment value determined by the current electrical angular frequency. The constraint condition corresponding to the two-dimensional adaptive inertia determination strategy is that the inertia value range is [1s, 20s].

[0027] In one embodiment, such as Figure 4 As shown, step S120 includes: S121. Obtain the absolute value of the difference between the current instantaneous frequency corresponding to the current electrical angular frequency of the current virtual synchronizer and the preset rated instantaneous frequency, and multiply it by the preset frequency deviation adjustment coefficient to obtain the first inertia adjustment value; wherein, the rated instantaneous frequency is obtained by dividing the rated angular frequency of the current virtual synchronizer by 2π. S122. Obtain the current frequency change rate and multiply it by a preset frequency change rate adjustment coefficient to obtain the second inertia adjustment value; S123. Add the preset reference inertia, the first inertia adjustment value, and the second inertia adjustment value to obtain the current initial inertia; S124. If it is determined that the current initial inertia satisfies the constraint conditions corresponding to the two-dimensional adaptive inertia determination strategy, then the current initial inertia is used as the current inertia.

[0028] In this embodiment, the current instantaneous frequency is denoted as f (f=ω / 2π), the rated instantaneous frequency is denoted as f0 (f0=ω0 / 2π, ω0 is the rated angular frequency of the current virtual synchronizer, and ω0=314.16rad / s), and the frequency deviation adjustment coefficient is denoted as k. j1 The frequency change rate adjustment coefficient is denoted as k. j2 Let the reference inertia be J0, the current rate of change of frequency be |f-f0|, and the current inertia be represented by J. Then the calculation formula for the two-dimensional adaptive inertia determination strategy is J = J0 + k j1 *|f-f0|+ k j2 *|df / dt|. The constraint condition corresponding to the dual-dimensional adaptive inertia determination strategy is generally set to 1s≤J≤20s. When the determined current initial inertia is within the value range corresponding to the above constraint condition, it means that the current initial inertia meets the constraint condition corresponding to the dual-dimensional adaptive inertia determination strategy, and the current initial inertia can be directly used as the current inertia. Through the above method, the adaptive adjustment of the rotor inertia of the simulated virtual synchronous machine is realized. When the frequency changes rapidly, the inertia is automatically increased to provide strong support, and when the frequency is steady, the inertia is automatically decreased to improve the response speed, thus achieving a balance between dynamic performance and steady-state performance.

[0029] It should be noted that after step S123, if it is determined that the current initial inertia does not meet the constraint conditions corresponding to the two-dimensional adaptive inertia determination strategy, then it is also necessary to determine the specific situation of the current initial inertia. Specifically, if the current initial inertia J is less than 1 second for 10 seconds, then a first prompt message (the system is in steady state, but in the state of minimum inertia) is generated and sent to the receiving end; if the current initial inertia J is greater than 20 seconds for 100 milliseconds, then a second prompt message (the inertia support capacity has reached its upper limit) is generated and sent to the receiving end; if the current initial inertia J is greater than 20 seconds for 1 second, then emergency control is triggered to coordinate other energy storage units to provide additional inertia support.

[0030] S130. Determine the current voltage phase based on the current instantaneous power, the current inertia, and the preset power direct self-synchronization mechanism.

[0031] In this embodiment, based on the power angle characteristics of the virtual synchronizer, the power deviation directly corresponds to the phase deviation. After determining the current instantaneous power and the current inertia, it is not necessary to use a phase-locked loop. Instead, the current voltage phase can be determined directly by combining the power direct self-synchronization mechanism, thereby fundamentally eliminating the risk of weak grid instability caused by the phase-locked loop.

[0032] In one embodiment, such as Figure 5 As shown, step S130 includes: S131. Obtain the current instantaneous active power, the preset active power command value corresponding to the network control command, the historical cumulative active power difference, the preset power loop proportional adjustment coefficient, the preset power loop integral adjustment coefficient, the preset rated angular frequency, the historical voltage phase of the previous control cycle, and the preset control cycle from the current instantaneous power. S132. Obtain the first power difference between the active power command value and the current instantaneous active power and multiply it by the power loop proportional adjustment coefficient to obtain the first adjustment value. Then, obtain the sum of the historical cumulative active power difference and the first power difference and multiply it by the power loop integral adjustment coefficient to obtain the second adjustment value. S133. The sum of the rated angular frequency, the first adjustment value, and the second adjustment value is multiplied by the preset control period and then added to the historical voltage phase of the previous control period to obtain the current voltage phase.

[0033] In this embodiment, if the current instantaneous active power is denoted as P, and the preset active power command value corresponding to the grid control command is denoted as P... ref The historical cumulative active power difference is denoted as P. his Let the power loop proportional control coefficient be denoted as kp, the power loop integral control coefficient as ki, the rated angular frequency as ω0, the historical voltage phase of the previous control cycle as θ(k-1), and the preset control cycle as Ts, with the current voltage phase represented by θ(k). The calculation formula for the direct power self-synchronization mechanism is as follows: θ(k) = θ(k-1) + [ω0 + kp × (P) ref -P)+ki×(P) ref -P+P his )]×Ts; That is, the current voltage phase can be obtained through the detailed calculation process in steps S132 to S133, realizing direct control of the output voltage phase based on the instantaneous active power deviation, and completely eliminating the phase-locked loop.

[0034] S140. The current three-phase voltage and the current three-phase current are transformed by Parker transformation based on the current voltage phase angle in the current voltage phase and then input to the preset current-free inner loop direct voltage control model to obtain the current adjusted two-phase voltage.

[0035] In this embodiment, since the current three-phase voltage and current are neither in the two-phase stationary coordinate system in the αβ domain nor in the dq domain, they need to be projected to the αβ domain by Clarke transform and Park transform in sequence before being input into the current-free inner loop direct voltage control model to obtain the current regulating voltage phase. The current-free inner loop direct voltage control model eliminates the current inner loop in the traditional dual-loop control, and the voltage control loop output is directly used as the modulation wave command to eliminate the delay and bandwidth limitation of the current inner loop and improve the dynamic response speed of the system.

[0036] In one embodiment, such as Figure 6 As shown, step S140 includes: S141. Obtain the current voltage phase angle in the current voltage phase and determine the current Parker transformation matrix based on the current voltage phase angle; S142. Transform the current three-phase voltage and the current three-phase current sequentially through the Clark transformation matrix and the current Park transformation matrix to obtain the third-transformed two-phase voltage and the third-transformed two-phase current in the dq domain. S143. The two-phase voltage after the third transformation and the obtained current voltage reference value are used as input parameters of the current-free inner loop direct voltage control model to obtain the current adjusted two-phase voltage; wherein, the current-free inner loop direct voltage control model includes a voltage reference feedforward control term, a voltage proportional-integral adjustment control term, a virtual impedance voltage drop control term, a dq axis decoupling control term, and a load current feedforward control term.

[0037] In this embodiment, if the current three-phase voltages are still denoted as ua, ub, and uc (i.e., the current corrected three-phase voltages are considered as the current three-phase voltages by default) and the current three-phase currents are denoted as ia, ib, and ic (i.e., the current corrected three-phase currents are considered as the current three-phase currents by default), the result is obtained by performing a Clark transformation using the Clark transformation matrix: [uα;uβ]=(2 / 3)*T Clarke *[ua;ub;uc],[iα;iβ]=(2 / 3)*T Clarke *[ia; ib; ic] (where [uα;uβ] still represents the projection of the current three-phase voltage in the two-phase stationary coordinate system of the αβ domain, and [iα;iβ] still represents the projection of the current three-phase current in the two-phase stationary coordinate system of the αβ domain); then, by performing a Parker transformation based on the current Parker transformation matrix determined by the current voltage phase angle, the two-phase voltage [ud';uq']=T after the third transformation in the dq domain is obtained. park '*[uα;uβ]=[uαcosθv+ uβsinθv;-uαsinθv+ uβcosθv], and the two-phase currents after the third transformation [id';iq']=T park'*[iα;iβ]= [iαcosθv+ iβsinθv;-iαsinθv+ iβcosθv], where in the current Parker transformation matrix T park In ', θv is the current voltage phase angle in the current voltage phase.

[0038] Knowing the two-phase voltages after the third transformation, these voltages are combined with the obtained current voltage reference value as input parameters to the current-free inner-loop direct voltage control model to obtain the current adjusted two-phase voltages [ud*;uq*]. The current-free inner-loop direct voltage control model is as follows: ud*=ud ref +(Kpu+Kiu / s)(ud ref -ud')-Rv*id'+ωLf•iq'+(1 / (sCf)) *id'; uq*=uq ref +(Kpu+Kiu / s)(uq ref -uq')-Rv*iq'-ωLf*id'+(1 / (sCf)) *iq'; In the two equations above, ud ref and UQ ref The current voltage reference values ​​are the values ​​of the d-axis and q-axis in the dq domain, respectively, and together they form the voltage reference feedforward control term; (Kpu+Kiu / s)(ud) ref -ud') and (Kpu+Kiu / s)( uq ref -uq') forms the voltage proportional-integral regulation control term, Kpu represents the voltage loop proportional coefficient (its value range is 0.5~2, the preferred value is 0.8), Kiu represents the voltage loop integral coefficient (its value range is 50~250, the preferred value is 200), and s represents the Laplace complex frequency operator; Rv*id' and Rv*iq' form the virtual impedance voltage drop control term, and Rv represents the equivalent series resistance of the current virtual synchronous machine; ωLf•iq' and -ωLf*id' form the dq axis decoupling control term, ω is the current electrical angular frequency corresponding to the current virtual synchronous machine, Lf=0.5mH and is the inverter-side filter inductance of the current virtual synchronous machine; (1 / (sCf)) *id' and (1 / (sCf)) *iq' form the load current feedforward control term, Cf=120μF and is the machine-side filter capacitor of the current virtual synchronous machine. By using the above-mentioned direct voltage control model without a current inner loop, the current inner loop in the traditional dual-loop control is eliminated, thus removing the delay and bandwidth limitations of the current inner loop, significantly improving the dynamic response speed, and the output of the voltage control loop can be directly used as a modulation wave command.

[0039] S150. Perform inverse Parker transform and inverse Clarke transform on the current adjusted two-phase voltage in sequence to obtain the current three-phase modulated wave.

[0040] In this embodiment, the obtained current adjusted two-phase voltage [ud*;uq*] cannot be directly converted into a three-phase modulated wave. It is necessary to perform inverse Parker transform and inverse Clarke transform on it in sequence to obtain the current three-phase modulated wave.

[0041] In one embodiment, such as Figure 7 As shown, step S150 includes: S151. Perform an inverse Parker transformation on the current adjusted two-phase voltage using the inverse Parker transformation matrix to obtain the inverse transformed two-phase voltage. S152. Perform an inverse Clarke transformation on the inverse-transformed two-phase voltage using an inverse Clarke transformation matrix to obtain the current three-phase modulated wave.

[0042] In this embodiment, during the above transformation process, the inverse Parker transformation matrix is ​​the Parker transformation matrix T. park The inverse matrix, and the inverse Clarke transformation matrix is ​​the Clarke transformation matrix T. Clarke The inverse matrix of the equation is used to convert the current adjusted two-phase voltage into the current three-phase modulated wave through these two transformations.

[0043] S160. The current three-phase modulation wave is generated into a current pulse width modulation signal through space vector pulse width modulation, and then input to the connected insulated gate bipolar transistor driver board; wherein, the insulated gate bipolar transistor driver board is located in the energy storage converter connected to the energy storage power station control system.

[0044] In this embodiment, in the transformation model corresponding to the current three-phase modulation wave, the input is the three-phase modulation wave command ua*, ub*, uc* corresponding to the current three-phase modulation wave, and the output is 6 channels (two-level) or 12 channels (three-level) PWM pulse signals (PWM stands for Pulse Width Modulation). By inputting the output current pulse width modulation signal to the connected insulated gate bipolar transistor driver board, grid control of the energy storage converter can be realized.

[0045] In one embodiment, such as Figure 8 As shown, step S160 includes: S161. Perform Clark transformation on the current three-phase modulated wave through the Clark transformation matrix to obtain the current target voltage vector; S162. Determine the target sector number of the current target voltage vector in the preset six sectors; S163. Obtain the first target base vector duration and the second target base vector duration corresponding to the target sector number from the preset sector and base vector duration mapping relationship, and subtract the first target base vector duration and the second target base vector duration from the switching cycle of the insulated gate bipolar transistor to obtain the current zero vector action time. S164. According to the preset seven-segment allocation method, the current zero vector action time, the duration of the first target base vector and the duration of the second target base vector, the zero vector, the first base vector and the second base vector are allocated in seven segments to obtain the current switch sequence. S165. Generate the current pulse width modulation signal according to the current switch sequence.

[0046] In this embodiment, after sequentially performing Clark transformation, obtaining the target sector number of the current target voltage vector, obtaining the current zero vector action time, seven-segment allocation, and generating the current pulse width modulation signal from the current three-phase modulated wave, the signal can be input to the connected insulated gate bipolar transistor driver board to realize grid-based control of the energy storage converter.

[0047] It is evident that the implementation of this method can employ a direct power synchronization mechanism without a phase-locked loop (PLL), directly controlling the output voltage phase based on the instantaneous active power deviation. This eliminates the need for a PLL at the architectural level and also avoids the problems of PLL instability in weak grids and the inability to dynamically adjust the fixed virtual inertia.

[0048] Figure 9 This is a schematic block diagram of a phase-locked loop-free adaptive inertia network control system provided in an embodiment of the present invention. Figure 9 As shown, corresponding to the above-described phase-locked loop-free adaptive inertia netting control method, the present invention also provides a phase-locked loop-free adaptive inertia netting control system 100. This phase-locked loop-free adaptive inertia netting control system 100 includes units for executing the above-described phase-locked loop-free adaptive inertia netting control method. Please refer to... Figure 9 The phase-locked loop-free adaptive inertia network control system 100 includes: a sampling unit 110, an inertia acquisition unit 120, a voltage phase acquisition unit 130, an adjustable voltage phase acquisition unit 140, a three-phase modulation wave acquisition unit 150, and a modulation signal generation unit 160.

[0049] The sampling unit 110 is used to obtain the current three-phase voltage and current three-phase current in response to the grid control command generated according to the preset control cycle, and to obtain the corresponding current instantaneous power according to the preset instantaneous power calculation strategy.

[0050] In this embodiment, the power storage station control system 10 can be pre-set with a preset control cycle (e.g., 50 μs; however, the value can be flexibly adjusted according to actual needs and is not limited to the listed values). Whenever a grid connection control command is detected, the current three-phase voltage and current are collected from the power grid via the current sampling module and voltage sampling module connected to the power storage station control system. Then, after a series of transformations, data processing, and calculations using the instantaneous power calculation strategy, the current instantaneous power is obtained.

[0051] In one embodiment, the sampling unit 110 is specifically used for: The current three-phase voltage and the current three-phase current are both subjected to first-order recursive low-pass filtering and sampling correction processing to obtain the current corrected three-phase voltage and the current corrected three-phase current; The Clarke transformation matrix in the instantaneous power calculation strategy is obtained, and the current corrected three-phase voltage and the current corrected three-phase current are both transformed by the Clarke transformation matrix to obtain the first transformed two-phase voltage and the first transformed two-phase current in the αβ domain; wherein, the αβ domain corresponds to the two-phase stationary coordinate system; Obtain the Parker transformation matrix in the instantaneous power calculation strategy, and transform the first transformed two-phase voltage and the first transformed two-phase current through the Parker transformation matrix to obtain the second transformed two-phase voltage and the second transformed two-phase current in the dq domain; wherein, the dq domain corresponds to an orthogonal rotating coordinate system that rotates synchronously with the grid voltage vector; The two-phase voltage and the two-phase current after the second transformation are both used as input parameters for the three-phase instantaneous active power acquisition model and the three-phase instantaneous reactive power acquisition model in the instantaneous power calculation strategy to obtain the current instantaneous active power and the current instantaneous reactive power, and then form the current instantaneous power.

[0052] In this embodiment, when the current three-phase voltage and the current three-phase current are both processed by first-order recursive low-pass filtering and sampling correction, specifically, first-order recursive low-pass filtering (i.e., first-order IIR low-pass filtering, IIR stands for Infinite Impulse Response) is used to filter out high-frequency noise in the current three-phase voltage and the current three-phase current. Then, DC bias correction (subtracting the sampling zero-point drift from both the filtered output three-phase voltage and three-phase current, for example, subtracting the sampling zero-point drift voltage from the filtered output three-phase voltage and subtracting the sampling zero-point drift current from the filtered output three-phase current) or phase synchronization correction (introducing phase offset into both the filtered output three-phase voltage and three-phase current for correction, so as to synchronize with the ADC sampling time of the three-phase voltage and the ADC sampling time of the three-phase current) is performed to obtain the current corrected three-phase voltage and the current corrected three-phase current.

[0053] Then, the current corrected three-phase voltages and currents are transformed using the Clarke transform matrix, converting the voltages or currents of phases a, b, and c into α and β components in a two-phase stationary coordinate system in the αβ domain. If the three-phase voltages in the current corrected three-phase voltages are denoted as ua, ub, and uc, and the current corrected three-phase currents as ia, ib, and ic, respectively, the Clarke transform matrix is ​​represented by T. Clarke Let the two-phase voltages after the first transformation be denoted as [uα;uβ], and the two-phase currents after the first transformation be denoted as [iα;iβ], then [uα;uβ] = (2 / 3) * T Clarke *[ua;ub;uc],[iα;iβ]=(2 / 3)*T Clarke *[ia; ib; ic].

[0054] Subsequently, both the first transformed two-phase voltages and the first transformed two-phase currents are transformed using the Parker transformation matrix. This transforms the α and β components in the two-phase stationary coordinate system of the αβ domain into a dq coordinate system rotating at an angular frequency using a virtual synchronizing machine. The dq domain in the dq coordinate system corresponds to an orthogonal rotating coordinate system that rotates synchronously with the grid voltage vector. If the second transformed two-phase voltages are denoted as ud and uq, and the second transformed two-phase currents as id and iq, the Parker transformation matrix is ​​represented by T. park This means that [ud;uq]=T park *[uα;uβ]=[uαcosθ+ uβsinθ;-uαsinθ+ uβcosθ], [id;iq]=T park *[iα;iβ]= [iαcosθ+ iβsinθ;-iαsinθ+ iβcosθ], where the Parker transformation matrix T park In this context, θ represents the voltage phase angle output by the virtual synchronous machine.

[0055] Finally, since the calculation formula for the three-phase instantaneous active power acquisition model in the instantaneous power calculation strategy is P=(ud*id+uq*iq)*3 / 2, and the calculation formula for the three-phase instantaneous reactive power acquisition model is Q=(uq*id-ud*iq)*3 / 2, the two-phase voltages and two-phase currents after the second transformation are used as input parameters for the three-phase instantaneous active power acquisition model and the three-phase instantaneous reactive power acquisition model in the instantaneous power calculation strategy to obtain the current instantaneous active power P and the current instantaneous reactive power Q. The current instantaneous power is composed of the above two powers, and the obtained current instantaneous power is used to determine the current inertia and the current voltage phase in the subsequent process.

[0056] The inertia acquisition unit 120 is used to acquire the current electrical angular frequency corresponding to the current virtual synchronizer, and determine the current inertia according to the preset two-dimensional adaptive inertia determination strategy, the current electrical angular frequency and the corresponding current frequency change rate.

[0057] In this embodiment, the acquisition of the current instantaneous power previously incorporated the angular frequency of the virtual synchronous machine (i.e., the current electrical angular frequency ω corresponding to the current virtual synchronous machine, which is directly given by the active-frequency control loop without relying on the phase-locked loop to measure the grid phase). Now, it is also necessary to combine the current rate of change of frequency (ROCOF) and the calculation model corresponding to the two-dimensional adaptive inertia determination strategy to jointly determine the current inertia. The two-dimensional adaptive inertia determination strategy is used to determine the current inertia based on a first inertia adjustment value determined by the difference between the current electrical angular frequency and a preset rated instantaneous frequency, and a second inertia adjustment value determined by the current electrical angular frequency. The constraint condition corresponding to the two-dimensional adaptive inertia determination strategy is that the inertia value range is [1s, 20s].

[0058] In one embodiment, the inertia acquisition unit 120 is specifically used for: The absolute value of the difference between the current instantaneous frequency corresponding to the current electrical angular frequency of the current virtual synchronizer and the preset rated instantaneous frequency is obtained and multiplied by the preset frequency deviation adjustment coefficient to obtain the first inertia adjustment value; wherein, the rated instantaneous frequency is obtained by dividing the rated angular frequency of the current virtual synchronizer by 2π. The current frequency change rate is obtained and multiplied by a preset frequency change rate adjustment coefficient to obtain the second inertia adjustment value; The current initial inertia is obtained by adding the preset reference inertia, the first inertia adjustment value, and the second inertia adjustment value. If it is determined that the current initial inertia satisfies the constraint conditions corresponding to the two-dimensional adaptive inertia determination strategy, then the current initial inertia is used as the current inertia.

[0059] In this embodiment, the current instantaneous frequency is denoted as f (f=ω / 2π), the rated instantaneous frequency is denoted as f0 (f0=ω0 / 2π, ω0 is the rated angular frequency of the current virtual synchronizer, and ω0=314.16rad / s), and the frequency deviation adjustment coefficient is denoted as k. j1 The frequency change rate adjustment coefficient is denoted as k. j2 Let the reference inertia be J0, the current rate of change of frequency be |f-f0|, and the current inertia be represented by J. Then the calculation formula for the two-dimensional adaptive inertia determination strategy is J = J0 + k j1 *|f-f0|+ k j2*|df / dt|. The constraint condition corresponding to the dual-dimensional adaptive inertia determination strategy is generally set to 1s≤J≤20s. When the determined current initial inertia is within the value range corresponding to the above constraint condition, it means that the current initial inertia meets the constraint condition corresponding to the dual-dimensional adaptive inertia determination strategy, and the current initial inertia can be directly used as the current inertia. Through the above method, the adaptive adjustment of the rotor inertia of the simulated virtual synchronous machine is realized. When the frequency changes rapidly, the inertia is automatically increased to provide strong support, and when the frequency is steady, the inertia is automatically decreased to improve the response speed, thus achieving a balance between dynamic performance and steady-state performance.

[0060] It is important to note that if the current initial inertia does not meet the constraints of the dual-dimensional adaptive inertia determination strategy, the specific situation of the current initial inertia needs to be determined. Specifically, if the current initial inertia J is less than 1 second for 10 seconds, a first prompt message (the system is in steady state, but in the state of minimum inertia) is generated and sent to the receiving end; if the current initial inertia J is greater than 20 seconds for 100 milliseconds, a second prompt message (the inertia support capacity has reached its limit) is generated and sent to the receiving end; if the current initial inertia J is greater than 20 seconds for 1 second, an emergency control is triggered to coordinate other energy storage units to provide additional inertia support.

[0061] The voltage phase acquisition unit 130 is used to determine the current voltage phase based on the current instantaneous power, the current inertia, and a preset power direct self-synchronization mechanism.

[0062] In this embodiment, based on the power angle characteristics of the virtual synchronizer, the power deviation directly corresponds to the phase deviation. After determining the current instantaneous power and the current inertia, it is not necessary to use a phase-locked loop. Instead, the current voltage phase can be determined directly by combining the power direct self-synchronization mechanism, thereby fundamentally eliminating the risk of weak grid instability caused by the phase-locked loop.

[0063] In one embodiment, the voltage phase acquisition unit 130 is specifically used for: The system acquires the current instantaneous active power, the preset active power command value corresponding to the network control command, the historical cumulative active power difference, the preset power loop proportional adjustment coefficient, the preset power loop integral adjustment coefficient, the preset rated angular frequency, the historical voltage phase of the previous control cycle, and the preset control cycle. The first adjustment value is obtained by obtaining the first power difference between the active power command value and the current instantaneous active power and multiplying it by the power loop proportional adjustment coefficient; and the second adjustment value is obtained by multiplying the sum of the historical cumulative active power difference and the first power difference by the power loop integral adjustment coefficient. The current voltage phase is obtained by multiplying the sum of the rated angular frequency, the first adjustment value, and the second adjustment value by the preset control period and then adding it to the historical voltage phase of the previous control period.

[0064] In this embodiment, if the current instantaneous active power is denoted as P, and the preset active power command value corresponding to the grid control command is denoted as P... ref The historical cumulative active power difference is denoted as P. his Let the power loop proportional control coefficient be denoted as kp, the power loop integral control coefficient as ki, the rated angular frequency as ω0, the historical voltage phase of the previous control cycle as θ(k-1), and the preset control cycle as Ts, with the current voltage phase represented by θ(k). The calculation formula for the direct power self-synchronization mechanism is as follows: θ(k) = θ(k-1) + [ω0 + kp × (P) ref -P)+ki×(P) ref -P+P his )]×Ts; That is, the current voltage phase can be obtained through the detailed calculation process in steps S132 to S133, realizing direct control of the output voltage phase based on the instantaneous active power deviation, and completely eliminating the phase-locked loop.

[0065] The voltage phase acquisition unit 140 is used to input the current three-phase voltage and the current three-phase current into a preset current-free inner loop direct voltage control model after performing Parker transformation based on the current voltage phase angle in the current voltage phase, so as to obtain the current adjusted two-phase voltage.

[0066] In this embodiment, since the current three-phase voltage and current are neither in the two-phase stationary coordinate system in the αβ domain nor in the dq domain, they need to be projected to the αβ domain by Clarke transform and Park transform in sequence before being input into the current-free inner loop direct voltage control model to obtain the current regulating voltage phase. The current-free inner loop direct voltage control model eliminates the current inner loop in the traditional dual-loop control, and the voltage control loop output is directly used as the modulation wave command to eliminate the delay and bandwidth limitation of the current inner loop and improve the dynamic response speed of the system.

[0067] In one embodiment, the voltage phase acquisition unit 140 is specifically used for: Obtain the current voltage phase angle in the current voltage phase and determine the current Parker transformation matrix based on the current voltage phase angle; The current three-phase voltages and currents are transformed sequentially through the Clarke transformation matrix and the current Parker transformation matrix to obtain the third-transformed two-phase voltages and third-transformed two-phase currents in the dq domain. The two-phase voltages after the third transformation and the obtained current voltage reference value are used as input parameters of the current-free inner loop direct voltage control model to obtain the current adjusted two-phase voltages; wherein, the current-free inner loop direct voltage control model includes a voltage reference feedforward control term, a voltage proportional-integral adjustment control term, a virtual impedance voltage drop control term, a dq axis decoupling control term, and a load current feedforward control term.

[0068] In this embodiment, if the current three-phase voltages are still denoted as ua, ub, and uc (i.e., the current corrected three-phase voltages are considered as the current three-phase voltages by default) and the current three-phase currents are denoted as ia, ib, and ic (i.e., the current corrected three-phase currents are considered as the current three-phase currents by default), the result is obtained by performing a Clark transformation using the Clark transformation matrix: [uα;uβ]=(2 / 3)*T Clarke *[ua;ub;uc],[iα;iβ]=(2 / 3)*T Clarke *[ia; ib; ic] (where [uα;uβ] still represents the projection of the current three-phase voltage in the two-phase stationary coordinate system of the αβ domain, and [iα;iβ] still represents the projection of the current three-phase current in the two-phase stationary coordinate system of the αβ domain); then, by performing a Parker transformation based on the current Parker transformation matrix determined by the current voltage phase angle, the two-phase voltage [ud';uq']=T after the third transformation in the dq domain is obtained. park '*[uα;uβ]=[uαcosθv+ uβsinθv;-uαsinθv+ uβcosθv], and the two-phase currents after the third transformation [id';iq']=T park '*[iα;iβ]= [iαcosθv+ iβsinθv;-iαsinθv+ iβcosθv], where in the current Parker transformation matrix T park In ', θv is the current voltage phase angle in the current voltage phase.

[0069] Knowing the two-phase voltages after the third transformation, these voltages are combined with the obtained current voltage reference value as input parameters to the current-free inner-loop direct voltage control model to obtain the current adjusted two-phase voltages [ud*;uq*]. The current-free inner-loop direct voltage control model is as follows: ud*=ud ref +(Kpu+Kiu / s)(ud ref -ud')-Rv*id'+ωLf•iq'+(1 / (sCf)) *id'; uq*=uq ref +(Kpu+Kiu / s)(uq ref -uq')-Rv*iq'-ωLf*id'+(1 / (sCf)) *iq'; In the two equations above, ud ref and UQ ref The current voltage reference values ​​are the values ​​of the d-axis and q-axis in the dq domain, respectively, and together they form the voltage reference feedforward control term; (Kpu+Kiu / s)(ud) ref -ud') and (Kpu+Kiu / s)( uq ref -uq') forms the voltage proportional-integral regulation control term, Kpu represents the voltage loop proportional coefficient (its value range is 0.5~2, the preferred value is 0.8), Kiu represents the voltage loop integral coefficient (its value range is 50~250, the preferred value is 200), and s represents the Laplace complex frequency operator; Rv*id' and Rv*iq' form the virtual impedance voltage drop control term, and Rv represents the equivalent series resistance of the current virtual synchronous machine; ωLf•iq' and -ωLf*id' form the dq axis decoupling control term, ω is the current electrical angular frequency corresponding to the current virtual synchronous machine, Lf=0.5mH and is the inverter-side filter inductance of the current virtual synchronous machine; (1 / (sCf)) *id' and (1 / (sCf))*iq' form the load current feedforward control term, Cf=120μF and is the machine-side filter capacitor of the current virtual synchronous machine. By using the above-mentioned direct voltage control model without a current inner loop, the current inner loop in the traditional dual-loop control is eliminated, thus removing the delay and bandwidth limitations of the current inner loop, significantly improving the dynamic response speed, and the output of the voltage control loop can be directly used as a modulation wave command.

[0070] The three-phase modulation wave acquisition unit 150 is used to sequentially perform inverse Parker transform and inverse Clarke transform on the current adjusted two-phase voltage to obtain the current three-phase modulation wave.

[0071] In this embodiment, the obtained current adjusted two-phase voltage [ud*;uq*] cannot be directly converted into a three-phase modulated wave. It is necessary to perform inverse Parker transform and inverse Clarke transform on it in sequence to obtain the current three-phase modulated wave.

[0072] In one embodiment, the three-phase modulation wave acquisition unit 150 is specifically used for: The current adjusted two-phase voltages are subjected to an inverse Parker transformation using an inverse Parker transformation matrix to obtain the inverse-transformed two-phase voltages. The inverse-transformed two-phase voltages are subjected to an inverse Clarke transform using an inverse Clarke transform matrix to obtain the current three-phase modulated wave.

[0073] In this embodiment, during the above transformation process, the inverse Parker transformation matrix is ​​the Parker transformation matrix T. park The inverse matrix, and the inverse Clarke transformation matrix is ​​the Clarke transformation matrix T. Clarke The inverse matrix of the equation is used to convert the current adjusted two-phase voltage into the current three-phase modulated wave through these two transformations.

[0074] The modulation signal generation unit 160 is used to generate a current pulse width modulation signal from the current three-phase modulation wave through space vector pulse width modulation, and input it to the connected insulated gate bipolar transistor driver board; wherein the insulated gate bipolar transistor driver board is located in the energy storage converter connected to the energy storage power station control system.

[0075] In this embodiment, in the transformation model corresponding to the current three-phase modulation wave, the input is the three-phase modulation wave command ua*, ub*, uc* corresponding to the current three-phase modulation wave, and the output is 6 channels (two-level) or 12 channels (three-level) PWM pulse signals (PWM stands for Pulse Width Modulation). By inputting the output current pulse width modulation signal to the connected insulated gate bipolar transistor driver board, grid control of the energy storage converter can be realized.

[0076] In one embodiment, the modulation signal generation unit 160 is specifically used for: The current three-phase modulated wave is subjected to Clark transformation through the Clark transformation matrix to obtain the current target voltage vector; Determine the target sector number of the current target voltage vector within the preset six sectors; The first target base vector duration and the second target base vector duration corresponding to the target sector number are obtained from the preset sector and base vector duration mapping relationship. The first target base vector duration and the second target base vector duration are subtracted from the switching cycle of the insulated gate bipolar transistor to obtain the current zero vector action time. Based on the preset seven-segment allocation method, the current zero vector's action time, the duration of the first target base vector, and the duration of the second target base vector, the zero vector, the first base vector, and the second base vector are allocated in seven segments to obtain the current switching sequence; The current pulse width modulation signal is generated according to the current switch sequence.

[0077] In this embodiment, after sequentially performing Clark transformation, obtaining the target sector number of the current target voltage vector, obtaining the current zero vector action time, seven-segment allocation, and generating the current pulse width modulation signal from the current three-phase modulated wave, the signal can be input to the connected insulated gate bipolar transistor driver board to realize grid-based control of the energy storage converter.

[0078] It is evident that the implementation of this system can adopt a direct power synchronization mechanism without a phase-locked loop (PLL), directly controlling the output voltage phase based on the instantaneous active power deviation. This eliminates the need for a PLL at the architectural level and also avoids the problems of PLL instability in weak grids and the inability to dynamically adjust the fixed virtual inertia.

[0079] The aforementioned phase-locked loop-free adaptive inertia network control system can be implemented as a computer program, which can be used in, for example... Figure 10 It runs on the computer device shown.

[0080] Please see Figure 10 , Figure 10 This is a schematic block diagram of a computer device provided in an embodiment of the present invention. The computer device integrates any of the phase-locked loop-free adaptive inertia network control systems provided in this embodiment of the present invention.

[0081] See Figure 10 The computer device 400 includes a processor 402, a memory, and a network interface 405 connected via a system bus 401. The memory may include a storage medium 403 and internal memory 404.

[0082] The storage medium 403 may store an operating system 4031 and a computer program 4032. The computer program 4032 includes program instructions that, when executed, cause the processor 402 to perform a phase-locked loop-free adaptive inertia network control method.

[0083] The processor 402 provides computing and control capabilities to support the operation of the entire computer device.

[0084] The internal memory 404 provides an environment for the computer program 4032 in the storage medium 403 to run. When the computer program 4032 is executed by the processor 402, the processor 402 can execute the above-mentioned phase-locked loop-free adaptive inertia network control method.

[0085] This network interface 405 is used for network communication with other devices. Those skilled in the art will understand that... Figure 10 The structure shown is merely a block diagram of a portion of the structure related to the present invention and does not constitute a limitation on the computer device to which the present invention is applied. A specific computer device may include more or fewer components than those shown in the figure, or combine certain components, or have different component arrangements.

[0086] The processor 402 is used to run the computer program 4032 stored in the memory to implement the phase-locked loop-free adaptive inertia network control method as described above.

[0087] It should be understood that, in this embodiment of the invention, the processor 402 may be a Central Processing Unit (CPU), or it may be other general-purpose processors, digital signal processors (DSPs), application-specific integrated circuits (ASICs), field-programmable gate arrays (FPGAs), or other programmable logic devices, discrete gate or transistor logic devices, discrete hardware components, etc. The general-purpose processor may be a microprocessor or any conventional processor.

[0088] It will be understood by those skilled in the art that all or part of the processes in the methods of the above embodiments can be implemented by a computer program instructing related hardware. The computer program includes program instructions and can be stored in a storage medium, which is a computer-readable storage medium. The program instructions are executed by at least one processor in the computer system to implement the process steps of the embodiments of the above methods.

[0089] Therefore, the present invention also provides a computer-readable storage medium. This computer-readable storage medium stores a computer program, wherein the computer program includes program instructions. When executed by a processor, the program instructions cause the processor to perform the phase-locked loop-free adaptive inertia netting control method as described above.

[0090] The storage medium can be any computer-readable storage medium that can store program code, such as a USB flash drive, external hard drive, read-only memory (ROM), magnetic disk, or optical disk.

[0091] Those skilled in the art will recognize that the units and algorithm steps of the various examples described in conjunction with the embodiments disclosed herein can be implemented in electronic hardware, computer software, or a combination of both. To clearly illustrate the interchangeability of hardware and software, the components and steps of the various examples have been generally described in terms of functionality in the foregoing description. Whether these functions are implemented in hardware or software depends on the specific application and design constraints of the technical solution. Those skilled in the art can use different methods to implement the described functions for each specific application, but such implementations should not be considered beyond the scope of this invention.

[0092] In the several embodiments provided by this invention, it should be understood that the disclosed apparatus and methods can be implemented in other ways. For example, the apparatus embodiments described above are merely illustrative. For example, the division of each unit is merely a logical functional division, and there may be other division methods in actual implementation. For example, multiple units or components may be combined or integrated into another system, or some features may be ignored or not executed.

[0093] The steps in the method of this invention can be adjusted, merged, or reduced in order according to actual needs. The units in the device of this invention can be merged, divided, or reduced according to actual needs. Furthermore, the functional units in the various embodiments of this invention can be integrated into one processing unit, or each unit can exist physically separately, or two or more units can be integrated into one unit.

[0094] If the integrated unit is implemented as a software functional unit and sold or used as an independent product, it can be stored in a storage medium. Based on this understanding, the technical solution of the present invention, in essence, or the part that contributes to the prior art, or all or part of the technical solution, can be embodied in the form of a software product. This computer software product is stored in a storage medium and includes several instructions to cause a computer device (which may be a personal computer, a terminal, or a network device, etc.) to execute all or part of the steps of the methods described in the various embodiments of the present invention.

[0095] The above description is merely a specific embodiment of the present invention, but the scope of protection of the present invention is not limited thereto. Any person skilled in the art can easily conceive of various equivalent modifications or substitutions within the technical scope disclosed in the present invention, and these modifications or substitutions should all be covered within the scope of protection of the invention. Therefore, the scope of protection of the present invention should be determined by the scope of the claims.

Claims

1. A phase-locked loop-free adaptive inertia network control method, characterized in that, include: In response to the grid control command generated according to the preset control cycle, the current three-phase voltage and current three-phase current are obtained, and the corresponding current instantaneous power is obtained according to the preset instantaneous power calculation strategy; The current electrical angular frequency corresponding to the current virtual synchronizer is obtained, and the current inertia is determined according to the preset two-dimensional adaptive inertia determination strategy, the current electrical angular frequency and the corresponding current frequency change rate; the two-dimensional adaptive inertia determination strategy is used to determine the current inertia based on the first inertia adjustment value determined by the difference between the current electrical angular frequency and the preset rated instantaneous frequency and the second inertia adjustment value determined by the current electrical angular frequency, and the constraint condition corresponding to the two-dimensional adaptive inertia determination strategy is that the inertia value range is [1s, 20s]; The current voltage phase is determined based on the current instantaneous power, the current inertia, and the preset power direct self-synchronization mechanism. The current three-phase voltage and the current three-phase current are transformed by Parker transformation based on the current voltage phase angle in the current voltage phase and then input to a preset direct voltage control model without current inner loop to obtain the current adjusted two-phase voltage; the direct voltage control model without current inner loop includes a voltage control loop but does not include a current inner loop; The current two-phase voltages after adjustment are subjected to inverse Parker transform and inverse Clarke transform in sequence to obtain the current three-phase modulated wave; The current three-phase modulated wave is generated into a current pulse width modulation signal through space vector pulse width modulation, and then input to the connected insulated gate bipolar transistor driver board; wherein, the insulated gate bipolar transistor driver board is located in the energy storage converter connected to the energy storage power station control system.

2. The method according to claim 1, characterized in that, The step of obtaining the corresponding current instantaneous power according to the preset instantaneous power calculation strategy includes: The current three-phase voltage and the current three-phase current are both subjected to first-order recursive low-pass filtering and sampling correction processing to obtain the current corrected three-phase voltage and the current corrected three-phase current; The Clarke transformation matrix in the instantaneous power calculation strategy is obtained, and the current corrected three-phase voltage and the current corrected three-phase current are both transformed by the Clarke transformation matrix to obtain the first transformed two-phase voltage and the first transformed two-phase current in the αβ domain; wherein, the αβ domain corresponds to the two-phase stationary coordinate system; Obtain the Parker transformation matrix in the instantaneous power calculation strategy, and transform the first transformed two-phase voltage and the first transformed two-phase current through the Parker transformation matrix to obtain the second transformed two-phase voltage and the second transformed two-phase current in the dq domain; wherein, the dq domain corresponds to an orthogonal rotating coordinate system that rotates synchronously with the grid voltage vector; The two-phase voltage and the two-phase current after the second transformation are both used as input parameters for the three-phase instantaneous active power acquisition model and the three-phase instantaneous reactive power acquisition model in the instantaneous power calculation strategy to obtain the current instantaneous active power and the current instantaneous reactive power, and then form the current instantaneous power.

3. The method according to claim 1, characterized in that, The step of determining the current inertia based on a preset two-dimensional adaptive inertia determination strategy, the current electric angular frequency, and the corresponding rate of change of the current frequency includes: The absolute value of the difference between the current instantaneous frequency corresponding to the current electrical angular frequency of the current virtual synchronizer and the preset rated instantaneous frequency is obtained and multiplied by the preset frequency deviation adjustment coefficient to obtain the first inertia adjustment value; wherein, the rated instantaneous frequency is obtained by dividing the rated angular frequency of the current virtual synchronizer by 2π. The current frequency change rate is obtained and multiplied by a preset frequency change rate adjustment coefficient to obtain the second inertia adjustment value; The current initial inertia is obtained by adding the preset reference inertia, the first inertia adjustment value, and the second inertia adjustment value. If it is determined that the current initial inertia satisfies the constraint conditions corresponding to the two-dimensional adaptive inertia determination strategy, then the current initial inertia is used as the current inertia.

4. The method according to claim 1, characterized in that, The step of determining the current voltage phase based on the current instantaneous power, the current inertia, and a preset power direct self-synchronization mechanism includes: The system acquires the current instantaneous active power, the preset active power command value corresponding to the network control command, the historical cumulative active power difference, the preset power loop proportional adjustment coefficient, the preset power loop integral adjustment coefficient, the preset rated angular frequency, the historical voltage phase of the previous control cycle, and the preset control cycle. The first adjustment value is obtained by obtaining the first power difference between the active power command value and the current instantaneous active power and multiplying it by the power loop proportional adjustment coefficient; and the second adjustment value is obtained by multiplying the sum of the historical cumulative active power difference and the first power difference by the power loop integral adjustment coefficient. The current voltage phase is obtained by multiplying the sum of the rated angular frequency, the first adjustment value, and the second adjustment value by the preset control period and then adding it to the historical voltage phase of the previous control period.

5. The method according to claim 1, characterized in that, The process of inputting the current three-phase voltage and the current three-phase current into a preset current-free inner-loop direct voltage control model after undergoing Parker transformation based on the current voltage phase angle in the current voltage phase, to obtain the current adjusted two-phase voltage, includes: Obtain the current voltage phase angle in the current voltage phase and determine the current Parker transformation matrix based on the current voltage phase angle; The current three-phase voltage and the current three-phase current are transformed sequentially through the Clark transformation matrix and the current Park transformation matrix to obtain the third-transformed two-phase voltage and the third-transformed two-phase current in the dq domain; The two-phase voltages after the third transformation and the obtained current voltage reference value are used as input parameters of the current-free inner loop direct voltage control model to obtain the current adjusted two-phase voltages; wherein, the current-free inner loop direct voltage control model includes a voltage reference feedforward control term, a voltage proportional-integral adjustment control term, a virtual impedance voltage drop control term, a dq axis decoupling control term, and a load current feedforward control term.

6. The method according to claim 1, characterized in that, The step of sequentially performing inverse Parker transform and inverse Clarke transform on the currently adjusted two-phase voltages to obtain the current three-phase modulated wave includes: The current adjusted two-phase voltages are subjected to an inverse Parker transformation using an inverse Parker transformation matrix to obtain the inverse-transformed two-phase voltages. The inverse-transformed two-phase voltages are subjected to an inverse Clarke transform using an inverse Clarke transform matrix to obtain the current three-phase modulated wave.

7. The method according to claim 1, characterized in that, The step of generating a current pulse width modulation signal from the current three-phase modulated wave through space vector pulse width modulation includes: The current three-phase modulated wave is subjected to Clark transformation through the Clark transformation matrix to obtain the current target voltage vector; Determine the target sector number of the current target voltage vector within the preset six sectors; The first target base vector duration and the second target base vector duration corresponding to the target sector number are obtained from the preset sector and base vector duration mapping relationship. The first target base vector duration and the second target base vector duration are subtracted from the switching cycle of the insulated gate bipolar transistor to obtain the current zero vector action time. Based on the preset seven-segment allocation method, the current zero vector's action time, the duration of the first target base vector, and the duration of the second target base vector, the zero vector, the first base vector, and the second base vector are allocated in seven segments to obtain the current switching sequence; The current pulse width modulation signal is generated according to the current switch sequence.

8. A phase-locked loop-free adaptive inertia network control system, characterized in that, include: The sampling unit is used to respond to the grid control command generated according to the preset control cycle, obtain the current three-phase voltage and current three-phase current, and obtain the corresponding current instantaneous power according to the preset instantaneous power calculation strategy; An inertia acquisition unit is used to acquire the current electrical angular frequency corresponding to the current virtual synchronizer, and determine the current inertia according to a preset two-dimensional adaptive inertia determination strategy, the current electrical angular frequency and the corresponding current frequency change rate; the two-dimensional adaptive inertia determination strategy is used to determine the current inertia based on a first inertia adjustment value determined by the difference between the current electrical angular frequency and the preset rated instantaneous frequency and a second inertia adjustment value determined by the current electrical angular frequency, and the constraint condition corresponding to the two-dimensional adaptive inertia determination strategy is that the inertia value range is [1s, 20s]; The voltage phase acquisition unit is used to determine the current voltage phase based on the current instantaneous power, the current inertia, and a preset power direct self-synchronization mechanism. The voltage phase acquisition unit is used to input the current three-phase voltage and the current three-phase current into a preset current-free inner-loop direct voltage control model after performing Parker transformation based on the current voltage phase angle in the current voltage phase, so as to obtain the current adjusted two-phase voltage; the current-free inner-loop direct voltage control model includes a voltage control loop but does not include a current inner loop; The three-phase modulation wave acquisition unit is used to sequentially perform inverse Parker transform and inverse Clarke transform on the current adjusted two-phase voltage to obtain the current three-phase modulation wave. The modulation signal generation unit is used to generate a current pulse width modulation signal from the current three-phase modulation wave through space vector pulse width modulation, and input it to the connected insulated gate bipolar transistor driver board; wherein the insulated gate bipolar transistor driver board is located in the energy storage converter connected to the energy storage power station control system.

9. The phase-locked loop-free adaptive inertia network control system according to claim 8, characterized in that, The inertia acquisition unit is specifically used for: The absolute value of the difference between the current instantaneous frequency corresponding to the current electrical angular frequency of the current virtual synchronizer and the preset rated instantaneous frequency is obtained and multiplied by the preset frequency deviation adjustment coefficient to obtain the first inertia adjustment value; wherein, the rated instantaneous frequency is obtained by dividing the rated angular frequency of the current virtual synchronizer by 2π. The current frequency change rate is obtained and multiplied by a preset frequency change rate adjustment coefficient to obtain the second inertia adjustment value; The current initial inertia is obtained by adding the preset reference inertia, the first inertia adjustment value, and the second inertia adjustment value. If it is determined that the current initial inertia satisfies the constraint conditions corresponding to the two-dimensional adaptive inertia determination strategy, then the current initial inertia is used as the current inertia.

10. The phase-locked loop-free adaptive inertia network control system according to claim 8, characterized in that, The voltage phase acquisition unit is specifically used for: The system acquires the current instantaneous active power, the preset active power command value corresponding to the network control command, the historical cumulative active power difference, the preset power loop proportional adjustment coefficient, the preset power loop integral adjustment coefficient, the preset rated angular frequency, the historical voltage phase of the previous control cycle, and the preset control cycle. The first adjustment value is obtained by obtaining the first power difference between the active power command value and the current instantaneous active power and multiplying it by the power loop proportional adjustment coefficient; and the second adjustment value is obtained by multiplying the sum of the historical cumulative active power difference and the first power difference by the power loop integral adjustment coefficient. The current voltage phase is obtained by multiplying the sum of the rated angular frequency, the first adjustment value, and the second adjustment value by the preset control period and then adding it to the historical voltage phase of the previous control period.

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