Network configuration type control method and system for on-board charger grid-connected resonance suppression

Abstract

The application relates to a grid-connected control method and system for grid-connected resonance suppression of an on-board charger, and relates to the technical field of on-board chargers, and comprises the following steps: collecting PCC three-phase voltages and grid-side currents, calculating actual active and reactive powers after coordinate transformation and low-pass filtering; adopting a power outer ring containing frequency / voltage droop and virtual inertia to generate phase and amplitude references of internal potential; introducing virtual inductance and resistance for d and q axis cross-decoupling compensation to obtain a modified voltage instruction; constructing a high-pass filter type transient virtual resistance ring to extract resonance components to generate a compensation voltage, and generating a PWM signal after voltage and current double-loop and overcurrent limiting. The application solves the technical problems that the grid-connected control of the on-board charger is prone to instability under a weak power grid, and that LCL resonance suppression and power decoupling are difficult to be considered.

Description

Technical Field

[0001] This application relates to the field of on-board charger technology, and in particular to a grid-type control method and system for suppressing grid-connected resonance in on-board chargers. Background Technology

[0002] In vehicle-to-grid (V2G) technology applications, on-board bidirectional chargers serve as the energy exchange interface between electric vehicles and low-voltage power distribution networks, requiring both grid-connected power generation and charging operation modes. On-board chargers typically employ LCL filters to filter out high-order harmonics generated by pulse-width modulation (PWM) of the switching frequency. However, LCL filters inherently possess resonant peaks, which can easily induce high-frequency oscillations when grid impedance changes or load fluctuates, leading to grid current distortion or even system instability.

[0003] In existing technologies, on-board chargers mainly employ grid-following control, using a phase-locked loop (PLL) to track the phase of the point of common coupling (PCC) voltage in real time and output current commands to control the inverter to follow the grid operation. To suppress the resonance peak of the LCL filter, physical damping resistors are typically connected in series or parallel on its filter branch, or a notch filter with fixed parameters is used to attenuate the signal at the resonant frequency. However, in weak grid conditions, due to the large and frequent fluctuations in the equivalent grid impedance, grid-following control makes it difficult for the PLL to stably track the voltage phase, easily leading to oscillations and instability in the control system. Physical damping resistors increase additional power losses and reduce system conversion efficiency; when the resonant frequency drifts due to changes in grid impedance, the center frequency of the notch filter with fixed parameters cannot follow the shift, significantly reducing the resonance suppression effect, and may even cause high-frequency oscillations or grid current distortion due to deterioration of phase characteristics. In addition, existing grid-type control technologies mostly directly draw on the droop characteristics of synchronous generators, lacking fine simulation of virtual inertia and frequency dynamic response. They also lack a systematic and coordinated design of virtual impedance decoupling, adaptive damping control and transient overcurrent protection, making it difficult for on-board chargers to simultaneously achieve resonance suppression, power decoupling and transient safety in the complex resistive environment of low-voltage distribution networks. Summary of the Invention

[0004] This application provides a grid-type control method and system for suppressing grid-connected resonance of on-board chargers. It improves the technical problems in the prior art where grid-type control relies on phase-locked loops to passively follow the grid, which is prone to instability due to impedance fluctuations under weak grid conditions. Furthermore, the use of physical resistors or fixed-parameter notch filters to suppress resonance introduces additional power losses, and the suppression effect decreases when the grid impedance changes and the resonant frequency drifts.

[0005] This application discloses the following technical solution: In a first aspect, this application provides a grid-type control method for suppressing grid-connected resonance in on-board chargers, the method comprising: The three-phase voltage at the point of common coupling and the three-phase current on the grid side are collected. After coordinate transformation and low-pass filtering, the actual active power and actual reactive power are extracted. Based on the actual active power and actual reactive power, a power control outer loop that includes frequency and voltage droop characteristics and virtual inertia simulation is used to generate the phase reference value of the internal potential and the voltage amplitude reference value. By introducing virtual inductors and virtual resistors, d-axis and q-axis cross-decoupling compensation is performed on the voltage amplitude reference value to generate corrected d-axis and q-axis voltage commands. A transient virtual resistance control loop based on a high-pass filter is constructed to extract the high-frequency resonant component in the grid-side current and generate a transient compensation voltage signal. A modulation wave signal is generated through voltage and current dual closed-loop control. When the output current exceeds the current limiting threshold, the current amplitude is limited by a current command limiting mechanism. Finally, a pulse width modulation signal is generated from the modulation wave signal.

[0006] Secondly, this application provides a grid-type control system for suppressing grid-connected resonance of on-board chargers, the system comprising: The data acquisition module is used to collect the three-phase voltage and grid-side three-phase current at the point of common coupling. After coordinate transformation and low-pass filtering, the actual active power and actual reactive power are extracted. The power control module is used to generate the phase reference value of the internal potential and the voltage amplitude reference value based on the actual active power and the actual reactive power, using a power control outer loop that includes frequency and voltage droop characteristics and virtual inertia simulation. The virtual compensation module is used to introduce virtual inductance and virtual resistance, perform d-axis and q-axis cross-decoupling compensation on the voltage amplitude reference value, and generate corrected d-axis and q-axis voltage commands; construct a transient virtual resistance control loop based on a high-pass filter, extract the high-frequency resonant component in the grid-side current, and generate a transient compensation voltage signal; The modulation generation module is used to generate a modulation wave signal through voltage and current dual closed-loop control, and when the output current exceeds the current limiting threshold, the current amplitude is limited by the current command limiting mechanism, and finally the modulation wave signal is used to generate a pulse width modulation signal.

[0007] One or more technical solutions provided in this application have at least the following technical effects or advantages: This application's technical solution provides a grid-based control method for suppressing grid-connected resonance in on-board chargers. First, by acquiring the three-phase voltage and grid-side three-phase current at the point of common coupling, the actual active and reactive power are extracted after synchronous rotating coordinate transformation and low-pass filtering, converting the raw electrical quantities into smooth and accurate power feedback signals. Coordinate transformation converts AC quantities into DC quantities for easier subsequent control processing. The low-pass filter adaptively sets the cutoff frequency based on the fundamental frequency, filtering out switching ripple and high-frequency noise while retaining the dynamic response capability required for power control. This process provides a stable and reliable power measurement basis for the power outer loop, solving the problem of large power feedback fluctuations and reduced control accuracy caused by directly using instantaneous power or fixed filter parameters in traditional methods.

[0008] Furthermore, by employing a power control outer loop that incorporates frequency and voltage droop characteristics as well as virtual inertia simulation, the rotor motion equation and excitation regulation characteristics of a synchronous generator are simulated. Frequency correction and inertia response compensation are generated based on the active power deviation using droop and virtual inertia coefficients, and then superimposed and integrated to obtain the phase reference value of the internal electromotive force. Simultaneously, voltage correction is generated based on the reactive power deviation using voltage droop coefficient, and this is superimposed on the reference voltage amplitude to obtain the voltage amplitude reference value of the internal electromotive force. This grid-based control method enables the on-board charger to autonomously establish voltage and frequency during grid-connected operation, and provides inertial support through virtual inertia to smooth frequency fluctuations during load changes. This solves the problems of phase-locking difficulties and frequency instability caused by lack of inertia in grid-connected control under weak grid conditions.

[0009] Next, by introducing virtual inductors and virtual resistors, d-axis and q-axis cross-decoupling compensation is performed on the voltage amplitude reference value, generating corrected d-axis and q-axis voltage commands. The virtual inductor makes the converter's equivalent output impedance inductive to achieve static decoupling of active and reactive power. The virtual resistor is set according to the desired damping coefficient at the LCL resonant frequency to enhance system damping. Simultaneously, a transient virtual resistor control loop based on a high-pass filter is constructed. High-frequency resonant components are extracted from the grid-side current and multiplied by the transient virtual resistor gain value to generate a transient compensation voltage signal, which is then superimposed on the voltage command. The high-pass filter's gain approaches zero at the fundamental frequency, generating compensation only in the high-frequency resonant band. It adaptively suppresses the LCL resonant peak without the need for physical damping resistors, avoiding additional power loss. This process combines virtual impedance decoupling with adaptive resonant damping, solving the problems of traditional fixed-parameter notch filters failing at resonant frequency drift and the additional losses introduced by physical resistors.

[0010] Finally, voltage commands are tracked and regulated through dual closed-loop control of voltage and current. The outer voltage loop generates current commands through proportional-integral (PI) calculations, while the inner current loop generates modulated voltages through PI calculations. When the output current amplitude exceeds the current limiting threshold determined by the ratio of rated capacity to rated voltage, the d-axis and q-axis current commands are scaled proportionally to limit the output current, automatically constraining the current amplitude to protect the equipment during transient overcurrent. Then, the dq-axis modulated voltage is transformed from a synchronous rotating coordinate system to a stationary coordinate system to obtain a three-phase modulated wave, which is compared with the carrier wave to generate a pulse width modulation signal to drive the inverter. This process connects voltage tracking, current limiting, and modulation generation into a complete underlying control link, solving the problem that traditional control methods lack an effective self-limiting protection mechanism under overcurrent transient conditions, which can easily lead to equipment damage or malfunctioning protection.

[0011] In summary, this application establishes a voltage and frequency reference by simulating the droop characteristics and virtual inertia of a synchronous generator in the outer loop of the grid-type power control. It also achieves decoupled control of active and reactive power by combining virtual impedance decoupling with adaptive transient damping compensation, thereby suppressing the resonance peak of the LCL filter. Finally, a transient current limiting protection mechanism is embedded in the dual closed-loop control, systematically solving the technical problems of easy instability of on-board chargers in weak grid conditions, fixed resonance suppression methods, and difficulty in simultaneously addressing power coupling. Attached Figure Description

[0012] To more clearly illustrate the technical solutions in the embodiments of this application, the accompanying drawings used in the description of the embodiments will be briefly introduced below. Obviously, the accompanying drawings described below are only some embodiments of this application. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0013] Figure 1 This is a schematic flowchart of a grid-type control method for suppressing grid-connected resonance of an on-board charger, provided in an embodiment of this application.

[0014] Figure 2 The flowchart shows the dual-closed-loop current limiting control and PWM generation process of the grid-type control method for suppressing grid resonance of on-board chargers provided in the embodiments of this application.

[0015] Figure 3 This is a schematic diagram of the grid-type control system for grid-connected resonance suppression of on-board chargers provided in an embodiment of this application.

[0016] In the attached diagram, Figure 3 The components represented by each number are described as follows: data acquisition module 11, power control module 12, virtual compensation module 13, and modulation generation module 14. Detailed Implementation

[0017] This application provides a grid-based control method and system for suppressing grid resonance in on-board chargers. It addresses the technical problems in the prior art where on-board chargers in low-voltage distribution network environments suffer from a lack of adaptability of the control system to changes in grid strength, resulting in harsh frequency and voltage regulation responses during load fluctuations, mutual interference between active and reactive power regulation, difficulty in adjusting the suppression of inherent filter resonance to follow the shift of the resonance point, and a lack of effective self-protection mechanisms for current under transient impacts. These issues lead to the charger's inability to operate stably and continuously under complex grid conditions.

[0018] The technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of this application, and not all embodiments. Based on the embodiments of this application, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of this application. It should be noted that the numerical values ​​in the embodiments are for illustrative purposes only and do not constitute a limitation on this application.

[0019] Example 1, as shown in the appendix Figure 1 As shown, this application provides a grid-type control method for suppressing grid-connected resonance in on-board chargers, the method comprising the following steps: S100: Collects the three-phase voltage and grid-side three-phase current at the point of common coupling, and extracts the actual active power and actual reactive power after coordinate transformation and low-pass filtering.

[0020] In this embodiment of the application, in a V2G application scenario, the on-board bidirectional charger is connected to the low-voltage distribution network via the AC side to provide accurate power feedback signals for grid-based control. This requires collecting voltage and current data from the point of common coupling and performing signal conditioning to convert the three-phase AC quantities into smooth DC power signals. Traditional methods directly employ instantaneous power calculation or fixed-cutoff-frequency filtering, failing to adaptively adjust the filtering parameters based on the relationship between the actual fundamental frequency and the switching frequency. This results in high-frequency switching ripple and fundamental components overlapping in the power feedback signal, affecting the steady-state accuracy and dynamic response speed of the subsequent power control outer loop.

[0021] Step S100 of the method provided in this application embodiment includes: acquiring the three-phase voltage and grid-side three-phase current at the common connection point using voltage and current sensors, wherein the common connection point is the connection point between the AC side of the on-board bidirectional charger and the low-voltage distribution network; transforming the acquired three-phase voltage and grid-side three-phase current to a synchronous rotating coordinate system to obtain d-axis voltage components, q-axis voltage components, d-axis current components, and q-axis current components; calculating the original active power and original reactive power based on the d-axis voltage components, q-axis voltage components, d-axis current components, and q-axis current components; obtaining the fundamental frequency of the three-phase voltage, determining the cutoff frequency of the low-pass filter based on the fundamental frequency, wherein the cutoff frequency is less than the fundamental frequency; passing the original active power and original reactive power through the low-pass filter respectively to filter out high-frequency switching ripple components, and outputting the actual active power and actual reactive power. Detailed explanation follows: In this embodiment, the common connection point is the electrical connection node between the AC side of the on-board bidirectional charger and the low-voltage distribution network, and it is also the installation location of the voltage and current sensors. The synchronous rotating coordinate system is a dq orthogonal coordinate system that rotates continuously at the fundamental angular frequency. In steady state, each voltage and current exhibits a constant DC component in this coordinate system, facilitating subsequent proportional-integral control processing. The d-axis voltage component and q-axis voltage component are the direct and quadrature axis voltage values ​​of the three-phase voltages after transformation in the synchronous rotating coordinate system. The d-axis current component and q-axis current component are the direct and quadrature axis current values ​​of the three-phase currents after transformation in the synchronous rotating coordinate system. The original active power is the active power value directly calculated based on the d-axis and q-axis voltage and current components according to instantaneous power theory, without filtering, and includes the high-frequency ripple component generated by the inverter switching action. Similarly, the original reactive power is the reactive power value calculated by multiplying the d-axis voltage by the q-axis current and subtracting the q-axis voltage multiplied by the d-axis current. A low-pass filter is a first- or second-order filter used to filter out high-frequency switching ripple components in a power signal. Its cutoff frequency is adaptively set according to the fundamental frequency and must be less than the fundamental frequency to ensure that the fundamental power component passes through completely while the high-frequency ripple is attenuated.

[0022] In this step, to obtain the accurate power feedback signal required by the subsequent power control outer loop, the instantaneous values ​​of the three-phase voltage and current are first synchronously acquired using voltage and current sensors installed at the common connection point. Then, the acquired three-phase AC quantities need to be transformed from the stationary coordinate system to the synchronous rotating coordinate system. The transformation process is performed in two steps: First, the three-phase voltage and current are transformed to a two-phase stationary coordinate system using Clark transformation, obtaining two-phase stationary voltage components and two-phase stationary current components; second, using the phase reference value output in real time by the power control outer loop as the rotation angle reference for the Park transformation, the two-phase stationary voltage and two-phase stationary current are transformed to the synchronous rotating coordinate system using Park transformation, obtaining the d-axis voltage component, q-axis voltage component, d-axis current component, and q-axis current component. The phase reference value is continuously updated by the power control outer loop to maintain the rotational angular velocity of the synchronous rotating coordinate system consistent with the fundamental angular frequency of the three-phase voltage, thereby ensuring that the d-axis and q-axis components are DC quantities in steady state.

[0023] Next, the original active power and original reactive power are calculated based on the d-axis and q-axis voltage and current components. The original active power is calculated as follows: multiply the d-axis voltage component by the d-axis current component, multiply the q-axis voltage component by the q-axis current component, and then add the two products to obtain the original active power. The original reactive power is calculated as follows: multiply the d-axis voltage component by the q-axis current component to obtain the first reactive power component, multiply the q-axis voltage component by the d-axis current component to obtain the second reactive power component, and then subtract the second reactive power component from the first reactive power component to obtain the original reactive power.

[0024] Then, to retain the dynamic response bandwidth required for power control while filtering out high-frequency switching ripple, the cutoff frequency of the low-pass filter needs to be adaptively determined. The fundamental frequency of the three-phase voltage is obtained, typically the rated fundamental frequency of 50Hz. The pulse width modulation switching frequency of the on-board bidirectional charger is obtained, and the initial proportional gain is determined based on the ratio of the switching frequency to the fundamental frequency; the higher the switching frequency, the smaller the initial proportional gain can be. Furthermore, the expected response time set by the power control outer loop during the design phase for load disturbances is obtained. This expected response time is greater than the fundamental period (20 milliseconds) and much larger than the switching period to ensure the dynamic response speed of the power outer loop. The initial proportional gain is corrected based on the expected response time, specifically using the formula: Preset proportional gain = Initial proportional gain × Fundamental period ÷ Expected response time. Here, the fundamental period is the reciprocal of the fundamental frequency. The physical meaning of this correction method is as follows: the shorter the desired response time, the larger the time correction factor (fundamental period divided by the desired response time), the larger the preset proportional coefficient, the higher the cutoff frequency, and the faster the dynamic response, but the smoothness of the filter slightly decreases; conversely, the longer the desired response time, the smaller the time correction factor, the smaller the preset proportional coefficient, the lower the cutoff frequency, the smoother the filter, but the slower the response. Through the above correction, the preset proportional coefficient simultaneously takes into account the requirements for switching ripple filtering and the dynamic response requirements of the power outer loop. Finally, the fundamental frequency is multiplied by the preset proportional coefficient to obtain the cutoff frequency of the low-pass filter. This cutoff frequency is less than the fundamental frequency, and its corresponding response period is less than the desired response time, thus satisfying the dual constraints of filtering effect and dynamic response.

[0025] Finally, the original active power and original reactive power are passed through the low-pass filter to filter out the superimposed high-frequency switching ripple components, and the smoothed actual active power and actual reactive power are output as the input feedback signal for the subsequent power control outer loop.

[0026] Furthermore, step S100 in the method provided in this application embodiment further includes: multiplying the d-axis voltage component and the q-axis voltage component with the d-axis current component and the q-axis current component on the corresponding axis, respectively, to obtain the active power component of each axis; and adding the active power components of each axis to obtain the original active power; multiplying the d-axis voltage component by the q-axis current component to obtain the first reactive power component; multiplying the q-axis voltage component by the d-axis current component to obtain the second reactive power component; and subtracting the second reactive power component from the first reactive power component to obtain the original reactive power. A detailed explanation follows: In this embodiment, the active power component refers to the product of each voltage axis component and its corresponding current axis component. Specifically, the d-axis active power component is obtained by multiplying the d-axis voltage component and the d-axis current component, and the q-axis active power component is obtained by multiplying the q-axis voltage component and the q-axis current component. The first reactive power component is the product of the d-axis voltage component and the q-axis current component, and the second reactive power component is the product of the q-axis voltage component and the d-axis current component. In a synchronously rotating coordinate system, the instantaneous value of reactive power is given by the difference between these two cross-axis terms.

[0027] In this step, to accurately calculate instantaneous power in a synchronous rotating coordinate system, the power calculation formula in the dq coordinate system is followed. In the synchronous rotating coordinate system, active power equals the sum of the products of each axis voltage and the coaxial current. Therefore, multiplying the d-axis voltage component by the d-axis current component yields the first active power component, and multiplying the q-axis voltage component by the q-axis current component yields the second active power component. Adding these two components gives the original active power. In the synchronous rotating coordinate system, reactive power equals the product of the d-axis voltage and the q-axis current minus the product of the q-axis voltage and the d-axis current. Therefore, multiplying the d-axis voltage component by the q-axis current component yields the first reactive power component, and multiplying the q-axis voltage component by the d-axis current component yields the second reactive power component. Subtracting the second reactive power component from the first reactive power component gives the original reactive power. The above calculations directly utilize DC quantities in the synchronous rotating coordinate system, avoiding the complex calculations in traditional three-phase instantaneous power calculations.

[0028] Furthermore, step S100 in the method provided in this application embodiment further includes: obtaining the rated fundamental frequency of the three-phase voltage and using the rated fundamental frequency as the fundamental frequency; obtaining the switching frequency of the on-board bidirectional charger and determining an initial proportional coefficient based on the ratio of the switching frequency to the fundamental frequency, wherein the switching frequency is greater than the fundamental frequency; obtaining the expected response time of the power control outer loop and correcting the initial proportional coefficient based on the expected response time to obtain a preset proportional coefficient; the expected response time is the adjustment time set by the power control outer loop for load disturbance during the design phase, and its value is greater than the fundamental period of the three-phase voltage and greater than the switching period of the on-board bidirectional charger; multiplying the fundamental frequency by the preset proportional coefficient to obtain the cutoff frequency of the low-pass filter, wherein the cutoff frequency is less than the fundamental frequency, and the response period corresponding to the cutoff frequency is less than the expected response time. Detailed explanation follows: In this embodiment, the fundamental frequency is taken as the rated fundamental frequency of the three-phase voltage, i.e., the standard frequency of the distribution network, 50Hz, as the calculation basis for determining the cutoff frequency. The switching frequency is the carrier frequency used by the power switching devices in the on-board bidirectional charger to perform pulse width modulation. It is usually in the range of several kHz to tens of kHz, much higher than the fundamental frequency, and is the source of high-frequency ripple in the power signal. The initial scaling factor is initially determined by the ratio of the switching frequency to the fundamental frequency. For example, when the switching frequency is 10kHz, this ratio is 200. The initial scaling factor can be a fraction less than 1, such as 0.2, so that the initial cutoff frequency is approximately 10Hz.

[0029] The expected response time is the anticipated time for the power control outer loop to adjust to load fluctuations and reach a steady state. This parameter is introduced to ensure that the low-pass filter does not excessively delay the transmission of the power feedback signal. The expected response time needs to be greater than the fundamental frequency period by approximately 20 milliseconds to ensure that the fundamental component is not falsely filtered out, and also much greater than the switching period to ensure that the filtering stage has sufficient smoothing time. The preset proportional coefficient is a value obtained by correcting the initial proportional coefficient based on the constraint of the expected response time. Its determination principle is to ensure that, under the condition that the cutoff frequency is less than the fundamental frequency, the response time constant corresponding to the cutoff frequency is less than the expected response time. The response period corresponding to the cutoff frequency is usually taken as several times the reciprocal of the cutoff frequency.

[0030] In this step, to adaptively determine the optimal cutoff frequency of the low-pass filter, the rated fundamental frequency of the three-phase voltage is first obtained as the fundamental frequency. Then, the switching frequency of the on-board bidirectional charger is obtained, and the ratio of the switching frequency to the fundamental frequency is calculated. A larger ratio indicates that the switching ripple is farther from the fundamental frequency and easier to filter and separate. Therefore, the initial proportional coefficient can be determined according to the reciprocal of this ratio, ensuring that the initial cutoff frequency is within a certain proportional range of the fundamental frequency. Next, the expected response time of the power control outer loop is obtained. This parameter reflects the system's requirements for dynamic power response. The expected response time is compared with the time constant corresponding to the initial cutoff frequency. If a faster response time is required, the cutoff frequency needs to be appropriately increased, i.e., the preset proportional coefficient is increased; if the response time requirement is more lenient, the cutoff frequency can be decreased to enhance the filtering effect. Finally, the fundamental frequency is multiplied by the corrected preset proportional coefficient to obtain the cutoff frequency. This cutoff frequency satisfies the constraint of being less than the fundamental frequency, and its corresponding response period is less than the expected response time, ensuring that the filtering process effectively attenuates the switching ripple without slowing down the dynamic adjustment speed of the power outer loop.

[0031] In summary, this step transforms the three-phase AC quantities into DC quantities in a synchronous rotating coordinate system through coordinate transformation. The low-pass filter cutoff frequency is adaptively determined with the fundamental frequency and switching frequency as references. This effectively filters out high-frequency switching ripple while retaining the dynamic response capability required for power control, providing stable and smooth feedback signals of actual active and reactive power for the subsequent power control outer loop.

[0032] S200: Based on the actual active power and actual reactive power, a power control outer loop including frequency and voltage droop characteristics and virtual inertia simulation is adopted to generate the phase reference value of the internal potential and the voltage amplitude reference value.

[0033] In this embodiment, based on the obtained actual active and reactive power, to establish an autonomous voltage and frequency reference for the on-board charger, it is necessary to simulate the active-frequency droop and reactive-voltage droop characteristics of the synchronous generator and provide inertial support to smooth frequency fluctuations during load abrupt changes. Traditional grid-following control passively follows the grid voltage phase through a phase-locked loop (PLL). Under weak grid conditions, due to the large and frequent fluctuations in the equivalent grid impedance, the PLL struggles to stably lock the phase, leading to system instability. Furthermore, the lack of simulation of the synchronous generator rotor inertia results in harsh frequency and voltage responses during load abrupt changes, failing to provide effective inertial support and frequency regulation capabilities for the grid.

[0034] Step S200 of the method provided in this application embodiment includes: obtaining a frequency correction amount by combining the difference between the actual active power and the active power reference command with a frequency droop coefficient, wherein the active power reference command is the target output active power value of the on-board bidirectional charger, and the frequency droop coefficient is a ratio determined based on the rated power of the on-board bidirectional charger and the allowable frequency deviation range; superimposing the frequency correction amount onto a reference angular frequency to obtain an initial angular frequency command value, wherein the reference angular frequency is the angular frequency value corresponding to the rated fundamental frequency of the three-phase voltage; and obtaining an inertia response compensation amount by combining the difference between the initial angular frequency command value and the rated angular frequency of the three-phase voltage with a virtual inertia coefficient and processing it through a first-order inertial element, wherein the virtual inertia coefficient is used to obtain an inertia response compensation amount. The pseudo-inertia coefficient is a ratio determined based on the rated capacity of the on-board bidirectional charger and the desired frequency change rate limit, and the virtual inertia coefficient is greater than zero. The inertia response compensation is superimposed on the initial angular frequency command value to output the angular frequency command value. The angular frequency command value is integrated to output the phase reference value of the internal potential. The difference between the actual reactive power and the reactive power reference command is multiplied by the voltage droop coefficient to obtain the voltage correction amount, where the voltage droop coefficient is a ratio determined based on the reactive power adjustment range and the allowable voltage deviation range of the on-board bidirectional charger. The voltage correction amount is superimposed on the reference voltage amplitude to output the voltage amplitude reference value of the internal potential, where the reference voltage amplitude is the rated voltage amplitude of the three-phase voltage. Detailed explanation follows: In this embodiment, the frequency droop coefficient is a ratio determined based on the rated active power of the on-board bidirectional charger and the allowable frequency deviation range, with units of Hz / W or rad / s / W. It represents the frequency adjustment caused by a unit active power deviation and is used to simulate the active-frequency droop characteristics of a synchronous generator. When the actual active power is less than the active power reference command, the frequency correction is positive, increasing the output frequency to increase active power output; when the actual active power is greater than the reference command, the frequency correction is negative, decreasing the output frequency to reduce active power output. The active power reference command is the target output active power value of the on-board bidirectional charger, given by the upper-level energy management system or charging scheduling strategy. The reference angular frequency is the angular frequency value corresponding to the rated fundamental frequency of the three-phase voltage; for example, the reference angular frequency corresponding to a 50Hz fundamental frequency is 314.16 rad / s. The initial angular frequency command value is the preliminary frequency command obtained by superimposing the frequency correction amount onto the reference angular frequency.

[0035] The virtual inertia coefficient is a ratio determined by the rated capacity of the on-board bidirectional charger to the desired frequency change rate limit. The specific calculation formula is: Virtual Inertia Coefficient = Rated Capacity ÷ Desired Frequency Change Rate Limit. Wherein, the rated capacity is the rated apparent power of the on-board bidirectional charger, measured in volt-amperes (VA); and the desired frequency change rate limit is the maximum permissible rate of change of the system's angular frequency, measured in radians per square second.

[0036] The physical meaning of this formula is as follows: the larger the rated capacity, the stronger the inertia support capability provided by the system, and the larger the virtual inertia coefficient; the smaller the desired frequency change rate limit, the higher the system's requirement for frequency stability, and the larger the required virtual inertia coefficient. This formula transforms the two quantifiable system parameters, rated capacity and frequency change rate limit, into specific values ​​for the virtual inertia coefficient, providing a clear parameterized basis for virtual inertia simulation. The value is greater than zero and is used to simulate the rotational inertia of the synchronous generator rotor. When a sudden change in active power causes a deviation between the initial angular frequency command value and the rated angular frequency, the virtual inertia coefficient converts this deviation into an inertia response compensation amount, smoothing frequency fluctuations by delaying the frequency change rate. The specific conversion method is: Inertia response compensation amount = Virtual inertia coefficient × Angular frequency deviation × Transfer function of the first-order inertial element. The first-order inertial element is used to further smooth the transition process of the inertia response compensation amount, and its time constant can be set according to the system's inertia time constant design requirements. After the inertia response compensation amount is superimposed on the initial angular frequency command value, the final angular frequency command value is output. The angular frequency command value is integrated using the formula θ=∫ωdt, and the phase reference value of the internal potential is output. This phase reference value is used for the inverse coordinate transformation when generating the three-phase modulated wave, and is also fed back to the aforementioned steps as the rotation angle reference for the Park transformation, forming a closed loop.

[0037] The voltage droop coefficient is a ratio determined based on the reactive power adjustment range of the on-board bidirectional charger to the allowable voltage deviation range. It represents the voltage amplitude adjustment caused by a unit reactive power deviation and is used to simulate the reactive power-voltage droop characteristics of a synchronous generator. When the actual reactive power is less than the reactive power reference command, the voltage correction is negative, reducing the voltage amplitude to decrease reactive power output; when the actual reactive power is greater than the reference command, the voltage correction is positive, increasing the voltage amplitude to increase reactive power output. The reactive power reference command is given by the upper-level dispatcher. The reference voltage amplitude is the rated voltage amplitude of the three-phase voltage; for example, the amplitude corresponding to a 220V phase voltage is approximately 311V. The voltage correction is superimposed on the reference voltage amplitude to output the reference value of the voltage amplitude of the internal potential.

[0038] In this step, to establish autonomous frequency regulation capability in the active power dimension, the deviation between the actual active power and the active power reference command is first calculated. This deviation is multiplied by the frequency droop factor to obtain the frequency correction. The value of the frequency droop factor is determined based on the rated active power and the maximum allowable frequency deviation range of the system. For example, if the rated power is 7kW and the allowable frequency deviation is 0.5Hz, then the frequency droop factor can be taken as 0.5÷7000≈7.14×10⁻ 5 Hz / W. The frequency correction is superimposed on the reference angular frequency of 314.16 rad / s to obtain the initial angular frequency command value.

[0039] To provide inertial support during sudden load changes, the difference between the initial angular frequency command value and the rated angular frequency is further calculated. This difference is multiplied by a virtual inertia coefficient and then processed by a first-order inertial element to obtain the inertia response compensation. For example, with a rated capacity of 7kVA and an allowable frequency change rate limit of 1Hz / s, the virtual inertia coefficient can be taken as 7000÷1=7000W·s / Hz. The inertia response compensation is then superimposed on the initial angular frequency command value to obtain the final angular frequency command value. The angular frequency command value is integrated over time, and the phase reference value of the internal potential is output. The integration process causes the phase to increase linearly with time, and the rotational angular velocity equals the angular frequency command value.

[0040] In terms of reactive power, the deviation between the actual reactive power and the reactive power reference command is calculated. This deviation is then multiplied by the voltage droop factor to obtain the voltage correction. The value of the voltage droop factor is determined based on the reactive power adjustment range and the allowable voltage deviation range. For example, if the reactive power adjustment range is ±3.5 kvar and the allowable voltage deviation is ±10%, approximately ±31 V, then the voltage droop factor can be taken as 31 ÷ 3500 ≈ 0.0089 V / var. The voltage correction is then superimposed on the reference voltage amplitude to output the voltage amplitude reference value of the internal potential.

[0041] For example, consider a 7kW on-board bidirectional charger operating in V2G mode. The current actual active power is 5kW, the reference active power command is 6kW, and the active power deviation is +1kW. The frequency droop factor is taken as 0.5Hz / 7kW ≈ 7.14 × 10⁻ 5 Hz / W, frequency correction = 1000 × 7.14 × 10⁻ 5The initial angular frequency command value is approximately 0.449 rad / s, corresponding to a frequency correction of approximately 0.0714 Hz. The initial angular frequency command value, after being superimposed from the reference angular frequency of 314.16 rad / s, is approximately 314.61 rad / s. The deviation between the initial angular frequency command value and the rated angular frequency is approximately 0.449 rad / s. The virtual inertia coefficient is taken as 7000 W·s / Hz, corresponding to approximately 43960 W·s / (rad / s). The inertia response compensation is smoothed by a first-order inertial element and then superimposed to obtain the final angular frequency command value. After integration, the phase reference value of the internal potential is output.

[0042] In summary, this step generates a frequency command by combining active power-frequency droop and virtual inertia, and obtains a phase reference value through integration. It also generates a voltage amplitude reference value by using reactive power-voltage droop, thus simulating the autonomous frequency and voltage regulation characteristics of a synchronous generator. This provides inertial support and autonomous frequency and voltage establishment capability for the on-board charger in weak grid conditions.

[0043] S300: Introduce virtual inductors and virtual resistors, perform d-axis and q-axis cross-decoupling compensation on the voltage amplitude reference value, and generate corrected d-axis and q-axis voltage commands; construct a transient virtual resistance control loop based on a high-pass filter, extract the high-frequency resonant component in the grid-side current, and generate a transient compensation voltage signal.

[0044] In this embodiment, based on the obtained reference value of the voltage amplitude of the internal potential, due to the inherent resonance peak of the LCL filter in the on-board charger and the coupling effect of line impedance, directly using this reference value for control will lead to mutual interference between active and reactive power. Furthermore, the resonant frequency drift caused by changes in grid impedance reduces the suppression effect of the fixed-parameter notch filter. Traditional methods use fixed virtual impedance decoupling parameters, which cannot adapt to changes in grid strength, and the power coupling problem is more severe under weak grid conditions. Simultaneously, LCL resonance peak suppression relies on physical resistance or a fixed-parameter notch filter. Physical resistance introduces additional losses, and the fixed notch filter cannot automatically follow the resonant frequency shift, significantly degrading the resonance suppression capability and even causing high-frequency oscillations. To solve these problems, this step uses a virtual inductor to make the equivalent output impedance inductive, achieving static power decoupling. A virtual resistor enhances the damping at the LCL resonant frequency, and a high-pass filter extracts the high-frequency resonant component, which is then used to generate a compensation voltage through the transient virtual resistor gain, achieving coordinated control of adaptive resonance suppression and power decoupling.

[0045] Step S300 of the method provided in this application embodiment includes: introducing virtual inductance and virtual resistance, performing d-axis and q-axis cross-decoupling compensation on the voltage amplitude reference value, and generating corrected d-axis and q-axis voltage commands, including: the value of the virtual inductance is determined based on the ratio of the line equivalent resistance to the equivalent reactance at the common connection point, and after dimensional conversion in combination with the fundamental frequency and the rated voltage amplitude of the three-phase voltage; the value of the virtual resistance is determined based on the expected damping coefficient at the resonant frequency of the LCL filter of the on-board bidirectional charger, the expected damping coefficient being a preset value greater than zero and less than 1, and the larger the expected damping coefficient, the larger the value of the virtual resistance; using the voltage amplitude reference value as the initial voltage command for the d-axis, and setting the initial voltage command for the q-axis to zero; determining the d-axis resistance compensation amount based on the virtual resistance and the d-axis current component, wherein the d-axis resistance compensation amount and the... The virtual resistance and the d-axis current component are both proportional. Based on the virtual inductance and the q-axis current component, a d-axis cross-compensation amount is determined, where the d-axis cross-compensation amount is proportional to both the virtual inductance and the q-axis current component. The d-axis resistance compensation amount is superimposed on the initial d-axis voltage command, and then the d-axis cross-compensation amount is subtracted to generate a corrected d-axis voltage command. Similarly, based on the virtual resistance and the q-axis current component, a q-axis resistance compensation amount is determined, where the q-axis resistance compensation amount is proportional to both the virtual resistance and the q-axis current component. Based on the virtual inductance and the d-axis current component, a q-axis cross-compensation amount is determined, where the q-axis cross-compensation amount is proportional to both the virtual inductance and the d-axis current component. The q-axis resistance compensation amount and the q-axis cross-compensation amount are superimposed on the initial q-axis voltage command to generate a corrected q-axis voltage command. Detailed explanation follows: In this embodiment, the virtual inductance is an equivalent inductance parameter introduced in the control algorithm, making the converter output impedance inductive in the fundamental frequency band. Its value is determined by dimensional conversion based on the ratio of the equivalent resistance to the equivalent reactance at the point of common coupling, combined with the fundamental frequency and rated voltage amplitude. When the ratio of line resistance to reactance is large (i.e., biased resistance), the virtual inductance value needs to be increased accordingly to make the inductive component of the equivalent output impedance dominant, thereby achieving static decoupling of active and reactive power. The virtual resistance is an equivalent resistance parameter introduced in the control algorithm, and its value is determined based on the desired damping coefficient at the resonant frequency of the LCL filter. The desired damping coefficient is a preset parameter characterizing the relative damping level that the system expects to achieve at the resonant frequency. Its value ranges from greater than zero to less than 1; a larger value indicates stronger suppression of the resonant peak, and the corresponding virtual resistance value is also larger.

[0046] The d-axis initial voltage command directly takes the voltage amplitude reference value, while the q-axis initial voltage command is set to zero to orient the internal potential vector towards the d-axis. The d-axis resistance compensation equals the virtual resistance multiplied by the d-axis current component, used to compensate for the voltage drop on the d-axis due to resistive effects. The d-axis cross compensation equals the virtual inductance multiplied by the q-axis current component, used to decouple the coupling effect of the q-axis current on the d-axis voltage. The corrected d-axis voltage command equals the d-axis initial voltage command plus the d-axis resistance compensation minus the d-axis cross compensation. The q-axis resistance compensation equals the virtual resistance multiplied by the q-axis current component, and the q-axis cross compensation equals the virtual inductance multiplied by the d-axis current component. The corrected q-axis voltage command equals the q-axis initial voltage command plus the q-axis resistance compensation plus the q-axis cross compensation.

[0047] In this step, to achieve static decoupling of active and reactive power in the fundamental frequency band, it is first necessary to determine the parameter values ​​of the virtual impedance. The value of the virtual inductance is determined as follows: The ratio of the line's equivalent resistance to its equivalent reactance at the point of common coupling is obtained. This ratio can be measured in real-time using an online grid impedance identification algorithm or estimated during the system design phase using typical parameters of the distribution network. Substituting the ratio of line resistance to reactance into the dimensional conversion formula, and combining it with the fundamental angular frequency and rated voltage amplitude, the virtual inductance value is calculated. The specific dimensional conversion formula is: Virtual Inductance = (Line Equivalent Resistance ÷ Line Equivalent Reactance) × (Line Equivalent Resistance ÷ Fundamental Angular Frequency).

[0048] The ratio of the line equivalent resistance to the line equivalent reactance is the inductive compensation demand coefficient. A larger ratio indicates stronger grid resistivity and a greater need for inductive compensation. The line equivalent resistance divided by the fundamental angular frequency serves as the conversion reference value, with dimensions in ohms divided by radians per second, equal to the Henry, used to convert the dimensionless ratio to the Henry dimension. The rated voltage amplitude serves as the system's rated parameter, used to determine the per-unit reference for the line equivalent resistance and equivalent reactance, ensuring that the value of the virtual inductance matches the system's rated voltage level. It does not participate in the dimension conversion calculations in the dimensional conversion formula.

[0049] The physical meaning of this formula is that when the equivalent resistance of the line is greater than the equivalent reactance of the line, the value of the virtual inductance is increased accordingly to provide a stronger inductive decoupling effect. The dimension conversion step maps the dimensionless ratio of the engineering to the Henry dimension through the fundamental angular frequency, and uses the rated voltage amplitude as the reference benchmark for impedance measurement, so that the virtual inductance can be directly applied to the d-axis and q-axis cross-coupling compensation calculation.

[0050] The value of the virtual resistance is determined based on the desired damping coefficient at the resonant frequency of the LCL filter. The desired damping coefficient is selected in the range of 0.2 to 0.8. For example, a value of 0.5 indicates a moderate level of damping at the resonant frequency. The larger the desired damping coefficient, the larger the virtual resistance value should be. Specifically, it can be determined by the following ratio: Virtual resistance = Proportional coefficient × Desired damping coefficient. The proportional coefficient can be set based on the characteristic impedance of the LCL filter.

[0051] After determining the virtual impedance parameters, cross-coupling compensation is performed on the d-axis and q-axis. For the d-axis, the d-axis resistance compensation equals the virtual resistance multiplied by the d-axis current component, and the d-axis cross-compensation equals the virtual inductance multiplied by the q-axis current component. Since the virtual inductance makes the output impedance inductive, the q-axis current generates a coupling voltage drop in the d-axis direction through the virtual inductance. This cross-compensation needs to be subtracted from the d-axis voltage command to eliminate coupling. The corrected d-axis voltage command equals the initial d-axis voltage command (voltage amplitude reference value) plus the d-axis resistance compensation and then minus the d-axis cross-compensation. For the q-axis, the q-axis resistance compensation equals the virtual resistance multiplied by the q-axis current component, and the q-axis cross-compensation equals the virtual inductance multiplied by the d-axis current component. These two are superimposed on the initial q-axis voltage command (zero value) to generate the corrected q-axis voltage command. Through these compensations, the d-axis voltage primarily controls active power, and the q-axis voltage primarily controls reactive power; the coupling between the two is canceled out by the virtual inductance compensation.

[0052] Furthermore, step S300 in the method provided in this application embodiment further includes: constructing a transient virtual resistance control loop based on a high-pass filter, extracting high-frequency resonant components from the grid-side current, and generating a transient compensation voltage signal, including: inputting the grid-side three-phase current into a high-pass filter to extract high-frequency resonant components, wherein the cutoff frequency of the high-pass filter is greater than the fundamental frequency of the three-phase voltage and less than the resonant frequency of the LCL filter of the on-board bidirectional charger; determining a transient virtual resistance gain value based on the desired damping ratio at the resonant frequency of the LCL filter, wherein the transient virtual resistance gain value is proportional to the desired damping ratio; multiplying the high-frequency resonant components by the transient virtual resistance gain value to obtain a transient compensation voltage signal; superimposing the transient compensation voltage signal onto the corrected d-axis voltage command and the corrected q-axis voltage command to generate updated d-axis voltage commands and updated q-axis voltage commands; wherein the gain of the high-pass filter at the fundamental frequency of the three-phase voltage tends to zero, so that the transient virtual resistance control loop does not generate compensation voltage in steady state. Detailed explanation follows: In this embodiment, the high-pass filter is a filtering stage used to extract high-frequency resonant components above the cutoff frequency from the three-phase current on the grid side. The cutoff frequency is greater than the fundamental frequency by 50Hz and less than the resonant frequency of the LCL filter, causing its gain at the fundamental frequency to approach zero while having unity gain near the resonant frequency. The high-frequency resonant component is an oscillating component in the grid-side current with a frequency near the resonant frequency of the LCL filter, generated by the inherent resonant characteristics of the LCL filter when the grid impedance is matched. The transient virtual resistance gain value is a proportionality coefficient determined based on the desired damping ratio at the resonant frequency of the LCL filter. It is proportional to the desired damping ratio; the larger the desired damping ratio, the larger the gain value and the stronger the suppression of resonance. The transient compensation voltage signal is the compensation voltage obtained by multiplying the high-frequency resonant component by the transient virtual resistance gain value, which is superimposed on the d-axis and q-axis voltage commands to generate a voltage compensation amount proportional to the high-frequency resonant current, equivalent to introducing a virtual resistance for damping in the resonant frequency band.

[0053] In this step, to adaptively suppress the resonance peak of the LCL filter without increasing physical power loss, the three-phase grid current is first input to a high-pass filter. The cutoff frequency of the high-pass filter is selected between the fundamental frequency and the LCL resonant frequency. For example, when the fundamental frequency is 50Hz and the LCL resonant frequency is approximately 1.5kHz, the cutoff frequency can be between 200Hz and 500Hz. The gain of the high-pass filter is almost zero at the fundamental frequency of 50Hz, so the steady-state fundamental current component will not pass through this filter branch and will not affect normal power control. When the grid impedance changes or the load suddenly triggers the LCL filter to resonate, high-frequency oscillation components with frequencies near the resonant point appear in the grid current. These components are extracted by the high-pass filter as high-frequency resonant components.

[0054] Then, the transient virtual resistance gain value is determined based on the expected damping ratio at the resonant frequency of the LCL filter. The expected damping ratio characterizes the relative damping level expected at the resonant frequency, typically ranging from 0.1 to 0.7. A larger value indicates a stronger expected resonance suppression effect, and the gain value is correspondingly larger. The transient compensation voltage signal obtained by multiplying the high-frequency resonant component by the transient virtual resistance gain value is superimposed on the aforementioned corrected d-axis voltage command and corrected q-axis voltage command to generate updated d-axis voltage commands and updated q-axis voltage commands, respectively. The updated d-axis voltage command is equal to the corrected d-axis voltage command plus the transient compensation voltage signal, and the updated q-axis voltage command is generated similarly. Since the gain of the high-pass filter tends to zero at the fundamental frequency, the transient compensation voltage signal is zero when the system is operating in steady state without resonance. The transient virtual resistance control loop does not generate any additional output for normal control; it only automatically generates a damping compensation voltage to suppress high-frequency oscillations when resonance occurs.

[0055] For example, taking a 7kW on-board bidirectional charger operating in V2G mode, continuing the S200 example, the equivalent resistance of the line at the point of common coupling is approximately 0.2Ω, the equivalent reactance is approximately 0.08Ω, the resistance-to-reactance ratio is 2.5, the fundamental angular frequency is 314.16 rad / s, the rated voltage amplitude is 311V, and the virtual inductance after dimensional conversion is approximately 3.2mH. The LCL filter resonant frequency is approximately 1.6kHz, the desired damping coefficient is 0.5, and the virtual resistance is approximately 0.25Ω. The current d-axis current component is 16A, and the q-axis current component is 3A. The d-axis resistance compensation is 0.25 × 16 = 4V, the d-axis cross compensation is 314.16 × 0.0032 × 3 ≈ 3.0V, and the corrected d-axis voltage command is 311 + 4 − 3.0 = 312V. The q-axis resistance compensation is 0.25 × 3 = 0.75V, the q-axis cross compensation is 314.16 × 0.0032 × 16 ≈ 16.1V, and the corrected q-axis voltage command is 0 + 0.75 + 16.1 = 16.85V.

[0056] Based on this, the high-pass filter cutoff frequency is set to 300Hz, which is 50Hz higher than the fundamental frequency and 1.6kHz lower than the resonant frequency. The transient virtual resistance gain is set to 0.8 multiplied by the desired damping ratio of 0.5, which equals 0.4. If a high-frequency resonant component with an amplitude of approximately 0.5A appears in the current grid-side current, then the transient compensation voltage signal = 0.4 × 0.5 = 0.2V. The updated d-axis voltage command = 312 + 0.2 = 312.2V, and the updated q-axis voltage command = 16.85 + 0.2 = 17.05V. In steady-state without resonance, the high-frequency resonant component is zero, the transient compensation voltage signal is zero, and the updated command is consistent with the corrected command.

[0057] In summary, this step achieves static decoupling of active and reactive power by making the equivalent output impedance inductive through virtual inductance, and enhances the damping of the system at the LCL resonant frequency through virtual resistance. At the same time, a high-pass filter extracts the high-frequency resonant component from the grid-side current and generates a compensation voltage through transient virtual resistance gain. No additional output is generated at the fundamental frequency, and damping is automatically provided only when resonance occurs, thus realizing the coordinated control of adaptive resonance suppression and power decoupling.

[0058] S400: A modulated wave signal is generated through voltage and current dual closed-loop control, and when the output current exceeds the current limiting threshold, the current amplitude is limited by the current command limiting mechanism, and finally the pulse width modulation signal is generated from the modulated wave signal.

[0059] In this embodiment, based on the updated d-axis and q-axis voltage commands, precise tracking of the voltage commands is achieved through dual voltage and current closed-loop control. Simultaneously, the current amplitude is automatically limited to protect the equipment when the output current exceeds a safety threshold. Finally, the modulated voltage is converted into a pulse-width modulation signal to drive the inverter. Traditional dual voltage and current closed-loop control lacks an effective limiting protection mechanism during transient overcurrent, potentially causing the current command to exceed the equipment's safe range and damage the power devices due to overcurrent. Furthermore, the fixed parameters of the dual closed-loop system make it difficult to balance tracking accuracy and stability when the grid strength changes.

[0060] Step S400 of the method provided in this application embodiment includes: performing a proportional-integral operation on the difference between the updated d-axis voltage command and the d-axis voltage component to output a d-axis initial current command; performing a proportional-integral operation on the difference between the updated q-axis voltage command and the q-axis voltage component to output a q-axis initial current command; calculating an initial current amplitude based on the d-axis initial current command and the q-axis initial current command; when the initial current amplitude exceeds a current limiting threshold, scaling the d-axis initial current command and the q-axis initial current command according to the ratio of the current limiting threshold to the initial current amplitude to obtain a limited d-axis current reference command and a q-axis current reference command, wherein the current limiting threshold is based on the rated capacity of the on-board bidirectional charger. The ratio of the initial current amplitude to the rated voltage at the common connection point is determined; when the initial current amplitude does not exceed the current limiting threshold, the d-axis initial current command and the q-axis initial current command are directly used as the d-axis current reference command and the q-axis current reference command; the difference between the d-axis current reference command and the d-axis current component is processed by proportional-integral operation to output the d-axis modulation voltage; the difference between the q-axis current reference command and the q-axis current component is processed by proportional-integral operation to output the q-axis modulation voltage; the d-axis modulation voltage and the q-axis modulation voltage are transformed from a synchronous rotating coordinate system to a stationary coordinate system to obtain a three-phase modulation wave signal; the three-phase modulation wave signal is compared with the carrier signal to generate the pulse width modulation signal. The process of dual closed-loop current limiting control and PWM generation in this step is as follows: Figure 2 As shown. A detailed explanation follows: In this embodiment, proportional-integral (PI) operation is a classic control algorithm in the field of automatic control. Its principle is as follows: the deviation between the given command and the actual feedback is processed proportionally and integrally. The proportional part outputs an adjustment amount proportional to the current deviation, enabling the system to respond quickly to changes; the integral part outputs an adjustment amount proportional to the cumulative deviation over time, used to eliminate steady-state error; the two results are added together to form the final control output. The d-axis initial current command is the current reference value output by the voltage outer loop d-axis channel after P / I operation, and the q-axis initial current command is the current reference value output by the voltage outer loop q-axis channel after P / I operation. The initial current amplitude is calculated by taking the square root of the sum of the squares of the d-axis and q-axis initial current commands, i.e., the initial current amplitude equals the square root of the square of the d-axis initial current command plus the square of the q-axis initial current command, used to compare with the current limiting threshold to determine whether to trigger amplitude limiting protection. The current limiting threshold is calculated by dividing the rated capacity of the on-board bidirectional charger by the rated voltage at the point of common coupling. In other words, the current limiting threshold equals the rated capacity divided by the rated voltage, representing the maximum current amplitude that the equipment is allowed to operate safely for extended periods. The d-axis current reference command and q-axis current reference command are the current command values ​​that are finally sent to the inner current loop after amplitude limiting judgment.

[0061] In this step, to convert the updated voltage command into an inverter-executable current control signal, a voltage outer loop is first constructed. The updated d-axis voltage command is subtracted from the current d-axis voltage component to obtain the d-axis voltage deviation. This deviation is then fed into a proportional-integral (PI) regulator, and after processing by proportional gain and integral gain, the initial d-axis current command is output. Similarly, the updated q-axis voltage command is subtracted from the current q-axis voltage component, and after proportional-integral (PI) calculation, the initial q-axis current command is output. The proportional gain and integral time constant of the voltage outer loop are set according to the system's bandwidth and stability margin requirements. A larger proportional gain results in a faster response but also increases overshoot, while a smaller integral time constant eliminates steady-state error more quickly.

[0062] To protect power devices from damage during transient overcurrent, the initial current amplitude output from the outer voltage loop is calculated. When the initial current amplitude exceeds the current limiting threshold, it indicates that the current required for the current condition exceeds the equipment's safe range, necessitating the activation of current limiting protection. At this point, the ratio of the current limiting threshold to the initial current amplitude is used as a scaling factor, multiplied by both the d-axis and q-axis initial current commands, proportionally reducing the current commands on both axes. This ensures that the current vector direction after limiting is consistent with the original command, while the amplitude is restricted within the current limiting threshold. When the initial current amplitude does not exceed the current limiting threshold, no current limiting is performed; the d-axis and q-axis initial current commands are directly used as the current reference commands. Through this current limiting mechanism, when sudden load changes or grid anomalies cause excessive current commands to be output from the outer voltage loop, the system can automatically constrain the current within a safe range, solving the problem of traditional dual-loop control lacking effective self-limiting protection during transient overcurrent.

[0063] Then, a current inner loop is constructed. The d-axis current deviation is obtained by subtracting the current d-axis current component from the limited d-axis current reference command, and then output as the d-axis modulated voltage after proportional-integral (PI) operation. The q-axis current modulated voltage is also output by subtracting the current q-axis current component from the limited q-axis current reference command and then output as the q-axis modulated voltage after PI operation. The proportional gain and integral time constant of the current inner loop are also set according to the requirements of system dynamic performance and disturbance rejection capability. Typically, the inner loop bandwidth is much larger than the outer loop bandwidth to achieve fast current tracking.

[0064] Furthermore, step S400 in the method provided in this application embodiment further includes: obtaining a phase reference value of the internal potential, using the phase reference value as the transformation angle for the transformation from the synchronous rotating coordinate system to the stationary coordinate system; performing an inverse coordinate transformation on the d-axis modulation voltage and the q-axis modulation voltage with the transformation angle as the reference to obtain a three-phase modulation wave signal in the stationary coordinate system; obtaining a carrier signal, wherein the frequency of the carrier signal is greater than the fundamental frequency; comparing the instantaneous value of each phase modulation wave in the three-phase modulation wave signal with the carrier signal, outputting a high level when the instantaneous value of the modulation wave is greater than or equal to the instantaneous value of the carrier signal, and outputting a low level when the instantaneous value of the modulation wave is less than the instantaneous value of the carrier signal, thereby generating the pulse width modulation signal. A detailed explanation follows: In this embodiment, the transformation angle is a phase reference value of the internal potential, obtained by integrating the angular frequency command value output from the power control outer loop in S200, and used as the rotation angle reference for the inverse transformation from the synchronous rotating coordinate system to the stationary coordinate system. The three-phase modulated wave signal is a three-phase sinusoidal modulated signal obtained by inverse Park and inverse Clark transformations of the d-axis and q-axis modulated voltages, and its frequency and phase are determined by the transformation angle. The carrier signal is a triangular wave or sawtooth wave signal with a frequency much higher than the fundamental frequency. The carrier frequency is usually equal to the switching frequency, ranging from several kHz to tens of kHz.

[0065] In this step, to convert the d-axis and q-axis modulated voltages output from the inner current loop into PWM signals that can drive the inverter's power switching devices, the phase reference value of the internal potential output from the outer power control loop in S200 is first obtained as the transformation angle. Using this transformation angle as a reference, the d-axis and q-axis modulated voltages undergo an inverse Park transform to obtain the modulated voltage components in a two-phase stationary coordinate system. Then, the modulated voltages in the two-phase stationary coordinate system are subjected to an inverse Clark transform to obtain a three-phase modulated wave signal in a three-phase stationary coordinate system. The three-phase modulated wave signal is a sine wave with a phase difference of 120 degrees between each phase; its amplitude and frequency are determined by the d-axis and q-axis modulated voltages, and the phase reference is determined by the internal potential phase reference value.

[0066] Then, a carrier signal is acquired. The carrier signal frequency is greater than the fundamental frequency, typically a switching frequency such as 10kHz. The instantaneous value of the modulation wave in each phase of the three-phase modulation wave signal is continuously compared with the instantaneous value of the carrier signal. When the instantaneous value of the modulation wave in a certain phase is greater than or equal to the instantaneous value of the carrier signal, the corresponding upper bridge arm power switch outputs a high-level turn-on signal, and the lower bridge arm outputs a low-level turn-off signal; when the instantaneous value of the modulation wave is less than the instantaneous value of the carrier signal, the upper bridge arm outputs a low-level turn-off signal, and the lower bridge arm outputs a high-level turn-on signal. Through this pulse width modulation method, the fundamental component of the modulation wave is equivalently amplified into the voltage waveform output by the inverter, driving the switching on and off of the inverter power switching devices to generate an AC output voltage proportional to the fundamental frequency of the modulation wave.

[0067] In summary, this step generates a current command through a voltage outer loop proportional-integral operation. When the output current amplitude exceeds the current limiting threshold determined based on the ratio of rated capacity to rated voltage, the current command is scaled proportionally to limit the current amplitude. A modulation voltage is generated through a current inner loop proportional-integral operation. Finally, a three-phase modulation wave is obtained by inverse coordinate transformation using the phase reference value and compared with the carrier wave to generate a PWM signal. This achieves a complete control link for accurate voltage tracking, transient overcurrent protection, and modulation signal generation.

[0068] The embodiments of this application, through the above specific implementation methods, achieve the following technical effects: This application proposes a grid-based control method for suppressing grid-connected resonance in on-board chargers. It collects three-phase voltage at the point of common coupling (PCC) and three-phase current on the grid side, extracting actual active and reactive power through coordinate transformation and adaptive low-pass filtering. Based on power deviation, it simulates internal potential phase and voltage amplitude reference values ​​using frequency droop, voltage droop, and virtual inertia, providing the system with autonomous frequency and voltage regulation capabilities and inertial support. A virtual inductor is introduced to achieve power decoupling, and a virtual resistor enhances damping at the LCL resonant frequency. A high-pass filter extracts high-frequency resonant components, which are then used to generate a compensation voltage through transient virtual resistor gain, adaptively suppressing the resonance peak without generating additional output at the fundamental frequency. Finally, in the voltage and current dual closed-loop control, when the output current exceeds a current-limiting threshold determined based on the ratio of rated capacity to rated voltage, the current command is scaled proportionally and uniformly, generating a PWM signal to drive the inverter through inverse coordinate transformation and carrier comparison. This organic combination of techniques effectively solves the technical problems of easy instability in grid-connected control of on-board chargers under weak grid conditions, fixed LCL resonance suppression methods, and difficulty in simultaneously addressing power coupling.

[0069] Example 2, as shown in the appendix Figure 3 As shown, based on the inventive concept of the grid-connected resonance suppression method for on-board chargers provided in Embodiment 1, this application also provides a grid-connected control system for on-board charger grid-connected resonance suppression, specifically including: Data acquisition module 11 is used to acquire the three-phase voltage and grid-side three-phase current at the point of common coupling. After coordinate transformation and low-pass filtering, the actual active power and actual reactive power are extracted. The power control module 12 is used to generate the phase reference value of the internal potential and the voltage amplitude reference value based on the actual active power and the actual reactive power, using a power control outer loop that includes frequency and voltage droop characteristics and virtual inertia simulation. The virtual compensation module 13 is used to introduce virtual inductance and virtual resistance, perform d-axis and q-axis cross-decoupling compensation on the voltage amplitude reference value, generate corrected d-axis and q-axis voltage commands; construct a transient virtual resistance control loop based on a high-pass filter, extract the high-frequency resonant component in the grid-side current, and generate a transient compensation voltage signal; The modulation generation module 14 is used to generate a modulation wave signal through voltage and current dual closed-loop control, and when the output current exceeds the current limiting threshold, the current amplitude is limited by the current command limiting mechanism, and finally the modulation wave signal is used to generate a pulse width modulation signal.

[0070] In one embodiment, the data acquisition module 11 is further configured to: acquire the three-phase voltage and grid-side three-phase current of the common connection point using voltage and current sensors, wherein the common connection point is the connection point between the AC side of the on-board bidirectional charger and the low-voltage distribution network; transform the acquired three-phase voltage and grid-side three-phase current to a synchronous rotating coordinate system to obtain d-axis voltage components, q-axis voltage components, d-axis current components, and q-axis current components; calculate the original active power and original reactive power based on the d-axis voltage components, q-axis voltage components, d-axis current components, and q-axis current components; acquire the fundamental frequency of the three-phase voltage, determine the cutoff frequency of the low-pass filter based on the fundamental frequency, wherein the cutoff frequency is less than the fundamental frequency; and pass the original active power and original reactive power through the low-pass filter respectively to filter out high-frequency switching ripple components, and output the actual active power and actual reactive power.

[0071] Furthermore, the data acquisition module 11 is also used to: multiply the d-axis voltage component and the q-axis voltage component with the d-axis current component and the q-axis current component on the corresponding axis, respectively, to obtain the active power component of each axis, and add the active power components of each axis to obtain the original active power; multiply the d-axis voltage component by the q-axis current component to obtain the first reactive power component; multiply the q-axis voltage component by the d-axis current component to obtain the second reactive power component; and subtract the second reactive power component from the first reactive power component to obtain the original reactive power.

[0072] Furthermore, the data acquisition module 11 is also used to: acquire the rated fundamental frequency of the three-phase voltage and use the rated fundamental frequency as the fundamental frequency; acquire the switching frequency of the on-board bidirectional charger and determine an initial proportional coefficient based on the ratio of the switching frequency to the fundamental frequency, wherein the switching frequency is greater than the fundamental frequency; acquire the expected response time of the power control outer loop and correct the initial proportional coefficient based on the expected response time to obtain a preset proportional coefficient; the expected response time is the adjustment time set by the power control outer loop for load disturbances during the design phase, and its value is greater than the fundamental period of the three-phase voltage and greater than the switching period of the on-board bidirectional charger; multiply the fundamental frequency by the preset proportional coefficient to obtain the cutoff frequency of the low-pass filter, wherein the cutoff frequency is less than the fundamental frequency, and the response period corresponding to the cutoff frequency is less than the expected response time.

[0073] In one embodiment, the power control module 12 is further configured to: obtain a frequency correction amount by combining the difference between the actual active power and the active power reference command with a frequency droop coefficient, wherein the active power reference command is the target output active power value of the on-board bidirectional charger, and the frequency droop coefficient is a ratio determined based on the rated power of the on-board bidirectional charger and the allowable frequency deviation range; superimpose the frequency correction amount onto a reference angular frequency to obtain an initial angular frequency command value, wherein the reference angular frequency is the angular frequency value corresponding to the rated fundamental frequency of the three-phase voltage; and obtain an inertia response compensation amount by combining the difference between the initial angular frequency command value and the rated angular frequency of the three-phase voltage with a virtual inertia coefficient and processing it through a first-order inertial element, wherein the virtual inertia coefficient is used to obtain an inertia response compensation amount. The inertia coefficient is a ratio determined based on the rated capacity of the on-board bidirectional charger and the desired frequency change rate limit, and the virtual inertia coefficient is greater than zero; the inertia response compensation is superimposed on the initial angular frequency command value to output the angular frequency command value; the angular frequency command value is integrated to output the phase reference value of the internal potential; the difference between the actual reactive power and the reactive power reference command is multiplied by the voltage droop coefficient to obtain the voltage correction amount, wherein the voltage droop coefficient is a ratio determined based on the reactive power adjustment range and the allowable voltage deviation range of the on-board bidirectional charger; the voltage correction amount is superimposed on the reference voltage amplitude to output the voltage amplitude reference value of the internal potential, wherein the reference voltage amplitude is the rated voltage amplitude of the three-phase voltage.

[0074] In one embodiment, the virtual compensation module 13 is further configured to: introduce virtual inductance and virtual resistance, perform d-axis and q-axis cross-decoupling compensation on the voltage amplitude reference value, and generate corrected d-axis and q-axis voltage commands, including: the value of the virtual inductance is determined based on the ratio of the line equivalent resistance to the equivalent reactance at the common connection point, and after dimensional conversion in conjunction with the fundamental frequency and the rated voltage amplitude of the three-phase voltage; the value of the virtual resistance is determined based on the desired damping coefficient at the resonant frequency of the LCL filter of the on-board bidirectional charger, the desired damping coefficient being a preset value greater than zero and less than 1, and the larger the desired damping coefficient, the larger the value of the virtual resistance; use the voltage amplitude reference value as the initial voltage command for the d-axis, and set the initial voltage command for the q-axis to zero; determine the d-axis resistance compensation amount based on the virtual resistance and the d-axis current component, wherein the d-axis resistance compensation amount and the virtual resistance are related. The simulated resistance and the d-axis current component are both proportional. Based on the virtual inductance and the q-axis current component, a d-axis cross-compensation amount is determined, wherein the d-axis cross-compensation amount is proportional to both the virtual inductance and the q-axis current component. The d-axis resistance compensation amount is superimposed on the initial d-axis voltage command, and then the d-axis cross-compensation amount is subtracted to generate a corrected d-axis voltage command. Based on the virtual resistance and the q-axis current component, a q-axis resistance compensation amount is determined, wherein the q-axis resistance compensation amount is proportional to both the virtual resistance and the q-axis current component. Based on the virtual inductance and the d-axis current component, a q-axis cross-compensation amount is determined, wherein the q-axis cross-compensation amount is proportional to both the virtual inductance and the d-axis current component. The q-axis resistance compensation amount and the q-axis cross-compensation amount are superimposed on the initial q-axis voltage command to generate a corrected q-axis voltage command.

[0075] Furthermore, the virtual compensation module 13 is also used to: construct a transient virtual resistance control loop based on a high-pass filter, extract high-frequency resonant components from the grid-side current, and generate a transient compensation voltage signal, including: inputting the grid-side three-phase current into a high-pass filter to extract high-frequency resonant components, wherein the cutoff frequency of the high-pass filter is greater than the fundamental frequency of the three-phase voltage and less than the resonant frequency of the LCL filter of the on-board bidirectional charger; determining the transient virtual resistance gain value according to the desired damping ratio at the resonant frequency of the LCL filter, wherein the transient virtual resistance gain value is proportional to the desired damping ratio; multiplying the high-frequency resonant components by the transient virtual resistance gain value to obtain a transient compensation voltage signal; superimposing the transient compensation voltage signal onto the corrected d-axis voltage command and the corrected q-axis voltage command to generate updated d-axis voltage commands and updated q-axis voltage commands; wherein the gain of the high-pass filter at the fundamental frequency of the three-phase voltage tends to zero, so that the transient virtual resistance control loop does not generate compensation voltage in steady state.

[0076] In one embodiment, the modulation generation module 14 is further configured to: output a d-axis initial current command by performing a proportional-integral operation on the difference between the updated d-axis voltage command and the d-axis voltage component; output a q-axis initial current command by performing a proportional-integral operation on the difference between the updated q-axis voltage command and the q-axis voltage component; calculate an initial current amplitude based on the d-axis initial current command and the q-axis initial current command; when the initial current amplitude exceeds a current limiting threshold, scale the d-axis initial current command and the q-axis initial current command according to the ratio of the current limiting threshold to the initial current amplitude to obtain a limited d-axis current reference command and a q-axis current reference command, wherein the current limiting threshold is based on the rated capacity of the on-board bidirectional charger. The ratio to the rated voltage at the common connection point is determined; when the initial current amplitude does not exceed the current limiting threshold, the d-axis initial current command and the q-axis initial current command are directly used as the d-axis current reference command and the q-axis current reference command; the difference between the d-axis current reference command and the d-axis current component is processed by proportional-integral operation to output the d-axis modulation voltage; the difference between the q-axis current reference command and the q-axis current component is processed by proportional-integral operation to output the q-axis modulation voltage; the d-axis modulation voltage and the q-axis modulation voltage are transformed from a synchronous rotating coordinate system to a stationary coordinate system to obtain a three-phase modulation wave signal; the three-phase modulation wave signal is compared with the carrier signal to generate the pulse width modulation signal.

[0077] Furthermore, the modulation generation module 14 is also used to: acquire the phase reference value of the internal potential, and use the phase reference value as the transformation angle for the transformation from the synchronous rotating coordinate system to the stationary coordinate system; perform inverse coordinate transformation on the d-axis modulation voltage and the q-axis modulation voltage with the transformation angle as the reference to obtain a three-phase modulation wave signal in the stationary coordinate system; acquire a carrier signal, wherein the frequency of the carrier signal is greater than the fundamental frequency; compare the instantaneous value of each phase modulation wave in the three-phase modulation wave signal with the carrier signal, and output a high level when the instantaneous value of the modulation wave is greater than or equal to the instantaneous value of the carrier signal, and output a low level when the instantaneous value of the modulation wave is less than the instantaneous value of the carrier signal, thereby generating the pulse width modulation signal.

[0078] The grid-connected control system for suppressing grid resonance in on-board chargers provided in this application enables stable grid-connected operation of on-board chargers under low-voltage power distribution networks in scenarios such as V2G grid connection of electric vehicles and bidirectional charging and discharging of charging piles. This system can be integrated into the on-board charger control unit or the charging pile power controller. It autonomously establishes a voltage frequency reference through droop characteristics and virtual inertia simulation, uses virtual impedance decoupling and high-pass filter-type transient virtual resistance to collaboratively suppress LCL resonance peaks, and embeds transient current limiting protection in the dual closed-loop control, effectively improving grid connection stability under weak grid conditions and reducing the risk of resonance instability caused by grid impedance fluctuations. For specific control methods and implementation details, please refer to Embodiment 1.

[0079] It should be noted that the order of the embodiments in this application is merely for descriptive purposes and does not represent the superiority or inferiority of the embodiments. Furthermore, the above description focuses on specific embodiments of this specification. Additionally, the processes depicted in the accompanying drawings do not necessarily require a specific or sequential order to achieve the desired results. In some embodiments, multitasking and parallel processing are possible or may be advantageous.

Claims

1. A grid-type control method for suppressing grid-connected resonance in on-board chargers, characterized in that, The method includes: The three-phase voltage at the point of common coupling and the three-phase current on the grid side are collected. After coordinate transformation and low-pass filtering, the actual active power and actual reactive power are extracted. Based on the actual active power and actual reactive power, a power control outer loop that includes frequency and voltage droop characteristics and virtual inertia simulation is used to generate the phase reference value of the internal potential and the voltage amplitude reference value. By introducing virtual inductors and virtual resistors, d-axis and q-axis cross-decoupling compensation is performed on the voltage amplitude reference value to generate corrected d-axis and q-axis voltage commands. A transient virtual resistance control loop based on a high-pass filter is constructed to extract the high-frequency resonant component in the grid-side current and generate a transient compensation voltage signal. A modulation wave signal is generated through voltage and current dual closed-loop control. When the output current exceeds the current limiting threshold, the current amplitude is limited by a current command limiting mechanism. Finally, a pulse width modulation signal is generated from the modulation wave signal.

2. The grid-type control method for grid-connected resonance suppression of on-board chargers according to claim 1, characterized in that, The three-phase voltage at the point of common coupling and the three-phase current on the grid side are collected. After coordinate transformation and low-pass filtering, the actual active power and actual reactive power are extracted, including: The three-phase voltage and grid-side three-phase current at the common connection point are collected using voltage and current sensors. The common connection point is the connection point between the AC side of the on-board bidirectional charger and the low-voltage distribution network. The collected three-phase voltages and grid-side three-phase currents are transformed into a synchronous rotating coordinate system to obtain the d-axis voltage component, q-axis voltage component, d-axis current component, and q-axis current component. The original active power and original reactive power are calculated based on the d-axis voltage component, the q-axis voltage component, the d-axis current component, and the q-axis current component. Obtain the fundamental frequency of the three-phase voltage, and determine the cutoff frequency of the low-pass filter based on the fundamental frequency, wherein the cutoff frequency is less than the fundamental frequency; The original active power and the original reactive power are respectively passed through the low-pass filter to filter out the high-frequency switching ripple component, and then the actual active power and the actual reactive power are output.

3. The grid-type control method for suppressing grid-connected resonance of on-board chargers according to claim 2, characterized in that, Based on the d-axis voltage component, the q-axis voltage component, the d-axis current component, and the q-axis current component, the original active power and the original reactive power are calculated, including: The d-axis voltage component and the q-axis voltage component are multiplied by the d-axis current component and the q-axis current component on the corresponding axis, respectively, to obtain the active power component of each axis. The active power components of each axis are then added together to obtain the original active power. Multiply the d-axis voltage component by the q-axis current component to obtain the first reactive power component; Multiply the q-axis voltage component by the d-axis current component to obtain the second reactive power component; The original reactive power is obtained by subtracting the second reactive power component from the first reactive power component.

4. The grid-type control method for suppressing grid-connected resonance of on-board chargers according to claim 2, characterized in that, Obtaining the fundamental frequency of the three-phase voltage and determining the cutoff frequency of the low-pass filter based on the fundamental frequency includes: Obtain the rated fundamental frequency of the three-phase voltage and use the rated fundamental frequency as the fundamental frequency; The switching frequency of the on-board bidirectional charger is obtained, and an initial proportional coefficient is determined based on the ratio of the switching frequency to the fundamental frequency, wherein the switching frequency is greater than the fundamental frequency. Obtain the expected response time of the power control outer loop, and correct the initial proportional coefficient based on the expected response time to obtain the preset proportional coefficient; The expected response time is the adjustment time set by the power control outer loop in the design phase for load disturbances, and its value is greater than the fundamental period of the three-phase voltage and greater than the switching period of the on-board bidirectional charger. The cutoff frequency of the low-pass filter is obtained by multiplying the fundamental frequency by the preset scaling factor, wherein the cutoff frequency is less than the fundamental frequency and the response period corresponding to the cutoff frequency is less than the expected response time.

5. The grid-type control method for grid-connected resonance suppression of on-board chargers according to claim 1, characterized in that, Based on the actual active power and actual reactive power, a power control outer loop incorporating frequency and voltage droop characteristics and virtual inertia simulation is used to generate phase reference values ​​for the internal potential and voltage amplitude reference values, including: The difference between the actual active power and the active power reference command is combined with the frequency droop coefficient to obtain the frequency correction amount, wherein the active power reference command is the target output active power value of the on-board bidirectional charger, and the frequency droop coefficient is the ratio determined based on the rated power of the on-board bidirectional charger and the allowable frequency deviation range. The frequency correction is superimposed on the reference angular frequency to obtain the initial angular frequency command value, wherein the reference angular frequency is the angular frequency value corresponding to the rated fundamental frequency of the three-phase voltage. The difference between the initial angular frequency command value and the rated angular frequency of the three-phase voltage is combined with the virtual inertia coefficient and processed by a first-order inertial element to obtain the inertia response compensation amount. The virtual inertia coefficient is a ratio determined based on the rated capacity of the on-board bidirectional charger and the expected frequency change rate limit, and the virtual inertia coefficient is greater than zero. The inertia response compensation is then added to the initial angular frequency command value, and the angular frequency command value is output. The angular frequency command value is integrated to output a phase reference value of the internal potential; The voltage correction amount is obtained by multiplying the difference between the actual reactive power and the reactive power reference command by the voltage droop coefficient, wherein the voltage droop coefficient is a ratio determined based on the reactive power adjustment range and the allowable voltage deviation range of the on-board bidirectional charger. The voltage correction is superimposed on the reference voltage amplitude to output a reference value for the voltage amplitude of the internal potential, wherein the reference voltage amplitude is the rated voltage amplitude of the three-phase voltage.

6. The grid-type control method for grid-connected resonance suppression of on-board chargers according to claim 1, characterized in that, By introducing virtual inductors and virtual resistors, d-axis and q-axis cross-decoupling compensation is performed on the voltage amplitude reference value to generate corrected d-axis and q-axis voltage commands, including: The value of the virtual inductance is determined by the ratio of the line equivalent resistance to the equivalent reactance at the common connection point, and by dimensional conversion of the fundamental frequency and the rated voltage amplitude of the three-phase voltage. The value of the virtual resistor is determined based on the desired damping coefficient at the resonant frequency of the LCL filter of the on-board bidirectional charger. The desired damping coefficient is a preset value that is greater than zero and less than 1. The larger the desired damping coefficient, the larger the value of the virtual resistor. Use the voltage amplitude reference value as the initial voltage command for the d-axis and set the initial voltage command for the q-axis to zero; Based on the virtual resistance and the d-axis current component, the d-axis resistance compensation amount is determined, wherein the d-axis resistance compensation amount is proportional to both the virtual resistance and the d-axis current component; Based on the virtual inductance and the q-axis current component, the d-axis cross compensation amount is determined, wherein the d-axis cross compensation amount is proportional to both the virtual inductance and the q-axis current component; The d-axis resistance compensation is superimposed on the d-axis initial voltage command, and then the d-axis cross compensation is subtracted to generate the corrected d-axis voltage command. The q-axis resistance compensation amount is determined based on the virtual resistance and the q-axis current component, wherein the q-axis resistance compensation amount is proportional to both the virtual resistance and the q-axis current component. Based on the virtual inductance and the d-axis current component, the q-axis cross compensation amount is determined, wherein the q-axis cross compensation amount is proportional to both the virtual inductance and the d-axis current component; The q-axis resistance compensation and the q-axis cross compensation are superimposed on the q-axis initial voltage command to generate the corrected q-axis voltage command.

7. The grid-type control method for grid-connected resonance suppression of on-board chargers according to claim 1, characterized in that, A transient virtual resistance control loop based on a high-pass filter is constructed to extract the high-frequency resonant component from the grid-side current and generate a transient compensation voltage signal, including: The three-phase current on the grid side is input into a high-pass filter to extract the high-frequency resonant component. The cutoff frequency of the high-pass filter is greater than the fundamental frequency of the three-phase voltage and less than the resonant frequency of the LCL filter of the on-board bidirectional charger. The transient virtual resistance gain value is determined based on the desired damping ratio at the resonant frequency of the LCL filter, wherein the transient virtual resistance gain value is proportional to the desired damping ratio; Multiply the high-frequency resonant component by the transient virtual resistance gain value to obtain the transient compensation voltage signal; The transient compensation voltage signal is superimposed on the corrected d-axis voltage command and the corrected q-axis voltage command to generate the updated d-axis voltage command and the updated q-axis voltage command; The high-pass filter has a gain of zero at the fundamental frequency of the three-phase voltage, so that the transient virtual resistance control loop does not generate a compensation voltage in steady state.

8. The grid-type control method for suppressing grid-connected resonance of on-board chargers according to claim 1, characterized in that, A modulated wave signal is generated through voltage and current dual closed-loop control. When the output current exceeds the current limiting threshold, the current amplitude is limited by a current command limiting mechanism. Finally, a pulse width modulation signal is generated from the modulated wave signal, including: The difference between the updated d-axis voltage command and the d-axis voltage component is used for proportional-integral calculation to output the initial d-axis current command. The difference between the updated q-axis voltage command and the q-axis voltage component is used for proportional-integral calculation to output the initial q-axis current command. Calculate the initial current amplitude based on the d-axis initial current command and the q-axis initial current command; When the initial current amplitude exceeds the current limiting threshold, the d-axis initial current command and the q-axis initial current command are scaled according to the ratio of the current limiting threshold to the initial current amplitude to obtain the limited d-axis current reference command and the q-axis current reference command. The current limiting threshold is determined based on the ratio of the rated capacity of the on-board bidirectional charger to the rated voltage at the common connection point. When the initial current amplitude does not exceed the current limiting threshold, the d-axis initial current command and the q-axis initial current command are directly used as the d-axis current reference command and the q-axis current reference command. The difference between the d-axis current reference command and the d-axis current component is processed by proportional-integral operation to output the d-axis modulated voltage. The difference between the q-axis current reference command and the q-axis current component is processed by proportional-integral operation to output the q-axis modulation voltage. The d-axis modulation voltage and the q-axis modulation voltage are transformed from a synchronous rotating coordinate system to a stationary coordinate system to obtain a three-phase modulation wave signal. The three-phase modulation wave signal is compared with the carrier signal to generate the pulse width modulation signal.

9. The grid-type control method for suppressing grid-connected resonance of on-board chargers according to claim 8, characterized in that, The d-axis modulation voltage and the q-axis modulation voltage are transformed from a synchronously rotating coordinate system to a stationary coordinate system to obtain a three-phase modulation wave signal. The three-phase modulation wave signal is compared with the carrier signal to generate the pulse width modulation signal, including: Obtain the phase reference value of the internal potential, and use the phase reference value as the transformation angle for the transformation from the synchronous rotating coordinate system to the stationary coordinate system; The d-axis modulation voltage and the q-axis modulation voltage are inversely transformed with the transformation angle as the reference to obtain a three-phase modulation wave signal in the stationary coordinate system. Acquire a carrier signal, wherein the frequency of the carrier signal is greater than the fundamental frequency; The instantaneous value of each phase of the three-phase modulated wave signal is compared with the carrier signal. When the instantaneous value of the modulated wave is greater than or equal to the instantaneous value of the carrier signal, a high level is output, and when the instantaneous value of the modulated wave is less than the instantaneous value of the carrier signal, a low level is output, thereby generating the pulse width modulation signal.

10. A grid-type control system for suppressing grid-connected resonance in on-board chargers, characterized in that, The system is used to execute the grid-type control method for grid-connected resonance suppression of on-board chargers as described in any one of claims 1 to 9, the system comprising: The data acquisition module is used to collect the three-phase voltage and grid-side three-phase current at the point of common coupling. After coordinate transformation and low-pass filtering, the actual active power and actual reactive power are extracted. The power control module is used to generate the phase reference value of the internal potential and the voltage amplitude reference value based on the actual active power and the actual reactive power, using a power control outer loop that includes frequency and voltage droop characteristics and virtual inertia simulation. The virtual compensation module is used to introduce virtual inductance and virtual resistance, perform d-axis and q-axis cross-decoupling compensation on the voltage amplitude reference value, and generate corrected d-axis and q-axis voltage commands; construct a transient virtual resistance control loop based on a high-pass filter, extract the high-frequency resonant component in the grid-side current, and generate a transient compensation voltage signal; The modulation generation module is used to generate a modulation wave signal through voltage and current dual closed-loop control, and when the output current exceeds the current limiting threshold, the current amplitude is limited by the current command limiting mechanism, and finally the modulation wave signal is used to generate a pulse width modulation signal.

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