Wireless charging transmission circuit and control method thereof, portable charging device

Abstract

The application provides a wireless charging transmitting circuit and a control method thereof and a portable charging device, wherein the wireless charging transmitting circuit comprises a wireless charging transmitting module and a resonance circuit; the wireless charging transmitting module comprises a full-bridge inverter, a pulse width modulation circuit and a controller; the resonance circuit and the pulse width modulation circuit are connected with the controller; the controller determines a zero-crossing point of a resonance current based on a change condition of the resonance current; in response to the zero-crossing point of the resonance current, a dead time length is adjusted based on a time difference between a first time corresponding to the zero-crossing point and a second time at which a next group of switch tubes is turned on; the dead time is a period in which all the switch tubes in the full-bridge inverter are in an off state; the pulse width modulation circuit is connected with the plurality of switch tubes; the pulse width modulation circuit adjusts a corresponding pulse width modulation signal based on the adjusted dead time length; and the pulse width modulation signal is used to drive the switch tubes.

Description

Technical Field

[0001] This application relates to the field of wireless charging technology, and in particular to a wireless charging transmitting circuit and its control method, and a portable charging device. Background Technology

[0002] Wireless charging technology has been widely applied in charging scenarios for consumer electronics products. The mainstream wireless charging standard uses magnetic induction coupling. The transmitter converts direct current (DC) into high-frequency alternating current (AC) via an inverter, driving the transmitting coil to generate an alternating magnetic field. The receiver then harvests energy through electromagnetic induction. In the inverter, the switching transistors incur switching losses during high-frequency switching. With the increasing power levels and switching frequencies of wireless charging, these switching losses have become a bottleneck limiting system efficiency and thermal design.

[0003] In some existing wireless charging transmitter solutions, a hard-switching method is used, which forces the switch to turn on before the drain-source voltage of the switching transistor has dropped to zero. This causes voltage and current to overlap during the switching transition, resulting in significant switching losses and electromagnetic interference (EMI). To address this issue, some wireless charging transmitter circuits employ zero-voltage switching (ZVS) schemes. However, these typically use a full-bridge drive with a fixed dead time, which limits the accuracy of ZVS state determination. It is difficult to accurately determine the turn-on time of the switching transistor to precisely achieve the ZVS state, resulting in the inability to achieve reliable ZVS operation across the entire operating range. Summary of the Invention

[0004] To address the existing technical problems, this application provides a wireless charging transmitter circuit and its control method that can improve ZVS control accuracy, as well as a portable charging device.

[0005] In a first aspect, embodiments of this application provide a wireless charging transmitting circuit, including: a wireless charging transmitting module and a resonant circuit; the wireless charging transmitting module includes a full-bridge inverter, a pulse width modulation circuit, and a controller; the full-bridge inverter includes multiple switching transistors; the multiple switching transistors are connected to the resonant circuit, and the resonant circuit generates a resonant current based on the signal output by the switching transistors, the resonant current being used to provide wireless charging power.

[0006] Both the resonant circuit and the pulse width modulation circuit are connected to the controller. The controller determines the zero-crossing point of the resonant current based on the changes in the resonant current. In response to the zero-crossing point of the resonant current, the controller adjusts the dead time based on the time difference between the first moment corresponding to the zero-crossing point and the second moment when the next set of switching transistors is turned on. The dead time is the period during which all the switching transistors in the full-bridge inverter are in the off state. The pulse width modulation circuit is connected to the multiple switching transistors. The pulse width modulation circuit adjusts the corresponding pulse width modulation signal based on the adjusted dead time. The pulse width modulation signal is used to drive the switching transistors.

[0007] Secondly, embodiments of this application provide a wireless charging transmitter circuit control method, applied to the wireless charging transmitter circuit described in any embodiment of the first aspect of this application. The wireless charging transmitter circuit control method includes: determining the zero-crossing point of the resonant current based on the change of the resonant current; responding to the zero-crossing point of the resonant current, adjusting the dead time based on the time difference between a first moment corresponding to the zero-crossing point and a second moment when the next set of switching transistors is turned on; the dead time is the period during which all the switching transistors in the full-bridge inverter are in the off state; and adjusting the corresponding pulse width modulation signal based on the adjusted dead time.

[0008] Thirdly, embodiments of this application provide a portable charging device, including the wireless charging transmitting circuit described in any embodiment of the first aspect of this application.

[0009] In the wireless charging transmitting circuit provided in the above embodiment, both the resonant circuit and the pulse width modulation circuit are connected to the controller. The controller determines the zero-crossing point of the resonant current based on the changes in the resonant current. In response to the zero-crossing point of the resonant current, the dead time is adjusted based on the time difference between the first moment corresponding to the zero-crossing point and the second moment when the next set of switching transistors is turned on. The dead time is the period during which all the switching transistors in the full-bridge inverter are in the off state. The pulse width modulation circuit is connected to the plurality of switching transistors, and the pulse width modulation circuit adjusts the corresponding pulse width modulation signal based on the adjusted dead time. The pulse width modulation signal is used to drive the switching transistors. Thus, within the dead time, because the resonant current of the resonant circuit cannot change abruptly and remains continuously flowing, it discharges the drain-source parasitic capacitance of the next set of switches that is about to be turned on. When the discharge is complete, that is, when the resonant current reaches zero, the drain-source voltage Vds of the next set of switches drops to 0. Therefore, by adjusting the dead time based on the above time difference, the dead time can be dynamically adjusted according to the time between the zero point of the resonant current and the turn-on time of the next set of switches, thereby ensuring that the turn-on time of the next set of switches is always aligned with the moment Vds=0, accurately achieving ZVS state operation. In this way, adaptive dead time control is achieved through real-time resonant current detection, ensuring that the full-bridge inverter can maintain ZVS state under all operating conditions such as input voltage changes, load changes, and temperature changes, significantly reducing switching losses and improving system efficiency.

[0010] The wireless charging transmitter circuit control method and portable charging device provided in the above embodiments belong to the same concept as the corresponding wireless charging transmitter circuit embodiments, and thus have the same technical effects as the corresponding wireless charging transmitter circuit embodiments, which will not be repeated here. Attached Figure Description

[0011] Figure 1 This is a schematic diagram of the structure of a wireless charging transmitter circuit provided in an embodiment of this application; Figure 2 This is a schematic diagram of the structure of a wireless charging transmitter circuit provided in another embodiment of this application; Figure 3 This is a flowchart illustrating a wireless charging transmitter circuit control method provided in another embodiment of this application.

[0012] Explanation of reference numerals in the attached figures 1. Wireless charging transmitter module; 2. Resonant circuit; 11. Full-bridge inverter; 12. Pulse width modulation circuit; 13. Controller; 14. Sampling resistor; 21. Transmitting coil; 22. Resonant capacitor. Detailed Implementation

[0013] The technical solution of this application will be further described in detail below with reference to the accompanying drawings and specific embodiments.

[0014] To make the objectives, technical solutions, and advantages of this application clearer, the following detailed description of this application will be provided in conjunction with the accompanying drawings. The described embodiments should not be considered as limitations on this application. All other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of this application.

[0015] In the following description, the phrase "some embodiments" refers to a subset of all possible embodiments. It should be noted that "some embodiments" can be the same subset or different subsets of all possible embodiments, and can be combined with each other without conflict.

[0016] In the following description, the terms "first," "second," and "third" are used merely to distinguish similar objects and do not represent a specific ordering of the objects. It is understood that "first," "second," and "third" can be interchanged in a specific order or sequence where permitted, so that the embodiments of this application described herein can be implemented in an order other than that illustrated or described herein. It should be noted that, unless otherwise specified, the embodiments and features in the embodiments of this application can be combined with each other.

[0017] In some embodiments, the switching transistor (e.g., a MOSFET) generates switching losses during high-frequency switching. The switching loss Psw can be approximated as: Psw = 1 / 2 × Vds × Id × (t_on + t_off) × f_sw, where Vds is the drain-source voltage of the switching transistor, Id is the switching current, t_on and t_off are the turn-on and turn-off transition times, respectively, and f_sw is the switching frequency. With the increase in wireless charging power levels and switching frequencies, switching losses have become a bottleneck limiting system efficiency and thermal design.

[0018] In some embodiments, although a soft-switching design is adopted, its dead time is usually a fixed value or determined by offline calibration, which cannot adapt to actual operating conditions such as input voltage fluctuations, changes in coil coupling state and load power changes. As a result, the ZVS condition cannot be met at some operating points, and the system efficiency deviates from the optimal value.

[0019] In some embodiments, the wireless charging transmitter circuit typically adopts a full-bridge drive mode with a fixed dead time, without real-time detection and adaptive dead-time adjustment of the resonant current; or, it relies only on current peak information and does not utilize the phase relationship between the current zero-crossing point and the voltage zero-crossing point, resulting in limited accuracy of ZVS state judgment.

[0020] In a resonant circuit, the dead time refers to the brief interval during which both the upper and lower bridge arm switches are turned off. This period is used to prevent shoot-through short circuits and to allow the junction capacitance to charge and discharge, thus achieving zero-voltage switching (ZVS). The purpose of this is to avoid a DC bus short circuit caused by simultaneous conduction of the upper and lower switches on the same bridge arm, and to create conditions for ZVS: the resonant current (mainly the magnetizing current) within the dead time charges and discharges the output capacitance (Coss) of the switch, causing its Vds to drop to near zero before turn-on.

[0021] In some embodiments, if the dead time is too short, the switch will be forcibly turned on before Vds drops to 0 due to the incomplete discharge of Coss, which is a hard switch and will generate a switching loss spike, leading to a sharp increase in losses or even tube failure. If the dead time is too long, the resonant current will flow in reverse after the discharge of Coss, Coss will be recharged, and Vds will rise again. Turning on the switch after the voltage rises will lead to increased reverse recovery losses, output waveform distortion, and decreased efficiency.

[0022] Based on this, embodiments of this application provide a wireless charging transmitting circuit, such as... Figure 1 As shown, it may include: a wireless charging transmitter module 1 and a resonant circuit 2; the wireless charging transmitter module 1 includes a full-bridge inverter 11, a pulse width modulation circuit 12 and a controller 13.

[0023] The full-bridge inverter 11 may include multiple switching transistors; the multiple switching transistors are connected to the resonant circuit 2, and the resonant circuit 2 generates a resonant current based on the signal output by the switching transistors, and the resonant current is used to provide wireless charging power.

[0024] The resonant circuit 2 and the pulse width modulation circuit 12 are both connected to the controller 13. The controller 13 determines the zero-crossing point of the resonant current based on the changes in the resonant current. In response to the zero-crossing point of the resonant current, the controller adjusts the dead time based on the time difference between the first moment corresponding to the zero-crossing point and the second moment when the next set of switching transistors is turned on. The dead time is the period during which all the switching transistors in the full-bridge inverter 11 are in the off state. The pulse width modulation circuit 12 is connected to the plurality of switching transistors. The pulse width modulation circuit 12 adjusts the corresponding pulse width modulation signal based on the adjusted dead time. The pulse width modulation signal is used to drive the switching transistors.

[0025] In one embodiment, the wireless charging transmitter module 1 can be a wireless charging transmitter chip, such as a system-on-a-chip (SoC). The wireless charging transmitter chip can integrate a full-bridge inverter 11, a pulse width modulation circuit 12, and a controller 13. For example, the controller 13 can be a microcontroller (MCU), such as a 32-bit microcontroller 13. The pulse width modulation (PWM) circuit can be an enhanced pulse width modulation (EPWM) module, for example, the clock frequency of the PWM signal output by the PWM circuit can be 288 MHz.

[0026] For example, the wireless charging transmitter chip can also integrate memory, current sensing amplifier, communication demodulator, linear regulator, and protection circuitry. For instance, the wireless charging transmitter chip can be packaged as a compact FCQFN24 (3mm × 4mm × 0.55mm).

[0027] In one embodiment, in the wireless charging transmitter module 1, the full-bridge inverter 11, the pulse width modulation circuit 12, and the controller 13 can also be independent devices.

[0028] In one embodiment, the full-bridge inverter 11 includes multiple switching transistors, such as MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors). For example, the full-bridge inverter 11 includes four switching transistors, which are paired up to form upper and lower bridge arms. The upper and lower bridge arms are respectively connected to the resonant circuit 2 to form a resonant current flow loop.

[0029] In one embodiment, the wireless charging transmitter module 1 is connected to an external power signal via a power voltage input pin. The power signal is input to the full-bridge inverter 11 via the power voltage input pin and output to the resonant circuit 2 through the switched transistor in the inverter to generate a resonant current.

[0030] In one embodiment, the resonant circuit 2 may include a transmitting coil 21, such as a transmitting coil 21 and a resonant capacitor 22 connected in series with the transmitting coil 21. The transmitting coil 21 is an inductor that radiates energy outward based on the resonant current generated by the signal output from the switching transistor, in order to charge the electrical load.

[0031] In one embodiment, the plurality of switching transistors includes a first switching transistor group and a second switching transistor group; the full-bridge inverter 11 further includes a first switching node and a second switching node, wherein adjacent switching transistors in the first switching transistor group are connected to form a first switching node, and adjacent switching transistors in the second switching transistor group are connected to form a second switching node. For example, the first switching node is between two switching transistors in the first switching transistor group, connected to a first switching pin SW1; the second switching node is between two switching transistors in the second switching transistor group, connected to a second switching pin SW2.

[0032] like Figure 1 As shown, the resonant circuit 2 may include a transmitting coil 21 and a resonant capacitor 22, which are connected in series between the first switching node and the second switching node.

[0033] For example, the wireless charging transmitter module 1 may include a first switch pin SW1 and a second switch pin SW2. The first switch pin is connected to a first switch node, and the second switch pin is connected to a second switch node. The transmitter coil 21 and the resonant capacitor 22 are connected in series between the first switch pin and the second switch pin.

[0034] In one embodiment, the full-bridge inverter 11 may include four switching transistors: Q1, Q2, Q3, and Q4. For example, Q1 and Q4 form a first switching transistor group, and Q2 and Q3 form a second switching transistor group. In each switching cycle of the full-bridge inverter 11, there are two dead zones: a first dead zone (from when Q1 and Q4 are turned off until when Q2 and Q3 are turned on) and a second dead zone (from when Q2 and Q3 are turned off until when Q1 and Q4 are turned on).

[0035] For example, the ZVS process, taking the first dead time period as an example, includes the following stages: (a) Before the dead zone begins, Q1 and Q4 are turned on, and the resonant current i_L flows in the forward direction along the path Q1→transmitting coil 21→resonant capacitor 22→Q4. The voltage at node SW1 is approximately equal to the power signal input voltage V_PVIN, and the voltage at node SW2 is approximately equal to the power ground PGND (0V). (b) After Q1 and Q4 are turned off, due to the inductive characteristics of the transmitting coil 21, the resonant current i_L cannot change abruptly and continues to flow in its original direction. This resonant current discharges the equivalent capacitance of node SW1 (including the Coss of Q1 and the Coss of Q2, for example, about 200pF each) and charges the equivalent capacitance of node SW2 (including the Coss of Q3 and the Coss of Q4). (c) When the voltage at node SW1 drops from the V_PVIN resonance to 0V (corresponding to Vds=0V for Q2), the body diode of Q2 is forward-biased, clamping SW1 to the PGND level within the range of one diode voltage drop; similarly, when the voltage at node SW2 rises to the V_PVIN resonance, Vds=0V for Q3. (d) During the conduction period of the body diode, if the gate drive signals of Q2 and Q3 are triggered, Q2 and Q3 will turn on under the condition that their respective Vds≈0V, realizing zero-voltage switching. The conduction time of the body diode is the delay window between the resonant voltage crossing zero and the gate turn-on, and the length of this window is the optimal dead time.

[0036] The condition for achieving ZVS is that, within the dead zone, the resonant current i_L must have sufficient energy to complete the charging and discharging of the parasitic capacitance of the switching transistor (i.e., the output capacitance Coss). This condition can be expressed as: (1 / 2) × L2 × i_L² ≥ (1 / 2) × (n × Coss) × V_PVIN².

[0037] Where n is the equivalent capacitance that each switching node needs to charge and discharge (for example, n=2 in the above symmetrical full bridge). The above formula shows that the energy stored in the resonant inductor, i.e., the transmitting coil 21, must be greater than or equal to the total energy required to charge the parasitic capacitance of n switching transistors from 0 to the power supply voltage.

[0038] In one embodiment, the resonant frequency of the resonant circuit 2 can be expressed as: Where L2 is the inductance value of the transmitting coil 21, Here is the capacitance value of the resonant capacitor 22. For example, the inductance L2 of the transmitting coil 21 can be 24 μH, the capacitance C_res of the resonant capacitor 22 can be 68 nF, and the resonant frequency is designed to be approximately 125 kHz. The selection of the resonant capacitor 22 needs to comprehensively consider its voltage rating, temperature coefficient, and Q value.

[0039] In one embodiment, the resonant capacitor 22 may be composed of one or more capacitors connected in series and / or in parallel.

[0040] In one embodiment, the controller 13 is further configured to detect the rate of change of the power signal input voltage and the resonant current of the wireless charging transmitter module 1, and to stop adjusting the dead time in response to the rate of change of the power signal input voltage and / or the resonant current reaching a preset threshold.

[0041] In one embodiment, the controller 13 is further configured to detect the current temperature of the wireless charging transmitter circuit and the peak value of the resonant current, and to stop adjusting the dead time in response to the current temperature reaching a preset upper temperature limit and / or the peak value of the resonant current reaching a preset upper current limit. For example, a temperature detection circuit connected to the controller 13 may also be included, which is used to detect the current temperature of the wireless charging transmitter circuit. Exemplarily, the current temperature of the wireless charging transmitter circuit could be the current temperature of the wireless charging transmitter module 1, the current temperature of the transmitter coil 21, etc.

[0042] In one embodiment, the controller 13 determines the zero-crossing point of the resonant current based on changes in the resonant current and adjusts the dead time. This can be performed once every N PWM clock cycles, where N is a positive integer, such as 1, 2, 4, etc. That is, the controller 13 determines the zero-crossing point of the resonant current and adjusts the dead time every unit number (N) of PWM clock cycles based on changes in the resonant current. Determining the zero-crossing point of the resonant current means determining whether the resonant current value becomes 0. Turning on the next group of switches refers to turning on the next group of switches, for example, turning on the second group of switches or the first group of switches.

[0043] Thus, during the dead time, since the resonant current of resonant circuit 2 cannot change abruptly and remains continuously flowing, it discharges the drain-source parasitic capacitance of the next set of switches that is about to be turned on. When the discharge is complete, that is, when the resonant current reaches zero, the drain-source voltage Vds of the next set of switches drops to 0. Therefore, by adjusting the dead time based on the above time difference, the dead time can be dynamically adjusted according to the time between the zero point of the resonant current and the turn-on time of the next set of switches, thereby ensuring that the turn-on time of the next set of switches is always aligned with the moment Vds=0, accurately achieving ZVS state operation. In this way, adaptive dead time control is achieved through real-time resonant current detection, ensuring that the full-bridge inverter 11 can maintain ZVS state under all operating conditions such as input voltage changes, load changes, and temperature changes, significantly reducing switching losses and improving system efficiency.

[0044] In some embodiments, the wireless charging transmitter module 11 may further include: a power voltage input pin and a current detection pin; the power voltage input pin is connected to and grounded by the plurality of switching transistors, and the power voltage input pin provides a power signal for generating the resonant current.

[0045] For example, such as Figure 2As shown, the power voltage input pins may include a first power voltage input pin PVIN1 and a second power voltage input pin PVIN2 for inputting external power signals. For example, PVIN1 and PVIN2 can also be bypassed to power ground PGND through 22μF ceramic capacitors, providing a low-impedance power path for the full-bridge inverter 11.

[0046] For example, the current detection pin may include a first current detection pin and a second current detection pin. For instance, the first current detection pin is a positive input pin SNSP, and the second current detection pin is a negative input pin SNSN. The first and second current detection pins can be connected across the sampling resistor 14 to collect the differential voltage across the sampling resistor 14 and transmit it to the controller 13 for real-time extraction of the amplitude and phase information of the resonant current.

[0047] like Figure 2 As shown, the wireless charging transmitter circuit 1 further includes: a sampling resistor 14, which is connected between the power voltage input pin and ground; a current detection pin is connected to both ends of the sampling resistor 14 and connected to the controller 13, which collects the resonant current and outputs it to the controller 13.

[0048] In one embodiment, the wireless charging transmitter module 1 further includes a current sense amplifier and an analog-to-digital converter (ADC). SNSP and SNSN can collect the differential voltage across the sampling resistor 14, amplify it in the current sense amplifier, and then transmit it to the ADC for sampling. The sampled current signal is then output to the controller 13.

[0049] In one embodiment, the controller 13 calculates the product of the resonant current collected at the current moment and the resonant current collected at the previous moment; in response to the product of the current values ​​being less than or equal to 0, it determines that the zero-crossing point of the resonant current has been detected, that is, it determines that the resonant current has passed through the zero point.

[0050] For example, the wireless charging transmitter module 1 also includes an analog comparator. The controller 13 acquires the resonant current i_sense through an ADC or analog comparator. The controller 13 may include a digital low-pass filter and a zero-crossing comparator. The resonant current is first filtered by the digital low-pass filter to remove switching noise and glitches before being fed into the zero-crossing comparator. The zero-crossing comparator is used to determine the zero-crossing point of the resonant current based on the changes in the resonant current. When i_sense[k-1]×i_sense[k]≤0, it is determined that the zero-crossing point of the resonant current has been detected, that is, it is determined that the resonant current has passed through the zero point, and the current time is recorded as the first time t_zc corresponding to the current zero-crossing point. i_sense[k] is the resonant current acquired at the current time, and i_sense[k-1] is the resonant current acquired at the previous time. Zero-crossing detection can simultaneously detect positive zero-crossing (negative → positive) and negative zero-crossing (positive → negative), which correspond to different half-bridge arm actions.

[0051] In one embodiment, the sampling resistor 14 can be a resistor with a fixed resistance value, such as 10mΩ.

[0052] In one embodiment, sampling resistor 14 is connected between the power voltage input pin and power ground.

[0053] Thus, the resonant current generated in the resonant circuit can be collected in real time through the sampling resistor 14, thereby obtaining the changes in the resonant current in real time, so as to determine the current zero-crossing point and adjust the dead time.

[0054] In some embodiments, the controller 13 detects the quality factor of the resonant circuit 2 after power-on and determines the initial dead time based on the quality factor; the pulse width modulation circuit 12 outputs a pulse width modulation signal based on the initial dead time after power-on.

[0055] For example, the wireless charging transmitting circuit may further include: a pulse transmitting circuit and a quality factor detection circuit; the pulse transmitting circuit is connected to the resonant circuit, and the pulse transmitting circuit outputs a pulse excitation current to the resonant circuit after power-on; wherein, the pulse transmitting circuit may be integrated inside the wireless charging transmitting module 1, and the quality factor detection circuit may include a series resistor and a parallel ground resistor.

[0056] The quality factor detection circuit is connected to the first switching node and the controller 13, and is used to acquire the resonant attenuation envelope generated by the resonant circuit 2 based on the pulse excitation current through the first switching node; the controller 13 calculates the quality factor of the resonant circuit 2 based on the resonant attenuation envelope, and determines the initial dead time based on the quality factor; the pulse width modulation circuit 12 outputs a pulse width modulation signal based on the initial dead time after power-on.

[0057] For example, the wireless charging transmitter module 1 may include a quality factor detection pin (QDET). The QDET can be connected to the quality factor detection circuit and the detection node VCOIL of the resonant circuit 2, respectively. It collects the resonant attenuation envelope generated by the resonant circuit 2 based on the pulse excitation current, which can reflect the attenuation envelope of the LC resonant attenuation oscillation of the transmitter coil 21 in the resonant circuit 2. The controller 13 can calculate the Q value of the resonant circuit by measuring the attenuation ratio of the amplitude of adjacent cycles. The Q value detection result can be used for foreign object detection (an abnormal drop in Q value indicates the presence of a metallic foreign object) and for pre-configuring the initial dead time of ZVS (the Q value determines the attenuation rate of the resonant current in the dead time).

[0058] For example, the controller 13 continuously samples the attenuation waveform through the resonant attenuation envelope, calculates the natural logarithm of the amplitude ratio of adjacent periods to obtain the logarithmic attenuation rate ρ, and then calculates the quality factor Q of the resonant circuit according to Q = π / ρ.

[0059] In one embodiment, the dead time corresponding to the first PWM clock cycle after power-on is the initial dead time, and the PWM signal is output based on the initial dead time to drive the switching transistor.

[0060] In one embodiment, a coil voltage demodulation circuit may also be included, which may be integrated into the wireless charging transmitter module 1. For example, the wireless charging transmitter module 1 may also include a coil voltage demodulation pin (VDM). The coil voltage demodulation circuit can be connected to the resonant circuit 2 through a high-pass filter, coupled to the switching voltage signal of the first switching node, and extract the AC voltage envelope generated by the transmitting coil 21 in the resonant circuit 2.

[0061] The high-pass filter can be connected to the VDM pin and may include a resistor divider network and an AC coupling capacitor. The AC voltage envelope can be fed back to the controller 13 to assist in determining the ZVS state. For example, the controller 13 can determine the node voltage of the first switching node based on the AC voltage envelope and correct the adjusted dead time based on the node voltage.

[0062] In this way, by using pulse excitation to detect and calculate the Q value of the current resonant circuit 2, an initial dead time that precisely matches the performance of the resonant circuit 2 can be pre-configured after power-on, so that the switching transistor drive can meet the precise operation of the ZVS state throughout the entire working cycle from start to finish.

[0063] In some embodiments, the initial dead time is configured to a preset fixed value, which is used by default during initial power-on and under no-load conditions. Setting the initial dead time to a fixed initial value helps reduce the computational complexity before adjusting the dead time.

[0064] The preset fixed value ranges from 50ns to 150ns. This range is determined by comprehensively considering factors such as chip driving capability, MOS junction capacitance charging characteristics, and the superposition of upper and lower bridge arm switching timings, so that the initial dead time covers the interval between 50ns and 150ns. The higher the overall conversion efficiency of the chip, the shorter the preset fixed value can be set, but the manufacturing cost of the chip will increase accordingly; 50ns is a relatively optimal value that balances conversion efficiency and manufacturing cost. If the initial dead time is longer than 150ns, additional conduction losses will be introduced, leading to a decrease in the overall conversion efficiency of the system.

[0065] Preferably, the initial dead time is set to 100ns. Using 100ns as the baseline value for the initial dead time can cover the vast majority of application scenarios.

[0066] In some embodiments, such as Figure 2 As shown, the wireless charging transmitter module 1 further includes: a first bootstrap pin BST1 and a second bootstrap pin BST2; the first bootstrap pin is connected to the first switching node through a series capacitor, and the second bootstrap pin is connected to the second switching node through a series capacitor.

[0067] For example, the first bootstrap pin BST1 is connected to the first switch pin SW1 through a series capacitor, and the second bootstrap pin BST2 is connected to the second switch pin SW2 through a series capacitor.

[0068] In one embodiment, the capacitance of the series capacitor (also known as the bootstrap capacitor) can be 0.1 μF, etc. When one switch group is turned on, the bootstrap capacitor charges from the power supply pin VDD (5.0V) of the wireless charging transmitter module 1 through an internal diode; when another switch group needs to be turned on, the charge stored in the bootstrap capacitor provides a floating power rail for the high-side gate driver. The value of the bootstrap capacitor must meet the requirement that the voltage drop during the maximum on-time does not exceed the minimum gate drive voltage requirement.

[0069] In this way, the bootstrap circuit turns on the switching transistor through the bootstrap capacitor to meet the driving voltage requirements, ensuring the normal and rapid turn-on of the switching transistor, which is conducive to precise alignment with the zero-crossing moment of the resonant current.

[0070] This application also provides a wireless charging transmitter circuit control method, applicable to the wireless charging transmitter circuit provided in any of the foregoing embodiments, such as... Figure 3 As shown, the wireless charging transmitter circuit control method may include: S10: Determine the zero-crossing point of the resonant current based on the changes in the resonant current.

[0071] S20: In response to the zero-crossing point of the resonant current, adjust the dead time based on the time difference between the first moment corresponding to the zero-crossing point and the second moment when the next set of switching transistors is turned on.

[0072] The dead zone is the period during which all the switches in the full-bridge inverter are in the off state.

[0073] S30: The pulse width modulation signal corresponding to the adjusted dead time.

[0074] In one embodiment, the variation of the resonant current can include the current values ​​collected in two consecutive samples. For example, the resonant current is positive when flowing in a first direction and negative when flowing in a second direction, with the first and second directions being opposite. Determining the zero-crossing point of the resonant current can refer to determining whether the current value of the resonant current passes through zero.

[0075] For example, the zero-crossing point of the resonant current is determined by whether the product of the resonant current value collected at the current moment and the resonant current value collected at the previous moment is less than or equal to 0. When the product is less than 0, it means that the resonant current values ​​collected at the current moment and the resonant current values ​​collected at the previous moment are one positive and one negative, respectively. When the product is equal to 0, it means that at least one of the resonant current values ​​collected at the current moment and the resonant current values ​​collected at the previous moment is 0. In this case, the zero-crossing point of the resonant current can be determined to have been detected.

[0076] In one embodiment, steps S10 and S20 can be executed by the controller 13, and step S30 can be executed by the pulse width modulation circuit 12.

[0077] In one embodiment, before adjusting the dead time based on the time difference between the first moment corresponding to the zero-crossing point and the second moment of turning on the next group of switches, the method may further include: determining the second moment of turning on the next group of switches by reading the timestamp of the current pulse width modulation signal period. For example, the timestamp of the current pulse width modulation signal period can be read from the pulse width modulation circuit 12.

[0078] It should be noted that the relevant definitions, explanations, optional implementation methods and beneficial effects in the method embodiments of this application can be referred to the description in the foregoing circuit embodiments, and will not be repeated here.

[0079] In some embodiments, adjusting the dead time based on the time difference between the first moment corresponding to the zero-crossing point and the second moment when the next set of switching transistors is turned on may include: The second moment to turn on the next set of switching transistors is determined by reading the timestamp of the current pulse width modulation signal cycle. The time difference between the first moment and the second moment corresponding to the zero crossing point is compared with a preset reference value; In response to the time difference exceeding a preset range, the dead time is adjusted.

[0080] In one embodiment, the turn-on time of the corresponding side switch is obtained, i.e., the second time t_on. The time difference Δt = t_on is calculated. t_zc. In an ideal ZVS state, Δt should be equal to the body diode conduction time, that is, the body diode conduction time between the completion of Coss discharge and the turn-on of the next set of switches. For example, Δt can be 0~20 ns.

[0081] In one embodiment, the time difference Δt is compared with a preset reference value Δtar (e.g., Δtar = 10 ns), where the preset reference value is a pre-set time difference corresponding to an ideal ZVS state. Exceeding the preset range can mean greater than a preset threshold, such as a preset threshold δ which can be two PWM clock cycles, approximately 7 ns at a 288 MHz clock. Here, the difference between the time difference Δt and the preset reference value Δtar refers to the absolute value of the difference.

[0082] In one embodiment, adjusting the dead time in response to the difference between the time difference and the preset reference value exceeding a preset range includes: In response to the time difference being greater than the preset reference value, and the difference between the time difference and the preset reference value exceeding a preset range, the dead time is reduced; In response to the time difference being less than the preset reference value and the difference between the time difference and the preset reference value exceeding a preset range, the dead time is increased.

[0083] That is, if Δt > Δtar + δ, it means that the dead time is too long (the conduction time of the body diode is too long, and the loss increases), and the dead time needs to be reduced. In this case, the dead time can be reduced by Δt_step.

[0084] If Δt < Δtar δ: This indicates that the dead time is too short (Coss discharge is not fully completed), and the dead time needs to be increased. In this case, the dead time can be increased by Δt_step.

[0085] If |Δt Δtar| ≤ δ: This indicates that the dead zone is in the optimal range, and the current value should be maintained.

[0086] In one embodiment, δ is the dead-time adjustment threshold, i.e., the aforementioned preset threshold. Δt_step is the single adjustment step size, which is the adjustment amount of the dead-time duration, for example, it can be 1 PWM clock cycle ≈ 3.5 ns, etc.

[0087] In one embodiment, adjusting the corresponding pulse width modulation signal based on the adjusted dead time may include: writing the adjusted dead time into the dead time rising edge and falling edge registers in the pulse width modulation circuit 12, and updating the dead time counter value; the pulse width modulation circuit 12 generates a PWM signal using the adjusted dead time in the next PWM clock cycle. For example, four complementary PWM signals are directly output to the four switching transistors of the full-bridge inverter 11.

[0088] In this way, the time difference can be used to intuitively determine whether the current dead time is too long or too short, so that the dead time can be adjusted in time to meet the precise operation of ZVS. Moreover, using the PWM clock cycle as the single adjustment step size can avoid the dead time becoming too long or too short due to excessive adjustment amplitude in a single adjustment, thus improving the adjustment accuracy.

[0089] In some embodiments, before determining the zero-crossing point of the resonant current based on changes in the resonant current, the method may further include: After power-on is detected, the quality factor of the resonant circuit is detected, and the initial dead time is determined based on the quality factor. The pulse width modulation signal is output based on the initial dead time to drive the switching transistor.

[0090] For example, detecting the quality factor of the resonant circuit may include: outputting a pulse excitation current to the resonant circuit; acquiring the resonant attenuation envelope generated by the resonant circuit based on the pulse excitation current; and calculating the quality factor of the resonant circuit based on the resonant attenuation envelope.

[0091] In one embodiment, after detecting power-on, the pulse transmitting circuit outputs a pulse excitation current to the resonant circuit 2, and the Q-value detection circuit acquires the resonant attenuation envelope generated by the resonant circuit 2 based on the pulse excitation current. The controller 13 calculates the quality factor of the resonant circuit 2 based on the resonant attenuation envelope, and determines the initial dead time based on the quality factor. The pulse width modulation circuit 12 outputs a pulse width modulation signal based on the initial dead time to drive the switching transistor.

[0092] The acquisition of the resonant attenuation envelope generated by the pulsed excitation current of the resonant circuit can be achieved by acquiring the resonant attenuation envelope generated by the pulsed excitation current of the resonant circuit through the first switching node.

[0093] In one embodiment, calculating the quality factor of the resonant circuit 2 based on the resonant attenuation envelope may include: calculating the natural logarithm of the amplitude ratio of adjacent periods based on the resonant attenuation envelope to obtain the logarithmic attenuation rate ρ; and calculating the quality factor Q of the resonant circuit 2 according to Q = π / ρ.

[0094] In one embodiment, outputting a pulse width modulation signal based on the initial dead time may include: writing the initial dead time into the dead time rising edge and falling edge registers in the pulse width modulation circuit 12, and updating the dead time counter value; the pulse width modulation circuit 12 generates a PWM signal using the initial dead time in the next PWM clock cycle.

[0095] In one embodiment, determining the initial dead time based on the quality factor may include: Acquire the power signal input voltage and temperature data of the wireless charging transmitter module; The initial dead time is calculated based on the quality factor, the power signal input voltage, and the temperature data.

[0096] For example, the wireless charging transmitter circuit described above may also include a temperature detection circuit connected to the controller 13. The temperature detection circuit can be used to detect the temperature data of the wireless charging transmitter module 1.

[0097] In one embodiment, the initial dead time t_dead_init can be calculated as: t_dead_init = T_res / 4 + t_margin, where T_res = 1 / f_res is the resonant period, and t_margin is a preset time margin (e.g., 10~20 ns). After obtaining t_dead_init, it can be corrected based on the power supply signal input voltage and the temperature data.

[0098] In one embodiment, the power signal input voltage refers to the voltage value of the power signal input to the power voltage input pin, and the temperature data refers to the temperature data of the wireless charging transmitter module 1.

[0099] In this way, the initial dead time ensures that the first switch after startup is close to the optimal dead time, reducing the ZVS convergence time.

[0100] In one embodiment, the initial dead time is configured to a preset fixed value, the preset fixed value ranging from 50ns to 150ns. Preferably, the initial dead time is set to 100ns.

[0101] In some embodiments, determining the zero-crossing point of the resonant current based on changes in the resonant current may include: Calculate the product of the resonant current sampled at the current moment and the resonant current sampled at the previous moment; In response to the product of the current values ​​being less than or equal to 0, the zero-crossing point of the resonant current is determined to be detected.

[0102] In one embodiment, the interval between the current time and the previous time should be less than a preset interval, such as less than 50ns, 100ns, etc.

[0103] In one embodiment, the following may be included after step S30: The rate of change of the power signal input voltage and the resonant current of the wireless charging transmitter module are detected. When the rate of change of the power signal input voltage and / or the resonant current reaches a preset threshold, the adjustment of the dead time is stopped.

[0104] In one embodiment, the above steps can be performed by controller 13. The power signal input voltage is the voltage of the power signal input to the PVIN power voltage input pin. The rate of change can be the rate of change within one PWM clock cycle, or the rate of change over multiple PWM clock cycles.

[0105] In one embodiment, the adjustment of the dead time is stopped, that is, steps S10 to S30 are stopped. When a step change in input voltage is detected (such as a fast charging adapter switching from 5V to 9V or 12V, with a voltage step exceeding 1V) or a jump in load power is detected (such as the receiver switching from constant current to constant voltage charging mode, with a power jump exceeding 3W), the protection mode is entered: first, the current dead time is locked, the adaptive adjustment of the dead time is stopped, and the dead time after the last adjustment is maintained.

[0106] In one embodiment, after stopping the adjustment of the dead time, the method may further include: in response to the rate of change of the power signal input voltage and / or the resonant current being lower than a preset threshold during a series of PWM clock cycles, restarting the adjustment of the dead time, that is, restarting the execution of steps S10 to S30.

[0107] For example, by monitoring the rate of change of PVIN voltage and resonant current, when the voltage and current fluctuation amplitudes are less than the preset thresholds (voltage < 100 mV, current < 50 mA) for M consecutive clock cycles (M is configurable, for example, 16), it is determined that the system has entered a steady state, the protection lock is released, and the adaptive dead time adjustment is restarted.

[0108] In this way, when the system experiences an overload anomaly, the dead time adjustment can be stopped in time, avoiding inaccurate adjustment that leads to wasted power consumption.

[0109] In one embodiment, the following may be included after step S30: The current temperature of the wireless charging transmitter circuit and the peak value of the resonant current are detected. In response to the current temperature reaching a preset upper temperature limit and / or the peak value of the resonant current reaching a preset upper current limit, the adjustment of the dead time is stopped.

[0110] In one embodiment, the above steps can be performed by the controller 13. The current temperature of the wireless charging transmitter circuit can refer to the current temperature of the wireless charging transmitter module 1, or the current temperature of the transmitter coil 21, etc. The preset upper limit of temperature can be 100℃, 110℃, 120℃, etc., and the preset upper limit of current can be 2A, 1.9A, etc.

[0111] In one embodiment, after stopping the adjustment of the dead time, the method may further include: in response to the current temperature being lower than a preset upper temperature value and the peak value of the resonant current being lower than a preset upper current value for a consecutive number of PWM clock cycles, restarting the adjustment of the dead time, that is, restarting the execution of steps S10 to S30.

[0112] In this way, when abnormal events such as over-temperature or over-current occur, the dead time adjustment can be stopped in time to avoid inaccurate adjustment leading to wasted power consumption.

[0113] In some embodiments, the method further includes: in standby mode or under light load conditions (such as when the wireless charging receiver or the electrical load is fully charged and enters trickle mode), the controller 13 can switch the full-bridge inverter 11 to at least one of the following low-power modes: ① Asymmetric PWM mode: only two diagonally opposite switching transistors are alternately turned on, reducing the switching frequency and duty cycle, and reducing drive losses; ② Burst mode: the full-bridge inverter 11 operates intermittently in the form of short pulse trains, and continues to execute the above steps S10~S30 during operation. When it is determined by Q value detection or communication demodulation decoding that the receiver has been reconnected and the load current exceeds the wake-up threshold, the light load mode is exited and the full-bridge ZVS continuous operation is restored.

[0114] In addition, the circuit and control method used in this application can simultaneously suppress EMI from the following four levels: (1) The switching node dv / dt is determined by the resonant characteristics, not by the switching speed of the switching transistor.

[0115] Under ZVS conditions, the voltage at switching nodes SW1 / SW2 is naturally established by the sinusoidal charging and discharging of the parasitic capacitance Coss of the switching transistor by the resonant current. The voltage rise / fall waveform is a segment of a sinusoidal curve determined by the LC resonance, and its peak value of dv / dt is determined by the resonant parameter: dv / dt_max = I_L_peak / (2×Coss). Here, dv / dt refers to the rate of voltage change, representing how quickly the voltage changes per unit time; a larger value indicates a more drastic voltage change.

[0116] Where I_L_peak is the peak resonant current during the dead zone. Calculated with typical parameters (I_L_peak≈0.5A, Coss≈200 pF), dv / dt_max≈1.25 V / ns, which is much lower than dv / dt (which can reach 10~20 V / ns) determined by the gate drive capability of the switching transistor in hard-switching mode. Since the switching node voltage waveform is no longer a rectangular wave but a smooth sine wave segment, its higher harmonic energy is significantly reduced. According to the spectral characteristics of a sine wave, harmonic components above the fifth resonant frequency have attenuated to below -20 dB of the fundamental frequency. This means that the spectral energy of the switching node voltage is concentrated near the resonant frequency f_res (100~150 kHz) and its lower harmonics (300~750 kHz), while noise energy has been fundamentally suppressed in the conducted EMI band above 30 MHz and the radiated EMI band of 30~1000 MHz.

[0117] (2) The di / dt of the resonant current is limited by the transmitting coil 21.

[0118] In hard-switching mode, at the instant the switch is turned on, PVIN directly charges the switching node capacitor through the on-channel resistance of the switch (only tens of mΩ). Di / dt is limited only by parasitic inductance and the turn-on speed of the switch. In ZVS mode, however, Vds is already 0 before the switch is turned on (the body diode is already conducting). There is almost no sudden current change at the moment of turn-on because the resonant current i_L is smoothly adjusted by the transmitting coil 21, i.e., inductance L2 (typically 24μH). The peak value of di / dt, di / dt_max = V_PVIN / L2 ≈ 9V / 24μH = 0.375 A / μs. This di / dt value is 2-3 orders of magnitude lower than that of hard switching, and the parasitic ringing amplitude and frequency under this condition can be ignored. Here, di / dt refers to the rate of change of current, representing how quickly the current changes per unit time; a larger value indicates a more drastic current change.

[0119] (3) Eliminate ringing at the switch node.

[0120] In hard-switching mode, LC parasitic ringing at the switching node after the switching transistor is turned off is one of the main sources of radiated EMI. Typical ringing frequencies are 50–200 MHz, falling within sensitive frequency bands such as FM broadcasting (88–108 MHz) and aviation communication (118–137 MHz), easily leading to excessive radiated EMI testing. In the ZVS mode of this application, the voltage change at the switching node is dominated by the main resonant mode of the series resonant circuit (f_res = 100–150 kHz). This frequency is much lower than the self-resonant frequency of PCB parasitic effects. The parasitic ringing mechanism is covered and absorbed by the powerful energy of the main resonant mode; therefore, no significant high-frequency ringing is observed on the switching node waveform.

[0121] (4) Suppression of common-mode noise paths.

[0122] In the 11-topology of the full-bridge inverter, switching nodes SW1 and SW2 operate in a complementary manner. Ideally, the common-mode currents generated by the dv / dt of the two switching nodes are canceled out by the parasitic capacitance of the PCB to ground (typically 5~20 pF). However, in hard-switching mode, due to nanosecond-level asymmetry in the switching moments of the two half-bridges, the dispersion of switching transistor parameters, and the asymmetry of PCB traces, the actual common-mode currents cannot be completely canceled out. The residual common-mode noise is conducted through the input cables to the line impedance stabilization network and is measured. In the ZVS mode provided in this application, the voltage waveform of the switching nodes is no longer a steep rectangular edge, but a smooth sinusoidal transition determined by resonance. The rate of change (dv / dt) of the voltages of the two switching nodes during the transition period is significantly reduced and becomes more symmetrical. Therefore, the amplitude and asymmetry of the common-mode current are significantly reduced. According to measured data, the common-mode conducted EMI in the ZVS mode is reduced by 10~20 dBμV in the 150 kHz~30 MHz frequency band compared to the hard-switching mode.

[0123] This application also provides a portable charging device, including the wireless charging transmitting circuit provided in any of the foregoing embodiments of this application.

[0124] Here, portable charging devices can be devices, equipment, or systems that have wireless charging capabilities and act as wireless charging transmitters, such as power banks, chargers, power adapters, or devices, equipment, or systems that have multiple of the above functions.

[0125] For example, a portable charging device could be a wireless charger that wirelessly charges at least one device such as a mobile phone, watch, or tablet.

[0126] As one possible implementation, a ZVS control method based on resonant current detection is provided, comprising the following steps: Step A: During the system power-on startup phase, the quality factor of the resonant circuit is determined by the Q-value detection circuit, and the initial dead time is generated based on the quality factor; Step B: During the normal power transmission phase, continuously acquire the resonant current detection signal and determine the time of current zero crossing using a comparator; Step C: Calculate the time difference Δt between the zero-crossing moment of the resonant current and the current turn-on moment of the switching transistor; Step D: Compare Δt with the preset target time difference Δtar: If Δt > Δtar + δ, then reduce the dead time of the next cycle; if Δt < Δtar If δ increases, then the dead time of the next cycle is increased; if |Δt If Δtar|≤δ, then maintain the current dead zone duration; Step E: Write the adjusted dead time into the PWM control register to generate four complementary PWM drive signals with the updated dead time, so that the full-bridge inverter continues to operate in the zero-voltage switching state.

[0127] It should be noted that, in this document, the terms "comprising," "including," or any other variations thereof are intended to cover non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements includes not only those elements but also other elements not expressly listed, or elements inherent to such a process, method, article, or apparatus. Unless otherwise specified, an element defined by the phrase "comprising one..." does not exclude the presence of other identical elements in the process, method, article, or apparatus that includes that element.

[0128] The above are merely specific embodiments of this application, but the scope of protection of this application is not limited thereto. Any variations or substitutions that can be easily conceived by those skilled in the art within the technical scope disclosed in this application should be included within the scope of protection of this application. Therefore, the scope of protection of this application should be determined by the scope of the claims.

Claims

1. A wireless charging transmitter circuit, characterized in that, include: A wireless charging transmitter module and a resonant circuit; the wireless charging transmitter module includes a full-bridge inverter, a pulse width modulation circuit, and a controller; the full-bridge inverter includes multiple switching transistors. The plurality of switching transistors are connected to the resonant circuit, and the resonant circuit generates a resonant current based on the signal output by the switching transistors. The resonant current is used to provide wireless charging power. Both the resonant circuit and the pulse width modulation circuit are connected to the controller. The controller determines the zero-crossing point of the resonant current based on the changes in the resonant current. In response to the zero-crossing point of the resonant current, the controller adjusts the dead time based on the time difference between the first moment corresponding to the zero-crossing point and the second moment when the next set of switching transistors is turned on. The dead time is the period during which all the switching transistors in the full-bridge inverter are in the off state. The pulse width modulation circuit is connected to the plurality of switching transistors, and the pulse width modulation circuit adjusts the corresponding pulse width modulation signal based on the adjusted dead time. The pulse width modulation signal is used to drive the switching transistor; The wireless charging transmitter module further includes: a power voltage input pin and a current detection pin; the power voltage input pin is connected to the plurality of switching transistors, and the power voltage input pin provides a power signal for generating the resonant current; The wireless charging transmitter circuit further includes: a sampling resistor connected between the power voltage input pin and ground; a current detection pin connected across the sampling resistor and connected to the controller, the current detection pin acquiring the resonant current and outputting it to the controller.

2. The wireless charging transmitting circuit according to claim 1, characterized in that, The plurality of switching transistors includes a first switching transistor group and a second switching transistor group; adjacent switching transistors in the first switching transistor group are connected to form a first switching node, and adjacent switching transistors in the second switching transistor group are connected to form a second switching node. The resonant circuit includes a transmitting coil and a resonant capacitor, which are connected in series between the first switching node and the second switching node.

3. The wireless charging transmitting circuit according to claim 2, characterized in that, After power-on, the controller detects the quality factor of the resonant circuit and determines the initial dead time based on the quality factor; after power-on, the pulse width modulation circuit outputs a pulse width modulation signal based on the initial dead time.

4. The wireless charging transmitting circuit according to claim 3, characterized in that, The wireless charging transmitting circuit further includes: a pulse transmitting circuit and a quality factor detection circuit; the pulse transmitting circuit is connected to the resonant circuit, and the pulse transmitting circuit outputs a pulse excitation current to the resonant circuit after power-on; Both the controller and the first switching node are connected to the quality factor detection circuit. The quality factor detection circuit acquires the resonant attenuation envelope generated by the resonant circuit based on the pulse excitation current through the first switching node. The controller calculates the quality factor of the resonant circuit based on the resonant attenuation envelope and determines the initial dead time based on the quality factor.

5. The wireless charging transmitting circuit according to claim 2, characterized in that, The wireless charging transmitter module further includes: a first bootstrap pin and a second bootstrap pin; the first bootstrap pin is connected to the first switching node through a series capacitor, and the second bootstrap pin is connected to the second switching node through a series capacitor.

6. A wireless charging transmitter circuit control method, applied to the wireless charging transmitter circuit according to any one of claims 1 to 5, characterized in that, The wireless charging transmitter circuit control method includes: The zero-crossing point of the resonant current is determined based on the changes in the resonant current; In response to the zero-crossing point of the resonant current, the dead time is adjusted based on the time difference between the first moment corresponding to the zero-crossing point and the second moment when the next set of switching transistors is turned on; the dead time is the period during which all the switching transistors in the full-bridge inverter are in the off state. The pulse width modulation signal is adjusted based on the adjusted dead time.

7. The wireless charging transmitter circuit control method according to claim 6, characterized in that, The adjustment of the dead time based on the time difference between the first moment corresponding to the zero-crossing point and the second moment of turning on the next group of switching transistors includes: The second moment to turn on the next set of switching transistors is determined by reading the timestamp of the current pulse width modulation signal cycle. The time difference between the first moment and the second moment corresponding to the zero crossing point is compared with a preset reference value; In response to the time difference exceeding a preset range, the dead time is adjusted.

8. The wireless charging transmitter circuit control method according to claim 7, characterized in that, The response to the time difference exceeding a preset reference value, adjusting the dead time, includes: In response to the time difference being greater than the preset reference value, and the difference between the time difference and the preset reference value exceeding a preset range, the dead time is reduced; In response to the time difference being less than the preset reference value and the difference between the time difference and the preset reference value exceeding a preset range, the dead time is increased.

9. The wireless charging transmitter circuit control method according to claim 7, characterized in that, The adjustment amount of the dead time is equal to the length of the pulse width modulation clock cycle corresponding to the current pulse width modulation signal.

10. The wireless charging transmitter circuit control method according to claim 6, characterized in that, Before determining the zero-crossing point of the resonant current based on its variation, the process also includes: After power-on is detected, the quality factor of the resonant circuit is detected, and the initial dead time is determined based on the quality factor. The pulse width modulation signal is output based on the initial dead time to drive the switching transistor.

11. The wireless charging transmitter circuit control method according to claim 6, characterized in that, Also includes: The rate of change of the power signal input voltage and the resonant current of the wireless charging transmitter module are detected. When the rate of change of the power signal input voltage and / or the resonant current reaches a preset threshold, the adjustment of the dead time is stopped.

12. A portable charging device, characterized in that, Includes the wireless charging transmitting circuit as described in any one of claims 1 to 5.

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