A coordinate transformation-free digital control method for three-phase grid-connected converters

Abstract

The present application relates to the technical field of digital control of power electronic converters, and particularly relates to a coordinate transformation-free digital control method for a three-phase networked converter, comprising the following steps: collecting or obtaining three-phase phase variables; directly calculating instantaneous active power and instantaneous reactive power based on the three-phase phase variables, and obtaining average active power and average reactive power through filtering; generating a virtual angle frequency and a power angle based on an active power synchronization ring of a virtual synchronous generator; generating a voltage amplitude reference based on a reactive power-voltage ring; generating a three-phase bridge arm side electromotive force reference according to the voltage amplitude reference and the power angle; generating a three-phase filter inductance current reference by using a phase variable virtual admittance; generating a three-phase inner loop control action according to a phase current error between the three-phase filter inductance current reference and a three-phase filter inductance current; adding the three-phase output voltage and the three-phase inner loop control action to generate a three-phase bridge arm voltage instruction, and generating a modulation signal and a PWM driving signal according to the three-phase bridge arm voltage instruction.

Description

Technical Field

[0001] This invention relates to the field of digital control technology for power electronic converters, and more specifically, to a coordinateless transformation digital control method for a three-phase grid converter. Background Technology

[0002] Network-based control enables voltage source converters to establish AC voltage and frequency in a manner similar to synchronous generators, and provides black start, inertial support, voltage source characteristics, and weak grid operation capabilities.

[0003] Existing three-phase network control is usually implemented in a synchronously rotating dq coordinate system or a stationary alpha-beta coordinate system.

[0004] Among these, the dq coordinate system scheme typically requires Clarke and Park transformations to convert three-phase AC quantities into approximate DC quantities before PI regulation. This type of scheme is relatively effective under balanced fundamental frequency conditions, but it introduces the following technical problems: First, the control path depends on the synchronization angle, coordinate transformation, and decoupling compensation, increasing the complexity of digital implementation and computation time; second, even after output voltage feedforward, the dq coordinate system current inner loop object still contains cross-coupling terms related to the angular frequency, and the control effect depends on the synchronization variable and parameter accuracy; third, asynchronous components and harmonic components exhibit oscillations in the dq coordinate system, making direct phase current waveform adjustment and harmonic processing more complex.

[0005] The stationary alpha-beta coordinate system QPR current loop avoids the Park transformation, but still requires the Clarke transformation to construct orthogonal control variables. In real digital control, the phase current sampling chain before the Clarke transformation inevitably contains non-ideal factors such as gain error, sampling timing deviation, and bias error. These factors are mapped to the alpha-beta axis feedback disturbance via the selected Clarke basis and propagate through the current closed loop, thus forming phase-to-phase asymmetric disturbances when reconstructing the three-phase modulation reference during the inverse transformation. Therefore, a new network-based control implementation method is needed. Summary of the Invention

[0006] This invention aims to provide a coordinate-transform-free three-phase grid control method for real digital controller implementation. It addresses the synchronous-dependent reconfiguration problem of the traditional dq current loop and the basis projection / redistribution phenomenon that may be caused by non-ideal factors in the sampling chain before Clarke transformation in the alpha-beta current loop. The method unifies the outer grid loop, phase variable interface, and inner current loop within physical three-phase variables, thereby reducing the computational burden of digital control and suppressing the amplification and redistribution of sampling chain mismatch by the coordinate transformation path. It is applicable to the digital control implementation of grid-connected inverters, grid-type converters, weak grid access devices, new energy power generation converters, energy storage converters, and high-proportion power electronic AC systems.

[0007] The objective of this invention is achieved through the following technical solution:

[0008] A coordinateless digital control method for a three-phase network converter includes the following steps:

[0009] Collect or acquire three-phase phase variables, including the three-phase filter inductor current i f,abc and three-phase output voltage u o,abc , where abc represents the physical three-phase phase variables;

[0010] The instantaneous active power p is calculated directly based on the three-phase phase variables. e and instantaneous reactive power q e The average active power P is obtained after filtering. e and average reactive power Q e ;

[0011] Based on the active power synchronization loop of the virtual synchronous generator, the virtual angular frequency ω and power angle θ are generated according to the active power reference value, average active power, virtual inertia and damping coefficient.

[0012] Based on the reactive power-voltage loop, a voltage amplitude reference E is generated according to the reactive power reference value, average reactive power, rated voltage amplitude, and regulator parameters.

[0013] The three-phase bridge arm side electromotive force reference is directly generated based on the voltage amplitude reference E and the power angle θ. ;

[0014] Using phase variable virtual admittance, based on the three-phase bridge arm side electromotive force reference... With three-phase output voltage u o,abc The difference generates a three-phase filter inductor current reference. ;

[0015] The CTF-QPR is a coordinate-transform-free quasi-proportional resonant current regulator that operates with phase-channel consistency and directly references the three-phase filter inductor current. With the three-phase filter inductor current i f,abc The phase current error generates the three-phase inner loop control action u. c,abc ;

[0016] The three-phase output voltage u o,abc With the three-phase inner loop control action u c,abc Add them together to generate the three-phase bridge arm voltage command. Based on this, modulation signals and PWM drive signals are generated.

[0017] Furthermore, in the three-phase phase variable model, the model of the converter LC output filter is as follows:

[0018] (1)

[0019] (2)

[0020] Among them, L f For filter inductance, R f C is the parasitic resistance of the filter inductor. f For the filter capacitor, i f,abc For the three-phase filter inductor current, u e,abc For the bridge arm voltage, u o,abc For the three-phase output voltage, i l,abc This is the alternating current.

[0021] Furthermore, the instantaneous active power p e Calculate using the following formula:

[0022] (3);

[0023] Under three-phase three-wire conditions, the instantaneous reactive power q e Calculate using the following formula:

[0024] (4);

[0025] The instantaneous active power p e and instantaneous reactive power q e The average active power P is obtained after passing through a first-order low-pass filter. e and average reactive power Q e .

[0026] Furthermore, in real digital control scenarios where the phase current sampling chain inevitably has non-ideal factors such as gain error, sampling timing error, or bias error, the digital controller is configured to adopt a physical three-phase coordinate-free digital control path.

[0027] The digital controller dynamically generates a virtual angular frequency ω based on the active power-frequency of the virtual synchronous generator.

[0028] (5)

[0029] And the work angle θ is obtained by integration:

[0030] (6)

[0031] Where J is the virtual inertia, and D p ω is the damping coefficient, and ω0 is the rated angular frequency.

[0032] Furthermore, the voltage amplitude reference E is generated by the reactive power regulator:

[0033] (7)

[0034] Where E0 is the rated voltage amplitude, K pv and K iv These are the proportional gain and integral gain of the reactive power regulator, respectively.

[0035] Based on the voltage amplitude reference E and the power angle θ, the bridge arm side electromotive force reference is directly generated from the three-phase phase variables:

[0036] (8)

[0037] in, , , This electromotive force reference represents the internal voltage state before virtual admittance shaping and is not equivalent to the final bridge arm voltage command.

[0038] Furthermore, the virtual admittance is defined as:

[0039] (9)

[0040] in, For virtual resistance, The three-phase filter inductor current is referenced as a virtual inductor. It is generated by the following formula:

[0041] (10).

[0042] Furthermore, the inner current loop employs a coordinate-transformation-free QPR regulator to directly adjust the three-phase phase current error phase by phase:

[0043] (11)

[0044] Three-phase bridge arm voltage command for:

[0045] (12)

[0046] The continuous-domain transfer function of the QPR regulator is:

[0047] (13)

[0048] For proportional gain, For resonant gain, For cutoff bandwidth, It is the rated electrical angular frequency or the QPR resonant center frequency.

[0049] Furthermore, adopting the following approach For a Tustin method discrete QPR regulator with predistortion frequency, let:

[0050] (14)

[0051] And adopt:

[0052] (15)

[0053] get:

[0054] (16)

[0055] in:

[0056] (17)

[0057] (18)

[0058] (19)

[0059] In a digital controller, the above can be... It is implemented using a second-order IIR structure or an equivalent difference equation, and the same discrete controller coefficients are used for the three-phase error channels a, b, and c. Each phase channel maintains independent historical inputs, historical outputs, or state variables, and completes state updates within the same sampling period or control interruption period.

[0060] Compared with the prior art, the beneficial effects of the present invention are:

[0061] ① By removing Clarke and Park coordinate transformations from the actual control path, synchronization, voltage formation, virtual admittance, and current regulation are all completed within the physical three-phase variables, thereby reducing the complexity of the digital control path.

[0062] ② By avoiding synchronization angle-dependent cross-coupling terms in the inner current loop of the dq coordinate system, the dependence on synchronization variables and parameter decoupling accuracy is reduced.

[0063] ③ By avoiding the Clarke transform pre-sampling chain mismatch projection path in the QPR current loop of the alpha-beta coordinate system, the possibility of sampling timing deviation, channel gain mismatch, and bias error being redistributed as modulation asymmetry through the coordinate base is reduced. The internal voltage reference is converted into a three-phase current reference through phase variable virtual admittance, giving the interface between voltage formation and current regulation a clear physical meaning.

[0064] ④ By using the phase-by-phase consistent CTF-QPR current inner loop, high gain is provided near the fundamental frequency, enabling direct tracking of the three-phase phase current reference.

[0065] ⑤ By reducing coordinate transformation matrix operations, synchronous rotation related calculations, and dq decoupling compensation calculations, the execution time of the current loop within the control interrupt is shortened, reserving computational margin for improving the sampling rate, adding protection logic, or expanding diagnostic tasks.

[0066] The present invention also provides a coordinateless transformation digital control system for a three-phase grid converter that performs the above-described control method, comprising:

[0067] The sampling unit is used to acquire three-phase output voltage, three-phase filter inductor current, DC bus voltage, and optional temperature, protection, and status parameters.

[0068] The sampling chain non-ideal parameter recording unit stores at least one of the gain error, sampling timing error, and bias error of the phase current sampling chain, serving as a reference for the configuration of the coordinate-transform-free digital control path. The phase variable power calculation unit directly calculates instantaneous active power and instantaneous reactive power based on the three-phase output voltage and the three-phase filter inductor current.

[0069] The network synchronization unit is used to generate virtual angular frequency and power angle based on the active power reference value and the average active power;

[0070] The voltage amplitude generation unit is used to generate a voltage amplitude reference based on the reactive power reference value and the average reactive power.

[0071] The three-phase electromotive force reference generation unit is used to generate a three-phase bridge arm side electromotive force reference based on the power angle and voltage amplitude reference.

[0072] The phase variable virtual admittance unit is used to generate a three-phase current reference based on the three-phase bridge arm side electromotive force reference and the three-phase output voltage.

[0073] The phase-by-phase CTF-QPR current regulation unit is used to generate three-phase control actions based on the three-phase current reference and the three-phase filter inductor current.

[0074] The bridge arm voltage command generation unit is used to add the three-phase output voltage feedforward and the three-phase control action to generate the three-phase bridge arm voltage command.

[0075] The modulation unit is used to generate PWM drive signals according to the three-phase bridge arm voltage command. Attached Figure Description

[0076] The accompanying drawings are provided to further illustrate the invention and form part of the specification. They are used in conjunction with embodiments of the invention to explain the invention and do not constitute a limitation thereof. In the drawings:

[0077] Figure 1 This is a schematic diagram of the main circuit and digital controller interface of a three-phase network-type VSC.

[0078] Figure 2 This is a schematic diagram of the phase variable CTF-GFM control link.

[0079] Figure 3 This is a schematic diagram of the phase-by-phase CTF-QPR current inner loop structure.

[0080] Figure 4 This is a schematic diagram of the comparison structure of the PI current inner loop in the dq coordinate system.

[0081] Figure 5 This is a schematic diagram of the QPR current inner loop comparison structure in the alpha-beta coordinate system.

[0082] Figure 6 This is a comparison chart of the execution time of different current inner loops in a digital controller.

[0083] Figure 7 This is a comparison diagram of the harmonic spectrum of the current in phase A and phase C of the experimental prototype in the embodiment. Among them, (a) is CTF-QPR, phase A; (b) is alpha-beta QPR, phase A; (c) is dq PI, phase A; (d) is CTF-QPR, phase C; (e) is alpha-beta QPR, phase C; and (f) is dq PI, phase C.

[0084] Figure 8 This is a schematic diagram of the three-phase VSC experimental prototype and test platform in the embodiment. Detailed Implementation

[0085] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0086] The coordinate-free digital control method for a three-phase grid converter in this invention includes the following steps:

[0087] In real digital control scenarios where the phase current sampling chain inevitably has non-ideal factors such as gain error, sampling timing error, or bias error, the digital controller is configured to adopt a physical three-phase coordinate-less digital control path.

[0088] Collect or acquire three-phase phase variables, including the three-phase filter inductor current i f,abc and three-phase output voltage u o,abc , where abc represents the physical three-phase phase change.

[0089] In a three-phase phase variable converter, the model of the LC output filter is as follows:

[0090] (1)

[0091] (2)

[0092] Among them, L f For filter inductance, R f C is the parasitic resistance of the filter inductor. f For the filter capacitor, i f,abc For the three-phase filter inductor current, u e,abc For the bridge arm voltage, u o,abc For the three-phase output voltage, i l,abc This is the alternating current.

[0093] The instantaneous active power p is calculated directly based on the three-phase phase variables. e and instantaneous reactive power q e The average active power P is obtained after filtering. e and average reactive power Q e。

[0094] Instantaneous active power p e Calculate directly using the following formula:

[0095] (3);

[0096] Under three-phase three-wire conditions, the instantaneous reactive power q e Calculate using the following formula:

[0097] (4);

[0098] Instantaneous active power p e and instantaneous reactive power q e The average active power P is obtained after passing through a first-order low-pass filter. e and average reactive power Q e The cutoff frequency of the low-pass filter can be determined based on the power loop response speed, noise suppression requirements, and control cycle.

[0099] Based on the active power synchronization loop of the virtual synchronous generator, the virtual angular frequency ω and power angle θ are generated according to the active power reference value, average active power, virtual inertia and damping coefficient.

[0100] In real digital control scenarios where the phase current sampling chain inevitably has non-ideal factors such as gain error, sampling timing error or bias error, the digital controller is configured to adopt a physical three-phase coordinate-less digital control path.

[0101] The digital controller dynamically generates a virtual angular frequency ω based on the active power-frequency of the virtual synchronous generator.

[0102] (5)

[0103] And the work angle θ is obtained by integration:

[0104] (6)

[0105] Where J is the virtual inertia, and D p Here, ω is the damping coefficient, and ω0 is the rated angular frequency. This step enables the converter to achieve self-synchronization without relying on a phase-locked loop.

[0106] Based on the reactive power-voltage loop, a voltage amplitude reference E is generated according to the reactive power reference value, average reactive power, rated voltage amplitude, and regulator parameters.

[0107] The voltage amplitude reference E is generated by the reactive power regulator:

[0108] (7)

[0109] Where E0 is the rated voltage amplitude, K pv and K iv These are the proportional gain and integral gain of the reactive power regulator, respectively.

[0110] Based on the voltage amplitude reference E and the power angle θ, a three-phase bridge arm side electromotive force reference is directly generated in the three-phase phase variables. :

[0111] (8)

[0112] in, , , This electromotive force reference represents the internal voltage state before virtual admittance shaping and is not equivalent to the final bridge arm voltage command.

[0113] Using phase variable virtual admittance, based on the three-phase bridge arm side electromotive force reference... With three-phase output voltage u o,abc The difference generates a three-phase filter inductor current reference. .

[0114] Virtual admittance is defined as:

[0115] (9)

[0116] in, For virtual resistance, For virtual inductance, the three-phase filter inductor current reference It is generated by the following formula:

[0117] (10).

[0118] The virtual admittance acts on the phase-by-phase consistency of the three-phase channels, which can be equivalent to... With this setup, the interface from the internal voltage to the current reference remains within the physical three-phase variables, without embedding an auxiliary transformation coordinate base.

[0119] The inner current loop employs a coordinate-transformation-free quasi-proportional resonant current regulator, CTF-QPR, which operates with consistent phase-path behavior and directly references the three-phase filter inductor current. With the three-phase filter inductor current i f,abc The phase current error generates the three-phase inner loop control action u. c,abc：

[0120] (11)

[0121] The three-phase output voltage u o,abc With the three-phase inner loop control action u c,abc Add them together to generate the three-phase bridge arm voltage command. Based on this, modulation signals and PWM drive signals are generated.

[0122] Three-phase bridge arm voltage command for:

[0123] (12)

[0124] The continuous-domain transfer function of the QPR regulator is:

[0125] (13)

[0126] For proportional gain, For resonant gain, For cutoff bandwidth, This is the rated electrical angular frequency or the QPR resonant center frequency. The regulator provides high gain near the fundamental frequency while maintaining a wideband response to non-fundamental components through a proportional term.

[0127] The coordinateless transformation digital control method for a three-phase grid converter in this invention can be implemented in a digital controller composed of DSP, MCU, FPGA, SoC, or a combination thereof. The digital controller uses the sampling period Ts or control interrupt period as the basic execution unit to perform sequential or pipelined scheduling of sampled quantity reading, power filtering, grid synchronization state, reactive power-voltage regulation state, virtual admittance discrete state, QPR discrete state, and PWM register updates. The above processing can be implemented using floating-point arithmetic, fixed-point arithmetic, or hardware logic; the specific processor model, data word length, and scheduling method do not constitute a limitation on the scope of protection of this invention.

[0128] In this invention, the following method is adopted: For a Tustin method discrete QPR regulator with predistortion frequency, let:

[0129] (14)

[0130] And adopt:

[0131] (15)

[0132] get:

[0133] (16)

[0134] in:

[0135] (17)

[0136] (18)

[0137] (19)

[0138] In a digital controller, the above can be... It is implemented using a second-order IIR structure or equivalent difference equations, and the same discrete controller coefficients are used for the three-phase error channels a, b, and c. Each phase channel maintains independent historical inputs, historical outputs, or state variables, and the state update is completed within the same sampling period or control interruption period. The specific implementation form of the difference equations can be determined according to the target controller code.

[0139] The present invention also provides a coordinateless transformation digital control system for a three-phase grid converter that performs the above-described control method, comprising:

[0140] The sampling unit is used to acquire three-phase output voltage, three-phase filter inductor current, DC bus voltage, and optional temperature, protection, and status parameters.

[0141] The sampling chain non-ideal parameter recording unit stores at least one of the gain error, sampling timing error, and bias error of the phase current sampling chain, serving as a reference for the configuration of the coordinate-transform-free digital control path. The phase variable power calculation unit directly calculates instantaneous active power and instantaneous reactive power based on the three-phase output voltage and the three-phase filter inductor current.

[0142] The network synchronization unit is used to generate virtual angular frequency and power angle based on the active power reference value and the average active power;

[0143] The voltage amplitude generation unit is used to generate a voltage amplitude reference based on the reactive power reference value and the average reactive power.

[0144] The three-phase electromotive force reference generation unit is used to generate a three-phase bridge arm side electromotive force reference based on the power angle and voltage amplitude reference.

[0145] The phase variable virtual admittance unit is used to generate a three-phase current reference based on the three-phase bridge arm side electromotive force reference and the three-phase output voltage.

[0146] The phase-by-phase CTF-QPR current regulation unit is used to generate three-phase control actions based on the three-phase current reference and the three-phase filter inductor current.

[0147] The bridge arm voltage command generation unit is used to add the three-phase output voltage feedforward and the three-phase control action to generate the three-phase bridge arm voltage command.

[0148] The modulation unit is used to generate PWM drive signals according to the three-phase bridge arm voltage command.

[0149] Example 1: CTF-GFM control of a three-phase three-wire VSC:

[0150] In this embodiment, the three-phase voltage source converter includes a DC-side power supply, two-level or multi-level three-phase bridge arms, an LC output filter, and three-phase AC terminals. The digital controller includes an ADC sampling module, a power calculation module, a VSG synchronization module, a reactive power-voltage regulation module, a virtual admittance module, a CTF-QPR current regulation module, and a PWM update module.

[0151] The controller uses the sampling period Alternatively, the interrupt cycle can be controlled to periodically execute the following steps:

[0152] 1. The ADC samples the three-phase filter inductor current and the three-phase output voltage to obtain... and .

[0153] 2. Directly from the three-phase filter inductor current and three-phase output voltage Calculate instantaneous active power and instantaneous reactive power The average active power is obtained after low-pass filtering. and average reactive power .

[0154] 3. The VSG synchronization module is based on... Update virtual angular frequency Harmony and angle .

[0155] 4. Reactive power-voltage regulation module according to Update voltage amplitude reference .

[0156] 5. The reference generation module uses voltage amplitude as a reference. , angle The three-phase bridge arm side electromotive force reference is generated by the fixed phase offset. .

[0157] 6. Virtual admittance module phase-by-phase processing The reference current of the three-phase filter inductor is obtained. .

[0158] 7. CTF-QPR Current Regulation Module Phase-by-Phase Processing The three-phase inner loop control action was obtained. .

[0159] 8. The voltage feedforward module generates three-phase bridge arm voltage commands. .

[0160] 9. The PWM module operates according to the three-phase bridge arm voltage command. The DC bus voltage generates a three-phase bridge arm drive signal.

[0161] Within the aforementioned control cycle, the actual control path does not execute Clarke transform, Park transform, inverse Park transform, or inverse Clarke transform. If the PWM modulation strategy itself requires zero-sequence injection, common-mode modulation, or space vector implementation, this modulation stage can be used as an optional modulation implementation after the bridge arm voltage command, without changing the characteristics of the coordinate transformation-free network control path of this invention. Sampling, discrete state update, bridge arm voltage command generation, and PWM update are all within the same digital control task chain.

[0162] Example 2, Implementation of a digital QPR controller:

[0163] In this embodiment, the QPR regulator operates at the rated electrical angular frequency. The resonant center is used, and the model is discretized using the Tustin pre-distortion method. The controller maintains second-order state variables with the same structure for the three-phase error channels a, b, and c. Within each sampling period, for phase k:

[0164] in, Depend on The calculations show that phases a, b, and c use the same coefficients, but their state variables are independent of each other.

[0165] The advantage of this embodiment is its simplicity; it does not require coordinate rotation matrices, trigonometric function transformations, or synchronization angles to participate in the current inner loop calculation, thereby reducing the current loop calculation time.

[0166] Example 3:

[0167] In one experimental prototype implementation, the three-phase VSC prototype uses a Texas Instruments F29H85x C2000 64-bit real-time MCU as its digital controller and IMZA120R040M1 SiC MOSFETs as its power devices. The main parameters of the experimental platform include:

[0168] Rated active power: 500W.

[0169] Rated voltage: 100V.

[0170] Rated frequency: 50Hz.

[0171] Filter inductor: approximately 450uH at 0A and approximately 310uH at 21.5A.

[0172] Switching frequency: 50kHz.

[0173] Control interrupt frequency and ADC sampling frequency: 100kHz.

[0174] The nominal current loop gain is consistent with the simulation parameters, namely, QPR proportional gain 0.5, resonant gain 5, cutoff bandwidth 5 rad / s; dq PI proportional gain 0.5, integral gain 5.

[0175] During the experiment, the interrupt-level execution time was measured by GPIO switching. The control cycle was divided into three time periods: pre-processing, current loop calculation, and modulation / PWM update. The measured CTF-QPR current loop calculation time was 0.165µs, and the total control cycle time was 1.680µs. This embodiment demonstrates that the present invention can be implemented on a real-time MCU and has the effect of reducing the burden of current loop calculation.

[0176] The experimental platform measured the current loop calculation time of CTF-QPR to be 0.165 μs, alpha-beta QPR to be 0.219 μs, and dq PI to be 0.319 μs. The current loop calculation time of CTF-QPR was reduced by approximately 24.7% and 48.3% compared to alpha-beta QPR and dq PI, respectively.

[0177] The experimental platform measured that, within the range of the 2nd to the 50th harmonic, the THD of the A-phase and C-phase currents of CTF-QPR were 0.484% and 0.689%, respectively, while those of alpha-beta QPR were 0.740% and 0.953%. CTF-QPR was about 34.6% and 27.7% lower than alpha-beta QPR, respectively.

[0178] Simulation results show that, under ideal synchronous sampling conditions, the SCR step dynamics of CTF-QPR and alpha-beta QPR are approximately coincident, with the maximum difference in peak values ​​between the two being approximately 1.7e-3 pu. Compared to dq PI, CTF-QPR reduces peak current tracking error under multiple SCR step conditions.

[0179] Finally, it should be noted that the above descriptions are merely preferred embodiments of the present invention and are not intended to limit the present invention. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art can still modify the technical solutions described in the foregoing embodiments or make equivalent substitutions for some of the technical features. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the protection scope of the present invention.

Claims

1. A coordinateless digital control method for a three-phase network converter, characterized in that, Includes the following steps: Collect or acquire three-phase phase variables, including the three-phase filter inductor current i f,abc and three-phase output voltage u o,abc , where abc represents the physical three-phase phase change; The instantaneous active power p is calculated directly based on the three-phase phase variables. e and instantaneous reactive power q e The average active power P is obtained after filtering. e and average reactive power Q e ; Based on the active power synchronization loop of the virtual synchronous generator, the virtual angular frequency ω and power angle θ are generated according to the active power reference value, average active power, virtual inertia and damping coefficient. Based on the reactive power-voltage loop, a voltage amplitude reference E is generated according to the reactive power reference value, average reactive power, rated voltage amplitude, and regulator parameters. The three-phase bridge arm side electromotive force reference is directly generated based on the voltage amplitude reference E and the power angle θ. ; Using phase variable virtual admittance, based on the three-phase bridge arm side electromotive force reference... With three-phase output voltage u o,abc The difference generates a three-phase filter inductor current reference. ; The CTF-QPR is a coordinate-transform-free quasi-proportional resonant current regulator that operates with phase-channel consistency and directly references the three-phase filter inductor current. With the three-phase filter inductor current i f,abc The phase current error generates the three-phase inner loop control action u. c,abc ; The three-phase output voltage u o,abc With the three-phase inner loop control action u c,abc Add them together to generate the three-phase bridge arm voltage command. Based on this, modulation signals and PWM drive signals are generated.

2. The coordinate-free digital control method for a three-phase network converter according to claim 1, characterized in that, In a three-phase phase variable converter, the model of the LC output filter is as follows: (1) (2) Among them, L f For the filter inductor, R f C is the parasitic resistance of the filter inductor. f For the filter capacitor, i f,abc For the three-phase filter inductor current, u e,abc For the bridge arm voltage, u o,abc For the three-phase output voltage, i l,abc This is the alternating current.

3. The coordinate-free digital control method for a three-phase network converter according to claim 2, characterized in that, The instantaneous active power p e Calculate using the following formula: (3)； Under three-phase three-wire conditions, the instantaneous reactive power q e Calculate using the following formula: (4)； The instantaneous active power p e and instantaneous reactive power q e The average active power P is obtained after passing through a first-order low-pass filter. e and average reactive power Q e .

4. The coordinate-free digital control method for a three-phase network converter according to claim 3, characterized in that, In real digital control scenarios where the phase current sampling chain inevitably has non-ideal factors such as gain error, sampling timing error or bias error, the digital controller is configured to adopt a physical three-phase coordinate-less digital control path. The digital controller dynamically generates a virtual angular frequency ω based on the active power-frequency of the virtual synchronous generator. (5) And the work angle θ is obtained by integration: (6) Where J is the virtual inertia, and D p ω is the damping coefficient, and ω0 is the rated angular frequency.

5. The coordinate-free digital control method for a three-phase network converter according to claim 4, characterized in that, The voltage amplitude reference E is generated by the reactive power regulator: (7) Where E0 is the rated voltage amplitude, K pv and K iv These are the proportional gain and integral gain of the reactive power regulator, respectively. Based on the voltage amplitude reference E and the power angle θ, the bridge arm side electromotive force reference is directly generated from the three-phase phase variables: (8) in, , , This electromotive force reference represents the internal voltage state before virtual admittance shaping and is not equivalent to the final bridge arm voltage command.

6. The coordinate-free digital control method for a three-phase network converter according to claim 5, characterized in that, The virtual admittance is defined as: (9) in, For virtual resistance, The three-phase filter inductor current is referenced as a virtual inductor. It is generated by the following formula: (10)。 7. The coordinate-free digital control method for a three-phase network converter according to claim 6, characterized in that, The inner current loop uses a coordinate-transform-free QPR regulator to directly adjust the three-phase phase current error phase by phase. (11) Three-phase bridge arm voltage command for: (12) The continuous-domain transfer function of the QPR regulator is: (13) For proportional gain, For resonant gain, For cutoff bandwidth, It is the rated electrical angular frequency or the QPR resonant center frequency.

8. The coordinate-free digital control method for a three-phase network converter according to claim 7, characterized in that, Adopted For a Tustin method discrete QPR regulator with predistortion frequency, let: (14) And adopt: (15) get: (16) in: (17) (18) (19) In a digital controller, the above can be... It is implemented using a second-order IIR structure or an equivalent difference equation, and the same discrete controller coefficients are used for the three-phase error channels a, b, and c. Each phase channel maintains independent historical inputs, historical outputs, or state variables, and completes state updates within the same sampling period or control interruption period.

9. A coordinateless transformation digital control system for a three-phase network converter according to any one of claims 1-8, characterized in that, include: The sampling unit is used to acquire three-phase output voltage, three-phase filter inductor current, DC bus voltage, and optional temperature, protection, and status parameters. The sampling chain non-ideal parameter recording unit stores at least one of the gain error, sampling timing error, and bias error of the phase current sampling chain, serving as a reference for the configuration of the coordinate-transform-free digital control path. The phase variable power calculation unit directly calculates instantaneous active power and instantaneous reactive power based on the three-phase output voltage and the three-phase filter inductor current. The network synchronization unit is used to generate virtual angular frequency and power angle based on the active power reference value and the average active power; The voltage amplitude generation unit is used to generate a voltage amplitude reference based on the reactive power reference value and the average reactive power. The three-phase electromotive force reference generation unit is used to generate a three-phase bridge arm side electromotive force reference based on the power angle and voltage amplitude reference. The phase variable virtual admittance unit is used to generate a three-phase current reference based on the three-phase bridge arm side electromotive force reference and the three-phase output voltage. The phase-by-phase CTF-QPR current regulation unit is used to generate three-phase control actions based on the three-phase current reference and the three-phase filter inductor current. The bridge arm voltage command generation unit is used to add the three-phase output voltage feedforward and the three-phase control action to generate the three-phase bridge arm voltage command. The modulation unit is used to generate PWM drive signals according to the three-phase bridge arm voltage command.

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