Wideband phase flattening loaded line phase shifter

Abstract

The application discloses a wideband phase flattening loaded line phase shifter and belongs to the technical field of microwave and radio frequency devices. The phase shifter comprises a main transmission line, a quarter-wavelength short-circuit resonant branch loaded at the center position of the main transmission line, and two open-circuit branches symmetrically loaded at the two ends of the main transmission line; each branch is connected or disconnected through a switching device, so that the circuit is switched between a reference state and a phase shifting state, and a required differential phase shift is realized. The application further proposes a parameter design method based on group delay compensation, the characteristic impedance of the short-circuit resonant branch is adjusted, so that the phase shifter obtains a flat differential phase shift response in a target frequency band. Compared with a traditional multi-stage cascaded digital phase shifter, the application can reduce the number of phase shifting units and the complexity of a switching network, has the advantages of simple structure, compact size, low insertion loss and high wideband phase flatness, and is suitable for systems such as phased array antennas, beam forming networks and wideband radio frequency front ends.

Description

Technical Field

[0001] This invention discloses a broadband phase-flattened linear phase shifter, belonging to the field of microwave and radio frequency technology. Background Technology

[0002] Loaded linear phase shifters achieve the target phase shift by altering the equivalent electrical length of the transmission line through the addition of reactive elements or switchable stubs. They offer advantages such as simple structure and low insertion loss. However, traditional loaded linear phase shifters typically only satisfy the target phase shift and impedance matching conditions at the center frequency. When the operating frequency deviates from the center frequency, the dispersion characteristics between the reference state and the phase-shifted state differ, resulting in inconsistent group delay responses and phase errors that vary with frequency. Therefore, loaded linear phase shifters generally suffer from inherent bandwidth limitations. To improve the operating bandwidth of loaded linear phase shifters, existing technologies typically employ methods such as adding loaded stubs, using complex switching networks, or constructing special transmission line structures to improve phase response. While these methods can extend bandwidth or improve phase flatness to some extent, they often lead to increased circuit complexity, a larger number of switches, and increased insertion loss.

[0003] Therefore, achieving a flat differential phase shift response over a wide bandwidth while maintaining a simple and compact circuit structure remains a pressing technical problem in the design of loaded linear phase shifters. Furthermore, establishing an analytical design method for broadband phase flattening, and controlling the group delay mismatch between the reference state and the phase-shifted state through a limited number of key parameters to reduce phase error within the target bandwidth, is also a crucial issue that urgently needs to be addressed in this field. Summary of the Invention

[0004] The purpose of this invention is to address the shortcomings of the aforementioned background technology by providing a broadband phase-flattened loaded linear phase shifter and its design method. This aims to optimize the characteristic impedance of the short-circuit resonant branch through group delay compensation and obtain a flat differential phase shift response over a wide bandwidth. It also solves the technical problems of high circuit complexity and large broadband phase error caused by the reliance on repeated simulation optimization in traditional loaded linear phase shifters.

[0005] To achieve the above objectives, the present invention adopts the following technical solution:

[0006] A broadband phase-flattened linear phase shifter includes: a main transmission line, a quarter-wavelength short-circuit resonant branch, and two open-circuit branches. When the short-circuit resonant branch is applied at the center of the main transmission line, the phase shifter operates in the reference state. When the two open-circuit branches are symmetrically applied at both ends of the main transmission line, the phase shifter operates in the phase-shifting state. The electrical lengths of the two open-circuit branches are determined based on the target differential phase shift and impedance matching conditions. The characteristic impedance of the short-circuit resonant branch is determined based on the target of keeping the differential phase shift constant within the target frequency band.

[0007] As a further optimization scheme for the broadband phase flattening loaded linear phase shifter, the short-circuit resonant branch is loaded at the center of the main transmission line through the first switch, and the two open-circuit branches are symmetrically loaded at both ends of the main transmission line through the second and third switches, respectively.

[0008] As a further optimization scheme for the broadband phase-flattened linear phase shifter, the electrical lengths of the two open branches are determined based on the target differential phase shift and impedance matching conditions. Specifically, the target differential phase shift is substituted into... Calculate the electrical length of the main transmission line at the center frequency f0. ,Will Substitute the impedance matching condition Calculate the input susceptance of the open-circuit branch. Then, based on the given characteristic impedance of the open-circuit branch, the electrical length of the open-circuit branch is calculated, where, For differential phase shift, This is the port impedance.

[0009] As a further optimization scheme for the broadband phase-flattened loaded linear phase shifter, the characteristic impedance of the short-circuit resonant branch is determined based on the goal of keeping the differential phase shift constant within the target frequency band. Specifically, the optimal value of the characteristic impedance of the short-circuit resonant branch is calculated with the goal of minimizing the maximum phase error within the target frequency band.

[0010] As a further optimization scheme for the broadband phase-flattened loaded linear phase shifter, with the goal of minimizing the maximum phase error within the target frequency band, the optimal value of the characteristic impedance of the short-circuit resonant branch is calculated. Specifically, the candidate frequency points where the delay of the reference state group and the delay of the phase-shifted state group are equal are traversed within the target frequency band. The maximum phase error at the candidate frequency points under the given characteristic impedance of the short-circuit resonant branch is calculated. The given characteristic impedance of the short-circuit resonant branch when the maximum phase error is minimized is taken as the optimal value of the characteristic impedance of the short-circuit resonant branch.

[0011] Compared with the prior art, the present invention has the following beneficial effects:

[0012] 1. This invention introduces a quarter-wavelength short-circuit resonant branch in the reference state. This branch is equivalent to an open circuit at the center frequency and does not affect the target phase shift and impedance matching conditions at the center frequency. At frequencies deviating from the center frequency, the frequency-dependent input susceptance of this branch can regulate the group delay response of the reference state, thereby compensating for the dispersion mismatch between the reference state and the phase-shifted state, reducing the phase error in the wideband, and providing a new degree of freedom for wideband phase flattening.

[0013] 2. This invention can achieve broadband phase flattening without the need for multi-stage phase shifting unit cascades, complex switching networks, or additional broadband matching circuits. Therefore, it has the advantages of simple circuit structure, clear control logic, compact size, and low insertion loss.

[0014] 3. The design method proposed in this invention has a clear analytical design process. Based on group delay compensation design, it optimizes the characteristic impedance of the short-circuit resonant branch to enable the phase shifter to obtain a flat differential phase shift response in the target frequency band. It can determine the main circuit parameters according to the target phase shift, center frequency and target frequency band, reducing the dependence on repeated simulation trial and error. It is suitable for phase control circuits in phase array antennas, beamforming networks, radar front-ends and broadband radio frequency systems. Attached Figure Description

[0015] Figure 1 This is a topology diagram of the loaded linear phase shifter proposed in this invention.

[0016] Figure 2 This is a topology diagram of the loaded linear phase shifter proposed in this invention operating in the reference state.

[0017] Figure 3 This is a topology diagram of the loaded linear phase shifter proposed in this invention operating in the phase-shifting state.

[0018] Figure 4 This is a diagram showing the simulation and measured amplitude response results of a 22.5-degree phase shifter provided in one embodiment of the present invention.

[0019] Figure 5 This is a diagram showing the simulation and measured phase response results of a 22.5-degree phase shifter provided in one embodiment of the present invention.

[0020] Figure 6 This is a diagram showing the simulation and measured amplitude response results of a 45-degree phase shifter provided in one embodiment of the present invention.

[0021] Figure 7 This is a diagram showing the simulation and measured phase response results of a 45-degree phase shifter provided in one embodiment of the present invention.

[0022] Explanation of the labels in the diagram: TL m Main transmission line, TL tShort-circuit resonant branch, TL s1 First open branch road, TL s2 The second open branch circuit includes D1, the first switch, D2, the second switch, and D3, the third switch. Detailed Implementation

[0023] To make the objectives, technical solutions, and beneficial effects of this invention clearer, the loaded linear phase shifter of this invention will be described in detail below with reference to the accompanying drawings and specific embodiments. It should be understood that the specific embodiments described herein are for illustrative purposes only and do not constitute a limitation thereof.

[0024] I. Circuit Topology and Operating Status

[0025] See Figure 1 The broadband phase-flattened loading linear phase shifter provided by this invention includes a main transmission line TL. m A short-circuit resonant branch TL t First open branch TL s1 Second open branch TL s2 And switching devices for controlling the connection or disconnection of each branch.

[0026] Main transmission line TL m The characteristic impedance is The electric length is Short-circuit resonant branch TL t Connected to the main transmission line TL m At the center of , its characteristic impedance is The electric length is And at the center frequency The first open-circuit branch TL is designed to be a quarter wavelength. s1 Second open branch TL s2 Symmetrically connected to the main transmission line TL m At both ends of the position, the characteristic impedance is The electric length is .

[0027] The switching device includes a first switch D1, a second switch D2, and a third switch D3. The first switch D1 is connected to the short-circuit resonant branch TL. t With main transmission line TL m Between them, the second switch D2 is connected to the first open branch TL. s1 With main transmission line TL m Between, the third switch D3 is connected to the second open branch TL. s2 With main transmission line TL m Between. The three switches are controlled by a set of complementary bias voltages, causing the phase shifter to switch between reference state and phase-shifted state.

[0028] Reference status such as Figure 2 As shown. In this state, the first switch D1 is turned on, short-circuiting the resonant branch TL. t Loaded to main transmission line TL m The center position; the second switch D2 and the third switch D3 are open, and the first open branch TL s1 Second open branch TL s2 Not loaded onto the main transmission line TL m At this point, the circuit forms a T-shaped loading structure.

[0029] Phase shift state as Figure 3 As shown. In this state, the first switch D1 is open, short-circuiting the resonant branch TL. t Not loaded onto the main transmission line TL m The second switch D2 and the third switch D3 are turned on, and the first open branch TL is open. s1 Second open branch TL s2 Symmetrical loading onto main transmission line TL m Both sides. At this time, the circuit forms a similar shape. Type structure.

[0030] By switching between a reference state and a phase-shifting state, the phase shifter can generate the desired differential phase shift. If the insertion phase of the reference state is... The insertion phase of the phase-shifted state is Then the differential phase shift can be expressed as: .

[0031] II. Theoretical Analysis and Design Parameter Derivation

[0032] To clarify the working principle and establish a clear design methodology, the following theoretical analysis will be conducted on the two working states respectively.

[0033] (1) Reference state

[0034] The characteristics of the reference state can be described by its ABCD matrix and scattering parameters, and its input reflection coefficient S 11 With forward transmission coefficient S 21 The expression is as follows:

[0035]

[0036]

[0037] In equations (1a) and (1b), , , These are the elements in the ABCD matrix; The port impedance is 50 Ω. The electrical length of the main transmission line; For short-circuit resonant branch TL t The input susceptance has a value of B. t = -cotθ t / Z t θ t and Z t TL t The electric length and characteristic impedance. ABCD matrix parameters. , , , for:

[0038]

[0039]

[0040]

[0041] Furthermore, the return loss in the reference state Insertion loss and inserted phase It can be represented as:

[0042]

[0043]

[0044] In the formula, the auxiliary variables P and Q are defined as follows:

[0045]

[0046]

[0047] At the center frequency f0, the short-circuit resonant branch TL t electric length θ t Since the angle is 90°, its input susceptance B t = 0. Substituting this condition into equations (1a), (1b), and (3b), we can obtain the amplitude and phase response characteristics at f0:

[0048]

[0049]

[0050] Here, This represents the electrical length of the main transmission line at the center frequency f0. Two key conclusions can be drawn from formulas (5a) and (5b). First, at the center frequency f0, the circuit achieves perfect impedance matching and lossless transmission; second, the insertion phase in this state is entirely determined by the electrical length of the main transmission line.

[0051] (2) Phase shift state

[0052] The circuit structure in phase-shifted state is as follows Figure 3 As shown. Identical open-circuit stubs TL are symmetrically loaded at both ends of the main transmission line. s1 and TL s2 Its input susceptance B s = tan(θ s ) / Z s θ s and Z s They are respectively open branch TL s1 and TL s2 The electrical length and characteristic impedance of the structure are given. The ABCD matrix of this structure is:

[0053]

[0054] The reflection coefficient S derived from it 33 With transmission coefficient S 43 for:

[0055]

[0056]

[0057] The insertion phase φ of this state 43 From the transmission coefficient S 43 The argument is obtained as follows:

[0058]

[0059] To achieve perfect impedance matching at f0 in the phase-shifted state, there must be no reflection at its input port, i.e., the reflection coefficient S must be set to... 33 The value is zero. This leads to a key design constraint, establishing the required susceptance B for the open-circuit branch. s Electrical length θ of the main transmission line at center frequency f0 m0 Relationship:

[0060]

[0061] Select the characteristic impedance Z of the open branch s Then, its electric length θ at f0 can be determined. s0 To simplify design and analysis, Z s Typically, a fixed value of 30 Ω is used, representing the electrical length of the open branch at f0. for:

[0062]

[0063] Substituting the matching condition at f0, i.e., formula (9), into the transmission coefficient expression, i.e., formula (7b), the insertion phase φ of the phase-shifted state at f0 can be derived. 43 :

[0064]

[0065] (3) Broadband phase flattening principle

[0066] The principle of broadband phase flattening in this invention lies in introducing a quarter-wavelength short-circuit resonant branch TL into the reference state of the loaded linear phase shifter. t and the characteristic impedance of the short-circuit resonant branch. As a key parameter for regulating the time delay response of the reference state group. By analyzing... The design can compensate for the group delay mismatch between the reference state and the phase-shifted state, so that the differential phase shift between the two operating states remains flat over a wide bandwidth.

[0067] Since the reference state and the phase-shifted state have different equivalent transmission structures, their insertion phase changes with frequency are denoted as follows: and For broadband phase shifters, the ideal goal is to achieve differential phase shift. In the target frequency band The internal content should be kept as constant as possible, that is:

[0068]

[0069] In equation (12), For the center frequency, This refers to the design differential phase shift at the center frequency. If the operating frequency deviates from the center frequency, the insertion phase of the reference state and the phase-shifted state will exhibit different trends with frequency. Consequently, the differential phase shift will shift with frequency, resulting in a phase error. This phase error can be defined as:

[0070]

[0071] Therefore, the essence of broadband phase flattening is to reduce the phase error within the target frequency band, especially to reduce the maximum phase error within the target frequency band.

[0072] To determine a suitable characteristic impedance The present invention aims to minimize the maximum phase error within the target frequency band.

[0073]

[0074] In equation (14), The characteristic impedance value that minimizes the maximum phase error within the target frequency band.

[0075] Furthermore, when the phase error reaches an extreme value within the target frequency band, the corresponding frequency point satisfies that the first derivative of the differential phase shift with respect to frequency is zero, meaning that the group delay of the reference state and the phase-shifted state are equal:

[0076]

[0077] lower frequency and the above side frequency The phase error within the determined target frequency band should also consider the frequency points within the target frequency band that satisfy the above-mentioned group delay equality condition. By comparing the phase errors at these candidate frequency points, a given... The maximum phase error under the condition; then through optimization This allows us to obtain the optimal broadband phase flattening design parameters.

[0078] III. Specific Implementation Cases

[0079] 1. General Implementation Conditions

[0080] In the following exemplary embodiments, each phase shifter is designed, fabricated, and tested at a center frequency f0 = 2.4 GHz. A Rogers RO4003C substrate with a thickness of 0.508 mm and a relative permittivity of 3.55 is selected as the dielectric substrate for circuit fabrication. Surface-mount diodes are used as switching elements. To balance performance under different operating conditions, targeted selection is implemented: In the reference state, to minimize the impact of junction capacitance on the signal path, two SMP1345-079LF PIN diodes are connected in series to implement the first switch, aiming to improve the isolation of the switching elements. The junction capacitance of the two SMP1345-079LF PIN diodes connected in series is 0.15 pF. In the phase-shifted state, to reduce insertion loss, SMP1320-040LF PIN diodes with lower series resistance are selected to implement the second and third switches. The typical series resistance of the SMP1320-040LF PIN diode is 0.75 Ω @ 10 mA. The DC bias network includes an RF choke inductor L = 100 nH, a DC blocking capacitor C = 51 pF, a current-limiting resistor R = 100 Ω, and an RF ground path. This is achieved by applying a set of complementary DC bias voltages U and Where 0V represents low level cutoff and 3V represents high level conduction, the switching on and off of the corresponding diodes is controlled, thereby realizing reliable switching of the circuit between the reference state and the phase-shifting state.

[0081] Based on the design process established by the aforementioned theoretical analysis, the specific implementation of this invention is carried out according to the following steps:

[0082] First, based on the target performance indicators such as phase shift, center frequency, and bandwidth, the ideal parameters of each transmission line, such as electrical length and characteristic impedance, are calculated using the formulas in this paper.

[0083] Subsequently, a complete simulation model was established in ANSYS HFSS full-wave electromagnetic simulation software, which included the above-mentioned ideal transmission line model, PIN diode SPICE model and actual bias circuit.

[0084] Through simulation optimization, the theoretical dimensions are fine-tuned to compensate for the effects of distributed parameters, device parasitic parameters, and mutual coupling, and finally the physical dimensions of all transmission lines that meet the performance indicators are determined.

[0085] Finally, the circuit is fabricated and tested based on the optimized layout. Specific Implementation

[0086] Example 1: 22.5° Phase Shifter

[0087] To achieve the target differential phase shift φ of 22.5° s = 22.5°, so parameter design is performed first. Based on the relationship φ s = π - 2θ m0 The electrical length θ of the main transmission line at the center frequency f0 is calculated. m0 = 78.75°. To achieve perfect impedance matching at the center frequency for the phase-shifted Pi-type structure, θ is... m0 Substituting into the matching condition formula (9), the required input susceptance B for the open-circuit stub is calculated. s The characteristic impedance Z of the open-circuit stub is 0.008 S. s = 30 Ω, thus determining its electrical length θ m0 ≈13.5°. To achieve a flat phase response over a wide bandwidth, the short-circuit stub TL was optimized with the goal of minimizing the maximum phase error within the target bandwidth. t Characteristic impedance Z t ≈ 70 Ω.

[0088] Based on the above theoretical parameters, modeling and optimization were performed in electromagnetic simulation software, and the physical dimensions of each microstrip transmission line were determined as follows: W0 = 1.10 mm, L0 = 31.40 mm, W s = 2.35 mm, L s = 2.60 mm, W t = 0.45 mm, L t =15.28 mm. The overall circuit layout dimensions of a 22.5-degree phase shifter are 31.5 mm × 24.0 mm.

[0089] The prototype 22.5° phase shifter was tested, and the results are as follows: Figure 4 and Figure 5 As shown, within the frequency range of 1.6 GHz to 3.2 GHz, with a relative bandwidth of 66.7%, the return loss at its input and output ports is better than 9.5 dB. The average insertion losses in the reference state and phase-shifted state are 0.60 dB and 0.51 dB, respectively. The differential phase shift remains constant throughout the entire operating band, with a maximum phase error of only ±2.79°. The test results verify the good matching, low loss, and high phase flatness of this design in the ultra-wideband environment.

[0090] Example 2: 45° Phase Shifter

[0091] The target differential phase shift φ in this embodiment s It is 45°. Based on φ s = π - 2θ m0 θ is calculated m0 = 67.5°. Similarly, the open-circuit stub characteristic impedance Z is selected. s = 30 Ω, its electrical length θ is calculated according to the impedance matching condition. s0 ≈26.57°. To achieve a flat phase response over a wide bandwidth, the characteristic impedance Z of the short-circuit stub is obtained. t ≈ 29.25 Ω.

[0092] The physical dimensions of each microstrip transmission line were determined through simulation optimization as follows: W0 = 1.10 mm, L0 = 27.8 mm, W s =2.35 mm, L s = 4.88 mm, W t = 1.82 mm, L t = 15.06 mm. The core circuit dimensions of the designed and implemented 45° phase shifter are 28.0 mm × 24.0 mm.

[0093] Experimental results show that within the 1.8 GHz to 3.0 GHz frequency band, with a relative bandwidth of 50%, the average insertion loss of the phase shifter in both the reference and phase-shifted states is 0.7 dB, the return loss is better than 9.8 dB, and the in-band phase error is controlled within ±4.64°. Figure 6 and Figure 7 As shown.

[0094] The examples described above are merely preferred embodiments of the present invention, but the embodiments of the present invention are not limited to the above embodiments. For those skilled in the art, several improvements can be made without departing from the principle of the present invention, and these improvements should also fall within the protection scope of the present invention.

Claims

1. A broadband phase-flattened linear phase shifter, characterized in that, include: The system comprises a main transmission line, a quarter-wavelength short-circuit resonant branch, and two open-circuit branches. When the short-circuit resonant branch is applied at the center of the main transmission line, the phase shifter operates in a reference state. When the two open-circuit branches are symmetrically applied at both ends of the main transmission line, the phase shifter operates in a phase-shifting state. The electrical lengths of the two open-circuit branches are determined based on the target differential phase shift and impedance matching conditions. The characteristic impedance of the short-circuit resonant branch is determined based on the target of maintaining a constant differential phase shift within the target frequency band.

2. The broadband phase-flattening loaded linear phase shifter according to claim 1, characterized in that, The short-circuit resonant branch is applied to the center of the main transmission line via the first switch, and the two open-circuit branches are symmetrically applied to both ends of the main transmission line via the second and third switches, respectively.

3. The broadband phase-flattening loaded linear phase shifter according to claim 2, characterized in that, The electrical lengths of the two open-circuit branches are determined based on the target differential phase shift and impedance matching conditions, specifically by substituting the target differential phase shift into... Calculate the electrical length of the main transmission line at the center frequency f0. ,Will Substitute the impedance matching condition Calculate the input susceptance of the open-circuit branch. Then, based on the given characteristic impedance of the open-circuit branch, the electrical length of the open-circuit branch is calculated, where, For differential phase shift, This is the port impedance.

4. The broadband phase flattening loaded linear phase shifter according to claim 3, characterized in that, The characteristic impedance of the short-circuit resonant branch is determined based on the objective of keeping the differential phase shift constant within the target frequency band. Specifically, the optimal value of the characteristic impedance of the short-circuit resonant branch is calculated with the goal of minimizing the maximum phase error within the target frequency band.

5. The broadband phase-flattening loaded linear phase shifter according to claim 4, characterized in that, The optimization objective is to minimize the maximum phase error within the target frequency band. Specifically, the optimal value of the characteristic impedance of the short-circuit resonant branch is calculated by: traversing candidate frequency points where the delay of the reference state group and the delay of the phase-shifted state group are equal within the target frequency band; calculating the maximum phase error at the candidate frequency points under a given characteristic impedance of the short-circuit resonant branch; and taking the given characteristic impedance of the short-circuit resonant branch when the maximum phase error is minimized as the optimal value of the characteristic impedance of the short-circuit resonant branch.

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