Dual-frequency co-boresight phased array antenna and configuration method thereof

Abstract

The application discloses a dual-frequency common-aperture phased array antenna and a configuration method thereof, and belongs to the technical field of microwave antennas. The antenna comprises a dielectric substrate, radiation metal layers and ground metal layers separately arranged on two surfaces of the dielectric substrate, a feed network, and a plurality of X-band and Ku-band microstrip radiation units. The X-band radiation unit is a ring-shaped microstrip radiator with a loading slot, and the Ku-band radiation unit is a microstrip patch embedded in the center of the ring-shaped structure. An isolation gap is arranged between the two units to form a common-aperture nested layout. The configuration method comprises determining frequency band requirements, arranging a nested structure, determining parameters of the ring-shaped structure and the loading slot, determining isolation and feed parameters, and configuring a phase control network. The application adopts a double-layer PCB process, the size of the X-band radiation body is compressed to about 8mm-9mm, and the X-band radiation body can be arranged in a period of about 10mm array elements, thereby solving the problem of mismatching of the dual-frequency array element spacing, the port isolation is better than-15dB, and the application is suitable for radar detection, satellite communication and unmanned platform load.

Description

Technical Field

[0001] This invention belongs to the field of microwave antenna technology, specifically relating to the miniaturization and multi-band common aperture integration technology of phased array antennas, and particularly to a dual-band common aperture phased array antenna and its configuration method. Background Technology

[0002] In fields such as satellite communication, radar detection, and unmanned platform data links, to improve system functional density and adapt to limited installation space, antenna systems often need to simultaneously support X-band (7.9–9 GHz) and Ku-band (12.25–12.75 GHz) operation within the same aperture. Existing dual-band or multi-band phased array antenna implementations mainly suffer from the following shortcomings:

[0003] First, the large array size and difficulty in achieving a common aperture present challenges. X-band wavelengths are relatively long, requiring large conventional half-wavelength microstrip patches (approximately 18mm–21mm). In contrast, to avoid high-end grating lobes in the Ku-band, the element center-to-center distance needs to be controlled within approximately 11mm. If a separate dual-band array scheme is adopted, the total area of ​​the two independent arrays increases exponentially, making it difficult to meet the installation requirements of a miniaturized platform. If conventionally sized X-band patches are used and arranged coplanarly with Ku-band elements at equal intervals, the X-band patches would compress the space of adjacent elements, making proper arrangement impossible. How to simultaneously accommodate the radiation structures of two frequency bands within a compact period of approximately 10mm is the primary problem facing existing technologies.

[0004] Second, the bandwidth coverage is insufficient. The relative bandwidth of the X-band (7.9–9 GHz) is approximately 13%, and it also needs to cover the Ku-band (12.25–12.75 GHz). The impedance bandwidth of a typical single-layer microstrip patch antenna is usually only 3%–5%, and even with a thick substrate or simple loading, it is difficult to completely cover both of these widebands simultaneously.

[0005] Third, the processing technology is complex and costly. Multi-layer lamination solutions require multiple pressings and high precision in aligning interlayer vias, resulting in low yield. Metal cavity solutions require precision CNC machining, which is costly and not conducive to mass production. None of the above solutions are suitable for engineering applications requiring low cost, lightweight design, and mass production.

[0006] Fourth, the element spacing is constrained by the Ku band. The half-wavelength corresponding to the high end of the Ku band, 12.75 GHz, is 11.5 mm. To avoid grating lobes during scanning, the element center distance needs to be controlled within half a wavelength. However, the conventional patch size in the X band is close to or exceeds this spacing, creating a structural contradiction.

[0007] Fifth, there is a lack of ability to control the annular focal line. In near-field focusing applications, it is necessary to form an annular focal line or annular focal spot by the array radiation energy in a specified spatial plane. In existing technical solutions, phase compensation is usually for point focusing or conventional beam scanning, and there is a lack of a systematic configuration method for annular focal lines. Summary of the Invention

[0008] The purpose of this invention is to provide a dual-frequency common-aperture phased array antenna and its configuration method, which solves the problems of mismatched spacing between dual-frequency array elements, insufficient bandwidth of microstrip patches, complex common-aperture structure, large array thickness, high cost, and difficulty in achieving ring focal line control in the prior art.

[0009] A dual-frequency common-aperture phased array antenna has a basic structure including a dielectric substrate and a radiating metal layer and a grounding metal layer respectively covered on the first and second surfaces of the substrate. At the same time, a feed network is arranged on the surface of the substrate to realize the distribution and transmission of radio frequency signals.

[0010] On the radiating metal layer, several microstrip radiating elements operating in the X-band and several microstrip radiating elements operating in the Ku-band are arranged. The X-band radiating elements use a ring-shaped microstrip radiator as the core radiating structure. This ring-shaped radiator has an outer boundary and an inner boundary, forming a ring-shaped metal region between the outer and inner boundaries, which constitutes a ring-shaped current path. The Ku-band radiating elements use a microstrip patch as the radiating body, nested within the central region enclosed by the X-band ring-shaped radiator. An isolation gap is reserved between the ring-shaped radiator and the patch, ensuring that the radiators of the two frequency bands are physically coplanar but spatially separated. In terms of array arrangement, the X-band elements are arranged with a first element spacing, and the Ku-band elements are arranged with a second element spacing, with the second element spacing being smaller than the first element spacing, allowing the two frequency bands to coexist on the same array plane with different electrical spacings.

[0011] This nested common-aperture architecture allows radiators in two frequency bands to share the aperture area within the same physical period, reducing the total array area and providing a structural basis for the miniaturization of dual-frequency arrays.

[0012] Furthermore, based on the aforementioned annular radiator, the outer boundary side length is set to 7.0mm–9.8mm, and the inner boundary side length is set to 4.5mm–6.5mm, so that the perimeter of the annular current path satisfies the resonance condition near the center frequency of the X-band. To extend the impedance bandwidth of the X-band, loading slots are symmetrically opened on the four sides of the annular metal region. The slot shape can be rectangular, U-shaped, or E-shaped, with a slot width of 0.2mm–0.4mm and a slot length of 1.5mm–3.5mm.

[0013] The loading slot causes the surface current to diffract at the slot edge and extends the propagation path, thereby introducing an additional resonant mode with a frequency slightly higher than the dominant mode. The dominant mode and the additional mode overlap and superimpose to form a dual-resonance synthesis effect, enabling a continuous low return loss response in the X-band within the range of 7.9–9 GHz. This scheme achieves wideband coverage of the X-band while compressing the physical size of the radiator through the synergistic effect of the ring path extension and the dual-resonance synthesis of the loading slot.

[0014] Furthermore, the Ku-band patch nested in the central region of the ring radiator has a side length of 3.0mm to 5.0mm, used to generate half-wavelength resonance in the 12.25 to 12.75 GHz frequency band. To further extend the Ku-band bandwidth, perturbation structures such as cross slots, narrow central slots, chamfered edges, or symmetrical open slots are formed on the patch, with slot widths of 0.2mm to 0.5mm and slot lengths of 0.5mm to 2.0mm.

[0015] The perturbation structure creates additional current paths on the patch, generating a secondary resonant mode with a frequency close to but slightly offset from the primary mode. When the frequency spacing between the primary and secondary modes is appropriate, the two modes superimpose to form a wider impedance bandwidth. This scheme effectively extends the bandwidth coverage of the Ku band without increasing the physical size of the patch through the perturbation structure on the patch surface.

[0016] Furthermore, the isolation gap between the X-band ring radiator and the Ku-band patch has a width of 0.2mm to 0.8mm. This metal-free gap blocks the direct electromagnetic coupling path between the two radiators. In addition, metallized grounding vias are provided in the dielectric substrate between adjacent radiating elements. These vias extend from the grounding metal layer through the dielectric substrate to the radiating metal layer, forming a metal fence structure around each element to confine surface waves within the element and suppress their propagation to adjacent elements.

[0017] The synergistic effect of the isolation gap and the grounding via enables the port isolation of the two frequency bands to be better than -15dB, which meets the requirements for decoupling between channels when the dual frequencies are operating independently.

[0018] Furthermore, regarding power supply, the X-band power supply probe passes through the dielectric substrate from below the grounding metal layer, and the top of the inner conductor is electrically connected to the radiation edge of the X-band ring radiator. The offset distance of the power supply point relative to the center of the array element is 3.0mm to 4.2mm. The Ku-band power supply probe passes through the dielectric substrate in the same manner and is electrically connected to the Ku-band patch. The offset distance of the power supply point is 0.5mm to 1.4mm.

[0019] The offset position of the feed point is determined based on the impedance matching principle: the input impedance gradually decreases as it moves away from the center along the radiation edge. By adjusting the offset, the input impedance of each frequency band at the target frequency is made close to the 50-ohm system impedance. The two probe feeds are physically independent, providing mutually non-interfering excitation channels for the X-band and Ku-band respectively, laying the foundation for subsequent dual-frequency independent phase shifting and beam control.

[0020] Furthermore, the dielectric substrate is selected from low-loss high-frequency microwave dielectric materials with a dielectric constant of 4.0 to 10.5 and a loss tangent of 0.002 to 0.004, and the substrate thickness is 0.5 mm to 2.0 mm. Rogers RO3006 with a dielectric constant of 6.15 or Rogers RO3010 with a dielectric constant of 10.2 are preferred.

[0021] A high dielectric constant shortens the equivalent wavelength in the dielectric, resulting in a corresponding reduction in the physical size of the radiator; a low loss tangent ensures low dielectric loss and high radiation efficiency in the high-frequency band; and the substrate thickness achieves a balance between bandwidth and surface wave suppression. This scheme, through the rational selection of dielectric material parameters, effectively compresses the radiator size while ensuring radiation efficiency, allowing the X-band radiator to be accommodated within the compact element period defined by the Ku-band.

[0022] Furthermore, at the array level, the X-band radiating elements and the Ku-band radiating elements are arranged in a 2×2, 8×8, or M×N rectangular grid, with the center distance between the array elements in the row direction and the column direction being equal, ranging from 8.5mm to 10.3mm.

[0023] This spacing is approximately 0.97 half-wavelengths of free space at the high end of the Ku band (12.75 GHz), which can suppress grating lobes during large-angle scanning; at the X band (9 GHz), it is approximately 0.23 half-wavelengths, with an electrical size much smaller than the wavelength, which is beneficial for wide-angle scanning. This array layout scheme allows the dual-frequency array to simultaneously meet the requirements of Ku-band grating lobe suppression and X-band large-angle scanning with the same element spacing, resolving the structural contradiction caused by the different element spacing requirements of the two frequency bands.

[0024] Furthermore, in terms of RF feedthrough and beam control, the first phase control network is connected to the X-band radiating element, and the second phase control network is connected to the Ku-band radiating element. The two networks are respectively composed of a phase shifter and a power divider network, providing independent phase gradient excitation for the array elements of each frequency band.

[0025] By independently setting the phase difference between adjacent array elements in each of the two frequency bands, the scanning function of the beams in the two frequency bands pointing in the same or different directions can be realized. This scheme completely decouples the beam scanning of the X band and the Ku band, and the beam pointing of the two frequency bands can be adjusted independently, improving the beam scheduling flexibility of the dual-frequency system.

[0026] Furthermore, the control unit can calculate the required phase compensation value for each array element based on the annular focal line equation, enabling the array to form an annular focal line or annular focal spot at a specified spatial location. The annular focal line equation describes the geometry and position of the target focal line in space, calculates the distance from the m-th row and n-th column array element to each sampling point on the annular focal line, and converts the corresponding path difference into a phase compensation value.

[0027] By switching different phase codebooks through the control unit, the antenna can be switched between near-field focusing mode and far-field beam scanning mode without changing its hardware structure. This solution endows the array with multi-mode operation capabilities, expanding the antenna's applicability in various scenarios such as near-field microwave imaging, near-field radiative heating, and long-distance communication.

[0028] Furthermore, the present invention also provides a configuration method for the above-mentioned dual-frequency common-aperture phased array antenna, including five steps: S1, determining the operating frequency range and array size of the X band (7.9~9GHz) and Ku band (12.25~12.75GHz) to provide constraint boundaries for subsequent parameter determination;

[0029] S2. Arrange a nested common aperture structure, place the Ku-band patch in the central region of the X-band ring radiator, and determine the array element center distance according to the high half wavelength of the Ku-band to suppress scanning grating lobes.

[0030] S3. Determine the outer and inner boundary dimensions of the ring radiator, and open loading slots on the four sides of the ring structure to introduce additional resonant modes, so that the X-band forms a dual-resonant synthesis to cover the target frequency band.

[0031] S4. Determine the width of the isolation gap and the feed point positions of the X-band and Ku-band feed probes to spatially separate the electric field distributions of the two frequency bands.

[0032] S5. Configure independent first phase control network and second phase control network for the two frequency bands respectively, and calculate the phase compensation value of each array element according to the ring focal line equation to realize the switching control of near-field focusing and far-field beam scanning.

[0033] This configuration method integrates frequency band requirements determination, nested structure arrangement, miniaturization parameter determination, isolation and feeding configuration, and phase control configuration into a systematic process. All parameters of the dual-band common-aperture phased array antenna can be determined and fabricated using only a two-layer PCB process.

[0034] The beneficial effects of this invention are as follows:

[0035] (1) Dual-band common aperture, high array area utilization. The X-band radiator is designed as a ring structure, and the Ku-band radiator is nested in the central region of the ring. The radiators of the two bands are arranged coplanarly within the same physical unit period, without the need for each band to occupy a separate aperture area. Compared with the separate two independent array scheme, the present invention reduces the total array area.

[0036] (2) High miniaturization, adaptable to compact array element spacing. A high dielectric constant substrate is used to reduce the equivalent wavelength. At the same time, the X-band radiator is designed as a ring structure to extend the surface current path. Combined with the loading slot, the resonant size is further compressed so that the outer boundary length of the X-band radiator can be accommodated within the array element period defined by the high end of the Ku band.

[0037] (3) Superior wideband coverage capability. By setting a loading slot on the outer ring of the X-band radiator to introduce a secondary resonant mode, the main mode and the secondary mode of the X-band form a dual resonant synthesis, achieving wideband coverage of 7.9 to 9 GHz, and also taking into account the broadband coverage of the Ku band of 12.25 to 12.75 GHz.

[0038] (4) Simple structure, low cost, suitable for mass production. It can be achieved with only conventional double-layer PCB process. Compared with multi-layer LTCC, metal cavity or complex multi-layer stacking solutions, the processing difficulty and manufacturing cost are reduced. The thickness of the dielectric substrate cross section is 0.5mm to 2.0mm.

[0039] (5) High port isolation and independent controllability of dual frequencies. The X-band radiator and the Ku-band patch are spatially separated and the feed points are independent of each other. With the help of the intermediate isolation gap, the port isolation of the two frequency bands is better than -15dB, and the beam scanning of the two frequency bands can be controlled by the first phase control network and the second phase control network respectively.

[0040] (6) It has the ability to adjust the annular focal line. By using the phase compensation method based on the annular focal line equation in the control unit, an annular focal line or annular focal spot can be formed at a specified spatial position, and the near-field focusing and far-field beam scanning can be switched by switching the phase code table. Attached Figure Description

[0041] Figure 1 This is a schematic diagram of the nested dual-frequency radiation unit structure of the present invention;

[0042] Figure 2 This is a bottom view of the 4×4 array layout of the present invention;

[0043] Figure 3 This is a top view showing the detailed structure of the radiating unit of the present invention;

[0044] Figure 4 This is a schematic diagram of the cross-sectional structure of the double-layer PCB of the present invention.

[0045] In the diagram: 1. Dielectric substrate; 2. Radiation metal layer; 3. Grounding metal layer; 4. Feed network; 5. X-band microstrip radiating element; 6. X-band ring radiator; 7. Outer boundary; 8. Inner boundary; 9. Loading slot; 10. Ku-band microstrip radiating element; 11. Ku-band patch; 12. Cross slot; 13. Isolation gap; 14. Metallized grounding via; 15. X-band feed probe; 16. Ku-band feed probe; 17. Insulating via; 18. First phase control network; 19. Second phase control network; 20. Control unit; 21. Array. Detailed Implementation

[0046] The following is in conjunction with the appendix Figure 1 ~Attached Figure 4 The present invention will be further described in detail with reference to specific embodiments.

[0047] Example 1

[0048] like Figure 1 and Figure 4 As shown, the dual-band common-aperture phased array antenna of this embodiment adopts a double-layer PCB structure as its basic framework, including a dielectric substrate 1 and a radiating metal layer 2 and a grounding metal layer 3 respectively disposed on the first and second surfaces of the substrate. Radiation patterns, feed lines, and matching branches are fabricated on the radiating metal layer 2, while the grounding metal layer 3 is a continuous metal plane. A feed network 4 is arranged on the surface of the dielectric substrate 1 to realize the distribution and transmission of radio frequency signals. This double-layer PCB structure can be processed using only conventional board manufacturing and electroplating processes. Compared with multilayer LTCC stacked solutions or metal cavity solutions, the processing difficulty and manufacturing cost are reduced, and the cross-sectional thickness of the dielectric substrate 1 is only 0.5mm to 2.0mm, which is beneficial for the thin design of the antenna system.

[0049] On the radiating metal layer 2, several X-band microstrip radiating units 5 operating in the X-band (7.9–9 GHz) and several Ku-band microstrip radiating units 10 operating in the Ku-band (12.25–12.75 GHz) are disposed. The core radiating structure of the X-band radiating unit 5 is an X-band annular radiator 6, which has an outer boundary 7 and an inner boundary 8. An annular metal region is formed between the outer boundary 7 and the inner boundary 8, and the annular metal region forms a closed or open annular current path on the ring. The radiating body of the Ku-band radiating unit 10 is a Ku-band patch 11, which is nested in the central region enclosed by the X-band annular radiator 6. An isolation gap 13 is reserved between the annular radiator 6 and the patch 11, so that the radiators of the two frequency bands are physically coplanar but spatially separated.

[0050] The aforementioned nested common-aperture architecture allows the radiators of the two frequency bands to share the aperture area within the same physical period, eliminating the need for each frequency band to occupy a separate aperture area. Compared to the separate dual-band array scheme, this embodiment reduces the total array area. Since the X-band annular radiator 6 uses an annular shape instead of a solid patch, space is reserved in its central region to accommodate the Ku-band patch 11, thus achieving a common-aperture layout. At the array arrangement level, the X-band elements are arranged with a first element spacing, and the Ku-band elements are arranged with a second element spacing, with the second element spacing being smaller than the first element spacing, allowing the two frequency bands to coexist on the same array surface with different electrical spacings.

[0051] like Figure 1 and Figure 3 As shown, the X-band radiator 6 is located on the first surface of the dielectric substrate 1 and is a square annular microstrip structure surrounded by an outer boundary 7 and an inner boundary 8. The side length of the outer boundary 7 is 7.0 mm to 9.8 mm, preferably 8.6 mm to 9.2 mm; the side length of the inner boundary 8 is 4.5 mm to 6.5 mm, preferably 5.0 mm to 5.8 mm. This annular structure extends the effective path of surface current, enabling it to resonate in the X-band even with a physical size smaller than a conventional half-wavelength patch. The size of a conventional X-band half-wavelength patch is approximately 18 mm to 21 mm, while the side length of the outer boundary 7 of the annular radiator 6 in this embodiment is compressed to approximately 8 mm to 9 mm, which can be incorporated into an array element period of approximately 10 mm.

[0052] To achieve miniaturization of the X-band radiator 6, a high dielectric constant material is selected for the dielectric substrate 1. A higher dielectric constant results in a shorter equivalent wavelength in the medium, allowing for a corresponding reduction in the physical size of the radiator. For an 8 GHz signal in the X-band, the wavelength in free space is approximately 37.5 mm. In a Rogers RO3006 substrate with a dielectric constant of 6.15, the equivalent wavelength is shortened to approximately 15.1 mm; in a Rogers RO3010 substrate with a dielectric constant of 10.2, the equivalent wavelength is further shortened to approximately 11.7 mm. The dielectric constant of the dielectric substrate 1 ranges from 4.0 to 10.5, the thickness from 0.5 mm to 2.0 mm, and the loss tangent from 0.002 to 0.004. The thickness h affects the antenna's impedance bandwidth and radiation efficiency: a thicker substrate provides a wider impedance bandwidth but excites higher surface wave modes, while a thinner substrate is beneficial for suppressing surface waves and reducing the profile but results in a narrower impedance bandwidth. In a preferred embodiment, the dielectric substrate 1 has a thickness of 1.0 mm, which balances bandwidth and efficiency.

[0053] To extend the impedance bandwidth of the X-band, loading slots 9 are symmetrically formed on the four sides of the annular radiator 6. The loading slots 9 are rectangular, U-shaped, or E-shaped, with a width of 0.2mm–0.4mm and a length of 1.5mm–3.5mm. In operation, energy is transmitted to the X-band annular radiator 6 via the X-band feed probe 15, forming a surface current flowing along the annular path on the annular radiator 6. The total path length of the annular structure is designed to be approximately an integer multiple of the dielectric wavelength around 8 GHz, forming a primary resonance and radiating outwards. Simultaneously, the loading slots 9 on the four sides cause the surface current to diffract and extend its path at the slot edges, forming an additional secondary resonant mode around 9 GHz. The primary and secondary resonant modes overlap and superimpose, forming a continuous broadband low-return-loss response in the 7.9–9 GHz range.

[0054] The geometric parameters of the loading slot 9 have an adjusting effect on the bandwidth coverage and resonance depth of the X-band. The slot width determines the intensity of current diffraction; a larger slot width results in stronger coupling of the secondary resonant modes, but may worsen impedance matching. The slot length determines the frequency position of the secondary resonant modes; a longer slot length results in lower secondary mode frequencies. By optimizing the parameters of the loading slot 9, the resonant frequency spacing between the primary and secondary modes can be appropriately adjusted, forming a flat broadband response. In a preferred embodiment, the loading slot 9 is a rectangular slot with a width of 0.3 mm and a length of 2.5 mm. In this case, the X-band bandwidth coverage is complete with a return loss of less than -10 dB in the range of 7.9–9 GHz. In another embodiment, the loading slot 9 is a U-shaped slot, consisting of a bottom edge and two sides, providing more parameter freedom. By adjusting the side lengths and bottom edge positions, the position and intensity of the secondary resonant modes can be further optimized. In yet another embodiment, the loading slot 9 is an E-shaped slot, adding an intermediate branch to the U-shaped slot to form a more complex multimode synthesis, suitable for applications with higher bandwidth requirements.

[0055] This embodiment, through a nested common-aperture layout combined with a double-layer PCB process, simultaneously accommodates the radiating structures of two frequency bands within the same physical period, solving the problems of large array size, difficulty in achieving common aperture, and complex and costly processing in the prior art. By extending the annular current path and compressing the size of the high-dielectric-constant substrate 1, the outer boundary 7 of the X-band radiator 6 is compressed to approximately 8mm to 9mm, resolving the structural contradiction that conventional X-band patch sizes are too large to be coplanarized with Ku-band patches. At the same time, by introducing dual-resonance synthesis through the loading slot 9, the impedance bandwidth of the single-layer microstrip patch antenna is extended from the conventional 3% to 5% to a relative bandwidth covering approximately 13% of 7.9 to 9 GHz, solving the problem of insufficient broadband coverage.

[0056] Example 2

[0057] like Figure 3As shown, the Ku-band patch 11 is located in the central region of the X-band ring radiator 6. It is a square or rectangular microstrip patch with a side length of 3.0 mm to 5.0 mm, preferably 3.6 mm to 4.5 mm. This Ku-band patch 11 is used to generate a half-wavelength resonance in the Ku-band (12.25–12.75 GHz). In terms of working principle, for the Ku-band, the X-band ring radiator 6 is relatively large and exhibits high impedance characteristics in this frequency band, so energy cannot form an effective resonance on the ring. Electromagnetic energy is mainly fed directly to the Ku-band patch 11 through the Ku-band feed probe 16, forming a half-wavelength resonance on the Ku-band patch 11 and radiating the Ku-band signal.

[0058] To broaden the Ku-band bandwidth, perturbation structures such as cross-shaped slots 12, central narrow slots, edge chamfers, or symmetrical slots are incorporated into the Ku-band patch 11. The slot width ranges from 0.2mm to 0.5mm, and the slot length ranges from 0.5mm to 2.0mm. The cross-shaped slot 12 consists of two narrow slots orthogonally aligned at the center of the Ku-band patch 11. Its function is to create additional current paths on the patch, generating a secondary resonant mode with a frequency close to but slightly offset from the primary mode. When the frequency spacing between the primary and secondary modes is appropriate, the two resonant modes superimpose to form a wider impedance bandwidth. The central narrow slot is simply a narrow slit along one direction at the center of the patch, achieving a similar effect to the cross-shaped slot 12 but with a simpler structure. Edge chamfers involve cutting small triangular areas at two opposite corners of the patch, introducing degenerate mode separation by altering the current distribution at the patch edges, thus achieving circular polarization or broadband matching. Symmetrical slots are U-shaped or rectangular notches created at the symmetrical edges of the patch to introduce additional resonant paths.

[0059] In a preferred embodiment, the Ku-band patch 11 adopts a cross-shaped slot 12 structure with a slot width of 0.3 mm and a slot length of 1.5 mm. Under these parameters, the return loss of the Ku-band in the 12.25–12.75 GHz range is less than -10 dB. In another embodiment, the Ku-band patch 11 adopts an edge-chamfered structure with a chamfer side length of 0.8 mm. Similar bandwidth expansion effects can be achieved by fine-tuning the patch size. The Ku-band patch 11 is nested in the center of the X-band ring radiator 6, making full use of the space within the ring without requiring additional aperture area.

[0060] like Figure 1 and Figure 3 As shown, the isolation gap 13 is located between the X-band annular radiator 6 and the Ku-band patch 11. It is an annular gap region without metal coverage, with a width of 0.2mm to 0.8mm, preferably 0.3mm to 0.6mm. This isolation gap 13 blocks the direct electromagnetic coupling path between the two frequency band radiators, reduces the electromagnetic coupling between the X-band annular radiator 6 and the Ku-band patch 11, and improves port isolation.

[0061] A metallized grounding via 14 is provided in the dielectric substrate 1 between adjacent X-band radiation unit 5 and Ku-band radiation unit 10. For example... Figure 4 As shown, metallized grounding vias 14 extend from the grounding metal layer 3 through the dielectric substrate 1 to the radiating metal layer 2, and are arranged around the periphery of each array element, forming a structure similar to a metal fence. This fence structure confines surface waves within the array element, suppressing their propagation to adjacent array elements, thereby reducing inter-array coupling.

[0062] The synergistic effect of the isolation gap 13 and the metallized grounding via 14 achieves port isolation better than -15dB for the two frequency bands. Because the isolation gap 13 exists between the X-band ring radiator 6 and the Ku-band patch 11, and the feed points of the two frequency bands are independent, the electric field distribution regions of the X-band and Ku-band are naturally separated in space, thus achieving high port isolation. The number and density of the metallized grounding vias 14 affect the isolation effect: in one embodiment, eight metallized grounding vias 14 are evenly distributed around each element; in another embodiment, the number of vias is increased to 16, forming a denser fence structure, further improving isolation, but also increasing manufacturing complexity. The via diameter is generally 0.2mm to 0.4mm to balance manufacturing feasibility and electromagnetic shielding effect.

[0063] The technological advancement of this embodiment compared to the prior art lies in the following: by introducing a secondary resonant mode through a perturbation structure on the Ku-band patch 11, the bandwidth coverage capability of the Ku-band is effectively expanded without increasing the physical size of the patch, enabling the Ku-band to fully cover the communication frequency band from 12.25 to 12.75 GHz; simultaneously, through the coordinated decoupling design of the isolation gap 13 and the metallized grounding via 14, a port isolation better than -15dB is achieved in the common-aperture nested layout, ensuring that the signals of the two frequency bands do not interfere with each other, thus solving the problems of dual-frequency mutual coupling and insufficient broadband coverage capability in the common-aperture layout.

[0064] Example 3

[0065] like Figure 3 and Figure 4As shown, the X-band feed probe 15 is located on the radiation edge of the X-band annular radiator 6, with the feed point offset from the center of the array element by 3.0 mm to 4.2 mm, preferably 3.2 mm to 3.8 mm. The X-band feed probe 15 passes through the dielectric substrate 1 from below the ground metal layer 3, with the top of the inner conductor electrically connected to the X-band annular radiator 6 and the outer conductor electrically connected to the ground metal layer 3. The Ku-band feed probe 16 is located inside the Ku-band patch 11, with the feed point offset from the center of the array element by 0.5 mm to 1.4 mm, preferably 0.6 mm to 1.2 mm. The Ku-band feed probe 16 also passes through the dielectric substrate 1 from below the ground metal layer 3, with the top of the inner conductor electrically connected to the Ku-band patch 11.

[0066] An insulating via 17 is provided on the ground metal layer 3, corresponding to the location through which the feed probe passes. The diameter of the insulating via 17 is slightly larger than the diameter of the inner conductor of the probe to ensure electrical isolation between the inner conductor and the ground metal layer 3. After the inner conductor of the probe passes through the dielectric substrate 1, its top end is connected to the radiating metal layer 2 by soldering or conductive adhesive. The feed network 4 is responsible for distributing the radio frequency signal to the feed probes of each array element.

[0067] The determination of the feed point offset distance is based on the impedance matching principle. For microstrip patches or ring radiators, the input impedance gradually decreases as the feed point moves away from the center along the radiating edge. By adjusting the feed point offset distance, the input impedance of the radiator in the target frequency band can be matched with the impedance of a 50-ohm system, thereby achieving low return loss. The feed point offset of the X-band ring radiator 6 is relatively large because the radiating edge of the ring structure is far from the center; the feed point offset of the Ku-band patch 11 is relatively small because the patch size is small and the impedance matching point is closer to the center.

[0068] In one alternative implementation, both the X-band and Ku-band can use microstrip line side-feeding instead of probe feeding. The microstrip line is introduced from the edge of the array element and connected to the radiator through an impedance transformation section. The advantage of side-feeding is simpler fabrication, eliminating the need for vias in the dielectric substrate 1, but the feed line occupies additional array area. In another implementation, probe feeding is used for the X-band and microstrip line side feeding for the Ku-band, balancing feeding flexibility and fabrication convenience. In yet another alternative implementation, electromagnetic coupling feeding can be used. Instead of directly connecting the probe to the radiator, an open-circuit microstrip line is placed below the radiator, and energy is transferred to the radiator via electromagnetic coupling. The advantage of electromagnetic coupling feeding is that it eliminates the need for vias in the dielectric substrate 1, and the dielectric layer isolates the feed line from the radiator, reducing the impact of parasitic capacitance.

[0069] like Figure 2As shown, the dual-frequency radiating elements in array 21 are arranged in a rectangular periodic pattern, with equal center-to-center spacing between elements in the row and column directions, dx=dy=d, where d is 8.5–10.3 mm, preferably 9.5–10.0 mm, and even more preferably 10.0 mm. This spacing is approximately 0.97 half-wavelengths of free space (approximately 20.69 mm) at the high end of the Ku band (12.75 GHz), close to half a wavelength, which can suppress grating lobes during large-angle scanning; simultaneously, it is approximately 0.23 half-wavelengths (approximately 42.9 mm) at the X band (9 GHz), with an electrical size much smaller than the wavelength, which is beneficial for wide-angle scanning.

[0070] The selection of the element spacing d needs to comprehensively consider both Ku-band grating lobe suppression and X-band radiation efficiency. For the Ku-band, the element spacing needs to be less than half the wavelength corresponding to the highest operating frequency; otherwise, grating lobes will appear during large-angle scanning. The Ku-band 12.75 GHz corresponds to a free-space half-wavelength of 11.5 mm. Considering the equivalent wavelength shortening effect of the dielectric substrate 1, the element spacing can be appropriately widened to approximately 10 mm. The relationship between the maximum scanning angle theta_max and the element spacing d is: d < lambda_min / (1 + sin(theta_max)), where lambda_min is the wavelength corresponding to the highest operating frequency. For the Ku-band 12.75 GHz and a 45-degree scanning angle, d needs to be less than approximately 12 mm. In this embodiment, d is selected to be approximately 10 mm, which can support scanning angles above 45 degrees. For the X-band, 9 GHz corresponds to a free-space half-wavelength of 16.7 mm. A spacing of 10 mm is much smaller than this value and will not produce grating lobes.

[0071] For the element in the m-th row and n-th column of array 21, its center coordinates can be expressed as: x_m = [-1.5d, -0.5d, 0.5d, 1.5d] (along the X direction), y_n = [-1.5d, -0.5d, 0.5d, 1.5d] (along the Y direction). This coordinate definition applies to 4×4 arrays. The X-band radiating element 5 and the Ku-band radiating element 10 can also be expanded into general rectangular array arrangements of 2×2, 8×8, or M×N to accommodate different gain and scanning angle requirements. 2×2 arrays are suitable for low-gain, wide-beam short-range communication terminals. 8×8 arrays are suitable for medium-gain, medium-scanning-angle vehicle-mounted radars. For an 8×8 array, the theoretical gain in the Ku band is approximately 24–26 dBi. M×N arrays allow for flexible configuration of the number and arrangement of elements according to system gain and beamwidth requirements. In the implementation of non-uniform arrays, the element spacing can be sparsely arranged. While maintaining the average spacing to meet the grating lobe suppression condition, the element positions are randomized or deterministically sparsed to reduce the sidelobe level.

[0072] In this embodiment, the X-band feed probe 15 and the Ku-band feed probe 16 are independent of each other, providing excitation channels for the X-band and Ku-band respectively, without interfering with each other, thus laying the physical foundation for independent phase shifting and beam control of dual bands; the element center distance is designed to be 8.5mm to 10.3mm, which simultaneously meets the requirements of Ku-band grating lobe suppression and X-band large-angle scanning under the same element spacing, and solves the structural contradiction that the element spacing is constrained by the Ku-band and the conventional patch size of the X-band exceeds the spacing.

[0073] Example 4

[0074] like Figure 2 As shown, the first phase control network 18 is connected to the X-band radiating element 5 and is used to apply a phase gradient to each element in the X-band to achieve beam scanning. The second phase control network 19 is connected to the Ku-band radiating element 10 and is used to apply a phase gradient to each element in the Ku-band to achieve beam scanning. The beam scanning of the two bands is controlled independently and does not affect each other. The first phase control network 18 and the second phase control network 19 are composed of phase shifters and power dividers.

[0075] Phase shifters can be analog or digital. Analog phase shifters change the phase shift by continuously adjusting voltage or current, offering high resolution but complex control. Digital phase shifters achieve discrete phase shifts by switching delay lines of different lengths, commonly using 4 to 6 bits, corresponding to phase shift resolutions of 22.5 degrees and 5.625 degrees, respectively. In a preferred embodiment, a 6-bit digital phase shifter is used, with a phase shift resolution of 5.625 degrees, enabling fine phase control during beam scanning. A power divider network distributes the RF signal to each array element and can employ Wilkinson power dividers, branch-line couplers, or T-junction power dividers. Wilkinson power dividers offer high output port isolation and good amplitude balance, making them suitable for arrays 21 with high consistency requirements. In one embodiment, independent Wilkinson power divider networks are used for the X-band and Ku-band to achieve equal-amplitude, in-phase feeding.

[0076] In phased array operation mode, each element in array 21 receives an excitation signal with a specific phase gradient from the phase-shifting network. The phase difference between adjacent elements determines the scanning direction of the beam. By independently setting the phase gradient in the feed networks 4 of the X-band and Ku-band respectively, independent beam scanning of the two bands can be achieved. The beams of the two bands can point in the same direction or in different directions, realizing dual-frequency independent beam tracking.

[0077] The control unit 20 can calculate the required phase compensation value for each array element according to the annular focal line equation, so that the array 21 forms an annular focal line or annular focal spot at a specified spatial position. An annular focal line refers to the focus of the radiated energy of the array 21 onto a certain plane in space as a circular curve, rather than the traditional point focusing. For an annular focal line with radius R located at a distance z in front of the array 21, the distance from the m-th row and n-th column element to the point on the focal line with angle phi is: r_mn(phi) = sqrt((x_m - R*cos(phi))^2 + (y_n - R*sin(phi))^2 + z^2). The corresponding phase compensation value is: phi_mn = -2*pi*r_mn(phi_0) / lambda, where phi_0 is the angle of the reference point on the annular focal line, and lambda is the operating wavelength. By traversing multiple sampling points on the annular focal line and averaging or optimizing the phase compensation value, the array 21 can form a uniform energy distribution on the annular focal line.

[0078] By switching different phase codebooks through the control unit 20, various operating modes such as ring focusing, point focusing, and far-field beam scanning can be achieved. In near-field focusing mode, the energy of array 21 is concentrated on the ring focal line at close range in front of the array, suitable for applications such as near-field microwave imaging and near-field radiation heating. In far-field scanning mode, the energy of array 21 forms a narrow beam pointing towards the far field, suitable for long-distance communication and radar detection. Switching between the two modes is achieved by loading different phase codebooks through the control unit 20, without changing the antenna hardware structure.

[0079] The technological advancements of this embodiment are as follows: the first phase control network 18 and the second phase control network 19 completely decouple the beam scanning of the X-band and the Ku-band, and the beam pointing of the two bands can be adjusted independently, thereby improving the beam scheduling flexibility of the dual-band system; the control unit 20, based on the phase compensation method of the annular focal line equation, endows the array 21 with the ability to switch between near-field focusing and far-field beam scanning, thus solving the problem of lacking annular focal line control capability.

[0080] Example 5

[0081] Based on embodiments 1 to 4 above, this embodiment also provides a configuration method for a dual-band common-aperture phased array antenna. This method integrates frequency band requirement determination, nested structure arrangement, miniaturization parameter determination, isolation and feeding configuration, and phase control configuration into a systematic process. All parameters of the dual-band common-aperture phased array antenna can be determined and fabricated using only a two-layer PCB process. The configuration method includes the following steps:

[0082] Step S1: Determine the operating frequency ranges of the X-band and Ku-band, and the size of array 21. The X-band is 7.9 to 9 GHz, and the Ku-band is 12.25 to 12.75 GHz. The size of array 21 is determined based on gain and scanning angle requirements. When determining the frequency range, the system's requirements for bandwidth, gain, sidelobe level, and other parameters need to be considered. The size of array 21 is determined based on the gain formula: G = 4*pi*A_eff / lambda^2, where A_eff is the effective aperture area of ​​the array. For a 4×4 array, the theoretical gain in the Ku-band is approximately 18 to 20 dBi; for an 8×8 array, the theoretical gain is approximately 24 to 26 dBi. This step clarifies the design objectives and provides constraints for subsequent structural parameter design.

[0083] Step S2: Arrange a nested common-aperture structure, placing the Ku-band patch 11 in the central region of the X-band ring radiator 6. The element center spacing is determined based on the high-half wavelength of the Ku-band to suppress scanning grating lobes. For example... Figure 1 and Figure 2 As shown, the nested structure allows two frequency bands to share the same physical aperture. When determining the element center distance, it is necessary to ensure that no grating lobes appear in the Ku band at the maximum scanning angle. For the Ku band at 12.75 GHz and a 45-degree scanning angle, d needs to be less than approximately 12 mm. This invention selects d to be approximately 10 mm, which can support scanning angles greater than 45 degrees. This step solves the problems of large array size and difficulty in achieving a shared aperture in the prior art.

[0084] Step S3: Determine the dimensions of the outer boundary 7 and inner boundary 8 of the X-band ring radiator 6, and create loading slots 9 on the four sides of the ring structure to introduce additional resonant modes. For example... Figure 3 As shown, the X-band ring radiator 6 extends the current path, and the dual-resonance synthesis introduced by the loading slot 9 broadens the bandwidth. When determining the size of the X-band ring radiator 6, the equivalent half-wavelength is first calculated based on the dielectric constant of the substrate 1 and the center frequency of the X-band. Then, the dimensions of the outer boundary 7 and inner boundary 8 of the ring structure are adjusted to ensure that the total length of the ring current path satisfies the resonance condition. The parameters of the loading slot 9 are optimized using electromagnetic simulation software, aiming to achieve wideband coverage with return loss below -10dB in the 7.9 to 9 GHz range. This step solves the problems of the X-band patch size being too large to be incorporated into a compact cycle and insufficient wideband coverage.

[0085] Step S4: Determine the width of the isolation gap 13 and the feed point locations of the X-band feed probe 15 and the Ku-band feed probe 16. For example... Figure 3 and Figure 4As shown, the width of the isolation gap 13 is 0.2mm to 0.8mm, and the feed point offset distance is determined based on impedance matching. When determining the width of the isolation gap 13, a balance needs to be struck between isolation and radiator size: a wider gap results in higher isolation, but reduces the available space for the Ku-band patch 11; a narrower gap reduces isolation, but allows for a larger Ku-band patch 11. The feed point positions of the X-band feed probe 15 and the Ku-band feed probe 16 are determined through input impedance simulation, with the goal of making the input impedance at the center frequency of each band close to 50 ohms. This step spatially separates the electric field distribution of the two frequency bands, achieving higher port isolation.

[0086] Step S5: Configure independent first phase control networks 18 and second phase control networks 19 for the X-band and Ku-band respectively, and have the control unit 20 calculate the phase compensation value of each array element according to the annular focal line equation. For example... Figure 2 As shown, the beam scanning of the two frequency bands is independently controllable and can switch between near-field focusing and far-field beam scanning. When configuring the phase control network, it is necessary to determine the number of phase shifters, the topology of the power divider network, and the feeding scheme. The number of phase shifters is determined based on the beam pointing accuracy requirements, generally selected as 4 to 6 bits. The calculation of the annular focal line phase compensation value is performed according to the method described in Example 4. By averaging or optimizing the phase compensation values ​​of multiple sampling points on the annular focal line, array 21 forms a uniform energy distribution on the target focal line. This step endows array 21 with multi-mode operating capability.

[0087] The technological advancement of this embodiment lies in the fact that the configuration method integrates frequency band requirement determination, nested structure arrangement, miniaturization parameter determination, isolation and feeding configuration, and phase control configuration into a systematic and standardized process. There are clear logical dependencies and parameter constraint boundaries between each step. Only a two-layer PCB process is required to realize the determination and fabrication of all parameters of the dual-frequency common-aperture phased array antenna, which solves the problems of complex processing technology and high cost in the background technology.

[0088] Example 6

[0089] Based on the above embodiments, combined with Figures 1 to 4 The structure shown also provides the following alternative implementations and parameter variations to suit different application requirements.

[0090] like Figure 1 and Figure 3As shown, in variations of the radiator shape, the X-band ring radiator 6 is not limited to a square ring; it can also be a circular ring, a polygonal ring, or a rounded square ring. The circular ring structure avoids current concentration at right-angle corners during fabrication, which helps reduce conductor losses, but requires higher precision in pattern generation and photolithography. Polygonal rings, such as hexagons or octagons, maintain near-circular performance, but require lower fabrication precision than circular rings. A rounded square ring involves rounding the four corners of a square ring, preserving the structural simplicity of the square ring while reducing field concentration at the corners.

[0091] In variations of the Ku-band radiator, the Ku-band patch 11 is not limited to a square shape; it can also be circular, elliptical, or polygonal. A circular patch offers better axisymmetry and is suitable for circular polarization applications. An elliptical patch can generate resonant frequencies in different orthogonal directions, making it suitable for dual-polarization or circular polarization designs.

[0092] like Figure 3 and Figure 4 As shown, in variations of the feeding method, in addition to probe feeding via X-band feed probe 15 and Ku-band feed probe 16, electromagnetic coupling feeding can also be used. Electromagnetic coupling feeding does not directly connect the probe to the radiator; instead, an open-circuit microstrip line is placed below the radiator, and energy is transferred to the radiator via electromagnetic coupling. The advantages of electromagnetic coupling feeding are that it eliminates the need for drilling holes in the dielectric substrate 1, simplifying fabrication, and the dielectric layer between the feed line and the radiator reduces the impact of parasitic capacitance.

[0093] like Figure 2 As shown, in the variations of array 21, besides the 4×4 array, larger-scale arrays of 21 can be expanded according to application requirements. For example, a 16×16 array (256 elements) can achieve a gain of approximately 30 dBi, suitable for long-range communication and radar detection. A 32×32 array (1024 elements) can achieve a gain of approximately 36 dBi, suitable for satellite communication ground stations. Expanding the array size to 21 requires a corresponding increase in the number of phase shifters and the complexity of the power divider network.

[0094] In variations of the polarization scheme, the X-band radiating element 5 and the Ku-band radiating element 10 of this invention can employ single-polarization or dual-polarization designs. In a single-polarization design, each element has only one feed probe, generating a linearly polarized beam. In a dual-polarization design, each element has two orthogonal feed probes, generating horizontal and vertically polarized beams respectively. By controlling the phase relationship between the two feed ports, linear, circular, or elliptical polarization can be achieved. The dual-polarization design requires integrating two independent feed networks 4 in each element, correspondingly increasing the system complexity.

[0095] In variations of application scenarios, the antenna of this invention can be used on a variety of platforms. In automotive applications, the antenna is mounted on the roof of a vehicle for broadband communication between vehicles and collision avoidance radar. In airborne applications, the antenna is mounted on the fuselage surface of a drone or aircraft for data link communication and synthetic aperture radar imaging. In spaceborne applications, the antenna is mounted on the surface of a low-Earth orbit satellite for satellite-to-ground communication and remote sensing. In portable terminals, the antenna is integrated inside the portable device for satellite phone or broadband access.

[0096] This embodiment demonstrates that the nested common aperture layout and double-layer PCB structure of the present invention have good scalability and adaptability. By adjusting parameters such as radiator shape, patch shape, feeding method, array size and polarization, the requirements of different application scenarios for gain, polarization, processing cost and installation platform can be met, thus expanding the applicability of the antenna.

[0097] In summary, according to Embodiments 1 to 6, the present invention has the following technical effects compared to the prior art:

[0098] First, by using a nested common aperture layout, the X-band ring radiator 6 is designed as a ring structure, and the Ku-band patch 11 is nested inside the ring, which solves the structural contradictions in the background technology, such as large array size, difficulty in achieving common aperture, and the constraint of array element spacing on the Ku-band.

[0099] Secondly, by extending the ring current path and compressing the size of the high dielectric constant substrate 1, and by introducing dual resonance synthesis through the loading groove 9 to broaden the bandwidth, the problem of insufficient broadband coverage in the background technology is solved.

[0100] Third, the use of a double-layer PCB process only requires conventional board manufacturing and electroplating processes, which solves the problems of complex processing and high cost of multi-layer stacking or metal cavity solutions in the background technology.

[0101] Fourth, the problem of dual-frequency mutual coupling in the common aperture layout is solved by using isolation gap 13, independent power supply probes 15 and 16 and metallized grounding via 14, and the port isolation is better than -15dB.

[0102] Fifth, by using the phase compensation method based on the annular focal line equation in the control unit 20, the array 21 is given the ability to switch between near-field focusing and far-field scanning, thus expanding the application range of the antenna.

[0103] Sixth, through a systematic configuration method, the frequency band requirement determination, nested structure arrangement, miniaturization parameter determination, isolation and feeding configuration, and phase control configuration are integrated into one, realizing a standardized configuration process for dual-band common-aperture phased array antennas.

[0104] The above description is merely a preferred embodiment of the present invention and is not intended to limit the scope of protection of the present invention. Any variations or substitutions that can be easily conceived by those skilled in the art within the scope of the technology disclosed in the present invention should be included within the scope of protection of the present invention.

Claims

1. A dual-frequency common-aperture phased array antenna, characterized in that: The device includes a dielectric substrate (1), a first surface of which is provided with a radiating metal layer (2), and a second surface of which is provided with a grounding metal layer (3); a power supply network (4) is provided on the first and / or second surfaces of the dielectric substrate (1); and it is also provided with a plurality of X-band microstrip radiating units (5) and a plurality of Ku-band microstrip radiating units (10). The X-band radiation unit (5) is an annular microstrip radiator (6), which is arranged on the first surface of the dielectric substrate (1). The annular microstrip radiator (6) has an outer boundary (7) and an inner boundary (8), and an annular metal region is formed between the outer boundary (7) and the inner boundary (8). The annular metal region constitutes an annular current path. The Ku-band radiating unit (10) is a microstrip patch (11), which is nested in the center of the annular metal region of the X-band radiator (6); an isolation gap (13) is provided between the X-band radiator (6) and the Ku-band patch (11). The X-band radiation unit (5) and the Ku-band radiation unit (10) are nested together with the same aperture. The X-band array elements are arranged with a first array element spacing, and the Ku-band array elements are arranged with a second array element spacing. The second array element spacing is smaller than the first array element spacing.

2. The dual-frequency common-aperture phased array antenna according to claim 1, characterized in that: The outer boundary (7) has a side length of 7.0 mm to 9.8 mm, and the inner boundary (8) has a side length of 4.5 mm to 6.5 mm. Loading grooves (9) are symmetrically arranged on the four sides of the annular metal region. The loading grooves (9) are rectangular grooves, U-shaped grooves or E-shaped grooves, with a groove width of 0.2 mm to 0.4 mm and a groove length of 1.5 mm to 3.5 mm. The loading grooves (9) are used to extend the current path and introduce additional resonant modes so that the X-band radiator (6) forms a double resonant synthesis. According to claim 1, the dual-frequency common-aperture phased array antenna is characterized in that: the side length of the Ku-band patch (11) is 3.0mm to 5.0mm; the Ku-band patch (11) is provided with a cross slot (12), a central narrow slot, an edge chamfered slot or a symmetrical opening slot, the slot width is 0.2mm to 0.5mm, and the slot length is 0.5mm to 2.0mm. By opening a slot structure on the Ku-band patch (11), a secondary resonant mode is introduced, thereby widening the Ku-band operating bandwidth.

3. The dual-frequency common-aperture phased array antenna according to claim 1, characterized in that: The width of the isolation gap (13) is 0.2mm to 0.8mm; a metallized grounding via (14) is provided on the dielectric substrate (1) between the adjacent X-band radiation unit (5) and Ku-band radiation unit (10). The metallized grounding via (14) is used to suppress the propagation of surface waves on the substrate and reduce the mutual coupling interference between the dual-band array elements.

4. The dual-frequency common-aperture phased array antenna according to claim 1, characterized in that: It also includes an X-band feed probe (15) and a Ku-band feed probe (16); the X-band feed probe (15) passes through the dielectric substrate (1) from below the ground metal layer (3), and the top of the inner conductor of the X-band feed probe (15) is electrically connected to the X-band radiator (6), and the offset distance of the feed point relative to the center of the array element is 3.0 mm to 4.2 mm; the Ku-band feed probe (16) passes through the dielectric substrate (1) from below the ground metal layer (3), and the top of the inner conductor of the Ku-band feed probe (16) is electrically connected to the Ku-band patch (11), and the offset distance of the feed point relative to the center of the array element is 0.5 mm to 1.4 mm.

5. The dual-frequency common-aperture phased array antenna according to claim 1, characterized in that: The dielectric substrate (1) has a dielectric constant of 4.0 to 10.5, a thickness of 0.5 mm to 2.0 mm, and a loss tangent of 0.002 to 0.

004. The dielectric substrate (1) is made of low-loss high-frequency microwave dielectric material, specifically Rogers RO3006 with a dielectric constant of 6.15 or Rogers RO3010 with a dielectric constant of 10.

2.

6. The dual-frequency common-aperture phased array antenna according to claim 1, characterized in that: The X-band radiation unit (5) and the Ku-band radiation unit (10) are arranged in a 2×2, 8×8 or M×N rectangular grid, with the center distance between the array elements in the row direction and the column direction being equal, and the center distance between the array elements being 8.5mm to 10.3mm.

7. The dual-frequency common-aperture phased array antenna according to claim 1, characterized in that: It also includes a first phase control network (18) and a second phase control network (19); the first phase control network (18) is connected to the X-band radiation unit (5) and is used to apply a phase gradient to each X-band array element to achieve beam scanning; the second phase control network (19) is connected to the Ku-band radiation unit (10) and is used to apply a phase gradient to each Ku-band array element to achieve beam scanning; the first phase control network (18) and the second phase control network (19) are independent of each other, and the two frequency band beams are scanned independently.

8. The dual-frequency common-aperture phased array antenna according to claim 1, characterized in that: It also includes a control unit (20), which is used to calculate the phase compensation value of each array element according to the annular focal line equation, so that the array (21) forms an annular focal line or annular focal spot at a specified spatial position; the control unit (20) realizes the switching of the array between near-field focusing and far-field beam scanning modes by switching the phase code table.

9. A method for configuring a dual-frequency common-aperture phased array antenna, employing the dual-frequency common-aperture phased array antenna as described in any one of claims 1 to 9, characterized in that, Includes the following steps: S1. Determine the operating frequency range and array (21) size of the X-band and Ku-band, where the X-band is 7.9 to 9 GHz and the Ku-band is 12.25 to 12.75 GHz; S2. Arrange a nested common aperture structure, place the Ku-band patch (11) in the central region of the X-band annular radiator (6), and determine the center distance of the array elements to suppress the Ku-band scanning grating lobe. S3. Determine the dimensions of the outer boundary (7) and inner boundary (8) of the X-band ring radiator (6), and open loading slots (9) on the four sides of the ring structure to introduce additional resonant modes, so that the X-band forms a double resonant synthesis to cover the target frequency band. S4. Determine the width of the isolation gap (13) and the feed point positions of the X-band feed probe (15) and the Ku-band feed probe (16) so that the electric field distribution of the two bands is spatially separated. S5. Configure independent first phase control network (18) and second phase control network (19) for X-band and Ku-band respectively, and calculate the phase compensation value of each array element according to the ring focal line equation to realize the switching control of near-field focusing and far-field beam scanning.

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