Reduced-order model predictive control method for full-bridge LLC resonant converter

Abstract

The application provides a reduced-order model predictive control method of a full-bridge LLC resonant converter. The method selects a fixed-frequency phase-shift modulation as a basic modulation mode, and establishes an accurate mapping relationship between a phase-shift ratio and output power through a time-domain analysis method, so as to provide an accurate model basis for high-performance control. In view of the core bottleneck that a controller contains a transcendental equation with a trigonometric function and is difficult to solve in real time in the model predictive control, a global optimal reduced-order method of a nonlinear term based on a cubic Chebyshev polynomial is provided, the complex transcendental equation is converted into a monomial cubic algebraic equation which can be solved in a closed form, the calculation complexity is greatly reduced while the control accuracy is ensured, and fast dynamic response and high voltage stabilization accuracy of the LLC resonant converter under wide working conditions are realized.

Description

Technical Field

[0001] This invention belongs to the field of model predictive control technology for LLC converters, specifically a reduced-order model predictive control method for a full-bridge LLC resonant converter. Background Technology

[0002] With the rapid development of high-power DC fast charging technology for new energy vehicles, LLC resonant converters have become the mainstream topology for DC / DC conversion in high-power DC charging equipment due to their wide-range soft-switching, high power density, and high-efficiency energy conversion characteristics. However, the inherent strong nonlinearity and multi-mode operation of LLC resonant converters mean that traditional PI control relies on linearized small-signal models. Under strong disturbance conditions such as input voltage fluctuations and large-scale load switching, it suffers from defects such as dynamic response lag, large overshoot, and oscillation, making it difficult to meet the high-performance control requirements of fast charging scenarios. Existing model predictive control schemes for LLC converters involve complex transcendental equations in their controller solution process, making it impossible to achieve closed-loop real-time solutions. These schemes suffer from problems such as large online computation, high difficulty in implementing digital controllers, and difficulty in ensuring real-time control under high-frequency conditions, which seriously restricts the large-scale application of model predictive control in the engineering of LLC resonant converters. Summary of the Invention

[0003] This invention proposes a reduced-order model predictive control method for a full-bridge LLC resonant converter.

[0004] The technical solution to achieve the objective of this invention is: a reduced-order model predictive control method for a full-bridge LLC resonant converter, comprising:

[0005] Step 1: Determine the main circuit topology of the full-bridge LLC resonant converter, select fixed-frequency phase-shift modulation as the basic modulation method, use time-domain analysis to divide 10 continuous operating modes within one switching cycle, determine the time-domain expressions of resonant current, excitation current and resonant capacitor voltage under each mode, and establish the accurate mapping relationship between phase shift D and output power P.

[0006] Step 2: Construct a cost function with the goal of minimizing the output voltage tracking error, transforming the model predictive control problem into an extremum problem of the cost function, and obtain the relationship between the output voltage and the shift ratio;

[0007] Step 3: Use cubic Chebyshev polynomials to perform global optimal approximation of the nonlinear trigonometric function terms within the shift ratio working interval, transforming the complex transcendental equation into a univariate cubic algebraic equation that can be solved in closed form.

[0008] Step 4: In each control cycle, the input voltage, output voltage, and load current of the full-bridge LLC resonant converter are detected in real time. The optimal phase shift ratio for the current cycle is solved by substituting the values ​​into a cubic algebraic equation, and the corresponding phase shift modulation signal is generated to drive the power switching transistors, thus completing the closed-loop rolling optimization control.

[0009] Compared with the prior art, the significant advantages of this invention are:

[0010] This invention addresses the inherent limitations of traditional PI control for LLC resonant converters, which relies on linearized small-signal models, suffers from lag in dynamic response under strong disturbance conditions, large overshoot, and oscillations. It also addresses the inherent limitations of traditional model predictive control, such as the difficulty in solving complex transcendental equations in real time, the large amount of online computation, and the high engineering difficulty in high-frequency operation. Based on fixed-frequency phase-shift modulation, this invention constructs an accurate predictive model for LLC resonant converters using time-domain analysis. A reduced-order model predictive control architecture based on Chebyshev polynomials is proposed, achieving a synergistic improvement in the converter's dynamic response performance and steady-state control accuracy. Under typical operating conditions such as output voltage step changes, input voltage fluctuations, and sudden changes in load power, the dynamic response speed is several times faster than traditional PI control, the output voltage overshoot is significantly reduced, and it also exhibits extremely low steady-state output ripple and extremely high voltage regulation accuracy.

[0011] This invention designs a nonlinear term reduction approximation method based on cubic Chebyshev polynomials, which performs global optimal approximation of trigonometric function terms in the control model, transforming complex transcendental equations into univariate cubic algebraic equations that can be solved in closed form. Under the premise of ensuring that the approximation error is much smaller than the inherent error of the system and that the control accuracy is not affected, the computational burden of the digital controller is greatly reduced. The algorithm has simple hardware implementation and strong robustness, providing reliable technical support for high-performance control of high-power DC charging equipment.

[0012] The present invention will now be described in further detail with reference to the accompanying drawings. Attached Figure Description

[0013] Figure 1 This is a topology diagram of a full-bridge LLC resonant converter.

[0014] Figure 2 This is a schematic diagram of the voltage and current waveforms during a switching cycle.

[0015] Figure 3 This is the equivalent circuit for the tenth mode corresponding to the end of the previous negative half-cycle.

[0016] Figure 4 This is the first operating mode of the full-bridge LLC resonant converter.

[0017] Figure 5 This is the second operating mode of the full-bridge LLC resonant converter.

[0018] Figure 6 This is the third mode of operation for the full-bridge LLC resonant converter.

[0019] Figure 7 This is the fourth mode of operation for the full-bridge LLC resonant converter.

[0020] Figure 8 This is the fifth mode of operation for the full-bridge LLC resonant converter.

[0021] Figure 9 This is a schematic diagram of the converter output side circuit.

[0022] Figure 10 This is the model prediction control block diagram.

[0023] Figure 11 The simulated waveform obtained by this invention is shown below. Detailed Implementation

[0024] A reduced-order model predictive control method for LLC resonant converters addresses the inherent limitations of traditional PI control in LLC resonant converters under strong disturbance conditions such as input voltage fluctuations and large load switching. These limitations include dynamic response lag, large overshoot, and oscillation. Furthermore, traditional model predictive control suffers from complex transcendental equations that are difficult to solve in real time, resulting in high online computational burden and significant engineering implementation challenges at high frequencies. Based on fixed-frequency phase-shift modulation, this method establishes a precise mapping relationship between the phase shift and output power through time-domain analysis, constructing a single-step predictive model for the output voltage of the LLC resonant converter. A reduced-order approximation method based on Chebyshev polynomials is proposed, transforming the complex transcendental equations into a closed-form cubic algebraic equation. This significantly reduces the computational burden on the controller while maintaining control accuracy, achieving a synergistic improvement in both the converter's dynamic response speed and steady-state voltage regulation accuracy. The specific steps are as follows:

[0025] Step 1: Set the preset value V of the output voltage of the LLC resonant converter. o-ref And the range of the shift ratio D, and the design parameters of each component of the LLC resonant converter, including the resonant cavity inductance L. rr inductance value The resonant cavity capacitance C rr capacitance value Magnetizing inductance L mm inductance value Transformer turns ratio n, output filter capacitor C ff The capacitance value is C f ;

[0026] The full-bridge LLC resonant converter topology is as follows: Figure 1 As shown, the voltage and current waveforms within one switching cycle are illustrated in the diagram below. Figure 2 .

[0027] The full-bridge LLC resonant converter includes a DC power supply, an input inverter network, a resonant cavity, a high-frequency transformer, a rectifier filter network, and a load. The input terminal of the input inverter network is connected to the DC power supply, and its output terminal is connected to the input terminal of the resonant cavity. The output terminal of the resonant cavity is connected to the primary side of the high-frequency transformer. The secondary side of the high-frequency transformer is connected to the input terminal of the rectifier filter network, wherein the turns ratio of the high-frequency transformer is n. The output terminal of the rectifier filter network is connected to the load. The input inverter network is a full-bridge inverter structure, including a first switch (Q1), a second switch (Q2), a third switch (Q3), and a fourth switch (Q4). 4) The source of the first switch (Q1) is connected to the positive terminal of the DC voltage source, and the drain of the first switch (Q1) is connected to the source of the second switch (Q2) and leads out to the midpoint A of the bridge arm; the drain of the second switch (Q2) is connected to the negative terminal of the DC voltage source; the source of the third switch (Q3) is connected to the positive terminal of the DC voltage source, and the drain of the third switch (Q3) is connected to the source of the fourth switch (Q4) and leads out to the midpoint B of the bridge arm; the drain of the fourth switch (Q4) is connected to the negative terminal of the DC voltage source; wherein (Q1) and (Q3) constitute the leading bridge arm, and (Q2) and (Q4) constitute the lagging bridge arm; the resonant cavity is composed of a resonant inductor (L rr ), resonant capacitor (C) rr ) and magnetizing inductance (L mm ) series connection; resonant inductor (L rr One end of the resonant inductor (L) is connected to the midpoint A of the bridge arm, and the resonant inductor (L) rr The other end of the capacitor is connected to the resonant capacitor (C). rr One end of the capacitor is connected; the resonant capacitor (C) is connected to the other end of the capacitor. rr The other end of the magnetizing inductor is connected to one end of the primary winding of the high-frequency transformer; mm The two ends of the high-frequency transformer's primary winding are connected in parallel, and the other end of the high-frequency transformer's primary winding is connected to the midpoint B of the bridge arm; in engineering, the inherent leakage inductance of the high-frequency transformer is equivalent to the resonant inductance (L) in parallel. rr The magnetizing inductance of the high-frequency transformer is directly reused as the magnetizing inductance (L). mm The rectifier-filter network includes a full-bridge rectifier bridge and an output filter capacitor (C). ff The full-bridge rectifier bridge consists of the first rectifier diode (D). R1 ), second rectifier diode (D) R2 ), third rectifier diode (D) R3 ) and the fourth rectifier diode (D R4 Composed of; the first rectifier diode (D) R1 The negative terminal of the first rectifier diode (D) is connected to the second rectifier diode (D). R2 The negative terminal of the third rectifier diode (D) is connected. R3 The positive terminal of ) and the fourth rectifier diode (D) R4The positive terminal of the first rectifier diode (D) is connected; R1 The positive terminal of ) and the fourth rectifier diode (D) R4 The negative terminal of the first rectifier diode (D) is connected to and led out to the input terminal C of the rectifier bridge; the second rectifier diode (D) R2 The positive terminal of ) and the third rectifier diode (D) R3 The negative terminal of the rectifier is connected to and led out to the input terminal D of the rectifier bridge; the input terminals C and D of the rectifier bridge are respectively connected to the two ends of the secondary side of the high-frequency transformer; the output filter capacitor (C) ff ) are connected in parallel across the output ends of the full-bridge rectifier bridge, and the load (R) Ld ) connected in parallel to the output filter capacitor (C) ff The two ends of ).

[0028] The full-bridge LLC resonant converter has ten modes:

[0029] The equivalent circuit before time t0, corresponding to the tenth mode at the end of the previous negative half-cycle, is as follows: Figure 3 Before time t0, the first switch (Q1) and the second switch (Q2) are in the on state, while the third switch (Q3) and the fourth switch (Q4) remain off. The voltage V between points A and B in the bridge arm is... AB =0, L rr C rr and L mm All three elements participate in resonance, and the resonant current i Lr With excitation current i Lm When the current is equal, no current flows through either the primary or secondary side of the transformer; all the energy of the load is supplied by the output filter capacitor C. ff Available separately.

[0030] First mode such as Figure 4 For the time interval t0 to t1, at time t0, the second switch (Q2) is turned off, and at this time the resonant current i Lr The parasitic capacitance C2 of the second switch (Q2) is charged, while the parasitic capacitance C4 of the fourth switch (Q4) is discharged. Since this stage lasts for a very short time, the resonant current i can be approximated. Lr and excitation current i Lm All remain constant, and the load is still provided by the output filter capacitor C. ff powered by.

[0031]

[0032]

[0033] in, It is the voltage across the parasitic capacitance C2 of the second switching transistor (Q2). This is the peak value of the magnetizing inductor current. It is the capacitance value of the parasitic capacitance of the switching transistor. It is the voltage of the parasitic capacitance C4 of the fourth switching transistor (Q4).

[0034] Second mode such as Figure 5 For the time interval t1 to t2, at time t1, the voltage across the parasitic capacitance C2 of the second switch (Q2) rises to the input voltage V. in When the voltage across the parasitic capacitance C4 of the fourth switch (Q4) drops to zero, the anti-parallel diode D4 of the fourth switch (Q4) naturally conducts, achieving zero-voltage turn-on (ZVS) for the fourth switch (Q4). At this point, the voltage between points A and B becomes the input voltage V. in Resonant current i Lr Excitation current i Lm Start increasing, the first rectifier diode (D) R1 ) and the fourth rectifier diode (D R4 When the transformer is in forward conduction, the primary voltage is clamped at nV. o V o Output voltage, excitation current i Lm It increases linearly. The resonant current i Lr Resonant capacitor voltage v Cr and excitation current i Lm The expression is as follows:

[0035]

[0036]

[0037]

[0038] Wherein, the resonant angular frequency is Characteristic impedance , For resonant inductance L rr inductance value, For the resonant capacitor C rr The capacitance value, For the magnetizing inductor L mm The inductance value.

[0039] Third mode such as Figure 6 The time interval is from t2 to t3. At time t2, the first switch (Q1) is turned off, and the resonant current i Lr The parasitic capacitance C1 of the first switching transistor (Q1) is charged, while the parasitic capacitance C3 of the third switching transistor (Q3) is discharged. The first rectifier diode (D...) R1) and the fourth rectifier diode (D R4 It is still conducting, nV o Added to the magnetizing inductor L mm Above, excitation current i Lm It continues to show a linear upward trend.

[0040] Fourth mode such as Figure 7 During the time interval t3 to t4, at time t3, the voltage across the parasitic capacitance C1 of the first switching transistor (Q1) rises to V. in When the voltage across the parasitic capacitance C3 of the third switch (Q3) drops to zero, the anti-parallel diode D3 of the third switch (Q3) naturally conducts, and the third switch (Q3) achieves ZVS. At this time, the voltage between points A and B is 0, and this switching mode is called the zero state. The first rectifier diode (D... R1 ) and the fourth rectifier diode (D R4 The transformer remains conductive, and the primary voltage of the transformer is still clamped at nV. o Excitation current i Lm The resonant current i continues to increase linearly. Lr Resonant capacitor voltage v Cr and excitation current i Lm The expression is as follows:

[0041]

[0042]

[0043]

[0044] in, For the resonant current i Lr Peak value.

[0045] Fifth mode such as Figure 8 For the time interval t4 to t5, at time t4, the resonant current i Lr With excitation current i Lm The primary current i of the transformer is equal to the primary current i. p Reduced to zero, the first rectifier diode (D) R1 ) and the fourth rectifier diode (D R4 The forward current of the output filter capacitor C also naturally crosses zero, thus achieving zero-current turn-off (ZCS) and completely eliminating reverse recovery losses. The energy required by the load is supplied by the output filter capacitor C. ff Provided separately. During this period, the resonant inductor L rr With excitation inductance L mm After being connected in series with the resonant capacitor C rrCommon resonance, due to the excitation inductance L mm The inductance value is much larger than the resonant inductance L. rr The resonant current i can be approximated. Lr The resonant capacitance C remains constant. rr Under constant current charging, the resonant current i Lr Resonant capacitor voltage v Cr and excitation current i Lm The expression is as follows:

[0046]

[0047]

[0048]

[0049] At time t5, the converter enters the negative half-cycle and also has five modes, which are completely symmetrical with the positive half-cycle, and will not be elaborated here.

[0050] Next, the average power P is calculated based on the expression of the time-domain modes. The phase shift ratio between the inverter arms is defined as D, and the phase shift time is T. ps =D / 2f s , where f s Let D be the switching frequency, and 0 < D < 1. From the modal analysis of phase-shift modulation, if the dead time is ignored (i.e., t0~t1 and t2~t3 are considered as dead times), then:

[0051]

[0052] Modal analysis reveals that the resonant current, excitation current, and resonant capacitor voltage exhibit periodic symmetry, therefore:

[0053]

[0054] Based on the time-domain formulas for each mode, the voltage and current values ​​at different time points can be obtained. With appropriate simplification, assuming that the duration of t3~t4 is very short relative to the dead time, it can be considered as the excitation current i at the end of the dead time. Lm With resonant current i Lr If they are equal, then:

[0055]

[0056]

[0057]

[0058] Wherein, the auxiliary constants a, b, c are defined as:

[0059]

[0060] When converter losses are ignored and the output voltage V in The average output power P of the LLC converter can be calculated from the time-domain equations (1) to (11) of each mode, which remain constant during the switching cycle:

[0061]

[0062] Step 2: Determine the single-step prediction equation for the output voltage of the LLC converter. Define the cost function with the goal of minimizing the output voltage tracking error as shown in equation (23). The model predictive control problem can be transformed into the problem of finding the extremum of the cost function, and finally the output voltage is obtained. Compared with the shift, the relationship of D is as follows (25);

[0063] The average output current on the rectifier side of the LLC converter can be calculated according to equation (18). for:

[0064]

[0065] Based on Kirchhoff's current law, the equations for nodal currents can be derived:

[0066]

[0067] The schematic diagram of the converter output side circuit is shown below. Figure 9 , For output current, For the charging current of the output capacitor, For the output filter capacitor C ff The capacitance value.

[0068] To solve for the predicted output voltage value of the next control cycle, based on the basic principle of model predictive control, one switching cycle of the converter is selected as the control cycle. The derivative term in equation (20) is discretized to obtain:

[0069]

[0070] Based on equations (19) to (21), the prediction model for the output voltage of the LLC converter can be obtained:

[0071]

[0072] To achieve model predictive control of the LLC converter, it is essential to ensure that the output voltage can quickly and accurately track the reference value. Therefore, its cost function J(t) k )for:

[0073]

[0074] in This is the reference value for the output voltage.

[0075] Cost function J(t) k The smaller the value of J(t), the smaller the deviation between the output voltage and the reference voltage in the next control cycle. Therefore, model predictive control is transformed into solving for the optimal control quantity that minimizes the cost function. Let the cost function J(t) be... k The optimal shift ratio D(t) can be obtained by making the partial derivative of the shift ratio D zero. k+1 The expression is:

[0076]

[0077] Operating switching frequency Designed at the resonant frequency The output voltage can be obtained by continuing to solve according to (19)~(24). And the control expression for the shift ratio D:

[0078]

[0079] in It is the resonant frequency.

[0080] Step 3: To address the difficulty in finding the extrema of the transcendental equation containing trigonometric functions, i.e., equation (25), in real time, a cubic Chebyshev polynomial is used to perform a high-precision global approximation of the nonlinear terms of the trigonometric functions within the shift ratio working interval, transforming the complex transcendental equation into a cubic equation that can be solved in a closed form. The specific method is as follows:

[0081] Introducing Chebyshev polynomials to approximate equation (25) is an important means of achieving fast solution.

[0082] Utilizing the properties of Chebyshev's order reduction method, and based on actual conditions, the actual operating range of the LLC resonant converter is shifted within the range of D∈(0, 0.5). The trigonometric function terms in the control expression are expanded as follows:

[0083]

[0084] Where the polynomial coefficients , , , as follows:

[0085]

[0086] Similarly, we can conclude that:

[0087]

[0088] Where the polynomial coefficients , , , as follows:

[0089]

[0090] Similarly, we can conclude that:

[0091]

[0092] Where the polynomial coefficients , , , as follows:

[0093]

[0094] Substituting the expanded polynomial into the expression (25) and combining like terms, we can obtain a cubic equation in one variable:

[0095]

[0096] The coefficients of each degree of the cubic equation are:

[0097]

[0098] Wherein, auxiliary constants are defined. , for:

[0099]

[0100] Step 4: In each control cycle, the input and output voltages and load current of the LLC converter are acquired in real time, substituted into a cubic equation to solve for the optimal phase shift, and the output phase-shift modulation signal drives the switching transistor to achieve closed-loop rolling optimization control. The specific process is as follows:

[0101] In any discrete control period t k At the start time, the input voltage V of the full-bridge LLC resonant converter for the current cycle is first obtained through a sensor. in (t k Output voltage V o (t k and load current I o (t k The algorithm then takes the discrete values ​​of the sampled data and proceeds to the prediction and cost evaluation stage.

[0102] Substitute the above state variables and main circuit parameters into equations (30) and (31) to calculate the auxiliary constant K. c K r The coefficients a0~a3 of the equation are used to solve the cubic equation (29) and select the optimal phase shift ratio D. This optimal phase shift ratio instruction will be updated to the underlying phase shift modulator before the end of the current operation cycle, thereby driving the LLC resonant converter to the target reference voltage V in the next switching cycle. o_ref Convergence, completing closed-loop rolling optimization, the model predictive control block diagram is as follows: Figure 10 .

[0103] Figure 10 The simulation waveforms obtained by this invention are shown in two sub-figures, which are the waveforms under the conditions of output voltage step and load power sudden change, respectively. The designed reduced-order model predictive control scheme can achieve sub-millisecond fast dynamic response under conditions such as output voltage step and load power sudden change, which is several times faster than the response speed of traditional PI control. At the same time, it greatly suppresses output voltage overshoot and has excellent steady-state voltage regulation accuracy and adaptability to all operating conditions.

Claims

1. A reduced-order model predictive control method for a full-bridge LLC resonant converter, characterized in that, include: Step 1: Determine the main circuit topology of the full-bridge LLC resonant converter, select fixed-frequency phase-shift modulation as the basic modulation method, use time-domain analysis to divide 10 continuous operating modes within one switching cycle, determine the time-domain expressions of resonant current, excitation current and resonant capacitor voltage under each mode, and establish the accurate mapping relationship between phase shift D and output power P. Step 2: Construct a cost function with the goal of minimizing the output voltage tracking error, transforming the model predictive control problem into an extremum problem of the cost function, and obtain the relationship between the output voltage and the shift ratio; Step 3: Use cubic Chebyshev polynomials to perform global optimal approximation of the nonlinear trigonometric function terms within the shift ratio working interval, transforming the complex transcendental equation into a univariate cubic algebraic equation that can be solved in closed form. Step 4: In each control cycle, the input voltage, output voltage, and load current of the full-bridge LLC resonant converter are detected in real time. The optimal phase shift ratio for the current cycle is solved by substituting the values ​​into a cubic algebraic equation, and the corresponding phase shift modulation signal is generated to drive the power switching transistors, thus completing the closed-loop rolling optimization control.

2. The reduced-order model predictive control method for a full-bridge LLC resonant converter according to claim 1, characterized in that, The full-bridge LLC resonant converter includes a DC power supply, an input inverter network, a resonant cavity, a high-frequency transformer, a rectifier filter network, and a load. The input terminal of the input inverter network is connected to the DC power supply, and its output terminal is connected to the input terminal of the resonant cavity. The output terminal of the resonant cavity is connected to the primary side of the high-frequency transformer. The secondary side of the high-frequency transformer is connected to the input terminal of the rectifier filter network, wherein the turns ratio of the high-frequency transformer is n. The output terminal of the rectifier filter network is connected to the load. The input inverter network is a full-bridge inverter structure, including a first switch (Q1), a second switch (Q2), a third switch (Q3), and a fourth switch (Q4). The first switch (Q1)... 1) The source of the first switch (Q1) is connected to the positive terminal of the DC voltage source; the drain of the first switch (Q1) is connected to the source of the second switch (Q2) and leads out to the midpoint A of the bridge arm; the drain of the second switch (Q2) is connected to the negative terminal of the DC voltage source; the source of the third switch (Q3) is connected to the positive terminal of the DC voltage source; the drain of the third switch (Q3) is connected to the source of the fourth switch (Q4) and leads out to the midpoint B of the bridge arm; the drain of the fourth switch (Q4) is connected to the negative terminal of the DC voltage source; wherein the first switch (Q1) and the third switch (Q3) form the leading bridge arm; the second switch (Q2) and the fourth switch (Q4) form the lagging bridge arm; the resonant cavity is composed of a resonant inductor (L rr ), resonant capacitor (C) rr ) and magnetizing inductance (L mm ) are connected in series to form a resonant inductor (L rr One end of the resonant inductor (L) is connected to the midpoint A of the bridge arm, and the resonant inductor (L) rr The other end of the capacitor is connected to the resonant capacitor (C). rr One end of the capacitor is connected; the resonant capacitor (C) is connected to the other end of the capacitor. rr The other end of the magnetizing inductor is connected to one end of the primary winding of the high-frequency transformer; mm The high-frequency transformer's primary winding is connected in parallel across its two ends, with the other end connected to the midpoint B of the bridge arm. The inherent leakage inductance of the high-frequency transformer is equivalent to the resonant inductance (L) in parallel. rr The magnetizing inductance of the high-frequency transformer is directly reused as the magnetizing inductance (L). mm The rectifier-filter network includes a full-bridge rectifier bridge and an output filter capacitor (C). ff ), output filter capacitor (C) ff ) are connected in parallel across the output ends of the full-bridge rectifier bridge, and the load (R) Ld ) connected in parallel to the output filter capacitor (C) ff The two ends of ).

3. The reduced-order model predictive control method for a full-bridge LLC resonant converter according to claim 2, characterized in that, The full-bridge LLC resonant converter has ten modes: The first mode is within the time interval t0 to t1. At time t0, the second switch (Q2) is turned off, and at this time the resonant current i Lr The parasitic capacitance C2 of the second switch (Q2) is charged, while the parasitic capacitance C4 of the fourth switch (Q4) is discharged, resulting in a resonant current i. Lr and excitation current i Lm All remain constant, and the load is supplied by the output filter capacitor C. ff powered by; The second mode occurs within the time interval t1 to t2. At time t1, the voltage across the parasitic capacitance C2 of the second switch (Q2) rises to the input voltage V. in When the voltage across the parasitic capacitance C4 of the fourth switch (Q4) drops to zero, the anti-parallel diode D4 of the fourth switch (Q4) naturally conducts, and the fourth switch (Q4) achieves zero-voltage turn-on (ZVS). At this time, the voltage between points A and B becomes the input voltage V. in Resonant current i Lr Excitation current i Lm Start increasing, the first rectifier diode (D) R1 ) and the fourth rectifier diode (D R4 When the transformer is in forward conduction, the primary voltage is clamped at nV. o V o Output voltage, excitation current i Lm It increases linearly; The third mode occurs within the time interval t2 to t3. At time t2, the first switch (Q1) is turned off, and the resonant current i Lr The parasitic capacitance C1 of the first switching transistor (Q1) is charged, while the parasitic capacitance C3 of the third switching transistor (Q3) is discharged; the first rectifier diode (D) R1 ) and the fourth rectifier diode (D R4 It is still conducting, nV o Added to the magnetizing inductor L mm Above, excitation current i Lm It continues to show a linear upward trend; The fourth mode occurs within the time interval t3 to t4. At time t3, the voltage across the parasitic capacitance C1 of the first switching transistor (Q1) rises to V. in When the voltage across the parasitic capacitance C3 of the third switch (Q3) drops to zero, the anti-parallel diode D3 of the third switch (Q3) naturally turns on, achieving zero-voltage turn-on for the third switch (Q3). At this time, the voltage between points A and B is 0, and the first rectifier diode (D... R1 ) and the fourth rectifier diode (D R4 The transformer remains conductive, and the primary voltage of the transformer is still clamped at nV. o Excitation current i Lm Continue to increase linearly; The fifth mode occurs within the time interval t4 to t5. At time t4, the resonant current i Lr With excitation current i Lm The primary current i of the transformer is equal to the primary current i. p Reduced to zero, the first rectifier diode (D) R1 ) and the fourth rectifier diode (D R4 The forward current of the output filter capacitor C also naturally crosses zero, thus achieving zero-current turn-off and completely eliminating reverse recovery losses. The energy required by the load is supplied by the output filter capacitor C. ff Supplied separately; resonant inductor L rr With excitation inductance L mm After being connected in series with the resonant capacitor C rr Common resonance, resonant current i Lr The resonant capacitance C remains constant. rr Charged by constant current At time t5, the converter enters the negative half-cycle, which is completely symmetrical with the first to fifth modes of the positive half-cycle.

4. The reduced-order model predictive control method for a full-bridge LLC resonant converter according to claim 1, characterized in that, The cost function constructed with the goal of minimizing the output voltage tracking error is as follows: In the formula, This is the output voltage reference value. This is the predicted output voltage value. Let be the cost function.

5. The reduced-order model predictive control method for a full-bridge LLC resonant converter according to claim 4, characterized in that, The formula for calculating the predicted output voltage is: In the formula, For the current moment Average output current on the rectifier side of the LLC converter For the current moment LLC converter output current For the output filter capacitor C ff The capacitance value, This refers to the operating switching frequency.

6. The reduced-order model predictive control method for a full-bridge LLC resonant converter according to claim 4, characterized in that, The specific method for transforming the model predictive control problem into an extremum problem of the cost function, and obtaining the relationship between the output voltage and the shift ratio, is as follows: Let the cost function J(t) k The optimal shift ratio D(t) is obtained when the partial derivative of D with respect to the shift ratio is zero. k+1 The expression for ), that is: Operating switching frequency Designed at the resonant frequency At that point, the output voltage is obtained by solving. And the control expression for the shift ratio D: in, The resonant frequency, This is the output voltage reference value. For the output filter capacitor C ff The capacitance value, For the magnetizing inductor L mm inductance value, This is the input voltage.

7. The reduced-order model predictive control method for a full-bridge LLC resonant converter according to claim 6, characterized in that, The cubic algebraic equation in one variable is specifically as follows: In the formula, ; 。

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