A voltage quantum differential sampling method, system and computer readable storage medium for direct current metering of an electric energy meter

Abstract

The present application relates to a kind of voltage quantum difference sampling methods, systems and computer readable storage medium of direct current measurement of electric energy meter, belong to precision measurement technical field.The present application first initializes Josephson junction array parameter according to the amplitude of the direct current difference voltage to be measured, generates microwave drive frequency and quantum reference initial value;Then residual voltage is calculated according to quantum reference initial value and the direct current difference voltage to be measured, generates adjustment instruction, and through microwave drive frequency dynamic adjustment quantum reference initial value, generates lock quantum reference value;Again, lock quantum reference value is used as rubidium atomic clock frequency reference, generates phase keying carrier, and quantumized carrier signal is obtained based on this;Finally, based on the direct current difference voltage to be measured and quantumized carrier signal generates baseband signal.The present application realizes the physical definition and precision traceability of direct current voltage by Josephson junction array dynamic initialization quantum reference and closed loop lock.

Description

Technical Field

[0001] This invention relates to a voltage quantum differential sampling method, system, and computer-readable storage medium for DC metering of electricity, belonging to the field of precision measurement technology. Background Technology

[0002] In the field of DC power metering, traditional high-precision voltage sampling techniques mainly rely on analog-to-digital conversion (ADC) methods based on semiconductor devices, such as Σ-Δ modulators or successive approximation ADC architectures. These techniques discretize the input voltage using high-resolution quantizers and combine them with digital filtering algorithms to improve the signal-to-noise ratio, achieving measurement accuracy at the microvolt or even nanovolt level. In recent years, with the development of quantum metrology, Josephson junction array voltage reference (JAVS) technology has been gradually applied to high-precision DC measurement. Based on superconducting quantum interference effects, it can provide a highly stable quantized voltage reference.

[0003] While existing quantum voltage reference technologies perform well in static calibration scenarios, key challenges remain in dynamic DC differential voltage sampling. Since the quantum phase-locking of the Josephson junction relies on a strict match between the microwave drive frequency and the input voltage, when the amplitude of the voltage under test varies significantly, the fixed-parameter quantum reference is prone to detuning of the phase-locked loop, thus introducing nonlinear errors. Furthermore, traditional methods rely on rubidium atomic clocks to provide a frequency reference, but their phase noise is transmitted to the baseband signal through the carrier modulation process, affecting the accuracy of quadrature demodulation.

[0004] Currently, quantum sampling for dynamic voltage is designed for AC signals. For example, Chinese patent application CN110632387A discloses a method for measuring harmonic voltage based on AC quantum voltage. This method first generates a periodic signal containing harmonics, determines the sampling frequency and the number of steps (the sampling frequency should be determined based on the fundamental frequency to achieve full-cycle sampling), then generates a stepped AC quantum voltage signal, and samples the differential signals before and after commutation, processing them separately (data filtering and recovery), performing a weighted Fourier transform on the signal data, and calculating the amplitude and phase of the fundamental and harmonic waves in the measured signal. Finally, the average of the results before and after commutation is calculated. While this method can overcome the problem of dynamic changes in the measured signal, it is designed for AC voltage measurement and is not applicable to DC voltage scenarios. Summary of the Invention

[0005] The purpose of this invention is to provide a voltage quantum differential sampling method, system, and computer-readable storage medium for DC metering of electricity, so as to solve the problem of low accuracy of current voltage quantum sampling for DC metering when the amplitude of the voltage being measured varies greatly.

[0006] To address the aforementioned technical problems, this invention provides, in a first aspect, a voltage quantum differential sampling method for DC metering of electricity, comprising the following steps:

[0007] 1) Initialize the Josephson junction array parameters based on the amplitude of the DC differential voltage to be measured, and generate the microwave drive frequency and initial value of the quantum reference;

[0008] 2) Calculate the residual voltage based on the initial value of the quantum reference and the DC differential voltage to be measured, generate adjustment commands, and dynamically adjust the initial value of the quantum reference through the microwave drive frequency to generate a locked quantum reference value;

[0009] 3) Using the locked quantum reference value as the frequency reference of the rubidium atomic clock, a phase-keyed carrier wave is generated, and a quantized carrier signal is obtained from it;

[0010] 4) Generate a baseband signal based on the measured DC differential voltage and the quantized carrier signal;

[0011] 5) The baseband signal is orthogonally decomposed into in-phase and quadrature components, and the mean square value is calculated for each component. Quantum precision sampling values ​​are generated based on the principle of quadrature demodulation.

[0012] In one possible implementation, the process of generating the microwave driving frequency and the initial value of the quantum reference includes:

[0013] The positive and negative terminal potentials of the DC differential voltage to be measured are collected, and the amplitude of the DC differential voltage to be measured is obtained accordingly.

[0014] Based on the amplitude of the DC differential voltage to be measured, the initial number of junctions and the initial value of the microwave driving frequency in the Josephson junction array parameters are obtained. The initial value of the quantum reference is calculated through the voltage-frequency relationship, and the microwave driving frequency is generated.

[0015] In one possible implementation, the adjustment command in step 2) is generated when the residual voltage is greater than the quantum lock-in error tolerance.

[0016] In one possible implementation, the generation of the locked quantum benchmark includes:

[0017] The microwave driving frequency is corrected by the polarity of the residual voltage to obtain the corrected microwave driving frequency. The new initial value of the quantum reference is calculated using the corrected microwave driving frequency, and the new residual voltage is calculated using the new initial value of the quantum reference.

[0018] When the new residual voltage is less than or equal to the quantum locking error tolerance, the new initial quantum reference value is used as the locked quantum reference value.

[0019] One possible implementation involves locking the quantum reference value as the frequency reference for the rubidium atomic clock, including:

[0020] The quantum reference value is locked by matching the modulation range of the rubidium atomic clock frequency reference through voltage amplitude, and a reference voltage signal is output.

[0021] The error signal is extracted from the reference voltage signal, and the frequency of the feedback-adjusted microwave signal is collected. The error signal is demodulated by lock-in amplification and the frequency is driven to adjust the frequency, thereby generating the transition frequency.

[0022] In one possible implementation, the generation of the quantized carrier signal in step 3) includes:

[0023] Electromagnetic wave oscillation is controlled by the transition frequency to generate a carrier fundamental frequency signal;

[0024] The carrier base frequency signal is converted into a binary phase control sequence to generate a phase keying control signal;

[0025] Phase switching is performed based on the phase keying control signal to generate a quantized carrier signal.

[0026] In one possible implementation, the generation of the baseband signal in step 4) includes:

[0027] The time-domain superposition waveform is obtained from the DC differential voltage under test and the quantized carrier signal by electromagnetic field coupling;

[0028] Based on the time-domain superimposed waveform, the difference-frequency and frequency relationship between the analytical order sideband components and the carrier fundamental frequency signal is expanded using the trigonometric identity to generate the modulation product.

[0029] The modulation product is low-pass filtered to generate the baseband signal.

[0030] In one possible implementation, the generation of the quantum-precision sample value in step 5) includes:

[0031] The baseband signal is orthogonally decomposed into in-phase and quadrature components;

[0032] The in-phase component is converted into a power signal, and energy is integrated within a fixed time window controlled by a rubidium atomic clock to generate the mean square value of the in-phase component.

[0033] Based on the orthogonal components, the mean square value of the orthogonal components is obtained by square-law detection and time window integration;

[0034] Based on the mean square values ​​of the in-phase and quadrature components, and after amplitude reconstruction, a demodulated signal is generated.

[0035] The demodulated signal is combined with the reference amplitude of the quantized carrier signal, and then converted through a proportional relationship to output a quantum-precision sampled value.

[0036] To address the aforementioned technical problems, a second aspect of the present invention provides a voltage quantum differential sampling system for DC metering of an electricity meter, comprising a processor that executes a computer program to implement the steps of the method in any possible implementation of the first aspect of the present invention.

[0037] To address the aforementioned technical problems, a third aspect of the present invention provides a computer-readable storage medium having a computer program stored thereon, which, when executed by a processor, implements the steps of the method in any possible implementation of the first aspect of the present invention.

[0038] The beneficial effects of this invention are as follows: This invention achieves the physical definition and accuracy traceability of DC voltage by dynamically initializing the quantum reference and locking it in a closed loop using a Josephson junction array; it combines a rubidium atomic clock to generate a phase-keyed carrier wave, and utilizes spectral expansion and orthogonal demodulation to extract an anti-interference baseband signal, ultimately outputting a sampled value with dual time-frequency domain quantization. The entire process is centered on end-to-end quantization (voltage-frequency-time) and dynamic adaptive adjustment, achieving both sub-ppm accuracy and anti-interference capability over a wide measurement range, and maintaining relatively high measurement accuracy even when the amplitude of the voltage being measured varies significantly. Attached Figure Description

[0039] Figure 1 This is a flowchart of the voltage quantum differential sampling method for DC metering of the present invention;

[0040] Figure 2 This is a flowchart of the Josephson junction array parameter initialization process in the voltage quantum differential sampling method for DC metering of the present invention;

[0041] Figure 3 This is a flowchart of the rubidium atomic clock frequency reference synchronization process in the voltage quantum differential sampling method for DC metering of the present invention;

[0042] Figure 4 This is a flowchart of the orthogonal decomposition and mean square value calculation in the voltage quantum differential sampling method for DC metering of the present invention;

[0043] Figure 5 This is a schematic diagram of the voltage quantum differential sampling system for DC metering of the present invention. Detailed Implementation

[0044] The specific embodiments of the present invention will be further described below with reference to the accompanying drawings.

[0045] This invention achieves the physical definition and accuracy traceability of DC voltage by dynamically initializing the quantum reference using a Josephson junction array and locking it in a closed loop.

[0046] Implementation Method 1 of Voltage Quantum Differential Sampling Method for DC Metering of Electricity Meters

[0047] This method first acquires the DC differential voltage to be measured and initializes the Josephson junction array parameters based on the amplitude of the DC differential voltage to be measured, generating the microwave driving frequency and initial value of the quantum reference. Then, it locks the quantum reference value to the rubidium atomic clock frequency reference, generating a phase-keyed carrier wave and acquiring the quantized carrier signal. Next, based on the DC differential voltage to be measured and the quantized carrier signal, a baseband signal is generated through waveform transformation, trigonometric identity spectral expansion, and low-pass filtering. Finally, the baseband signal is decomposed into in-phase and quadrature components through phase-locked loop (PLL), and the mean square value is calculated for each component. Quantum precision sampling values ​​are then generated based on the quadrature demodulation principle. The implementation flow of this method is as follows: Figure 1 As shown below, a detailed explanation will follow.

[0048] S1. Initialize the Josephson junction array parameters based on the amplitude of the DC differential voltage to be measured, and determine the microwave drive frequency and initial value of the quantum reference. The specific implementation of this step is as follows: Figure 2 As shown, the process includes the following:

[0049] S1.1 Acquire the positive and negative terminal potentials of the DC differential voltage to be measured, and obtain the amplitude of the DC differential voltage to be measured.

[0050] This embodiment uses a high-impedance differential probe to acquire the DC differential voltage to be measured. The high-impedance differential probe is connected to the positive and negative terminals of the DC differential voltage to be measured, respectively. The acquired positive and negative terminal potential signals are first input to a low-temperature drift instrumentation amplifier for signal conditioning. Then, the amplified differential signal is synchronously sampled by a high-precision analog-to-digital converter. The difference between the positive and negative terminal potentials (i.e., voltage amplitude) is calculated in a digital signal processor. The moving average filtering process is performed on multiple consecutive sampling points, and finally the amplitude of the DC differential voltage to be measured is output.

[0051] S1.2. Based on the amplitude of the DC differential voltage to be measured, obtain the initial number of junctions and the initial value of the microwave drive frequency in the Josephson junction array parameters.

[0052] The amplitude of the DC differential voltage to be measured is matched against a pre-stored Josephson junction array parameter lookup table. This table contains mappings between the initial number of junctions and the initial microwave drive frequency for different voltage ranges. Based on the specific voltage range of the DC differential voltage amplitude to be measured, the corresponding initial number of junctions (e.g., 500 in this example) and the initial microwave drive frequency (e.g., 12 GHz in this example) are extracted from the lookup table. The Josephson junction array parameter lookup table is established based on empirical data of the voltage-frequency characteristics and junction configuration of the Josephson junction array, ensuring that each voltage range corresponds to a validated set of parameter combinations. The matching process can use a nearest neighbor interpolation algorithm to determine the closest voltage range boundary value. When the amplitude of the DC differential voltage to be measured is between the upper limit of the currently matched lower voltage range and the lower limit of the next higher voltage range, the parameter combination corresponding to the higher voltage range is selected to ensure measurement margin.

[0053] The determination of the voltage range boundary values ​​was obtained through a combination of calibration experiments and theoretical calculations of the Josephson junction array: First, based on the Josephson voltage-frequency relationship, the array was subjected to frequency sweep tests under low-temperature conditions to measure the quantized voltage steps of the array with different junction numbers; then, combined with the inter-junction coupling effect of the array and microwave power tolerance, Monte Carlo simulation was used to analyze the critical stability point of each voltage range; finally, the boundary values ​​of each voltage range were determined through three-parameter Pareto optimization, and an overlap margin was usually set for the boundaries of adjacent ranges to ensure measurement continuity.

[0054] S1.3. Based on the initial number of junctions and the initial value of the microwave driving frequency, the initial value of the quantum reference is calculated through the voltage-frequency relationship, and the microwave driving frequency is generated.

[0055] The initial value for calculating the quantum benchmark is expressed as follows:

[0056]

[0057] in Represents the initial value of the quantum benchmark. is Planck's constant. For elementary charge, This is the microwave drive frequency.

[0058] It should also be noted that the initial value of the microwave driving frequency can be obtained through a pre-established Josephson junction array parameter lookup table. That is, first match the corresponding voltage range according to the amplitude of the DC differential voltage to be measured, and then read the initial value of the microwave driving frequency corresponding to the voltage range from the Josephson junction array parameter lookup table. The Josephson junction array parameter lookup table is pre-determined by the calibration experiment of the Josephson junction array, the voltage-frequency relationship, and the frequency range that can be stably locked.

[0059] The specific calculation process is as follows: Substitute the initial number of junctions and the initial value of the microwave driving frequency into the Josephson voltage-frequency relationship (i.e., the above equation). The initial number of junctions represents the number of junctions connected in parallel in the Josephson junction array, and the initial value of the microwave driving frequency represents the initial frequency of the microwave signal applied to the Josephson junction array. Using the physical principle of the Josephson effect, the initial value of the quantum reference is calculated. The initial value of the quantum reference represents the standard quantum voltage generated by the Josephson junction array under given initial number of junctions and initial value of the microwave driving frequency. The initial value of the microwave driving frequency is output to the microwave input port of the Josephson junction array through a microwave signal source.

[0060] S2. Based on the initial value of the quantum reference and the DC differential voltage to be measured, calculate the residual voltage, generate adjustment commands, and dynamically adjust the initial value of the quantum reference through the microwave drive frequency to generate a locked quantum reference value.

[0061] S2.1 Calculate the residual voltage based on the initial value of the quantum reference and the DC differential voltage to be measured.

[0062] The residual voltage is calculated using the following expression:

[0063]

[0064] in, Represents the residual voltage. This represents the reference voltage signal, which is the actual voltage signal formed by the initial quantum reference value through a Josephson junction array and then buffered and conditioned. The initial quantum reference value is used to characterize the theoretical quantum voltage target value of the reference voltage signal. This indicates the DC differential voltage to be measured.

[0065] Specifically, the initial quantum reference value and the DC differential voltage to be measured are simultaneously fed into a high-precision voltage comparison circuit. A precision differential amplifier (a differential measurement device used to directly amplify the difference between two input voltages) performs real-time comparison. A difference detection circuit, constructed using an operational amplifier with extremely low temperature drift characteristics (an operational amplifier is the basic amplification device for building difference detection circuits), outputs the difference signal between the two, forming the residual voltage. The precision differential amplifier can be constructed by combining a low-temperature drift operational amplifier with a high-precision resistor network. During the measurement process, the input impedance is maintained above the example value of 10 GΩ to ensure that the original state of the DC differential voltage to be measured is undisturbed. The formed residual voltage signal is digitally acquired by a high-resolution analog-to-digital converter, and a multi-point moving average algorithm is used to eliminate random interference. A dedicated polarity identification circuit monitors the changing trend of the residual voltage in real time, providing directional guidance for subsequent adjustment. The dedicated polarity identification circuit is implemented using an existing zero-crossing comparator circuit or a differential comparator circuit to determine the polarity direction of the residual voltage relative to zero potential.

[0066] S2.2 When the residual voltage is greater than the quantum lock-in error tolerance, the adjustment command is triggered.

[0067] This invention compares the real-time monitored absolute value of the residual voltage with a preset quantum locking error tolerance threshold. When the absolute value of the residual voltage continuously exceeds the quantum locking error tolerance threshold, the adjustment command generation circuit is triggered. The absolute value of the residual voltage is read, and the quantum locking error tolerance threshold is subtracted from the absolute value of the residual voltage to obtain the residual over-limit. The residual over-limit is then input into the proportional adjustment circuit, which generates an amplitude control signal according to the linear proportional relationship between the residual over-limit and the frequency correction step size. The larger the residual over-limit, the larger the microwave drive frequency correction step size corresponding to the amplitude control signal; the closer the residual over-limit is to the quantum locking error tolerance threshold, the smaller the microwave drive frequency correction step size corresponding to the amplitude control signal. During the triggering process, a Schmitt trigger circuit is used to eliminate jitter interference (including a residual voltage sampling terminal, an error tolerance comparator, a Schmitt trigger, and an adjustment command generation circuit; the residual voltage sampling terminal is connected to the input of the error tolerance comparator, which compares the absolute value of the residual voltage with the quantum locking error tolerance and outputs a comparison signal; the output of the error tolerance comparator is connected to the input of the Schmitt trigger, which uses an upper action level and a lower reset level to hysteresis-shape the comparison signal, eliminating the repeated flipping caused by fluctuations in the residual voltage near the quantum locking error tolerance; the output of the Schmitt trigger is connected to the enable terminal of the adjustment command generation circuit, outputting a stable trigger signal to the adjustment command generation circuit), ensuring a stable output of the adjustment command. The adjustment command is transmitted to the microwave frequency through an opto-isolator, while an activation status indicator shows an abnormal locking status.

[0068] In this embodiment, the quantum locking error tolerance threshold is determined by the noise statistical analysis of the Josephson junction array under standard quantum voltage. Combined with the Allen variance calculation of short-term stability and temperature drift characteristics, and verified by Monte Carlo simulation, it is finally set as the quantum locking error tolerance threshold.

[0069] S2.3 Execute the adjustment command and acquire the polarity direction of the residual voltage. After receiving the adjustment command, the polarity direction of the residual voltage is captured in real time by a high-sensitivity polarity detection circuit. The differential comparator in the polarity detection circuit receives the residual voltage and zero potential, determines the offset direction of the residual voltage relative to the zero potential, and outputs a logic level indicating positive or negative polarity. The polarity detection circuit has built-in Schmitt trigger characteristics and sets a hysteresis voltage to prevent critical state jitter. The detection result is transmitted to the microwave frequency adjustment circuit via optocoupler isolation to control the increase or decrease direction of the microwave drive frequency. At the same time, the drive direction indicator shows the current polarity status. The polarity status is connected to the data input terminal of the edge-triggered latch, and the update pulse of the amplitude control signal is connected to the clock terminal of the edge-triggered latch, so that the polarity status and the amplitude control signal are synchronously latched at the same sampling time and synchronously output to the microwave frequency adjustment circuit within a fixed time window. The input impedance of the detection circuit avoids causing a load effect on the residual voltage measurement circuit.

[0070] S2.4. The microwave driving frequency is corrected by the polarity direction of the residual voltage to generate an updated microwave driving frequency, which is then converted into a new initial value for the quantum reference.

[0071] Specifically, the adjustment direction of the microwave frequency synthesizer is controlled according to the polarity of the residual voltage. When the polarity is positive, the microwave driving frequency is increased; when the polarity is negative, the microwave driving frequency is decreased. Precise frequency fine-tuning can be achieved through a digital phase-locked loop circuit. The updated microwave driving frequency is converted into a corresponding voltage quantity by the Josephson junction array, forming a new initial value for the quantum reference. During frequency adjustment, a settling time is used to ensure that the Josephson junction array reaches an equilibrium state, while the ambient temperature is monitored and maintained within the example value of 23 degrees Celsius ± 0.1 degrees Celsius. A high-precision voltage comparison circuit compares the output voltage of the Josephson junction array corresponding to the new initial value for the quantum reference with the DC differential voltage to be measured. If the new residual voltage obtained by the comparison meets the quantum locking error tolerance, the verification of the new initial value for the quantum reference is deemed successful.

[0072] It should also be noted that the settling time refers to the waiting time required for the Josephson junction array to transition from the frequency change state to the stable output state of the quantum voltage step after the microwave driving frequency has been corrected once. This is used to avoid calculating the initial value of the new quantum reference in advance before the array output has stabilized.

[0073] S2.5. Calculate the new residual voltage based on the new initial value of the quantum reference.

[0074] This invention simultaneously inputs a new initial quantum reference value and the DC differential voltage to be measured into a high-precision voltage comparison circuit. Real-time voltage comparison is performed via a differential amplifier. During the comparison process, an instrumentation amplifier with automatic zero-crossing function is used to ensure the measurement circuit is free from interference. The generated voltage difference signal is rate-sampled by a 24-bit analog-to-digital converter, and high-frequency noise is eliminated through a digital filtering algorithm, ultimately outputting a new residual voltage value. The polarity of the new residual voltage is determined in real time by a zero-crossing detection circuit. The quantum-locked residual voltage polarity determination threshold is set to ±50 nanovolts of the example value, and the response time is controlled within 1 microsecond of the example value. The entire measurement process remains synchronized with the previous adjustment cycle to ensure data continuity.

[0075] In this embodiment, the quantum-locked residual voltage polarity discrimination threshold is determined by noise background testing of the Josephson junction array at a low temperature of 4.2K. Combined with uncertainty analysis of the quantum voltage standard and temperature drift experiments, the quantum-locked residual voltage polarity discrimination threshold is set using the three sigma criterion. After verification by Monte Carlo simulation, it is written into the threshold comparator module of the DSP and integrated with real-time temperature compensation to generate the quantum-locked residual voltage polarity discrimination threshold.

[0076] S2.6 When the new residual voltage is less than or equal to the quantum locking error tolerance, the new initial quantum reference value is used as the locked quantum reference value.

[0077] The new absolute value of the residual voltage is compared in real time with the preset quantum locking error tolerance threshold using a window comparator. If the new absolute value of the residual voltage does not exceed the quantum locking error tolerance threshold within a set number (e.g., 10) consecutive sampling periods, the locking condition is considered met. At this point, the new initial value of the quantum reference is written to non-volatile memory via digital logic circuitry, and a status indicator light is activated to show successful locking. A redundancy check mechanism is used during storage to ensure data integrity. The locked quantum reference value is output to the reference voltage bus via an isolated digital-to-analog converter. During the locked state, the residual voltage change is continuously monitored, and a readjustment process is immediately triggered when it exceeds the quantum locking error tolerance threshold.

[0078] S3. Connect the locked quantum reference value to the rubidium atomic clock frequency reference, generate a phase-keyed carrier wave, and obtain the quantized carrier signal. The implementation of this process is as follows: Figure 3 As shown below, a detailed explanation will follow.

[0079] S3.1 The locked quantum reference value is matched with the modulation range of the rubidium atomic clock frequency reference by the voltage amplitude, and the reference voltage signal is output.

[0080] This invention inputs a locked quantum reference value into a programmable attenuator network. A precision resistor divider circuit adjusts the voltage amplitude to the input modulation range of the rubidium atomic clock frequency reference. The attenuation ratio of the programmable attenuator network is determined by the ratio of the locked quantum reference value to the rated input voltage of the rubidium atomic clock frequency reference. The adjusted reference voltage signal is driven by a low-noise buffer amplifier to ensure impedance matching with the rubidium atomic clock frequency reference. Signal transmission uses a twisted-pair shielded cable, with the shield grounding resistance less than the example value of 0.1 ohms to suppress interference. After matching is complete, an LED indicator displays the ready state, and simultaneously activates the voltage modulation input terminal of the rubidium atomic clock frequency reference.

[0081] It should also be noted that the programmable attenuator network is constructed from precision resistor voltage divider branches, analog switch arrays, and buffered output branches. First, multiple precision resistor branches with different voltage division ratios are connected to the analog switch array in parallel selection. Then, the control terminal of the analog switch array is connected to the attenuation level output terminal of the controller. The corresponding voltage divider branch is selected according to the input modulation range of the locked quantum reference value and the rubidium atomic clock frequency reference. Finally, the selected voltage divider is sent to the low-noise buffered output branch to generate a reference voltage signal that matches the input modulation range of the rubidium atomic clock frequency reference.

[0082] The low-noise buffer amplifier can adopt a voltage follower structure composed of a low-noise operational amplifier. Specifically, the reference voltage signal output from the programmable attenuator network is connected to the non-inverting input of the low-noise operational amplifier, and the output of the low-noise operational amplifier is fed back to the inverting input to form a unity-gain buffer structure. An isolation resistor and a low-pass filter network are connected in series at the output to improve the input impedance, reduce the output impedance, and suppress noise before driving the voltage modulation input of the rubidium atomic clock frequency reference.

[0083] S3.2 Extract the error signal from the reference voltage signal, collect the feedback adjustment microwave signal frequency, demodulate the error signal through lock-in amplification and drive frequency adjustment to generate an ultra-fine transition frequency.

[0084] This invention inputs the adjusted reference voltage signal into the servo control loop of the rubidium atomic clock frequency reference. A high-sensitivity phase detector detects the phase difference between the reference voltage signal and the local oscillation signal of the rubidium atomic clock, generating an error signal. The error signal is demodulated by a lock-in amplifier, outputting a DC error voltage component. This DC error voltage component generates a frequency correction signal through a proportional-integral regulator, driving a voltage-controlled oscillator to adjust the microwave signal frequency. The corrected microwave signal is output to the rubidium atom bubble via a frequency multiplier chain, exciting the hyperfine transition of rubidium atoms. When the absolute value of the error voltage is less than a set value (10 microvolts), (10 microvolts can be determined by the phase detector noise, voltage modulation sensitivity, and frequency stability requirements of the rubidium atomic clock frequency reference; specifically, by pre-calibrating the correspondence between the error voltage and the frequency offset, selecting the voltage boundary where the frequency offset change is below the allowable locking error when the error voltage continues to decrease, and using this voltage boundary as the set value for determining locking completion), locking is determined to be complete. At this point, the frequency of the output microwave signal is the hyperfine transition frequency of the rubidium atom.

[0085] S3.3. Based on ultra-fine transition frequency control of electromagnetic wave oscillation, a carrier fundamental frequency signal is generated.

[0086] The determined ultrafine transition frequency is input to a direct digital frequency synthesizer (DDS). A quartz crystal temperature-controlled oscillator provides the time base reference. After the sinusoidal signal is filtered to eliminate higher harmonics, it is input to a microwave frequency multiplier chain for frequency boosting. The frequency-multiplied signal is then stabilized by a dielectric resonator oscillator (composed of a quartz crystal temperature-controlled oscillator, DDS, low-pass filter, microwave frequency multiplier chain, and dielectric resonator connected in sequence; the quartz crystal temperature-controlled oscillator is connected to the clock input of the DDS, and the output of the dielectric resonator oscillator outputs the carrier fundamental frequency signal). The output is a carrier fundamental frequency signal with low fluctuation. The carrier fundamental frequency signal is transmitted to a phase keying circuit via a coaxial cable. The cable length is calibrated to an integer multiple of 1 / 4 wavelength of the example value to reduce the influence of standing waves.

[0087] S3.4 Convert the carrier base frequency signal into a binary phase control sequence to generate a phase keying control signal.

[0088] This invention first inputs the obtained carrier base frequency signal to a high-speed comparator circuit, compares it with a zero-potential reference level in real time, and generates a square wave clock signal. Then, the square wave clock signal is divided by a digital frequency divider. This embodiment uses a two-way frequency division to generate two orthogonal carrier signals, I and Q. The two carrier signals are respectively connected to the local oscillator port of a double-balanced mixer. The RF port of the mixer receives a digital modulation signal from a locked quantum reference value. The high-speed comparator, digital frequency divider, double-balanced mixer, operational amplifier adder circuit, and impedance matching output circuit are connected sequentially. The circuit consists of a high-speed comparator that receives the carrier baseband signal and outputs a square wave clock signal. A digital frequency divider generates a quadrature carrier signal based on the square wave clock signal. A double-balanced mixer outputs a mixed signal based on the quadrature carrier signal and the digital modulation signal. An operational amplifier adder circuit superimposes and shapes the mixed signal to generate a binary phase control sequence. This phase control sequence is input to a synchronous trigger, using the zero-crossing detection pulse of the carrier baseband signal as the trigger clock. This ensures that the transition edge of the phase control sequence aligns with the zero-crossing point of the carrier baseband signal. The resulting signal is then output as a phase-keying control signal via a level conversion circuit. This signal is output to the modulator through an impedance matching network. The phase-keying control signal is transmitted from the output of the level conversion circuit to the control input of the phase-keying modulator. The transmission path uses equal-length differential pairs to maintain phase consistency between the two control signals, with a length deviation less than 1 mm (example value) to ensure phase consistency. The rise time of the control signal is controlled within 1 nanosecond (example value) to ensure that the transient characteristics of the phase transition meet the BPSK modulation requirements.

[0089] S3.5. Perform phase transitions based on the phase keying control signal to generate a quantized carrier signal.

[0090] After the phase keying control signal is input to the microwave switching phase shifter, a high level selects the 0-degree phase path, and a low level selects the 180-degree phase path. The two phase paths are combined by a Wilkinson power divider and then filtered and attenuated by a dielectric resonator to obtain the quantized carrier signal. Therefore, the quantized carrier signal can be understood as the output signal synthesized by the power divider and then filtered and compensated by subsequent stages. In this embodiment, the set value is 10dBm ± 0.1dB (the reason for this value is to ensure that the quantized carrier signal meets the input power requirements of the subsequent modulation coupler and quadrature demodulation circuit, while avoiding a decrease in signal-to-noise ratio due to excessively low power or nonlinear distortion due to excessively high power; ±0.1dB is used to limit the output power fluctuation range and ensure stable carrier injection strength). The signal is output to the next stage circuit via a coaxial connector. The phase noise of the carrier signal is better than -105dBc / Hz at the example value of 1kHz offset, and the spurious suppression reaches more than 60dBc at the example value.

[0091] S4. Based on the DC differential voltage to be measured and the quantized carrier signal, a baseband signal is generated through waveform transformation, trigonometric identity spectrum expansion and low-pass filtering.

[0092] S4.1 Obtain the time-domain superimposed waveform from the DC differential voltage under test and the quantized carrier signal through electromagnetic field coupling.

[0093] In this embodiment, the DC differential voltage to be measured is connected to the main signal channel of the modulation coupler through a high-impedance probe, and the quantized carrier signal is connected to the secondary signal channel of the modulation coupler through a directional coupler. The modulation coupler performs electromagnetic field coupling on the DC differential voltage to be measured and the quantized carrier signal and outputs a hybrid coupled signal. The hybrid coupled signal is sequentially passed through a low-noise amplifier and a bandpass filter for amplitude enhancement and out-of-band noise suppression to form a time-domain superimposed waveform. At the same time, the quantized carrier signal is passed through a frequency divider circuit to generate a synchronous trigger signal, and the synchronous trigger signal is input to the trigger input terminal of an oscilloscope or a high-speed sampling circuit, so that the oscilloscope or high-speed sampling circuit acquires the time-domain superimposed waveform according to the phase starting point of the quantized carrier signal.

[0094] S4.2. Based on the time-domain superimposed waveform, apply the trigonometric identities to expand the difference frequency and frequency relationship between the analytical order sideband components and the carrier fundamental frequency signal to generate the modulation signal.

[0095] Expanding the trigonometric identities, the expression is:

[0096]

[0097] in, This represents the product of two cosine functions. and These represent two angular frequency variables with different frequencies or phases. Represents the sum and frequency components, This represents the difference frequency component.

[0098] The time-domain superimposed waveform is input to a digital signal processor (DSP). A Fast Fourier Transform (FFT) converts the time-domain signal into a frequency-domain spectrum. Frequency-domain analysis identifies the main spectral line and sideband components of the carrier fundamental frequency signal. This implementation uses digital quadrature demodulation to extract the phase relationship between the carrier fundamental frequency signal and the sideband components. The demodulated local oscillator frequency is locked to the carrier fundamental frequency, and the sideband components are down-converted to baseband using a digital mixer. The down-converted time-domain superimposed waveform is then filtered by a digital filter to separate the difference frequency component and the sum frequency component. The difference frequency component corresponds to the low-frequency information of the measured DC differential voltage, while the sum frequency component reflects the nonlinear characteristics of the modulation process. The demodulated difference frequency and sum frequency signals are output as analog modulation products via a digital-to-analog converter (DAC), achieving a conversion accuracy of 16 bits as shown in the example.

[0099] S4.3. Perform low-pass filtering on the modulation products to generate the baseband signal.

[0100] This invention inputs the modulation products into a Butterworth low-pass filter, the order of which is chosen to achieve a steep transition band characteristic. The filtering process employs a cascaded operational amplifier structure, with the gain of each stage set to 0.5 times the example value to ensure stability. The baseband modulation product signal is transmitted using twisted-pair shielded cable, with the shielding layer grounded at a single point to suppress common-mode interference. The final output baseband signal is digitally sampled using a 24-bit analog-to-digital converter.

[0101] S5. The baseband signal is decomposed into in-phase and quadrature components using phase-locked loop (PLL) and the mean square value is calculated for each component. Quantum precision sampling values ​​are then generated based on the quadrature demodulation principle. The process is as follows: Figure 4 As shown, the specific process includes the following steps.

[0102] S5.1 Decompose the baseband signal into in-phase and quadrature components using phase-locked loop (PLL) orthogonal decomposition.

[0103] The present invention first inputs the baseband signal to the digital quadrature demodulator. The local oscillator signal of the quadrature demodulator is provided by the quantized carrier signal after frequency division. The local oscillator signal generates two quadrature reference signals with phase difference through the digital phase shifter. The baseband signal is mixed with the two quadrature reference signals in the digital mixer. The two signals output by filtering are the in-phase component and the quadrature component.

[0104] It should also be noted that phase-locked quadrature decomposition refers to using the synchronization reference signal obtained by frequency division of the quantized carrier signal as the quadrature local oscillator to decompose the baseband signal into in-phase and quadrature components. Compared with ordinary quadrature decomposition, the reference phase of phase-locked quadrature decomposition is synchronized with the quantized carrier signal, thus reducing the impact of carrier phase drift on the demodulation results.

[0105] S5.2. Convert the in-phase component into a power signal, perform energy integration within a fixed time window controlled by a rubidium atomic clock, and generate the mean square value of the in-phase component.

[0106] The in-phase component is input to a precision square-law detector circuit. The squared signal is then passed through an integrating amplifier with input impedance. The integration time window is precisely controlled by a gate signal obtained by dividing the rubidium atomic clock frequency reference. The integration result is sampled by a 24-bit analog-to-digital converter with a conversion rate set to 10 s / s (example value) and an effective bit depth of 20 bits (example value). The mean square value of the in-phase component is processed by a digital smoothing filter with a filter time constant set to 10 seconds (example value) and an output data update interval of 1 second (example value).

[0107] S5.3. Based on the orthogonal components, the mean square value of the orthogonal components is obtained by square-law detection and time window integration.

[0108] This invention inputs the quadrature component to a square-law detector circuit that is completely symmetrical to the in-phase component processing channel. The detector uses Schottky diodes of the same specifications to ensure consistent characteristics between the two channels. The squared quadrature component voltage signal is then accumulated by an integrating amplifier with an input impedance of 1 MΩ (example value). The integration time window is synchronously controlled by a 1-second gate signal provided by a rubidium atomic clock frequency reference (example value). The integrating capacitor is the same polypropylene film capacitor used in the in-phase component channel, with a capacitance matching accuracy better than the example value by 0.1%. The quadrature component mean square value is digitized by a 24-bit analog-to-digital converter, with a conversion rate synchronized with the in-phase component channel at 10 s / s (example value). The measurement results are processed by the same digital smoothing filter, with filter parameters set to a 10-second time constant (example value), ensuring that the output data is updated synchronously with the in-phase component mean square value.

[0109] S5.4. Based on the mean square values ​​of the in-phase components and the quadrature components, and after amplitude reconstruction, a demodulated signal is generated.

[0110] This invention inputs the mean square values ​​of the in-phase and quadrature components into a digital signal processor (DSP). Amplitude reconstruction is achieved through a coordinate rotation digital calculation method. The amplitude reconstruction process utilizes existing coordinate rotation digital calculation methods, where the DSP performs sum-of-squares, square root, and phase inversion calculations on the mean square values ​​of the in-phase and quadrature components to obtain the amplitude and phase information of the demodulated signal. During reconstruction, a lookup table method is used to calculate the square root function, and the phase information of the demodulated signal is obtained through the arctangent function, achieving an angular resolution of 0.01 degrees (example value). The reconstructed demodulated signal is output via a 24-bit digital-to-analog converter with a conversion rate of 100 kS / s (example value) and a settling time of less than 1 microsecond (example value). The output signal is driven by a buffer amplifier with an output impedance of 50 ohms (example value), achieving an amplitude accuracy of 0.01% (example value) and long-term stability better than 1 ppm / day (example value).

[0111] S5.5 Combine the demodulated signal with the reference amplitude of the quantized carrier signal, and output the quantum precision sampled value through proportional conversion.

[0112] This invention inputs the reference amplitudes of the demodulated signal and the quantized carrier signal into a proportional operation circuit. The proportional operation circuit performs a proportional conversion on the demodulated signal according to the reference amplitude of the quantized carrier signal to obtain a sampled conversion signal. The sampled conversion signal is then input into a 24-bit digital-to-analog converter for digital-to-analog conversion. The 24-bit digital-to-analog converter reduces the integral nonlinearity error during the conversion process through a segmented resistor network and error calibration logic to generate a sampled analog signal. The sampled analog signal is then input into a voltage output stage. The voltage output stage uses a low-offset operational amplifier and a power follower output structure to buffer the sampled analog signal and reduces the output impedance through negative feedback to generate a quantum-precision sampled value. The path from the output of the 24-bit digital-to-analog converter to the voltage output stage uses a low thermoelectric potential connector to reduce the influence of the connection contact potential on the quantum-precision sampled value.

[0113] It should also be noted that the segmented resistor network is constructed from high-order coarse-adjustment resistor segments and low-order fine-adjustment resistor segments. First, precision resistors with consistent temperature drift are selected and arranged into multiple resistor segments according to binary weights or an R-2R trapezoidal structure. Then, the connection status of each resistor segment is controlled by analog switches, so that the digital input code selects different resistor combinations. Finally, the voltage or current of each resistor segment is combined to the output node to form an analog output signal corresponding to the digital input code.

[0114] By dynamically initializing the quantum reference using a Josephson junction array and locking it in a closed loop, the physical definition and accuracy traceability of DC voltage are achieved. A phase-keyed carrier wave is generated using a rubidium atomic clock, and anti-interference baseband signals are extracted through spectral expansion and orthogonal demodulation, ultimately outputting a sampled value with dual time-frequency domain quantization. This scheme, centered on end-to-end quantization (voltage-frequency-time) and dynamic adaptive adjustment, achieves both sub-ppm accuracy and anti-interference capabilities over a wide measurement range, providing an atomic clock-level stable and engineering-feasible solution for DC power trading metering in smart grids.

[0115] Implementation Method 2 of Voltage Quantum Differential Sampling Method for DC Metering of Electricity Meters

[0116] 1. Initialize the Josephson junction array parameters based on the amplitude of the DC differential voltage to be measured, and generate the microwave drive frequency and initial value of the quantum reference.

[0117] Specifically, firstly, the positive and negative terminal potentials of the DC differential voltage to be measured are acquired, and the amplitude of the DC differential voltage to be measured is obtained from these potentials. The amplitude of the DC differential voltage to be measured is equal to the difference between the positive and negative terminal potentials. Secondly, based on the amplitude of the DC differential voltage to be measured, the initial number of junctions and the initial value of the microwave driving frequency in the Josephson junction array parameters are obtained. The initial value of the quantum reference is calculated through the voltage-frequency relationship, and the microwave driving frequency is generated.

[0118] 2. Calculate the residual voltage based on the initial value of the quantum reference and the DC differential voltage to be measured, generate adjustment commands, and dynamically adjust the initial value of the quantum reference through microwave drive frequency to generate a locked quantum reference value.

[0119] The adjustment command is generated when the residual voltage is greater than the quantum locking error tolerance. When the residual voltage is greater than the quantum locking error tolerance, the adjustment command is generated, and the generation of the locking quantum reference value is controlled according to the adjustment command. Specifically, it includes: correcting the microwave driving frequency by the polarity direction of the residual voltage to obtain the corrected microwave driving frequency; calculating the new initial value of the quantum reference with the corrected microwave driving frequency; and using the new initial value of the quantum reference to calculate the new residual voltage; when the new residual voltage is less than or equal to the quantum locking error tolerance, the new initial value of the quantum reference is used as the locking quantum reference value.

[0120] 3. Using the locked quantum reference value as the frequency reference of the rubidium atomic clock, a phase-keyed carrier wave is generated, and a quantized carrier signal is obtained from it.

[0121] Using a locked quantum reference value as the frequency reference for a rubidium atomic clock includes:

[0122] The quantum reference value is locked by matching the modulation range of the rubidium atomic clock frequency reference through voltage amplitude, and a reference voltage signal is output.

[0123] Error signals are extracted from reference voltage signals, and feedback is collected to adjust the frequency of microwave signals. The error signals are demodulated by lock-in amplification and frequency adjustment to generate transition frequencies. Electromagnetic wave oscillation is controlled based on the transition frequencies to generate carrier fundamental frequency signals. The carrier fundamental frequency signals are converted into binary phase control sequences to generate phase keying control signals. Phase transitions are performed according to the phase keying control signals to generate quantized carrier signals.

[0124] 4. Generate a baseband signal based on the DC differential voltage to be measured and the quantized carrier signal.

[0125] Specifically, the generation of the baseband signal includes: obtaining a time-domain superimposed waveform from the DC differential voltage under test and the quantized carrier signal through electromagnetic field coupling; expanding the difference frequency and frequency relationship between the analytical order sideband components and the carrier fundamental frequency signal using trigonometric identities based on the time-domain superimposed waveform to generate modulation products; and performing low-pass filtering on the modulation products to generate the baseband signal.

[0126] 5. The baseband signal is orthogonally decomposed into in-phase and quadrature components, and the mean square value is calculated for each component. Quantum precision sampling values ​​are generated based on the principle of quadrature demodulation.

[0127] Specifically, the generation of quantum-precision sampled values ​​includes: orthogonally decomposing the baseband signal into in-phase and quadrature components; converting the in-phase component into a power signal and integrating the energy within a fixed time window controlled by a rubidium atomic clock to generate the mean square value of the in-phase component; obtaining the mean square value of the quadrature component through square-law detection and time window integration based on the quadrature component; generating a demodulated signal based on the mean square values ​​of the in-phase and quadrature components and performing amplitude reconstruction; and combining the demodulated signal with the reference amplitude of the quantized carrier signal and converting it through a proportional relationship to output the quantum-precision sampled value.

[0128] Implementation of Voltage Quantum Differential Sampling System for DC Metering of Electricity Meters

[0129] like Figure 5 As shown, the system specifically includes a memory, a processor, a system bus, and a computer program stored in the memory. The processor and memory communicate and interact with each other via the system bus. The processor executes the computer program to implement the steps of the voltage quantum differential sampling method for DC metering of the present invention, as described in Embodiment 1 of the voltage quantum differential sampling method for DC metering of the present invention. The processor is a microprocessor (MCU) or other device capable of providing computing and control capabilities. The memory can be any type of memory that stores information using electrical energy, such as non-volatile storage media (including computer programs, databases), or other types of memory. The communication interface is used for wired or wireless communication with external terminals. Wireless communication can be achieved through WIFI, carrier networks, NFC (Near Field Communication), or other technologies. A display screen can also be connected; the display screen can be an LCD screen or an e-ink screen. The input device of the system can be a touch layer covering the display screen, or it can be a button, trackball, or touchpad, or an external keyboard, touchpad, or mouse.

[0130] Computer-readable storage medium implementation

[0131] This invention also provides a computer-readable storage medium storing a computer program. This computer program is executed by a processor to implement the voltage quantum differential sampling method for DC metering of the present invention, as described in Embodiment 1 of the voltage quantum differential sampling method for DC metering of the present invention. The computer-readable storage medium can be implemented by any type of volatile or non-volatile storage device or a combination thereof, such as Static Random Access Memory (SRAM), Electrically Erasable Programmable Read-Only Memory (EEPROM), Erasable Programmable Read Only Memory (EPROM), Programmable Red-Only Memory (PROM), Read-Only Memory (ROM), magnetic storage, flash memory, magnetic disk, or optical disk. As one implementation, this computer-readable storage medium can be installed in an energy meter, i.e., the voltage quantum differential sampling method for DC metering of the present invention can be run in the energy meter.

[0132] In summary, this invention achieves the physical definition and accuracy traceability of DC voltage by dynamically initializing the quantum reference and locking it in a closed loop using a Josephson junction array. It combines a rubidium atomic clock to generate a phase-keyed carrier wave, and utilizes spectral expansion and orthogonal demodulation to extract an anti-interference baseband signal, ultimately outputting a sampled value with dual time-frequency domain quantization. This scheme, centered on end-to-end quantization (voltage-frequency-time) and dynamic adaptive adjustment, achieves both sub-ppm accuracy and anti-interference capability over a wide measurement range, providing an atomic clock-level stable and engineering-feasible solution for DC power trading metering in smart grids.

Claims

1. A voltage quantum differential sampling method for DC metering of electricity, characterized in that, The method includes the following steps: 1) Initialize the Josephson junction array parameters based on the amplitude of the DC differential voltage to be measured, and generate the microwave drive frequency and initial value of the quantum reference; 2) Calculate the residual voltage based on the initial value of the quantum reference and the DC differential voltage to be measured, generate adjustment commands, and dynamically adjust the initial value of the quantum reference through the microwave drive frequency to generate a locked quantum reference value; 3) Using the locked quantum reference value as the frequency reference of the rubidium atomic clock, a phase-keyed carrier wave is generated, and a quantized carrier signal is obtained from it; 4) Generate a baseband signal based on the measured DC differential voltage and the quantized carrier signal; 5) The baseband signal is orthogonally decomposed into in-phase and quadrature components, and the mean square value is calculated for each component. Quantum precision sampling values ​​are generated based on the principle of quadrature demodulation.

2. The voltage quantum differential sampling method for DC metering of electricity meters according to claim 1, characterized in that, The process of generating the microwave driving frequency and the initial value of the quantum reference includes: The positive and negative terminal potentials of the DC differential voltage to be measured are collected, and the amplitude of the DC differential voltage to be measured is obtained accordingly. Based on the amplitude of the DC differential voltage to be measured, the initial number of junctions and the initial value of the microwave driving frequency in the Josephson junction array parameters are obtained. The initial value of the quantum reference is calculated through the voltage-frequency relationship, and the microwave driving frequency is generated.

3. The voltage quantum differential sampling method for DC metering of electricity according to claim 1, characterized in that, The adjustment command in step 2) is generated when the residual voltage is greater than the quantum lock error tolerance.

4. The voltage quantum differential sampling method for DC metering of electricity according to claim 1, characterized in that, The generation of the locked quantum benchmark value includes: The microwave driving frequency is corrected by the polarity of the residual voltage to obtain the corrected microwave driving frequency. The new initial value of the quantum reference is calculated using the corrected microwave driving frequency, and the new residual voltage is calculated using the new initial value of the quantum reference. When the new residual voltage is less than or equal to the quantum locking error tolerance, the new initial quantum reference value is used as the locked quantum reference value.

5. The voltage quantum differential sampling method for DC metering of electricity according to claim 1, characterized in that, Using a locked quantum reference value as the frequency reference for a rubidium atomic clock includes: The quantum reference value is locked by matching the modulation range of the rubidium atomic clock frequency reference through voltage amplitude, and a reference voltage signal is output. The error signal is extracted from the reference voltage signal, and the frequency of the feedback-adjusted microwave signal is collected. The error signal is demodulated by lock-in amplification and the frequency is driven to adjust the frequency, thereby generating the transition frequency.

6. The voltage quantum differential sampling method for DC metering of electricity meters according to claim 5, characterized in that, The generation of the quantized carrier signal in step 3) includes: Electromagnetic wave oscillation is controlled by the transition frequency to generate a carrier fundamental frequency signal; The carrier base frequency signal is converted into a binary phase control sequence to generate a phase keying control signal; Phase switching is performed based on the phase keying control signal to generate a quantized carrier signal.

7. The voltage quantum differential sampling method for DC metering of electricity meters according to claim 1, characterized in that, The generation of the baseband signal in step 4) includes: The time-domain superposition waveform is obtained from the DC differential voltage under test and the quantized carrier signal by electromagnetic field coupling; Based on the time-domain superimposed waveform, the difference-frequency and frequency relationship between the analytical order sideband components and the carrier fundamental frequency signal is expanded using the trigonometric identity to generate the modulation product. The modulation product is low-pass filtered to generate the baseband signal.

8. The voltage quantum differential sampling method for DC metering of electricity according to claim 1, characterized in that, The generation of quantum precision sample values ​​in step 5) includes: The baseband signal is orthogonally decomposed into in-phase and quadrature components; The in-phase component is converted into a power signal, and energy is integrated within a fixed time window controlled by a rubidium atomic clock to generate the mean square value of the in-phase component. Based on the orthogonal components, the mean square value of the orthogonal components is obtained by square-law detection and time window integration; Based on the mean square values ​​of the in-phase and quadrature components, and after amplitude reconstruction, a demodulated signal is generated. The demodulated signal is combined with the reference amplitude of the quantized carrier signal, and then converted through a proportional relationship to output a quantum-precision sampled value.

9. A voltage quantum differential sampling system for DC metering of an electricity meter, comprising a processor, characterized in that, The processor executes a computer program to implement the steps of the voltage quantum differential sampling method for DC metering of the electricity meter according to any one of claims 1 to 8.

10. A computer-readable storage medium having a computer program stored thereon, characterized in that, When executed by a processor, the computer program implements the steps of the voltage quantum differential sampling method for DC metering of an electricity meter as described in any one of claims 1 to 8.

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