A dual-pulse radar detection method based on frequency domain multiplexing and digital separation

Abstract

The application provides a double-pulse radar detection method based on frequency domain multiplexing and digital separation, belongs to the technical field of radar signal processing, and comprises the following steps: transmitting a wide pulse and a narrow pulse combined waveform in the same pulse repetition interval, carrying out digital sampling, data copying, digital mixing, digital filtering, frequency domain separation and pulse compression on a return signal, carrying out coherent accumulation and constant false alarm detection to obtain a target track, carrying out spectrum splicing based on the distance of the target track to obtain a target point distance, and finally determining a single-side pulse angle of the target point according to channel data after the coherent accumulation, so that the measurement of the target point is realized.

Description

Technical Field

[0001] This invention belongs to the field of radar signal processing technology, and specifically relates to a dual-pulse radar detection method based on frequency domain multiplexing and digital separation. Background Technology

[0002] Pulse Doppler radar is a core sensor in modern airborne detection systems. It detects, tracks, and measures the Doppler frequency shift of target echoes by transmitting periodic pulse trains. In airborne radar waveform design, engineers face a mutually constraining technical challenge: the transmission energy required for long-range detection versus the limitation of the blind zone at close range. The blind zone size is directly determined by the pulse width. While wide pulses provide high transmission energy to ensure long-range detection, they introduce a blind zone of hundreds of meters or more, preventing the radar from detecting nearby targets. Narrow pulses, on the other hand, can compress the blind zone to a few hundred meters, but insufficient energy results in a loss of long-range detection capability. This fundamental contradiction has long constrained traditional single-pulse waveform modes, making it difficult to simultaneously meet the dual requirements of both long-range and short-range detection.

[0003] To alleviate this contradiction, two main improvement schemes have been developed in engineering practice.

[0004] The first type is the alternating transmission mode, which alternates between transmitting wide and narrow pulses in different frames or pulse repetition cycles. Although this scheme achieves a certain degree of coverage at both long and short ranges, the radar data rate is only 50% of that of a single waveform. For high-speed maneuvering targets and low-altitude fast-penetrating targets, the reduction in data rate directly weakens the radar's continuous tracking capability, resulting in a longer track update cycle, accumulated extrapolation errors, and a significant increase in the risk of track interruption.

[0005] The second method is sequential transmission mode. This scheme transmits a wide pulse and a narrow pulse sequentially within the same repetition period. Specifically, a wide pulse is transmitted first, then the transmitter switches to receiver mode via a transceiver switch to obtain long-range echoes. After the wide pulse echo reception window closes, the transmitter switches back to transmit a narrow pulse, and then switches back to receiver mode again to obtain short-range echoes. This mode maintains a constant repetition frequency, avoiding data rate loss. However, each switch requires a protection time (the delay required for the switch to fully return to receiver mode from transmit mode), which directly limits the minimum detection range achievable by the radar. More importantly, sequential transmission mode requires multiple cascaded RF switches, causing system losses. This not only worsens the noise figure of the receiver link but also reduces the transmission efficiency from transmitter to antenna, ultimately resulting in a maximum detection range loss of approximately 10% to 20%. In addition, the tight temporal coupling between wide and narrow pulses can cause serious conflicts in echo reception: the receiver is turned off during wide pulse transmission, making it impossible to receive any near-range echoes; conversely, narrow pulse transmission blocks the reception of echoes from far-range targets corresponding to wide pulses; within a specific range, two echoes may even arrive at the receiver simultaneously, and their cross-correlation sidelobes may produce false targets or mask weak targets. This temporal constraint makes the design of the receiving window extremely complex, making it difficult to achieve seamless transitions between near and far-range detection, resulting in detection blind spots or overlapping interference zones in the intermediate range.

[0006] Therefore, neither alternating nor sequential transmission modes can simultaneously solve the problem of effective short-range blind spot coverage without reducing the data rate. Thus, there is an urgent need for a radar technology solution for multi-scenario target detection that can simultaneously achieve high-sensitivity long-range detection and high-resolution short-range blind spot coverage within the same PRI, without relying on time-domain switching. This solution should be able to efficiently separate and optimize the frequency domain of overlapping wide and narrow pulse echoes, and achieve seamless coverage for both long and short range detection through spectrum stitching. Summary of the Invention

[0007] The purpose of this invention is to provide a dual-pulse radar detection method based on frequency domain multiplexing and digital separation, so as to solve or alleviate at least one of the problems in the background art.

[0008] The technical solution of this invention is: a dual-pulse radar detection method based on frequency domain multiplexing and digital separation, comprising:

[0009] Step 1: Modulate wide pulse transmission signals and narrow pulse transmission signals of different frequencies, wherein the wide pulse transmission signals and narrow pulse transmission signals are transmitted within the same pulse repetition interval;

[0010] Step 2: Receive the mixed echo signal of the wide pulse transmission signal and the narrow pulse transmission signal, perform signal processing on the mixed echo signal to obtain a digital mixed echo signal, and divide the digital mixed echo signal into two identical signals. The first digital mixed echo signal is digitally mixed at a first digital local oscillator frequency to obtain a wide pulse echo signal, and the second digital mixed echo signal is digitally mixed at a second digital local oscillator frequency to obtain a narrow pulse echo signal.

[0011] Step 3: Perform frequency domain separation and pulse compression on the wide pulse echo signal and the narrow pulse echo signal respectively to obtain the wide pulse echo compressed signal and the narrow pulse echo compressed signal.

[0012] Step 4: Perform coherent accumulation on the wide pulse echo compression signal and the narrow pulse echo compression signal respectively, and perform constant false alarm rate detection based on the accumulated wide pulse echo compression signal and narrow pulse echo compression signal to obtain the target point trace.

[0013] Step 5: Based on the distance to the target point, the spectra of the accumulated wide pulse echo compressed signal and the narrow pulse echo compressed signal are spliced ​​together to obtain the spliced ​​spectrum, and the distance to the target point is obtained based on the spliced ​​spectrum.

[0014] Step 6: Select the corresponding accumulated wide pulse echo compression signal or accumulated narrow pulse echo compression signal based on the distance to the target point, and determine the single pulse angle measurement of the target point based on the selected accumulated wide pulse echo compression signal or accumulated narrow pulse echo compression signal.

[0015] In a preferred embodiment of the present invention, a wide-pulse transmission signal is modulated by linear frequency modulation, and the time-domain expression of the wide-pulse transmission signal is: sw(t)=rect(t / Tw)*exp[j*(2π*f c1 [*t+π*K1*t²)], where T w K1 = B, where B is the pulse width of the wide-pulse transmitted signal. w / T w f is the linear frequency modulation slope. c1 The center frequency of the wide pulse transmission signal;

[0016] A narrow pulse transmission signal is modulated using a single-carrier rectangular pulse. The time-domain expression of the narrow pulse transmission signal is: sn(t) = rect((tT) d ) / Tn)*exp(j*2π*f c2 *t), where T n T is the pulse width of the narrow pulse transmitted signal. d For time delay, under the condition of overlapping transmission in the time domain, T is satisfied. d =0, f c2 The center frequency of the narrow pulse transmission signal;

[0017] The frequency interval Δf=|f c2 -f c1 The non-aliasing constraint condition for the spectrum is: Δf>=(B w +B n ) / 2+δf, where B w For the bandwidth of the wide pulse transmission signal, B n δf is the bandwidth of the narrow pulse transmit signal, and δf is the spectral guard interval.

[0018] The radio frequency coverage constraint is: Δf+B w / 2+B n / 2<=B RF Among them, B RF This refers to the passband width of the narrowband filter in the RF front end.

[0019] In a preferred embodiment of the present invention, the method for obtaining a wide-pulse echo signal by digitally mixing the first digital hybrid echo signal using a first digital local oscillator frequency is as follows: s d1 (t)=smix(t)*exp(-j*2π*f IF *t), f IF This is the first digital local oscillator frequency;

[0020] The method for obtaining a narrow pulse echo signal by digitally mixing the second digital hybrid echo signal using the second digital local oscillator frequency is as follows: s d2 (t)=smix(t)*exp(-j*2π*(f IF +Δf)*t).

[0021] In a preferred embodiment of the present invention, the spectrum splicing method is as follows:

[0022] Sout(f,R)=α(R)*Sn(f)+β(R)*Sw(f)

[0023] The clock, α(R) and β(R) are distance weighting coefficients that satisfy α(R) + β(R) = 1, Sn(f) is the narrow pulse spectrum after coherent accumulation, and Sw(f) is the wide pulse spectrum after coherent accumulation;

[0024] Where the distance to the target point R <= R stitch When α(R)≈1 and β(R)≈0, narrow pulse echo data is used; when the target distance R>R stitch When α(R)≈0, β(R)≈1, wide pulse echo data is used, R stitch This represents the distance between the splicing points.

[0025] In a preferred embodiment of the present invention, step six, determining the single-pulse angle measurement of the target point based on the accumulated wide-pulse echo compression signal or the accumulated narrow-pulse echo compression signal, includes:

[0026] Azimuth θ Az =arcsin(k*|S ΔAz | / |SΣ|*cosφ)

[0027] Pitch angle θ El =arcsin(k*|S ΔEl | / |SΣ|)

[0028] Where k is the single-pulse slope, φ is the current beam pointing angle, SΣ is the sum channel signal, and S ΔAz For the azimuth difference channel signal, S ΔE l represents the pitch difference channel signal.

[0029] The method of this invention sequentially transmits wide and narrow pulses within the same PRI without increasing the pulse repetition period. It has high frequency domain reuse efficiency, shares the RF front-end, IF channel and ADC, eliminates the need for dual-channel hardware, and has low hardware cost. Digital domain NCO mixing + LPF achieves complete separation, eliminating analog crosstalk. Spectrum splicing achieves seamless coverage of near-range high resolution and long-range high sensitivity, resulting in a wider coverage range. Attached Figure Description

[0030] To more clearly illustrate the technical solution provided by the present invention, the accompanying drawings will be briefly described below. Obviously, the drawings described below are merely some embodiments of the present invention.

[0031] Figure 1 This is a schematic diagram of the frequency domain multiplexing dual-pulse radar detection process of the present invention.

[0032] Figure 2 This is a schematic diagram of a time-domain overlapping transmission waveform of wide pulse and narrow pulse in one embodiment of the present invention.

[0033] Figure 3 This is a schematic diagram of a digital spectrum separation process according to an embodiment of the present invention.

[0034] Figure 4 This is a schematic diagram of spectrum splicing according to an embodiment of the present invention.

[0035] Figure 5 This is a schematic diagram of the detection range coverage according to an embodiment of the present invention. Detailed Implementation

[0036] To make the objectives, technical solutions, and advantages of this invention clearer, the technical solutions in the embodiments of this invention will be described in more detail below with reference to the accompanying drawings.

[0037] To address the technical problems in existing radar detection technologies, such as the difficulty in balancing long-range and short-range detection and the reduction in data rate caused by time-division multiplexing, this invention provides a dual-pulse radar detection method based on frequency domain multiplexing and digital separation. By transmitting a combination waveform of wide and narrow pulses separated by the radio frequency domain within the same pulse repetition interval, the radio frequency front-end is highly multiplexed, the system is simplified, and it is a radar detection scheme that does not reduce the data rate.

[0038] As shown in Figure 1, the dual-pulse radar detection method based on frequency domain multiplexing and digital separation provided by the present invention includes the following steps:

[0039] Step 1: Modulate wide-pulse transmission signals and narrow-pulse transmission signals of different frequencies, wherein the wide-pulse transmission signals and narrow-pulse transmission signals are transmitted within the same pulse repetition interval.

[0040] This invention transmits wide and narrow pulses within the same pulse repetition interval (PRI), such as... Figure 2 As shown, wide pulses and narrow pulses are transmitted in a time-domain overlapped sequence, with the wide pulses preceding the narrow pulses. The wide pulses use the first center frequency f. c1 With the first bandwidth Bw, the narrow pulse uses the second center frequency f. c2 The second bandwidth Bn, where f c1 with f c2 There is a frequency interval Δf between them.

[0041] The wide-pulse transmitted signal uses linear frequency modulation (LFM), and its time-domain expression is as follows:

[0042] sw(t)=rect(t / Tw)*exp[j*(2π*f c1 *t+π*K1*t²)]

[0043] In the formula, T w Wide pulse width (T) w ≥Pulse resolution), B w For a wide pulse bandwidth, K1=B w / T w This represents the linear frequency modulation slope.

[0044] The narrow pulse transmission signal uses single-carrier rectangular pulse modulation, and its time-domain expression is as follows:

[0045] sn(t) = rect((tT) d ) / Tn)*exp(j*2π*f c2 *t)

[0046] In the formula, T n Narrow pulse width (T) w ≥Pulse resolution) && (T n×c / 2≤external detection blind zone requirement), T d For time delay, under the condition of overlapping transmission in the time domain, T is satisfied. d =0.

[0047] The frequency interval Δf=|f c2 -f c1 The non-aliasing constraint condition for the spectrum is:

[0048] Δf>=(B w +B n ) / 2+δf

[0049] Where δf is the spectrum guard interval. The radio frequency coverage constraint is:

[0050] Δf+B w / 2+B n / 2<=B RF

[0051] Among them, B RF This refers to the passband width of the narrowband filter in the RF front end.

[0052] For example, in this embodiment of the present invention, the specific parameters of the transmission signal designed above are as follows:

[0053] The wide pulse uses the first center frequency f c1 =9.55GHz and first bandwidth B w =5MHz, the modulation method is linear frequency modulation (LFM), and the pulse width is T. w =10μs, used for target detection at long distances of 2~10km;

[0054] Narrow pulse uses the second center frequency f c2 =9.57GHz and second bandwidth B n =20MHz, modulation method uses single-carrier rectangular pulse, pulse width is T n =0.2μs, used for target detection at close range of 60m~2.5km;

[0055] First center frequency f c1 With the second center frequency f c2 There exists a frequency interval Δf=|f c2 -f c1 |=20MHz, satisfying: Δf≥(B w +B n ) / 2+δf=12.5MHz+7.5MHz=20MHz, where δf=7.5MHz is the spectrum guard interval, which ensures that the spectrum of the wide pulse and the narrow pulse are not mixed in the frequency domain and can be separated by subsequent digital filters;

[0056] Simultaneously satisfying: Δf + Bw / 2+B n / 2=20+2.5+10=32.5MHz≤B RF =40MHz, where B RF =40MHz is the passband width of the RF front-end narrowband filter, with a center frequency of approximately 9.56GHz, covering the 9.54~9.58GHz frequency band. This ensures that the spectrum of both wide and narrow pulses falls within the passband range of the RF front-end, eliminating the need to replace or switch the RF front-end.

[0057] Step 2: Receive the mixed echo signal of the wide pulse transmission signal and the narrow pulse transmission signal, perform signal processing on the mixed echo signal to obtain the digital mixed echo signal, and divide the digital mixed echo signal into two identical signals. The first digital mixed echo signal is digitally mixed at the first digital local oscillator frequency to obtain the wide pulse echo signal, and the second digital mixed echo signal is digitally mixed at the second digital local oscillator frequency to obtain the narrow pulse echo signal.

[0058] The system receives a mixed echo signal of wide and narrow pulses. The mixed echoes completely overlap in the time domain and cannot be separated by a time-domain switch. After being amplified by a low-noise LNA, the RF signal is converted to an intermediate frequency (IF) and filtered through two down-conversion processes. For example, in this embodiment of the invention, the two down-conversion processes are: the first down-conversion uses the first local oscillator frequency f. LO1 =7.95GHz, the wide pulse center frequency f c1 =9.55GHz downconverted to 1600MHz intermediate frequency, reducing the narrow pulse center frequency f c2 =9.57GHz downconversion to 1620MHz intermediate frequency; the second downconversion uses the second local oscillator frequency f LO2 =1520MHz, downconvert the 1600MHz intermediate frequency to the final intermediate frequency of 80MHz, and downconvert the 1620MHz intermediate frequency to the final intermediate frequency of 100MHz; after two downconversions, the center frequency of the wide pulse echo is 80MHz and the center frequency of the narrow pulse echo is 100MHz, and the two are kept 20MHz apart in the final intermediate frequency domain.

[0059] The filtered intermediate frequency signal is subjected to high-speed analog-to-digital conversion (ADC) at a sampling rate f. s It satisfies the Nyquist sampling theorem. For example, in this embodiment of the invention, the sampling rate f s =250MHz, satisfying the Nyquist sampling theorem f s ≥2×(100MHz+10MHz)=220MHz.

[0060] The sampled digitized mixed echo data is copied into two identical data streams, designated as the first and second streams, respectively. This data copying is performed in the digital domain without requiring additional RF hardware. For the first stream, digital mixing is performed at the first digital local oscillator frequency NCO1, and the wide-pulse echo is shifted to the baseband: s d1 (t)=smix(t)*exp(-j*2π*f IF *t), where f IF This is the first digital local oscillator frequency;

[0061] For the second data stream, the second digital local oscillator frequency NCO2=f IF +Δf performs digital mixing to shift the narrow pulse echo to baseband: s d2 (t)=smix(t)*exp(-j*2π*(f IF +Δf)*t).

[0062] Step 3: Perform frequency domain separation and pulse compression on the wide pulse echo signal and the narrow pulse echo signal respectively to obtain the wide pulse echo compressed signal and the narrow pulse echo compressed signal.

[0063] In this invention, the first data stream undergoes frequency domain separation and pulse compression via a first digital filter. The pulse compression matched filter coefficients are: hw(n) = s*(-n)*wT(n), where s(t) is the transmitted wide-pulse LFM signal, and wT(n) is the Taylor window weighted sequence; the distance resolution is: ΔRw = c / (2*B w ).

[0064] The second data stream undergoes frequency domain separation and low sidelobe matched filtering via a second digital filter. The matched filter coefficients are: hn(m) = p*(-m)*wK(m), where p(t) is the transmitted narrow pulse signal and wK(m) is the Kaiser window weighted sequence; the range resolution is: ΔRn = c / (2*B n ).

[0065] For example, in this embodiment of the invention, the first data stream is digitally mixed at a first digital local oscillator frequency NCO1 = 80MHz: s_w(t) = s_mix(t) × e^(-j × 2π × 80MHz × t), where s_mix(t) is the mixed echo digital signal. After mixing, the wide pulse echo component is shifted to the baseband (zero intermediate frequency), while the narrow pulse echo component is shifted to a frequency offset of 20MHz. Frequency domain separation and pulse compression are performed using a digital filter.

[0066] First stage: Bandpass filter, center frequency 0 (baseband), bandwidth B. w =5MHz, used to filter out narrow pulse components and out-of-band noise located at a frequency offset of 20MHz;

[0067] Second stage: Pulse compression matched filter, with coefficients hw(n)=s*(-n)×wT(n), where s(t) is the transmitted wide pulse LFM signal (10μs, 5MHz), s*(-n) is the conjugate time inversion of the discrete sequence of the transmitted signal s(n), and wT(n) is the Taylor window weighted sequence.

[0068] For the second data stream, digital mixing is performed at the second digital local oscillator frequency NCO2 = 100MHz: s_n(t) = s_mix(t) × e^(-j × 2π × 100MHz × t). After mixing, the narrow pulse echo component is shifted to the baseband, while the wide pulse echo component is shifted to a frequency offset of -20MHz. Frequency domain separation and matched filtering are then performed using a digital filter.

[0069] First stage: Bandpass filter, center frequency 0 (baseband), bandwidth B. n =20MHz, used to filter out wide pulse components and out-of-band noise located at a frequency offset of -20MHz;

[0070] Second stage: Low sidelobe matched filter, with coefficients hn(m)=p*(-m)×wK(m), where p(t) is the transmitted narrow pulse signal (0.2μs), p*(-m) is the conjugate time inversion of the discrete sequence of the transmitted signal p(m), and wK(m) is the Kaiser window weighted sequence.

[0071] like Figure 3 As shown, Figure 3 (a) is the spectrum of the intermediate frequency sampled signal, with the wide pulse center located at f. IF The narrow pulse center is located at f IF +Δf; Figure 3 (b) is NCO1=f IF After mixing, the spectrum (wide pulse channel) shows that the wide pulse is shifted to the baseband (0 frequency) and the narrow pulse is shifted to +Δf. Figure 3 (c) is NCO2=f IF After mixing with +Δf, the spectrum (narrow pulse channel) is shifted from the narrow pulse to the baseband (0 frequency), and the wide pulse is shifted to -Δf. Figure 3 (d) shows the separated spectrum after digital low-pass filtering, where the two signals are completely separated in the baseband.

[0072] Step four: Perform coherent accumulation on the wide pulse echo compression signal and the narrow pulse echo compression signal respectively, and obtain the target point trace by constant false alarm rate detection based on the accumulated wide pulse echo compression signal and narrow pulse echo compression signal.

[0073] For each channel's data after pulse compression or matched filtering, coherent accumulation is performed over N consecutive pulse repetition cycles: Y(n,k)=|Sum_{m=0}^{N-1}*r[m,n]*exp(-j*2π*k*m / N)|², where r[m,n] is the echo sample of the nth range cell of the m-th PRI, and k is the Doppler cell number. The coherent accumulation gain is: GCI=10*log(N).

[0074] The accumulated data is fed into a constant false alarm rate detector for adaptive threshold detection to extract target points.

[0075] Step 5: Based on the distance to the target point, the spectra of the accumulated wide pulse echo compressed signal and the narrow pulse echo compressed signal are spliced ​​together to obtain the spliced ​​spectrum, and the distance to the target point is obtained based on the spliced ​​spectrum.

[0076] Based on the target distance, the detection results of narrow pulses and wide pulses are spliced ​​together using the following method: Sout(f,R)=α(R)*Sn(f)+β(R)*Sw(f)

[0077] Where α(R) and β(R) are range weighting coefficients, satisfying α(R) + β(R) = 1. Sn(f) is the narrow pulse spectrum after coherent accumulation, and Sw(f) is the wide pulse spectrum after coherent accumulation. The splicing rule is: when the target distance R <= R stitch When α(R)≈1 and β(R)≈0, high-resolution data with narrow pulses is used; when the target distance R>R stitch When α(R)≈0 and β(R)≈1, wide-pulse long-distance data is used. At the splicing point distance R... stitch It enables seamless transition between narrow pulse data and wide pulse data.

[0078] For example, in this embodiment of the invention, the splicing point distance R stitch =2.5km, at this time the stitching rule is: when the target distance R <= R stitch When the target distance is 2.5km, α(R)≈1, β(R)≈0, and high-resolution data with narrow pulses is used (narrow pulse channel data is preferred for track updates because narrow pulses have higher range resolution); when the target distance R>R stitch =2.5km, α(R)≈0, β(R)≈1, using wide-pulse long-distance data. At the splicing point R stitch Achieving seamless transition between narrow-pulse and wide-pulse data at a distance of 2.5km, such as... Figure 4 As shown.

[0079] Step 6: Select the corresponding accumulated wide pulse echo compression signal or accumulated narrow pulse echo compression signal based on the distance to the target point, and determine the single pulse angle measurement of the target point based on the selected accumulated wide pulse echo compression signal or accumulated narrow pulse echo compression signal.

[0080] The single-pulse angular measurement angle of the target is calculated using the amplitude ratio of the sum channel and the difference channel, which includes:

[0081] Azimuth θ Az =arcsin(k*|S ΔAz | / |SΣ|*cosφ)

[0082] Pitch angle θ El =arcsin(k*|S ΔEl | / |SΣ|)

[0083] Where k is the single-pulse slope, φ is the current beam pointing angle, SΣ is the sum channel signal, and S ΔAz For the azimuth difference channel signal, S ΔE l represents the pitch difference channel signal.

[0084] The separated data is packaged into eight parallel data channels according to beam channels: long-range sum channel (CH1), long-range azimuth difference channel (CH2), long-range elevation difference channel (CH3), long-range protection channel (CH4), short-range sum channel (CH5), short-range azimuth difference channel (CH6), short-range elevation difference channel (CH7), and short-range protection channel (CH8). Parallel signal processing is performed on the eight channels, including pulse compression, coherent accumulation, constant false alarm rate (CFAR) detection, and single-pulse angle measurement. Seamless coverage of near and long-range detection is achieved through spectrum stitching, such as... Figure 5 As shown, a wide pulse covers a long-distance area, while a narrow pulse covers R. min ~R stitch Close range area.

[0085] Compared with the prior art, the method of the present invention has the following significant advantages:

[0086] 1) High frequency domain reuse efficiency: sequential transmission within the same PRI does not increase the pulse repetition period;

[0087] 2) Low hardware cost: It shares the same RF front-end, IF channel, and ADC, eliminating the need for dual-channel hardware;

[0088] 3) Thorough signal separation: Digital domain NCO mixing + LPF achieves complete separation, with no analog crosstalk;

[0089] 4) Wide coverage: Spectrum splicing achieves seamless coverage of near-field high resolution and long-field high sensitivity.

[0090] This invention is applicable to fields such as short-to-medium range surveillance radar, weather radar, UAV detection radar, and short-range air defense radar.

[0091] The above description is merely a specific embodiment of the present invention, but the scope of protection of the present invention is not limited thereto. Any variations or substitutions that can be easily conceived by those skilled in the art within the technical scope disclosed in the present invention should be included within the scope of protection of the present invention. Therefore, the scope of protection of the present invention should be determined by the scope of the claims.

Claims

1. A dual-pulse radar detection method based on frequency domain multiplexing and digital separation, characterized in that, include: Step 1: Modulate wide pulse transmission signals and narrow pulse transmission signals of different frequencies, wherein the wide pulse transmission signals and narrow pulse transmission signals are transmitted within the same pulse repetition interval; Step 2: Receive the mixed echo signal of the wide pulse transmission signal and the narrow pulse transmission signal, perform signal processing on the mixed echo signal to obtain a digital mixed echo signal, and divide the digital mixed echo signal into two identical signals. The first digital mixed echo signal is digitally mixed at a first digital local oscillator frequency to obtain a wide pulse echo signal, and the second digital mixed echo signal is digitally mixed at a second digital local oscillator frequency to obtain a narrow pulse echo signal. Step 3: Perform frequency domain separation and pulse compression on the wide pulse echo signal and the narrow pulse echo signal respectively to obtain the wide pulse echo compressed signal and the narrow pulse echo compressed signal. Step 4: Perform coherent accumulation on the wide pulse echo compression signal and the narrow pulse echo compression signal respectively, and perform constant false alarm rate detection based on the accumulated wide pulse echo compression signal and narrow pulse echo compression signal to obtain the target point trace. Step 5: Based on the distance to the target point, the spectra of the accumulated wide pulse echo compressed signal and the narrow pulse echo compressed signal are spliced ​​together to obtain the spliced ​​spectrum, and the distance to the target point is obtained based on the spliced ​​spectrum. Step 6: Select the corresponding accumulated wide pulse echo compression signal or accumulated narrow pulse echo compression signal based on the distance to the target point, and determine the single pulse angle measurement of the target point based on the selected accumulated wide pulse echo compression signal or accumulated narrow pulse echo compression signal.

2. The dual-pulse radar detection method based on frequency domain multiplexing and digital separation as described in claim 1, characterized in that, A wide-pulse transmission signal is transmitted using linear frequency modulation. The time-domain expression of the wide-pulse transmission signal is: sw(t)=rect(t / Tw)*exp[j*(2π*f c1 [*t+π*K1*t²)], where T w K1 = B, where B is the pulse width of the wide-pulse transmitted signal. w / T w f is the linear frequency modulation slope. c1 The center frequency of the wide pulse transmission signal; A narrow pulse transmission signal is modulated using a single-carrier rectangular pulse. The time-domain expression of the narrow pulse transmission signal is: sn(t) = rect((tT) d ) / Tn)*exp(j*2π*f c2 *t), where T n T is the pulse width of the narrow pulse transmitted signal. d For time delay, under the condition of overlapping transmission in the time domain, T is satisfied. d =0, f c2 The center frequency of the narrow pulse transmission signal; The frequency interval Δf=|f c2 -f c1 The non-aliasing constraint condition for the spectrum is: Δf>=(B w +B n ) / 2+δf, where B w For the bandwidth of the wide pulse transmission signal, B n δf is the bandwidth of the narrow pulse transmit signal, and δf is the spectral guard interval. The radio frequency coverage constraint is: Δf+B w / 2+B n / 2<=B RF Among them, B RF This refers to the passband width of the narrowband filter in the RF front end.

3. The dual-pulse radar detection method based on frequency domain multiplexing and digital separation as described in claim 2, characterized in that, The method for obtaining a wide-pulse echo signal by digitally mixing the first digital hybrid echo signal using the first digital local oscillator frequency is as follows: s d1 (t)=smix(t)*exp(-j*2π*f IF *t), f IF This is the first digital local oscillator frequency; The method for obtaining a narrow pulse echo signal by digitally mixing the second digital hybrid echo signal using the second digital local oscillator frequency is as follows: s d2 (t)=smix(t)*exp(-j*2π*(f IF +Δf)*t).

4. The dual-pulse radar detection method based on frequency domain multiplexing and digital separation as described in claim 2, characterized in that, The spectrum splicing method is as follows: Sout(f,R)=α(R)*Sn(f)+β(R)*Sw(f) The clock, α(R) and β(R) are distance weighting coefficients that satisfy α(R) + β(R) = 1, Sn(f) is the narrow pulse spectrum after coherent accumulation, and Sw(f) is the wide pulse spectrum after coherent accumulation; Where the distance to the target point R <= R stitch When α(R)≈1 and β(R)≈0, narrow pulse echo data is used; when the target distance R>R stitch When α(R)≈0, β(R)≈1, wide pulse echo data is used, R stitch This represents the distance between the splicing points.

5. The dual-pulse radar detection method based on frequency domain multiplexing and digital separation as described in claim 4, characterized in that, Step six involves determining the single-pulse angle measurement of the target point based on the accumulated wide-pulse echo compression signal or the accumulated narrow-pulse echo compression signal, including: Azimuth θ Az =arcsin(k*|S ΔAz | / |SΣ|*cosφ) Pitch angle θ El =arcsin(k*|S ΔEl | / |SΣ|) Where k is the single-pulse slope, φ is the current beam pointing angle, SΣ is the sum channel signal, and S ΔAz For the azimuth difference channel signal, S ΔE l represents the pitch difference channel signal.

Images