A full-soft-switching parallelized two-stage converter for formation and distribution and a control method thereof

Abstract

The application discloses a full-soft-switching parallelized two-stage forming and distribution converter, which comprises a high-voltage direct-current input module, a front-stage power conversion module, an intermediate bus capacitor, a rear-stage power conversion module and a control module; the output end of the high-voltage direct-current input module is connected with the input end of the front-stage power conversion module, the output end of the front-stage power conversion module is connected with the input end of the rear-stage power conversion module through the intermediate bus capacitor, and the output end of the rear-stage power conversion module is used for connecting a battery load; the control module is configured to adjust the switching frequency of the front-stage power conversion module through an adaptive frequency modulation strategy, so that the inductance current of the front-stage power conversion module is zero in each switching cycle, and the zero-voltage turn-on of all the switching tubes is realized; the rear-stage power conversion module is configured to operate in a direct-current transformer mode at a resonant point in a fixed-frequency open-loop mode, so that the forward voltage gain and the reverse voltage gain are both 1, the primary-side switching tube realizes the zero-voltage turn-on, and the secondary-side switching tube realizes the zero-current turn-off.

Description

Technical Field

[0001] This invention relates to the field of battery conversion into capacity and separation technology, specifically to a fully soft-switching parallel two-stage converter for conversion into capacity and separation and its control method. Background Technology

[0002] Formation and capacity testing of lithium batteries is a critical process in lithium battery manufacturing. Parallel testing equipment typically uses an independent charging and discharging power supply for each battery. Currently, in the industry, parallel charging and discharging testing equipment, especially in high-voltage DC bus power supply systems, generally adopts a two-stage conversion structure for the formation and capacity testing power supply: the front stage realizes the isolation conversion and voltage matching from the high-voltage bus to the intermediate DC bus, and the rear stage realizes the charging and discharging control from the intermediate bus to the battery load.

[0003] In existing technologies, a common two-stage breakdown converter architecture is "front-stage isolated LLC resonant converter + rear-stage non-isolated Buck / Boost converter". The front-stage LLC resonant converter utilizes the inherent characteristics of the resonant network to achieve zero-voltage turn-on (ZVS) for the primary-side switch and zero-current turn-off (ZCS) for the secondary-side rectifier, offering significant advantages such as high efficiency and low electromagnetic interference. The rear-stage Buck / Boost converter then performs precise closed-loop regulation of the battery's charging and discharging current / voltage. However, this architecture has the following shortcomings in practical engineering applications:

[0004] Firstly, the downstream Buck / Boost converter operates in a hard-switching state, with a significant voltage-current overlap during the turn-on and turn-off processes, resulting in substantial switching losses. Especially under the high-current charging and discharging conditions required by the formation and capacity testing process, these losses severely limit further improvements in overall efficiency.

[0005] Secondly, in multi-parallel output scenarios, the inductor current ripple of traditional Buck / Boost converters is relatively large, which is not conducive to the stringent current accuracy requirements of the capacity breaking process, and requires increasing the volume of the filter inductor to meet the ripple index, thus restricting the improvement of the power density of the equipment. Summary of the Invention

[0006] To overcome the above-mentioned technical defects, the present invention provides a fully soft-switching parallel-connected two-stage capacity converter and a control method thereof;

[0007] To solve the above problems, the present invention is implemented according to the following technical solution:

[0008] In a first aspect, the present invention provides a fully soft-switching parallel-converted two-stage converter, comprising: a high-voltage DC input module, a front-stage power conversion module, an intermediate bus capacitor, a rear-stage power conversion module, and a control module;

[0009] The output terminal of the high voltage DC input module is connected to the input terminal of the front-stage power conversion module. The output terminal of the front-stage power conversion module is connected to the input terminal of the rear-stage power conversion module via the intermediate bus capacitor. The output terminal of the rear-stage power conversion module is used to connect to the battery load.

[0010] The front-end power conversion module adopts a three-level bidirectional Buck / Boost topology. The control module is configured to adjust the switching frequency of the front-end power conversion module through an adaptive frequency modulation strategy so that the inductor current crosses zero in each switching cycle, thereby achieving zero-voltage turn-on of all switching transistors.

[0011] The subsequent power conversion module adopts a half-bridge LLC topology and is configured to operate in DC transformer mode at the resonant point with a fixed frequency in an open loop, so that both the forward voltage gain and the reverse voltage gain are 1, the primary-side switch achieves zero-voltage turn-on, and the secondary-side switch achieves zero-current turn-off.

[0012] In conjunction with the first aspect, the present invention provides a first specific embodiment of the first aspect, specifically, the front-end bidirectional DC-DC converter includes:

[0013] Voltage divider unit, energy storage unit, and three-level bridge arm unit;

[0014] The voltage divider unit includes a first voltage divider capacitor and a second voltage divider capacitor. One end of the first voltage divider capacitor is connected to one end of the high voltage DC input module, and one end of the second voltage divider capacitor is connected to the other end of the high voltage DC input module. The other end of the first voltage divider capacitor is connected to the other end of the second voltage divider capacitor.

[0015] The three-level bridge arm unit includes a first switch, a second switch, a third switch, and a fourth switch connected in series. The first switch and the second switch are complementary in conducting to form an upper half-bridge, and the third switch and the fourth switch are complementary in conducting to form a lower half-bridge.

[0016] The energy storage unit includes a first filter inductor and a second filter inductor and a filter capacitor. One end of the first filter inductor is connected to the upper half-bridge, and the other end of the first filter inductor is connected to the other end of the intermediate bus capacitor as the low-voltage positive terminal of the front-end power conversion module. One end of the second filter inductor is connected to the lower half-bridge, and the other end of the second filter inductor is connected to the other end of the intermediate bus capacitor as the low-voltage negative terminal of the front-end power conversion module.

[0017] In conjunction with the first aspect, the present invention provides a second specific embodiment of the first aspect, specifically, the control module includes a front-end control unit, which is configured to perform near-critical conduction mode control:

[0018] During the dead time of the switching transistor in the three-level bridge arm unit, the current of the filter inductor is kept at a preset small reverse current or small forward current. The inductor current is used to charge and discharge the parasitic capacitance of the switching transistor to complete the energy transfer and achieve zero-voltage turn-on of all switching transistors.

[0019] In conjunction with the first aspect, the present invention provides a third specific embodiment of the first aspect, specifically, the front-end control unit includes a frequency modulation calculation subunit, which is configured as follows:

[0020] Obtain the high-voltage DC bus voltage, the intermediate bus voltage, the average current of the filter inductor, and the preset zero-crossing current threshold.

[0021] Based on the relationship between the voltage of the high-voltage DC bus and the voltage of the intermediate bus, the corresponding switching frequency calculation model is selected to obtain the current switching frequency.

[0022] In conjunction with the first aspect, the present invention provides a fourth specific embodiment of the first aspect, wherein the switching frequency of the adaptive frequency modulation is calculated in the following manner:

[0023] When the intermediate bus voltage At that time, the switching frequency is calculated using the first frequency modulation formula: ;

[0024] When the intermediate bus voltage At that time, the switching frequency is calculated using the second frequency modulation formula: ;

[0025] in, This represents the peak-to-peak value of the inductor current ripple. The inductance value of the filter inductor. The reverse small current threshold, This is the voltage of the high-voltage DC bus. This is the intermediate bus voltage. This represents the average current of the filter inductor.

[0026] In conjunction with the first aspect, the present invention provides a fifth specific implementation of the first aspect, wherein the subsequent power conversion module includes a primary-side inverter unit, a resonant unit, and a secondary-side rectifier unit;

[0027] The primary-side inverter unit includes a first primary-side switch and a second primary-side switch that are complementary in conduction, forming a square wave generating circuit.

[0028] The resonant unit includes a resonant capacitor, a resonant inductor, and an excitation inductor connected in parallel across the primary winding of the transformer.

[0029] The secondary-side rectifier unit includes a transformer with a center tap and a full-wave rectifier circuit. The full-wave rectifier circuit includes a first synchronous rectifier tube and a second synchronous rectifier tube, which are respectively connected to both sides of the center tap.

[0030] In conjunction with the first aspect, the present invention provides a fifth specific embodiment of the first aspect, specifically, the control module includes a downstream control unit, which is configured to:

[0031] An open-loop fixed-frequency PWM modulation strategy is adopted to lock the operating frequency of the subsequent power conversion module at the resonant frequency. ,in The inductance value of the resonant inductor. The capacitance value of the resonant capacitor;

[0032] The duty cycle of the complementary PWM pulses on the primary and secondary sides is fixed at 50%, and the dead time is set according to the excitation current and the junction capacitance parameters of the switching transistors to improve the converter gain during forward operation. Converter gain during reverse operation .

[0033] Secondly, the present invention also provides a control method for a fully soft-switching parallel-converted two-stage converter, comprising:

[0034] Sample the high-voltage DC bus voltage, intermediate bus voltage, filter inductor current, and battery load current;

[0035] The intermediate bus voltage is compared with the half bus voltage. If the intermediate bus voltage is less than the half high voltage DC bus voltage, the first frequency modulation formula is used to calculate the switching frequency. If the intermediate bus voltage is greater than the half high voltage DC bus voltage, the second frequency modulation formula is used to calculate the switching frequency, and the calculation result is limited to a preset frequency range.

[0036] Closed-loop PI regulation is performed based on the battery load current, and a drive signal is output to the front-end power conversion module in combination with duty cycle feedforward control. Voltage equalization of the voltage divider capacitors is achieved by finely adjusting the differential duty cycle of the upper and lower half-bridges.

[0037] The downstream power conversion module is controlled to operate in DC transformer mode at the resonant point with a fixed frequency and open loop, locking the duty cycle of the primary and secondary sides to 50%, thereby achieving zero voltage turn-on of the primary side and zero current turn-off of the secondary side.

[0038] Compared with the prior art, the beneficial effects of the present invention are:

[0039] This invention breaks away from the conventional topological inertia of "pre-stage LLC + post-stage Buck / Boost," creatively placing a three-level bidirectional Buck / Boost converter in the pre-stage and a half-bridge LLC resonant converter in the post-stage. The pre-stage uses an adaptive frequency modulation strategy to ensure the inductor current crosses zero in each switching cycle, achieving zero-voltage turn-on (ZVS) for all switches. The post-stage half-bridge LLC converter operates at its resonant frequency, functioning as a DC-DC transformer (DCX), with the primary-side switches achieving ZVS and the secondary-side switches achieving ZCS. Both pre- and post-stages operate in soft-switching mode, eliminating voltage-current crossover losses during switching and ensuring high efficiency across the entire system's load range. Compared to conventional architectures, this invention effectively solves the inherent drawback of high hard-switching losses in the post-stage Buck / Boost converter, with a particularly significant efficiency improvement under high-current charging and discharging conditions.

[0040] The downstream half-bridge LLC converter of this invention operates at a fixed resonant frequency, running in open-loop DC transformer mode with a constant voltage gain of 1. Compared to conventional LLC converters that require wide-range frequency conversion to regulate output voltage, this invention completely avoids the design challenges of magnetic components (transformer, resonant inductor) across a wide frequency band. Resonant parameter optimization only needs to be performed on a single frequency point, and the transformer can operate within its optimal magnetic flux density range, thus improving both system efficiency and reliability. Furthermore, open-loop operation eliminates the need for complex voltage / frequency closed-loop regulation, reducing the complexity of the control software and its dependence on sampling accuracy, which is beneficial for parallel current sharing control between multiple modules. Attached Figure Description

[0041] The specific embodiments of the present invention will be further described in detail below with reference to the accompanying drawings, wherein:

[0042] Figure 1 This is a topology diagram of a two-stage converter for the fractionation and capacity application of the present invention.

[0043] Figure 2 This is a schematic diagram of the operating waveform of the front-end power conversion module of the present invention.

[0044] Figure 3 This is the forward operating gain curve of the power conversion module of the present invention.

[0045] Figure 4 This is the reverse operating gain curve of the power conversion module of the present invention.

[0046] Figure 5 This is a control block diagram of the front-end power conversion module of the present invention.

[0047] Figure 6 This is a control block diagram of the power conversion module of the present invention. Detailed Implementation

[0048] The preferred embodiments of the present invention will be described below with reference to the accompanying drawings. It should be understood that the preferred embodiments described herein are for illustration and explanation only and are not intended to limit the present invention.

[0049] This invention provides a parallel two-stage converter for forming and capacitating lithium batteries with full soft switching. By swapping the front-stage LLC + rear-stage Buck / Boost converter in the traditional two-stage architecture with the front-stage Buck / Boost converter + rear-stage LLC converter, and combining it with front-stage adaptive frequency modulation technology and rear-stage fixed-frequency open-loop resonant point operation strategy, it achieves full-range soft switching, thereby significantly improving system efficiency and reducing switching losses. It is suitable for the high efficiency and high reliability requirements of charging and discharging power supplies in the lithium battery forming and capacitating process.

[0050] The purpose of this invention is to provide a high-efficiency, low-loss two-stage fractional-capacitance power supply architecture and its control method. By placing the Buck / Boost converter in the high-voltage, low-current front stage and utilizing adaptive frequency modulation technology to operate it in a near-critical conduction mode, zero-voltage turn-on of all switches is achieved. The LLC converter is placed in the low-voltage, high-current rear stage, operating it at a constant frequency in DC transformer mode at the resonant point, achieving zero-voltage turn-on on the primary side and zero-current turn-off on the secondary side. Through this architectural innovation and optimized control strategy, the technical problem of high switching losses and limited system efficiency improvement caused by hard switching of the Buck / Boost converter in the rear stage of traditional two-stage fractional-capacitance converters is solved, thereby reducing equipment operating energy consumption and achieving the goal of energy saving and carbon reduction.

[0051] like Figures 1-6 As shown, this invention relates to a fully soft-switching parallel-connected two-stage converter and its control method.

[0052] Example 1

[0053] like Figure 1 As shown, this invention presents a preferred topology for a parallel-connected, two-stage, fully soft-switching converter.

[0054] In this invention, the fully soft-switching two-stage converter is used to provide charging and discharging power for the lithium battery formation and capacity testing process. The system includes a high-voltage DC input module, a front-stage power conversion module, an intermediate bus capacitor, a rear-stage power conversion module, and a control module. The control module is connected to both the front-stage and rear-stage power conversion modules, and it is equipped with the control system described below for executing the control method described below.

[0055] like Figure 1As shown, the fully soft-switching two-stage converter includes a high-voltage DC input module, a front-stage power conversion module, an intermediate bus capacitor, a rear-stage power conversion module and a control module, as well as related sampling circuits and drive circuits.

[0056] The output terminal of the high voltage DC input module is connected to the input terminal of the front-stage power conversion module. The output terminal of the front-stage power conversion module is connected to the input terminal of the rear-stage power conversion module via the intermediate bus capacitor. The output terminal of the rear-stage power conversion module is used to connect to the battery load.

[0057] (1) Regarding the high-voltage DC input module:

[0058] The high-voltage DC input module converts mains power into a stable high-voltage DC bus voltage after three-phase AC / DC rectification, providing input power to the preceding power conversion module. In specific implementations, the high-voltage DC input module includes a three-phase rectifier bridge, a power factor correction circuit, and a filter energy storage capacitor. The input terminal of the high-voltage DC input module is connected to a three-phase 380V AC mains power supply, and the output terminal provides... The module is connected to the input of the preceding three-level bidirectional Buck / Boost converter via a 650V or 750V high-voltage DC bus.

[0059] In practical implementation, the high-voltage DC input module also features overvoltage protection, overcurrent protection, and undervoltage protection to ensure safe system shutdown under abnormal operating conditions. The high-voltage DC bus voltage output by the high-voltage DC input module is fed back to the control module via a voltage sampling circuit, serving as one of the input parameters for the upstream adaptive frequency modulation calculation.

[0060] (2) Regarding the front-end power conversion module

[0061] The front-end power conversion module is responsible for converting the high-voltage DC bus voltage to a lower intermediate bus voltage and for handling the constant voltage and constant current process control functions required for battery formation and capacity testing. The front-end power conversion module adopts a three-level bidirectional Buck / Boost topology. Its input is connected to the high-voltage DC input module, and its output is connected to the subsequent power conversion module via the intermediate bus capacitor. This three-level bidirectional Buck / Boost topology allows the front-end power conversion module to operate in either buck mode (charging condition), converting high-voltage DC to a low-voltage DC suitable for subsequent processing, or in boost mode (discharging condition), feeding the energy released by the battery back to the high-voltage DC bus, thus achieving bidirectional energy flow.

[0062] It's worth noting that the advantages of using a three-level topology are: the power switching transistors only need to withstand half the bus voltage, thus allowing the use of MOSFETs or IGBTs with a withstand voltage of 650V. Compared to two-level topologies that require devices with a withstand voltage of over 1200V, the three-level topology offers greater advantages in device selection and lower switching losses. More importantly, placing the Buck / Boost converter in the front stage, where its operating current is a relatively small current on the high-voltage side (relative to the large current in the subsequent stages), facilitates the creation of a zero-crossing point for the inductor current through frequency modulation, achieving zero-voltage turn-on.

[0063] In practical implementation, the front-end three-level Buck / Boost converter can be regarded as a series two-level Buck / Boost converter. Taking the upper half-bridge as an example, when , When a complementary pulse is emitted, its operating waveform is as follows: Figure 2 As shown (the ripple of the lower half-bridge is completely symmetrical with that of the upper half-bridge, so it will not be described again): the inductor current changes with the conduction and turn-off of the switching transistor. If the inductor current is always positive during charging, it operates in CCM mode: time, The inductor current was positive before activation, indicating that the inductor current had already changed from positive to positive during the dead time. Circulation to The body diode freewheeling, The two ends are only the forward voltage drop of the diode, at which point it is turned on. This can be considered as achieving zero-voltage turn-on (ZVS). The inductor current remains positive at any given time, indicating that... During the dead time before activation, the inductor current passes through The body diode freewheeling, Under half bus voltage, turn on at this time. This is a hard turn-on scenario. The discharge condition is similar and will not be elaborated further. The above analysis shows that only one switch can achieve ZVS in CCM mode. However, if the inductor current has both positive and negative values ​​in each switching cycle, the converter can achieve ZVS for all switches, reducing turn-on losses. To reduce the additional conduction losses caused by ripple while maintaining the advantages of ZVS, this invention employs an approximately critical turn-on mode. This involves actively adjusting the switching frequency to ensure the ripple current at the switching moment has a small reverse value, during which energy transfer of the switch's parasitic capacitance is completed, thus achieving ZVS.

[0064] By placing the Buck / Boost converter before the breakout converter, it operates under high voltage and low current conditions, which facilitates frequency modulation operation in near-critical conduction mode. In contrast, the traditional two-stage breakout converter Buck / Boost architecture is placed after the breakout converter, resulting in a larger DC current and making it difficult to construct a current ripple zero-crossing point through frequency modulation. This is the advantage of this invention.

[0065] like Figure 1 As shown, the front-end power conversion module includes a voltage divider unit, an energy storage unit, and a three-level bridge arm unit.

[0066] Specifically, the voltage divider unit includes a first voltage divider capacitor. Second voltage divider capacitor First voltage divider capacitor One end is connected to the positive output terminal of the high-voltage DC input module, and the second voltage divider capacitor. One end is connected to the negative output terminal of the high-voltage DC input module, and the first voltage divider capacitor. The other end is connected to the second voltage divider capacitor. The other end is connected. First voltage divider capacitor. Second voltage divider capacitor The connection point forms the neutral point, and the voltage at this neutral point is half the bus voltage. First voltage divider capacitor Second voltage divider capacitor The capacitors have the same value to evenly bear the high-voltage DC bus voltage. The function of the voltage-reducing unit is to divide the high-voltage DC bus voltage into two halves, so that each switching transistor only bears half of the bus voltage, thereby allowing the use of switching devices with a withstand voltage of 650V.

[0067] Specifically, the three-level bridge arm unit includes a first switching transistor connected in series. Second switching transistor Third switching transistor and the fourth switching transistor First switching transistor With the second switching transistor Complementary conduction forms the upper half-bridge, with the third switching transistor... With the fourth switching transistor Complementary conduction forms the lower half-bridge. The first switching transistor... The drain of the transistor is connected to the positive output terminal of the high-voltage DC input module, and the first switching transistor... The source is connected to the second switching transistor. The drain of the second switching transistor The source is connected to the third switch. The drain of the third switching transistor The source is connected to the fourth switch. The drain of the fourth switching transistor The source of the transistor is connected to the negative output terminal of the high-voltage DC input module. The second switching transistor... The source and the third switch The connection point of the drain terminal serves as the output midpoint of the three-level bridge arm unit, and this output midpoint is connected to the filter inductor of the energy storage unit.

[0068] It should be noted that the first switching transistor To the fourth switching transistor All are power switches with anti-parallel diodes, and each switch has parasitic capacitance, providing the necessary condition for zero-voltage turn-on. Under the complementary conduction control strategy, the upper half-bridge ( and ) and lower half bridge ( and They complement each other, with the driving pulses of the upper and lower half-bridges differing by 180° in phase.

[0069] Specifically, the energy storage unit includes a first filter inductor. Second filter inductor, first filter inductor One end is connected to the first switching transistor With the second switching transistor Connection point, first filter inductor The other end is connected to the intermediate bus capacitor. One end of the first inductor serves as the low-voltage positive terminal of the pre-conversion module; one end of the second filter inductor is connected to the third switch transistor. With the fourth switching transistor The connection point of the second filter inductor is connected to the intermediate bus capacitor. The other end serves as the low-voltage negative terminal of the pre-amplifier power conversion module.

[0070] It should be noted that the front-end three-level Buck / Boost converter can be considered as two two-level Buck / Boost converters connected in series, with the ripple of the upper and lower half-bridges being completely symmetrical. The principle is explained using the upper half-bridge as an example: when... , When a complementary pulse is emitted, the inductor current changes with the on and off states of the switching transistor. During charging, if the inductor current is always positive, the converter operates in continuous conduction mode (CCM). In CCM mode, only one switching transistor can achieve zero-voltage turn-on—specifically, when… The inductor current was positive before activation, indicating that the inductor current had already changed from positive to positive during the dead time. Circulation to The body diode freewheeling, The two ends are only the forward voltage drop of the diode, at which point it is turned on. It can achieve zero-voltage turn-on; and The inductor current was still positive before it was turned on. Under half bus voltage, turn on at this time. This is a hard turn-on. Similarly, under discharge conditions, only one switching transistor can achieve zero-voltage turn-on.

[0071] Therefore, the front-end power conversion module of this invention does not operate in the conventional DCM mode, but rather in an approximate critical conduction mode. By actively adjusting the switching frequency, the inductor current at each switching moment has a small reverse value, ensuring that the inductor current has both positive and negative values ​​within each switching cycle. This allows the inductor current to charge and discharge its parasitic capacitance at the turn-on moment of each switch, completing energy transfer and achieving zero-voltage turn-on for all switches. Simultaneously, the approximate critical conduction mode effectively reduces additional conduction losses caused by large ripple current, thus optimizing efficiency.

[0072] 3) Regarding the intermediate bus capacitor:

[0073] Intermediate bus capacitor It is connected between the output of the front-stage power conversion module and the input of the rear-stage power conversion module to couple the two power conversion modules, smooth the DC voltage output of the front-stage module, and provide a stable input power supply for the rear-stage LLC converter.

[0074] In specific implementation, the intermediate bus capacitor These are film capacitors or electrolytic capacitors, and their capacitance values ​​are selected based on the power matching and voltage ripple requirements of the preceding and following stages, typically ranging from hundreds to thousands of microfarads. Intermediate bus capacitors. The voltage across the two ends is This voltage is controlled by the preceding three-level Buck / Boost converter to adapt to the voltage levels required for battery charging and discharging processes (typically...). (Set within the range of 10V~15V).

[0075] It should be noted that the intermediate bus capacitor voltage It is one of the important input parameters for the front-end adaptive frequency modulation calculation, and the control module obtains it in real time through the voltage sampling circuit. The value.

[0076] (4) Regarding the subsequent power conversion module:

[0077] The subsequent power conversion module is responsible for converting the low-voltage DC power provided by the intermediate bus capacitor into a final voltage (below 5V) suitable for battery charging and discharging, and realizing bidirectional energy transfer between the battery and the load. In this invention, the subsequent power conversion module adopts a half-bridge LLC resonant converter topology and operates in open-loop mode at the resonant point with a fixed frequency, making both the forward voltage gain and the reverse voltage gain 1. The primary-side switch achieves zero-voltage turn-on, and the secondary-side switch achieves zero-current turn-off.

[0078] It should be noted that the advantages of using the LLC-DCX mode in the subsequent stage are as follows: when the LLC converter operates at the resonant point, the gain is constant at 1 and is not affected by the load, and it has good voltage regulation characteristics; the primary-side switching transistor can achieve zero-voltage turn-on in the full load range, and the secondary-side synchronous rectifier transistor can achieve zero-current turn-off, resulting in extremely low switching losses; the converter operates in an open-loop fixed-frequency state, eliminating the need for complex closed-loop control, and the system has high reliability.

[0079] like Figure 1 As shown, the subsequent power conversion module includes a primary-side inverter unit, a resonant unit, and a secondary-side rectifier unit.

[0080] Specifically, the primary-side inverter unit includes a first primary-side switch that is complementary in conduction. Second primary-side switch This forms a square wave generating circuit, using the intermediate bus capacitor. The DC voltage is inverted into a square wave voltage and applied to the resonant network. The first primary-side switch... The drain is connected to the intermediate bus capacitor. One end, the first primary-side switch transistor The source is connected to the second primary-side switch transistor. The drain of the second primary-side switching transistor The source is connected to the intermediate bus capacitor. The other end. The first primary-side switch transistor. With the second primary-side switch All are power switching transistors with anti-parallel diodes, which are complementary in conduction and have a fixed duty cycle of 50%.

[0081] Specifically, the resonant unit includes a resonant capacitor. Resonant inductor and the magnetizing inductance connected in parallel across the primary winding of the transformer Resonant capacitor Resonant inductor And excitation inductance Together, they form an LLC resonant network. One end of the resonant inductor is connected to the connection point between the source of the first primary-side switch and the drain of the second primary-side switch. The other end of the resonant inductor is connected to one end of the resonant capacitor. The other end of the resonant capacitor is connected to one end of the magnetizing inductor. The other end of the magnetizing inductor is connected to the source of the second primary-side switch. The magnetizing inductor is connected in parallel to both ends of the primary winding of the transformer.

[0082] Specifically, the secondary-side rectifier unit includes a transformer T with a center tap and a full-wave rectifier circuit, the full-wave rectifier circuit including a first synchronous rectifier diode. Second synchronous rectifier tube The primary winding of transformer T is connected to a resonant network, and the secondary winding has a center tap connected to one end of the battery load (e.g., the negative terminal of the battery). The two ends of the secondary winding are respectively connected to the first synchronous rectifier diode. The drain and the second synchronous rectifier tube The drain of the first synchronous rectifier diode. The source and the second synchronous rectifier diode The source terminals are connected together to the other end of the battery load (e.g., the positive terminal of the battery). The first synchronous rectifier diode... Second synchronous rectifier tube All are power switching transistors with anti-parallel diodes. The synchronous rectifier transistors are pulse-synchronized with the corresponding primary-side switching transistors to achieve efficient operation of synchronous rectification across the entire range.

[0083] In practical implementation, the transformer T is designed according to the power rating and voltage transformation ratio of the break-in converter, and the turns ratio n satisfies... The matching relationship between the LLC converter and the battery voltage ensures that the output voltage meets the battery charging and discharging requirements when the LLC converter is running at the resonant point.

[0084] (5) Regarding the control module:

[0085] The control module includes a front-end control unit and a back-end control unit.

[0086] In one implementation, the front-end control unit is configured to perform near-critical conduction mode control. Its core principle is: during the dead time of the switching transistors in the three-level bridge arm unit, the filter inductor... The current is kept at a preset small reverse current (charging condition) or small forward current (discharging condition). The inductor current is used to charge and discharge the parasitic capacitance of the switching transistor to complete the energy transfer and achieve zero-voltage turn-on of all switching transistors.

[0087] Specifically, such as Figure 2 As shown, taking the upper half-bridge as an example, when the inductor current alternates between positive and negative in each switching cycle, during the dead time before the switch is turned on, the inductor current will flow through the body diode of the switch that is about to be turned on, clamping the voltage across the switch to near zero. Turning on the switch at this point achieves ZVS (Zero-Voltage Switching). This invention actively adjusts the switching frequency to ensure that the ripple current at the switching moment has a small reverse value (in charging mode, the inductor current has a small negative current I-) or a small positive current (in discharging mode, the inductor current has a small positive current I+). This small current is used to complete the charging and discharging of the switch's parasitic capacitance during the dead time, thus achieving ZVS.

[0088] The front-end control unit includes a frequency modulation calculation subunit, which is configured to: acquire the high-voltage DC bus voltage. Intermediate bus voltage The filter inductor average current and preset zero-crossing current thresholds (negative small current I- or positive small current I+); based on the high-voltage DC bus voltage With intermediate bus voltage Based on the magnitude relationship, select the corresponding switching frequency calculation model to obtain the current switching frequency.

[0089] More specifically, the switching frequency of the adaptive frequency modulation is calculated as follows:

[0090] When the intermediate bus voltage At that time, the switching frequency is calculated using the first frequency modulation formula: ;

[0091] When the intermediate bus voltage At that time, the switching frequency is calculated using the second frequency modulation formula: ;

[0092] in, This represents the peak-to-peak value of the inductor current ripple. The inductance value of the filter inductor. The reverse small current threshold, This is the voltage of the high-voltage DC bus. This is the intermediate bus voltage. This represents the average current of the filter inductor.

[0093] It should be noted that the derivation of the above formula is based on the volt-second balance principle. Taking the charging condition as an example, when... At that time, within one switching cycle, the increase in inductor current when the upper transistor is turned on is equal to the decrease in inductor current when the upper transistor is turned off. Substituting this into the target ripple constraint (i.e., the inductor current at the switching moment is...), The required switching frequency can then be derived from this. Similarly, the following can be deduced: The formula for calculating the frequency at that time. If it is a discharge condition, the positive ripple current of the approximate critical conduction mode is: The above formula still applies.

[0094] In actual operation, upper and lower limits for the pre-amplifier frequency modulation are set: when the DC current decreases to a certain value, it operates at the upper limit of the switching frequency, and at this time, positive and negative values ​​of the inductor current naturally appear within the switching cycle; by reasonably designing the inductor value to ensure operation at the lower limit of the switching frequency under rated power conditions, the full range of ZVS can be met. In addition, the pre-amplifier also needs to include capacitor voltage balance control for CH1 and CH2, which is achieved by superimposing and finely adjusting the differential duty cycle of the upper and lower half-bridges to achieve voltage equalization.

[0095] The control module includes a downstream control unit, which is configured to:

[0096] An open-loop fixed-frequency PWM modulation strategy is adopted to lock the operating frequency of the subsequent power conversion module at the resonant frequency. ,in The inductance value of the resonant inductor. The capacitance value of the resonant capacitor;

[0097] The duty cycle of the complementary PWM pulses on the primary and secondary sides is fixed at 50%, and the dead time is set according to the excitation current and the junction capacitance parameters of the switching transistors to improve the converter gain during forward operation. Converter gain during reverse operation .

[0098] like Figure 3 As shown, during forward operation, the LLC gain curve shows a constant gain of 1 at the resonant point, unaffected by the load; as Figure 4 As shown, the LLC gain curve remains constant at the resonant point during reverse operation, with a gain of 1. This characteristic allows the LLC converter to function as an equivalent DC transformer (DCX) when operating at the resonant point, achieving isolated voltage transformation without introducing additional voltage gain changes.

[0099] (6) Regarding the working principle of the parallel-connected two-stage converter with fully soft switching of the present invention:

[0100] The fully soft-switching two-stage converter of this invention is essentially used for the formation, capacity testing, charging, and discharging of lithium batteries, addressing the need for high efficiency and low loss in the charging and discharging power supply during the formation and capacity testing process. Specifically, under the control of the control module, the system operation process of this invention is as follows:

[0101] Step 1, Pre-stage Voltage Conversion and Frequency Modulation Control: The 650V~750V high-voltage DC power output from the high-voltage DC input module is input to the pre-stage three-level Buck / Boost converter. The pre-stage control unit samples the high-voltage DC bus voltage. Intermediate bus voltage and filter inductor current ,according to and The switching frequency is calculated using the appropriate frequency modulation formula based on the magnitude relationship. Taking charging conditions as an example, when... At the same time, the switching frequency is calculated according to the first frequency modulation formula, so that the filter inductor current contains a small negative current I- in each switching cycle. This small negative current is used to charge and discharge the parasitic capacitance of the switching transistors during the dead time, thereby achieving ZVS for all switching transistors. Simultaneously, the front-end control unit fine-tunes the differential duty cycle of the upper and lower half-bridges to achieve voltage divider capacitance. and Voltage equalization.

[0102] Step 2, Stabilizing the intermediate bus voltage output: The front-end three-level Buck / Boost converter converts the high-voltage DC bus voltage into a stable low-voltage DC bus voltage (typically 10V~15V), which is then output via the intermediate bus capacitor. The filtered power supply powers the subsequent LLC converter.

[0103] Step 3, DCX mode operation of the subsequent stage: The subsequent LLC converter operates at the resonant point in DC transformer mode. The primary-side switching transistors SL1 and SL2 are turned on complementaryly with a 50% duty cycle, and the secondary-side synchronous rectifier transistor... and The drive pulse is synchronized with the primary-side pulse. The forward and reverse voltage gains of the LLC converter are both 1, achieving isolated conversion from the intermediate bus voltage to the battery-side voltage. The primary-side switch implements ZVS, and the secondary-side switch implements ZCS.

[0104] Step 4, Charge / Discharge Process Control: The front-end power conversion module uses the battery current output from the rear stage as a command value. The PI controller output, superimposed with a duty cycle feedforward component, serves as the basis for calculating the switching frequency. By adjusting the switching frequency, the output voltage and current are regulated to achieve the constant current and constant voltage charging / discharging processes required for battery formation and capacity control. During discharge, energy is transferred from the battery side to the intermediate bus capacitor via the rear-end LLC converter, and then back to the high-voltage DC bus via the front-end three-level Buck / Boost converter, achieving bidirectional energy flow. At this time, the front-end still achieves ZVS through adaptive frequency modulation.

[0105] Example 2

[0106] The method of this invention is based on the parallel-connected, two-stage, fully soft-switching, split-capacitance converter described in Embodiment 1. Specifically, the system is completely identical to that described in Embodiment 1. The system includes a front-stage three-level Buck / Boost converter, a rear-stage half-bridge LLC converter, and corresponding voltage and current sampling circuits; as shown... Figures 1 to 4 As shown, the control method includes:

[0107] S100: Samples the high-voltage DC bus voltage, intermediate bus voltage, filter inductor current, and battery load current.

[0108] In practice, the voltage of the high-voltage DC bus is collected by voltage Hall sensors and current Hall sensors, respectively. Intermediate bus voltage Filter inductor current and battery load current The sampled signal is processed by the conditioning circuit and then sent to the ADC interface of the control module for analog-to-digital conversion.

[0109] S200: Compare the intermediate bus voltage with the half bus voltage. If the intermediate bus voltage is less than the half high-voltage DC bus voltage, then use the first frequency modulation formula to calculate the switching frequency. If the intermediate bus voltage is greater than the half high-voltage DC bus voltage, then use the second frequency modulation formula to calculate the switching frequency and limit the calculation result to a preset frequency range.

[0110] S300: Based on the battery load current, a closed-loop PI regulation is performed, and a duty cycle feedforward control is combined to output a drive signal to the front-end power conversion module. The voltage equalization of the voltage divider capacitors is achieved by finely adjusting the differential duty cycle of the upper and lower half-bridges.

[0111] S400: Controls the subsequent power conversion module to operate in DC transformer mode at the resonant point with a fixed frequency open loop, locking the duty cycle of the primary and secondary sides to 50%, thereby achieving zero voltage turn-on of the primary side and zero current turn-off of the secondary side.

[0112] Specifically, this invention provides a detailed description of each step in the control method for a parallel-connected two-stage converter with fully soft switching.

[0113] S100: Samples the high-voltage DC bus voltage, intermediate bus voltage, filter inductor current, and battery load current.

[0114] In practice, the control module acquires the high-voltage DC bus voltage in real time through a voltage sampling circuit. and intermediate bus voltage The current of the filter inductor is collected in real time by a current sensor. and battery load current The sampling frequency should be much higher than the switching frequency of the power conversion module (typically more than 10 times the switching frequency) to ensure control accuracy and dynamic response performance. The sampled analog signal is converted into a digital quantity by an analog-to-digital converter for subsequent calculations by the control module. (High-voltage DC bus voltage) and intermediate bus voltage These are the direct input parameters for the pre-amplifier adaptive frequency modulation calculation, including the filter inductor current. Used to determine the zero-crossing state of inductor current and battery load current. Used for closed-loop regulation to precisely control the charging and discharging current.

[0115] S200: Compare the intermediate bus voltage with the half bus voltage. If the intermediate bus voltage is less than the half high-voltage DC bus voltage, then use the first frequency modulation formula to calculate the switching frequency. If the intermediate bus voltage is greater than the half high-voltage DC bus voltage, then use the second frequency modulation formula to calculate the switching frequency and limit the calculation result to a preset frequency range.

[0116] In practical implementation, based on the volt-second balance principle of the front-end three-level bidirectional Buck / Boost converter, when When the converter operates in the region where the buck ratio is less than 0.5, the switching frequency is calculated using the first frequency modulation formula; when When the converter operates in the region where the buck ratio is greater than 0.5, the switching frequency is calculated using the second frequency modulation formula.

[0117] Specifically, the switching frequency of the adaptive frequency modulation is calculated as follows:

[0118] When the intermediate bus voltage At that time, the switching frequency is calculated using the first frequency modulation formula: ;

[0119] When the intermediate bus voltage At that time, the switching frequency is calculated using the second frequency modulation formula: ;

[0120] in, The peak-to-peak value of the inductor current ripple is [value]. (Charging conditions) or (Discharge condition) The inductance value of the filter inductor. I+ is the reverse low-current threshold under charging conditions, and I+ is the forward low-current threshold under discharging conditions. This is the voltage of the high-voltage DC bus. This is the intermediate bus voltage. This represents the average current of the filter inductor. The values ​​of I- and I+ only need to be very small to meet the requirements for zero-voltage turn-on, and are usually set to 1% to 3% of the rated current.

[0121] It should be noted that the core idea of ​​the above frequency modulation formula is: within one switching cycle, by controlling the rise and fall of the inductor current, the inductor current at the switching moment is made exactly equal to the preset reverse small current I- (or forward small current I+). This current is then used to charge and discharge the parasitic capacitance of the switching transistor during the dead time, achieving zero-voltage turn-on. The formula for calculating the switching frequency is derived based on the volt-second balance principle, ensuring that the inductor current crosses zero in each switching cycle.

[0122] S300: Based on the battery load current, a closed-loop PI regulation is performed, and a duty cycle feedforward control is combined to output a drive signal to the front-end power conversion module. The voltage equalization of the voltage divider capacitors is achieved by finely adjusting the differential duty cycle of the upper and lower half-bridges.

[0123] In practical implementation, the control block diagram of the front-end three-level Buck / Boost converter is as follows: Figure 5As shown, the control module adopts a composite control strategy combining current closed-loop PI regulation and duty cycle feedforward: the battery load current is used as the feedback quantity, and the preset charging and discharging process current command value is used as the setpoint, and the error signal between the two is sent to the PI controller; at the same time, the feedforward component is calculated according to the steady-state duty cycle and superimposed on the PI output (after amplitude limiting processing), which is used as the basis for modulation of the switching frequency, thereby improving the dynamic response speed of the system while ensuring steady-state accuracy.

[0124] Furthermore, the control module generates drive pulses for the three-level bridge arm unit based on the calculated switching frequency. This is for units consisting of two half-bridges (the upper half-bridge contains...). , The lower half of the bridge includes , The four-switch topology, consisting of , has the following driving logic: the upper arm internally ( and Complementary conduction, inside the lower arm ( and The two half-bridges are designed for complementary conduction; the drive pulses of the upper and lower half-bridges are 180° out of phase to achieve interleaved control, effectively reducing input / output current ripple. In addition, reasonable dead time is set between the drive signals of each switch to avoid the risk of shoot-through short circuit.

[0125] In actual operation, the first voltage divider capacitor is affected by factors such as inconsistent device parameters and differences in drive delay. The voltage across the second voltage divider capacitor CH2 may become unbalanced, leading to uneven voltage stress on the switching transistors or a shift in the DC bus midpoint potential. To address this, the control module introduces a differential duty cycle voltage equalization control strategy: through real-time sampling... Voltage and Voltage Calculate the deviation ;Will The signal is fed into the equalizing PI controller, which outputs a small differential duty cycle correction. Subsequently, It is added to the duty cycle instructions of the upper and lower half-bridges in an equal and inverse manner (for example, increasing the duty cycle of the upper half-bridge). The duty cycle of the lower half-bridge will decrease accordingly. (or vice versa), thereby finely adjusting the distribution of transmission power between the upper and lower half-bridges, causing the voltages of the two voltage divider capacitors to tend towards dynamic balance, and ensuring balanced voltage stress on the switching transistors.

[0126] S400: Controls the subsequent power conversion module to operate in DC transformer mode at the resonant point with a fixed frequency open loop, locking the duty cycle of the primary and secondary sides to 50%, thereby achieving zero voltage turn-on of the primary side and zero current turn-off of the secondary side.

[0127] In practical implementation, the control block diagram of the subsequent half-bridge LLC converter is as follows: Figure 3 As shown. The control module adopts an open-loop fixed-frequency PWM modulation strategy, which does not rely on feedback adjustment, and locks the operating frequency of the subsequent power conversion module to the resonant frequency. The resonant frequency The expression is: ,in The inductance value of the resonant inductor. The capacitance value of the resonant capacitor is given.

[0128] Specifically, the control module will switch the first primary-side transistor... Second primary-side switch The PWM pulse duty cycle is fixed at 50%, and the two transistors conduct complementaryly, generating a symmetrical square wave voltage to excite the resonant network. Simultaneously, the first synchronous rectifier transistor on the secondary side... Second synchronous rectifier tube The drive pulse is synchronized with the pulse of the corresponding primary-side switching transistor, that is, when When conducting Conductive, When conducting It conducts, enabling synchronous rectification across the entire range.

[0129] In practical implementation, the control module sets an appropriate dead time based on the excitation current and the junction capacitance parameters of the switching transistor. The selection of the dead time must meet the following conditions: within the dead time, the parasitic capacitance of the switching transistor can be fully charged and discharged by the excitation current (or resonant current), causing the voltage across the switching transistor to drop to near zero, thus ensuring zero-voltage turn-on. A dead time that is too short will result in insufficient charging and discharging of the parasitic capacitance, failing to meet the zero-voltage turn-on condition; a dead time that is too long will increase the conduction time of the body diode, increasing conduction losses. Typically, the dead time is set between 50ns and 300ns, with the specific value optimized based on the switching transistor model and the magnitude of the excitation current.

[0130] Furthermore, when the subsequent LLC converter operates at the resonant point, both its forward and reverse voltage gains are 1. During forward operation (charging condition), the forward gain expression obtained based on the fundamental frequency analysis method is: ;

[0131] in, For normalized frequency, The switching frequency; Inductance ratio; The quality factor during forward operation; This is the equivalent load for positive AC communication; For transformer turns ratio; This is the forward load resistance. When... When the gain is 1, it is always 1 and is unaffected by the load, achieving the effect of a DC transformer, such as... Figure 3 As shown.

[0132] During reverse operation (discharge condition), the magnetizing inductance Always clamped by the secondary voltage, it does not participate in resonance, and the circuit degenerates into a series resonant converter (SRC). Its reverse gain expression is: ;in, The normalized frequency is defined the same as the positive frequency. This is the quality factor during reverse operation; This is the reverse AC equivalent load resistance. Similarly, when When = 1, the gain is always 1, such as Figure 4 As shown.

[0133] As can be seen from the forward and reverse gain curves, when the LLC-DCX operates at the resonant point, the gain is constant at 1 in both forward and reverse directions, unaffected by the load, achieving the effect of a DC transformer. Simultaneously, the primary-side switch can achieve zero-voltage turn-on across the entire load range, and the secondary-side synchronous rectifier achieves zero-current turn-off. Its soft-switching characteristics ensure that its efficiency is, in principle, superior to that of traditional Buck / Boost converters.

[0134] The above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention in any way. Therefore, any modifications, equivalent changes, and alterations made to the above embodiments based on the technical essence of the present invention without departing from the scope of the present invention shall still fall within the scope of the present invention.

Claims

1. A fully soft-switching parallel-conversion two-stage converter, characterized in that, include: High voltage DC input module, front-end power conversion module, intermediate bus capacitor, back-end power conversion module and control module; The output terminal of the high voltage DC input module is connected to the input terminal of the front-stage power conversion module. The output terminal of the front-stage power conversion module is connected to the input terminal of the rear-stage power conversion module via the intermediate bus capacitor. The output terminal of the rear-stage power conversion module is used to connect to the battery load. The front-end power conversion module adopts a three-level bidirectional Buck / Boost topology. The control module is configured to adjust the switching frequency of the front-end power conversion module through an adaptive frequency modulation strategy so that the inductor current crosses zero in each switching cycle, thereby achieving zero-voltage turn-on of all switching transistors. The subsequent power conversion module adopts a half-bridge LLC topology and is configured to operate in DC transformer mode at the resonant point with a fixed frequency in an open loop, so that both the forward voltage gain and the reverse voltage gain are 1, the primary-side switch achieves zero-voltage turn-on, and the secondary-side switch achieves zero-current turn-off.

2. The parallel-connected, two-stage, fully soft-switching converter according to claim 1, characterized in that, The front-end bidirectional DC-DC converter includes: Voltage divider unit, energy storage unit, and three-level bridge arm unit; The voltage divider unit includes a first voltage divider capacitor and a second voltage divider capacitor. One end of the first voltage divider capacitor is connected to one end of the high voltage DC input module, and one end of the second voltage divider capacitor is connected to the other end of the high voltage DC input module. The other end of the first voltage divider capacitor is connected to the other end of the second voltage divider capacitor. The three-level bridge arm unit includes a first switch, a second switch, a third switch, and a fourth switch connected in series. The first switch and the second switch are complementary in conducting to form an upper half-bridge, and the third switch and the fourth switch are complementary in conducting to form a lower half-bridge. The energy storage unit includes a first filter inductor and a second filter inductor and a filter capacitor. One end of the first filter inductor is connected to the upper half-bridge, and the other end of the first filter inductor is connected to the other end of the intermediate bus capacitor as the low-voltage positive terminal of the front-end power conversion module. One end of the second filter inductor is connected to the lower half-bridge, and the other end of the second filter inductor is connected to the other end of the intermediate bus capacitor as the low-voltage negative terminal of the front-end power conversion module.

3. A fully soft-switching parallel-connected two-stage converter with capacity division according to claim 2, characterized in that, The control module includes a front-end control unit configured to perform near-critical conduction mode control: During the dead time of the switching transistor in the three-level bridge arm unit, the current of the filter inductor is kept at a preset small reverse current or small forward current. The inductor current is used to charge and discharge the parasitic capacitance of the switching transistor to complete the energy transfer and achieve zero-voltage turn-on of all switching transistors.

4. A fully soft-switching parallel-connected two-stage converter with capacity division according to claim 3, characterized in that, The front-end control unit includes a frequency modulation calculation subunit, which is configured to: Obtain the high-voltage DC bus voltage, the intermediate bus voltage, the average current of the filter inductor, and the preset zero-crossing current threshold. Based on the relationship between the voltage of the high-voltage DC bus and the voltage of the intermediate bus, the corresponding switching frequency calculation model is selected to obtain the current switching frequency.

5. A fully soft-switching parallel-connected two-stage converter according to claim 3, characterized in that, The switching frequency of the adaptive frequency modulation is calculated as follows: When the intermediate bus voltage At that time, the switching frequency is calculated using the first frequency modulation formula: ; When the intermediate bus voltage At that time, the switching frequency is calculated using the second frequency modulation formula: ; in, This represents the peak-to-peak value of the inductor current ripple. The inductance value of the filter inductor. The reverse small current threshold, This is the voltage of the high-voltage DC bus. This is the intermediate bus voltage. This represents the average current of the filter inductor.

6. A fully soft-switching parallel-connected two-stage converter according to claim 1, characterized in that, The power conversion module includes a primary-side inverter unit, a resonant unit, and a secondary-side rectifier unit. The primary-side inverter unit includes a first primary-side switch and a second primary-side switch that are complementary in conduction, forming a square wave generating circuit. The resonant unit includes a resonant capacitor, a resonant inductor, and an excitation inductor connected in parallel across the primary winding of the transformer. The secondary-side rectifier unit includes a transformer with a center tap and a full-wave rectifier circuit. The full-wave rectifier circuit includes a first synchronous rectifier tube and a second synchronous rectifier tube, which are respectively connected to both sides of the center tap.

7. A fully soft-switching parallel-connected two-stage converter according to claim 1, characterized in that, The control module includes a downstream control unit, which is configured to: An open-loop fixed-frequency PWM modulation strategy is adopted to lock the operating frequency of the subsequent power conversion module at the resonant frequency. ,in The inductance value of the resonant inductor. The capacitance value of the resonant capacitor; The duty cycle of the complementary PWM pulses on the primary and secondary sides is fixed at 50%, and the dead time is set according to the excitation current and the junction capacitance parameters of the switching transistors to improve the converter gain during forward operation. Converter gain during reverse operation .

8. A control method for a fully soft-switching parallel-converted two-stage converter, applied to the fully soft-switching parallel-converted two-stage converter according to any one of claims 1 to 7, characterized in that, include: Sample the high-voltage DC bus voltage, intermediate bus voltage, filter inductor current, and battery load current; The intermediate bus voltage is compared with the half bus voltage. If the intermediate bus voltage is less than the half high voltage DC bus voltage, the first frequency modulation formula is used to calculate the switching frequency. If the intermediate bus voltage is greater than the half high voltage DC bus voltage, the second frequency modulation formula is used to calculate the switching frequency, and the calculation result is limited to a preset frequency range. Closed-loop PI regulation is performed based on the battery load current, and a drive signal is output to the front-end power conversion module in combination with duty cycle feedforward control. Voltage equalization of the voltage divider capacitors is achieved by finely adjusting the differential duty cycle of the upper and lower half-bridges. The downstream power conversion module is controlled to operate in DC transformer mode at the resonant point with a fixed frequency and open loop, locking the duty cycle of the primary and secondary sides to 50%, thereby achieving zero voltage turn-on of the primary side and zero current turn-off of the secondary side.

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