Intelligent controller for fuel pump and control method thereof

Abstract

The application relates to the technical field of fuel pumps, and provides a fuel pump intelligent controller and a control method thereof. The fuel pump control method comprises the following steps: acquiring an actual rotating speed and a bus current of a fuel pump motor at a current moment; in the case that a change rate of the actual rotating speed is greater than a first threshold value and the bus current is less than a steady-state load current base value corresponding to the current rotating speed, it is determined that there is a rotating speed overshoot risk, and a first negative correction amount is generated; the first negative correction amount is superimposed on a target current instruction output by a field-oriented control (FOC) speed loop to obtain a first target current instruction; and the output torque of the fuel pump motor is controlled based on the first target current instruction. The fuel pump control method provided by the application can improve the technical problem that a vehicle fuel pump motor is prone to rotating speed overshoot and overcharged water hammer effect under an urgent acceleration pressure building transient state in the related art.

Description

Technical Field

[0001] This application relates to the field of fuel pump technology, and in particular to a fuel pump intelligent controller and its control method. Background Technology

[0002] In the control scheme of brushless motor fuel pumps, the field-oriented control (FOC) dual closed-loop architecture is currently the most mainstream technical solution. This architecture takes the motor speed as the only closed-loop control target and contains two-stage regulators: a speed loop and a current loop. The speed loop takes the deviation between the target speed and the actual speed as input and outputs the target current. The current loop takes the deviation between the target current and the actual bus current as input and adjusts the PWM duty cycle to drive the three-phase inverter bridge to control the motor. Under stable operating conditions, the fluid load of the fuel pump is relatively stable, and the traditional FOC dual closed-loop can maintain the speed stability well.

[0003] However, during rapid acceleration and pressure build-up transients, traditional FOC control schemes immediately output a large current when the FOC speed loop receives a step target speed command. The motor can quickly increase its speed to the target value, but for fuel to enter the fuel rail from the pump outlet and establish stable pressure, it needs to overcome pipeline flow resistance, fuel rail volume expansion, and the dynamic response delay of the fuel pressure regulating valve. This means that in the initial stage of rapid acceleration, the motor speed rises sharply, but the fuel rail pressure has not yet been established, and the pump outlet is in a low-resistance state. This is because traditional FOC controllers only consider the target speed as the primary factor. The control target is to continuously drive the motor with a large current. When the motor speed reaches the target value, due to the extremely light load, the system may experience severe rotor overshoot. Subsequently, the high-speed rotating impeller encounters the suddenly established fluid resistance, converting kinetic energy into pressure energy, generating severe oil pressure overcharging and oscillation, i.e., water hammer effect. The pressure impact peak caused by this water hammer effect can be several times the normal oil pressure, which may generate impact loads on fuel line seals and oil pressure regulating valves that exceed their design tolerance, accelerating component fatigue aging. Under long-term repeated impacts, there is a risk of degradation of the oil return function and seal failure.

[0004] In the existing technology, one solution is to use fixed high-frequency PWM control to quickly adjust the speed, but under low load conditions, the excessively high PWM frequency will cause the motor to generate high-frequency electromagnetic noise; another solution is to directly limit the maximum acceleration or maximum current of the motor to avoid overshoot, but this will significantly reduce the fuel supply response speed during rapid acceleration and cannot meet the engine's need for rapid pressure build-up. Summary of the Invention

[0005] This application provides a fuel pump intelligent controller, which can improve the technical problems in related technologies where the fuel pump motor of an vehicle is prone to speed overshoot and oil pressure overfilling water hammer effect under transient conditions of rapid acceleration and pressure build-up.

[0006] In a first aspect, embodiments of this application provide a fuel pump intelligent controller, including a control circuit. The control circuit is connected to an engine ECU, a Hall position sensor installed inside the fuel pump motor, and a Hall current sensor connected in series with a DC bus. The control circuit receives a target speed or target pressure command output by the engine ECU, a rotor position signal output by the Hall position sensor, and a bus current sampling analog voltage output by the Hall current sensor. The fuel pump intelligent controller further includes: A gate drive circuit, connected to the control circuit, is used to receive six PWM signals output by the control circuit, amplify them according to a specified duty cycle and timing, and output corresponding gate drive signals; and The three-phase inverter power circuit is connected to the vehicle 12-volt battery and the gate drive circuit. It is used to receive the gate drive signal output by the gate drive circuit, and to invert the DC voltage input from the vehicle 12-volt battery through the DC bus into a three-phase AC voltage with adjustable amplitude and frequency, and then output it to the three-phase winding of the fuel pump motor to drive the motor rotor to rotate and drive the impeller to pump fuel.

[0007] The technical solutions described in this application embodiment have at least the following technical effects: The fuel pump intelligent controller provided in this application receives target speed or target pressure commands from the engine ECU, rotor position signals from the Hall position sensor, and bus current sampling analog voltages from the Hall current sensor via a control circuit. After processing, it outputs six PWM signals to the gate drive circuit. The gate drive circuit amplifies and times the six PWM signals and then outputs corresponding gate drive signals to the three-phase inverter power circuit. The three-phase inverter power circuit uses the DC voltage input from the vehicle's 12-volt battery via the DC bus as its energy source. Under the control of the gate drive signals, it inverts the DC voltage into a three-phase AC voltage with adjustable amplitude and frequency, which is then supplied to the three-phase windings of the fuel pump motor. This drives the motor rotor to rotate and drives the impeller to pump fuel. All parts cooperate to achieve intelligent operation control of the fuel pump.

[0008] Secondly, embodiments of this application provide a fuel pump control method, applied to the fuel pump intelligent controller described in the first aspect, the method comprising: Obtain the actual speed and bus current of the fuel pump motor at the current moment; wherein, the bus current is used to reflect the load state of the fuel pump motor, and the degree of deviation of the bus current from the steady-state load current base value reflects the deviation of the output torque of the fuel pump motor; If the rate of change of the actual rotational speed is greater than a first threshold and the bus current is less than the base value of the steady-state load current corresponding to the current rotational speed, it is determined that there is a risk of speed overshoot and a first negative correction amount is generated; wherein, the base value of the steady-state load current is the reference load current value required for the fuel pump motor to operate in steady state at the current rotational speed, and the first negative correction amount is a correction value used to reduce the target current command. The first negative correction amount is superimposed on the target current command output by the magnetic field orientation control (FOC) speed loop to obtain the first target current command; Based on the first target current command, the output torque of the fuel pump motor is controlled.

[0009] The technical solutions described in this application embodiment have at least the following technical effects: By acquiring the actual speed and bus current of the fuel pump motor to reflect the load state, and identifying the fluid pressure build-up hysteresis characteristics when the rate of change of the actual speed is greater than a first threshold and the bus current is less than the steady-state load current base value corresponding to the current speed, a first negative correction amount is generated to reduce the target current command. The first target current command is obtained by superimposing the first negative correction amount onto the target current command output by the field-oriented control (FOC) speed loop, and the output torque of the fuel pump motor is controlled based on the first target current command to reduce the drive torque in advance before speed overshoot and oil pressure overcharge occur. This reduces the speed overshoot amplitude caused by the time constant mismatch between the fast electrical response of the motor and the slow fluid pressure build-up response under the transient condition of rapid acceleration and pressure build-up, and suppresses the pressure impact peak of the water hammer effect during oil pressure overcharge. Attached Figure Description

[0010] To more clearly illustrate the technical solutions in the embodiments of this application, the drawings used in the description of the embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are only some embodiments of this application. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0011] Figure 1 A schematic diagram of the gate drive circuit provided in an embodiment of this application; Figure 2 A schematic diagram of a three-phase inverter power circuit provided in an embodiment of this application; Figure 3 This is a schematic flowchart of the fuel pump control method provided in an embodiment of this application.

[0012] The following are the labeling elements in the figure: 100. Intelligent controller for fuel pump; 10. Gate drive circuit; 20. Three-phase inverter power circuit. Detailed Implementation

[0013] To make the technical problems, technical solutions, and beneficial effects to be solved by this application clearer, the following detailed description is provided in conjunction with the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and are not intended to limit the scope of this application.

[0014] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs; the terminology used herein is for the purpose of describing particular embodiments only and is not intended to limit the application. The terms “comprising” and “having”, and any variations thereof, in the specification, claims, and foregoing description of the drawings are intended to cover non-exclusive inclusion.

[0015] In the control scheme of the brushless motor fuel pump, the field-oriented control (FOC) dual closed-loop architecture takes the motor speed as the closed-loop control target, and can maintain the speed stability well under stable operating conditions.

[0016] However, during rapid acceleration and pressure build-up, while the motor speed can reach the target value within hundreds of milliseconds, the establishment of fuel rail pressure requires overcoming pipeline flow resistance, fuel compressibility, and fuel rail filling delay. This mismatch in response speeds causes the FOC controller to overshoot under light load conditions, leading to fuel pressure overcharging and water hammer effects, which can cause impact damage to fuel line seals and fuel pressure regulating valves. Traditional fixed high-frequency PWM schemes or acceleration limiting schemes struggle to balance fuel supply response speed with the stability of both speed and fuel pressure during rapid acceleration and pressure build-up.

[0017] Based on this, in order to improve the technical problems of vehicle fuel pump motors being prone to speed overshoot and oil pressure overfilling water hammer effect under the transient conditions of rapid acceleration and pressure build-up in related technologies, the embodiments of this application provide the following solutions.

[0018] Please see Figure 1 and Figure 2 This application provides a fuel pump intelligent controller 100. The fuel pump intelligent controller 100 includes a control circuit, which is connected to the engine ECU, a Hall position sensor installed inside the fuel pump motor, and a Hall current sensor connected in series with the DC bus. The control circuit is used to receive the target speed or target pressure command output by the engine ECU, the rotor position signal output by the Hall position sensor, and the bus current sampling analog voltage output by the Hall current sensor. The fuel pump intelligent controller 100 also includes a gate drive circuit 10 and a three-phase inverter power circuit 20.

[0019] The gate drive circuit 10 is connected to the control circuit and is used to receive the six PWM signals output by the control circuit, amplify them according to the specified duty cycle and timing, and output the corresponding gate drive signals.

[0020] The three-phase inverter power circuit 20 is connected to the vehicle 12-volt battery and the gate drive circuit 10. It is used to receive the gate drive signal output by the gate drive circuit 10, and to invert the DC voltage input from the vehicle 12-volt battery through the DC bus into a three-phase AC voltage with adjustable amplitude and frequency, and then output it to the three-phase winding of the fuel pump motor to drive the motor rotor to rotate and drive the impeller to pump fuel.

[0021] It is understood that the control circuit is located inside the fuel pump intelligent controller 100 and is connected to the engine ECU, Hall position sensor, Hall current sensor and gate drive circuit 10 respectively. It is responsible for signal reception, logic operation and control command output. For example, the control circuit can be a vehicle-grade single-chip SoC such as TLE987x series or S12ZVM series.

[0022] The engine ECU is a high-level control component of the vehicle's power system. It is connected to the control circuit of the fuel pump intelligent controller 100 and is used to issue target speed or target pressure commands to the fuel pump. For example, the engine ECU can be the vehicle's original engine electronic control unit, the vehicle power domain controller, etc.

[0023] The Hall position sensor is installed inside the fuel pump motor and connected to the control circuit. It is used to detect the real-time position of the motor rotor and output the corresponding rotor position signal. Switch-type Hall sensors, linear Hall sensors, etc. can be used.

[0024] The Hall current sensor is connected in series in the DC bus line and connected to the control circuit. It is used to collect the bus current and convert it into a corresponding bus current sampling analog voltage output. Open-loop Hall current sensors, closed-loop Hall current sensors, etc. can be used.

[0025] The vehicle's 12-volt battery is a low-voltage power supply for the vehicle. It is connected to the three-phase inverter power circuit 20 via a DC bus to provide DC power for the operation of the fuel pump. For example, the vehicle's 12-volt battery can be a lead-acid battery, a lithium iron phosphate battery, etc.

[0026] As the power actuator of the fuel pump, the fuel pump motor has its three-phase windings connected to the output terminal of the three-phase inverter power circuit 20. It has a Hall position sensor installed inside and can drive the rotor and impeller to pump fuel under the drive of three-phase AC power. For example, the fuel pump motor can be a permanent magnet synchronous motor, a brushless DC motor, etc.

[0027] As can be seen from the above, the fuel pump intelligent controller 100 provided in this application embodiment receives the target speed or target pressure command output by the engine ECU, the rotor position signal output by the Hall position sensor, and the bus current sampling analog voltage output by the Hall current sensor through the control circuit. After processing, it outputs six PWM signals to the gate drive circuit 10. The gate drive circuit 10 amplifies and matches the timing of the six PWM signals and outputs the corresponding gate drive signal to the three-phase inverter power circuit 20. The three-phase inverter power circuit 20 uses the DC voltage input from the vehicle's 12-volt battery through the DC bus as its energy source. Under the control of the gate drive signal, it inverts the DC voltage into a three-phase AC voltage with adjustable amplitude and frequency, which is then delivered to the three-phase windings of the fuel pump motor to drive the motor rotor to rotate and drive the impeller to pump fuel. All parts cooperate with each other to realize the intelligent operation control of the fuel pump.

[0028] In some embodiments, please refer to Figure 1 The gate drive circuit 10 includes a gate drive chip U1, a power supply filter module, a bootstrap boost module, and an overcurrent feedback module.

[0029] Pins 2, 3, 4, 5, 6, and 7 of the gate driver chip U1 are connected to the control circuit to receive six PWM signals. Pins 14, 15, 16, 19, 23, and 27 of the gate driver chip U1 are connected to the three-phase inverter power circuit 20 to output gate drive signals. Pins 13, 18, 22, and 26 of the gate driver chip U1 are reference ground pins and are connected to the corresponding potential nodes of the three-phase inverter power circuit 20.

[0030] The power filter module's input terminal is connected to a 15V operating power supply, and its output terminal is connected to pin 1 of the gate driver chip U1, providing a stable 15V operating power supply for the gate driver chip U1 and its surrounding circuits.

[0031] The bootstrap boost module is connected to a 15V operating power supply at its input and to a gate driver chip U1 at its output, which provides floating power for the U / V / W three-phase upper bridge arm drives respectively.

[0032] The input terminal of the overcurrent feedback module is connected to the bus current sampling voltage input point, and the output terminal is connected to pin 9 of the gate driver chip U1.

[0033] It can be understood that the gate drive circuit 10 is located between the control circuit and the three-phase inverter power circuit 20. It is the intermediate link connecting the control signal and the power circuit in the fuel pump intelligent controller 100, and is used to complete the power amplification and timing adaptation of the six PWM signals.

[0034] The gate driver chip U1 has multiple pins that are connected to the control circuit, the three-phase inverter power circuit 20, and various auxiliary function modules, respectively, and are responsible for receiving control signals, processing internal logic, and outputting drive levels.

[0035] Power supply filtering modules are used to filter out ripple and interference on power lines and improve power supply stability. They can be implemented using a filter structure that combines electrolytic capacitors and ceramic capacitors, or an RC filter network.

[0036] The bootstrap boost module is used to provide floating power for the U / V / W three-phase upper bridge arm drive. It can be in the form of a classic bootstrap structure with diodes and energy storage capacitors, or a charge pump boost circuit.

[0037] The overcurrent feedback module is used to transmit the bus current sampling signal to the driver chip to support the overcurrent protection function. It can be in the form of a sampling structure with resistor voltage divider and filter capacitor, differential signal conditioning circuit, etc.

[0038] The 15V operating power supply is a DC power source that provides operating power to the gate drive circuit 10. It can be a regulated power supply obtained by DC-DC conversion from an on-board power supply, or a DC power supply output from a linear voltage regulator circuit, etc.

[0039] With this configuration, the power supply filtering module filters the input 15V operating power and then sends it to the power pin of the gate driver chip U1, providing a stable operating power supply for the gate driver chip U1 and its surrounding circuits. The bootstrap module takes the 15V operating power supply as input and provides a floating power supply for the three-phase upper bridge arm drive section of the gate driver chip U1, supporting the normal drive of the upper bridge arm power devices. The gate driver chip U1 receives six PWM signals output from the control circuit through corresponding pins, processes them internally, and outputs the gate drive signal to the three-phase inverter power circuit 20. At the same time, it establishes a potential reference with the corresponding potential node of the three-phase inverter power circuit 20 through the reference ground pin. The overcurrent feedback module sends the sampling signal of the bus current sampling voltage input point to the detection pin of the gate driver chip U1, providing a signal basis for the overcurrent protection action. All modules cooperate with each other to complete the conditioning, output, and overcurrent protection functions of the gate drive signal.

[0040] In some embodiments, please refer to Figure 1 The power filtering module includes capacitors C4, C5, and C6.

[0041] The 15V operating power supply is connected to the anode of capacitor C4, the anode of capacitor C5, and one end of capacitor C6. The cathodes of capacitor C4 and C5 are connected to power ground PGND, and the other end of capacitor C6 is connected to power ground PGND, which is the same ground as pin 13 of the gate driver chip U1.

[0042] It is understandable that capacitors C4 and C5 are connected in parallel to form a high-frequency decoupling unit, which is used to filter out high-frequency interference components in the power supply line and reduce the equivalent impedance in the high-frequency band. Multilayer ceramic capacitors, monolithic ceramic capacitors, and other types can be used.

[0043] Capacitor C6 is used to suppress low-frequency ripple in the power supply and provide instantaneous current storage. It can be an aluminum electrolytic capacitor, a solid electrolytic capacitor, or other types.

[0044] With this configuration, the three capacitors are connected in parallel to form a high- and low-frequency filter network. After decoupling and suppressing the ripple of the input 15V operating power supply, the network provides power to the gate driver chip U1. Pin 13 of the gate driver chip U1 shares a common ground with the power ground PGND, forming a complete power supply reference circuit, which provides a relatively stable operating power supply for the gate driver chip and surrounding circuits.

[0045] In some embodiments, please refer to Figure 1 The gate drive circuit 10 also includes capacitor C7, resistor R3, and resistor R4.

[0046] One end of resistor R3 is connected to a 15V operating power supply, and the other end is connected to one end of capacitor C7 and pin 11 of gate driver chip U1.

[0047] The other end of capacitor C7 is connected to power ground PGND and pin 12 of gate driver chip U1.

[0048] One end of resistor R4 is connected to a 5VA auxiliary power supply, and the other end is connected to pin 10 of the gate driver chip U1.

[0049] It is understandable that resistor R3 and capacitor C7 together form an RC timing network, which can be used to set the reset delay time under fault conditions. Combinations such as metal film resistors with multilayer ceramic capacitors and carbon film resistors with monolithic capacitors can be used.

[0050] Resistor R4 is used to limit the input current of the enable pin and provide a stable level bias. It can be a surface-mount precision resistor, a through-hole carbon film resistor, or other types.

[0051] With this configuration, resistor R3, capacitor C7, and resistor R4 work together with gate driver chip U1 to complete the reset delay configuration and enable level configuration, which helps to improve the stability of the drive circuit operation.

[0052] In some embodiments, please refer to Figure 1 The gate drive circuit 10 also includes a resistor R2 and a light-emitting diode D4.

[0053] One end of resistor R2 is connected to a 5VA auxiliary power supply, and the other end is connected to the anode of LED D4. The cathode of LED D4 is connected to pin 8 of gate driver chip U1.

[0054] It is understandable that resistor R2 and LED D4 are connected in series to form a status indicator branch, located at the fault output port of gate driver chip U1. This branch is used to intuitively reflect the operating status of gate driver chip U1 through its on / off state. It can be formed by combining surface-mount current limiting resistor with through-hole LED, surface-mount resistor with surface-mount LED, etc.

[0055] With this configuration, when pin 8 of the gate driver chip U1 outputs a low-level fault signal, the indicator branch is turned on and the LED D4 is lit, indicating to the outside that the circuit has entered a fault protection state. Resistor R2 also serves to limit the loop current, reducing the risk of LED overcurrent damage and assisting the gate driver circuit 10 in realizing visual feedback of the fault state.

[0056] In some embodiments, please refer to Figure 1 The bootstrap boost module includes resistor R1, diode D1, diode D2, diode D3, capacitor C1, capacitor C2, and capacitor C3.

[0057] One end of resistor R1 is connected to the 15V operating power supply and pin 1 of the gate driver chip U1, and the other end is connected to the anodes of diodes D1, D2, and D3.

[0058] The cathode of diode D3 is connected to pin 28 of gate driver chip U1 and the anode of capacitor C1. The cathode of capacitor C1 is connected to pin 26 of gate driver chip U1 and the three-phase inverter power circuit 20.

[0059] The cathode of diode D2 is connected to pin 24 of gate driver chip U1 and the anode of capacitor C2. The cathode of capacitor C2 is connected to pin 22 of gate driver chip U1 and the three-phase inverter power circuit 20.

[0060] The cathode of diode D1 is connected to pin 20 of gate driver chip U1 and the anode of capacitor C3. The cathode of capacitor C3 is connected to pin 18 of gate driver chip U1 and the three-phase inverter power circuit 20.

[0061] It is understandable that resistor R1, as a common current limiting device, is connected in series between the 15V working power supply and the anodes of the three bootstrap diodes to limit the charging current of the bootstrap capacitor and reduce current surges. It can be a surface-mount metal film resistor, a through-hole carbon film resistor, or other types.

[0062] Diode D3 and capacitor C1 together form the U-phase bootstrap energy storage branch, which is used to provide energy storage power for the floating drive of the upper arm of the U-phase bridge. Diode D3 can be a fast recovery diode, Schottky diode, or other types, and capacitor C1 can be a high-voltage ceramic capacitor, film capacitor, or other types.

[0063] Diode D2 and capacitor C2 together form the V-phase bootstrap energy storage branch, which is used to provide energy storage power for the floating drive of the V-phase upper bridge arm. Diode D2 can be a fast recovery diode, Schottky diode, or other types, and capacitor C2 can be a high-voltage ceramic capacitor, film capacitor, or other types.

[0064] Diode D1 and capacitor C3 together form the W-phase bootstrap energy storage branch, which is used to provide energy storage power for the floating drive of the upper arm of the W phase. Diode D1 can be a fast recovery diode, Schottky diode, or other types, and capacitor C3 can be a high-voltage ceramic capacitor, film capacitor, or other types.

[0065] With this configuration, diode D3 and capacitor C1, diode D2 and capacitor C2, and diode D1 and capacitor C3 form three independent bootstrap energy storage branches. Each branch is connected to the floating power supply pin and reference ground pin of the gate driver chip U1, and is connected to the corresponding node of the three-phase inverter power circuit 20. When the lower bridge arm is turned on, the bootstrap capacitor is charged, and when the upper bridge arm is turned on, the floating power supply is maintained by the energy storage of the capacitor.

[0066] In some embodiments, please refer to Figure 1 The overcurrent feedback module includes resistor R5, resistor R6, and capacitor C8.

[0067] One end of resistor R5 is connected to the bus current sampling voltage input point, and the other end is connected to one end of resistor R6. The other end of resistor R6 is connected to power ground PGND.

[0068] Pin 9 of the gate driver chip U1 is connected to the common node between resistors R5 and R6.

[0069] One end of capacitor C8 is connected to the common node between resistor R6 and power ground PGND, and the other end is connected to the common node between resistor R5 and resistor R6.

[0070] It is understandable that resistors R5 and R6 are connected in series to form a resistor voltage divider network, which is used to adjust the input sampling voltage proportionally to the voltage range of the detection port of the adapter chip. Surface mount metal film resistors, through-hole carbon film resistors, and other types can be used.

[0071] Capacitor C8 is connected in parallel across resistor R6, forming an RC low-pass filter network together with the voltage divider resistor. This network is used to filter out high-frequency switching interference in the sampling signal and reduce the probability of false triggering of protection. Multilayer ceramic capacitors, monolithic ceramic capacitors, and other types can be used.

[0072] With this configuration, resistors R5 and R6, along with capacitor C8, work together to condition and transmit the sampling signal, providing a suitable input signal for the overcurrent protection function of the gate driver chip.

[0073] In some embodiments, please refer to Figure 2 The three-phase inverter power circuit 20 includes an input protection module, a three-phase half-bridge inverter drive module, a reference ground interface module, and a detection and conditioning module.

[0074] The input protection module is connected to an external DC power supply and is used to output the DC bus voltage after fuse protection. The DC bus voltage includes the positive DC bus and the negative DC bus.

[0075] The input terminal of the three-phase half-bridge inverter drive module is connected to the DC positive bus and the gate drive circuit 10. It is used to receive the gate drive signal to output the corresponding three-phase AC voltage to the external load, as well as to output the lower bridge arm common node potential PV and the three-phase half-bridge midpoint potential.

[0076] The input terminal of the reference ground interface module is connected to the midpoint potential of the three-phase half-bridge and the common node potential PV of the lower bridge arm, and the output terminal is connected to the gate drive circuit 10.

[0077] The input terminal of the detection and conditioning module is connected to the DC negative bus, the 5VD auxiliary power supply and the 3.3VD auxiliary power supply, and the output terminal is connected to the ADC sampling port of the control circuit to output an analog sampling signal idcadc that is proportional to the bus current.

[0078] It is understood that the input protection module is located at the power input port of the three-phase inverter power circuit 20. The input terminal is connected to the external DC power supply, and the output terminal outputs the DC bus voltage including the DC positive bus and the DC negative bus to power the subsequent circuit. It is used to provide fuse protection to reduce the risk of fault propagation. It can be in the form of a fuse with terminal block structure, power input interface with overcurrent protection, etc.

[0079] The three-phase half-bridge inverter drive module is the power conversion execution unit of the three-phase inverter power circuit 20. Its input terminals are respectively connected to the DC positive bus and the gate drive signal output by the gate drive circuit 10. The power output terminal is connected to the external load, and the potential output terminal is connected to the reference ground interface module. It is used to output the corresponding three-phase AC voltage under the control of the gate drive signal, and at the same time output the lower bridge arm common node potential PV and the three-phase half-bridge midpoint potential. It can be a three-phase half-bridge structure built with metal oxide semiconductor field-effect transistors, a three-phase inverter bridge built with insulated gate bipolar transistors, etc.

[0080] The reference ground interface module is connected in series between the three-phase half-bridge inverter drive module and the gate drive circuit 10. The input terminal collects the midpoint potential of the three-phase half-bridge and the common node potential PV of the lower bridge arm. The output terminal is connected to the corresponding port of the gate drive circuit 10. It is used to transmit the power side node potential as a reference for the drive circuit, while suppressing power side interference from entering the drive circuit. It can be implemented using an interface structure with series damping resistors, a potential transmission circuit with a filter network, or other similar forms.

[0081] The detection and conditioning module is located on the DC negative bus side of the three-phase inverter power circuit 20. Its input terminals are connected to the DC negative bus, the 5VD auxiliary power supply, and the 3.3VD auxiliary power supply, respectively. Its output terminal is connected to the ADC sampling port of the control circuit. It is used to collect the bus current and output an analog sampling signal idcadc that is proportional to it. It can be in the form of a Hall current sensor with a RC filter network, a shunt resistor with a differential amplifier circuit, etc.

[0082] With this configuration, the external DC power supply is connected to the input protection module, and after fuse protection, it outputs the DC bus voltage, which includes both the positive and negative DC bus voltages. The positive DC bus voltage is fed into the three-phase half-bridge inverter drive module, and the gate drive signal output by the gate drive circuit 10 is simultaneously connected to the three-phase half-bridge inverter drive module, controlling it to convert DC power into the corresponding three-phase AC voltage and output it to the external load. At the same time, it outputs the lower bridge arm common node potential PV and the three-phase half-bridge midpoint potential. The node potentials output by the three-phase half-bridge inverter drive module are sent to the reference ground interface module, and after interface processing, they are transmitted to the gate drive circuit 10 as the drive reference potential. The detection and conditioning module collects current information from the negative DC bus, and with the support of the 5VD auxiliary power supply and the 3.3VD auxiliary power supply, it outputs an analog sampling signal idcadc proportional to the bus current to the ADC sampling port of the control circuit. The four sub-modules cooperate with each other to complete the functions of power input protection, DC-AC power conversion, drive reference potential transmission, and bus current sampling and conditioning, supporting the stable operation of the three-phase inverter power circuit 20.

[0083] In some embodiments, please refer to Figure 2 The three-phase half-bridge inverter drive module includes NMOS transistors Q1, Q2, Q3, Q4, Q5, and Q6, resistors R7, R8, R9, R10, R11, and R12, and diodes D5, D6, D7, D8, D9, and D10.

[0084] NMOS transistors Q1, Q2, and Q3 serve as the upper bridge arm MOS transistors, with their drains connected to the DC positive bus and then to the output of the input protection module.

[0085] NMOS transistors Q4, Q5, and Q6 serve as the lower bridge arm MOS transistors, with their sources connected to the common node potential PV of the lower bridge arm.

[0086] The source of NMOS transistor Q1 is connected to the drain of NMOS transistor Q4, forming the midpoint of the U-phase half-bridge. The midpoint of the U-phase half-bridge also leads out the U-phase power output terminal and the VS1 reference potential node. Resistor R7 and diode D5 are connected in reverse parallel, with their left ends connected to the pre-stage drive signal HO1 and their right ends connected to the gate of NMOS transistor Q1. The cathode of diode D5 is connected to the HO1 side, and the anode is connected to the gate side of NMOS transistor Q1. Resistor R10 and diode D8 are connected in reverse parallel, with their left ends connected to the pre-stage drive signal LO1 and their right ends connected to the gate of NMOS transistor Q4. The cathode of diode D8 is connected to the LO1 side, and the anode is connected to the gate side of NMOS transistor Q4.

[0087] The source of NMOS transistor Q2 is connected to the drain of NMOS transistor Q5, forming the midpoint of the V-phase half-bridge. The midpoint of the V-phase half-bridge also leads out the V-phase power output terminal and the VS2 reference potential node. Resistor R8 and diode D6 are connected in reverse parallel, with their left ends connected to the pre-stage drive signal HO2 and their right ends connected to the gate of NMOS transistor Q2. The cathode of diode D6 is connected to the HO2 side, and the anode is connected to the gate side of NMOS transistor Q2. Resistor R11 and diode D9 are connected in reverse parallel, with their left ends connected to the pre-stage drive signal LO2 and their right ends connected to the gate of NMOS transistor Q5. The cathode of diode D9 is connected to the LO2 side, and the anode is connected to the gate side of NMOS transistor Q5.

[0088] The source of NMOS transistor Q3 is connected to the drain of NMOS transistor Q6, forming the midpoint of the W-phase half-bridge. The midpoint of the W-phase half-bridge also leads out the W-phase power output terminal and the VS3 reference potential node. Resistor R9 and diode D7 are connected in reverse parallel, with their left ends connected to the pre-amplifier drive signal HO3 and their right ends connected to the gate of NMOS transistor Q3. The cathode of diode D7 is connected to the HO3 side, and the anode is connected to the gate side of NMOS transistor Q3. Resistor R12 and diode D10 are connected in reverse parallel, with their left ends connected to the pre-amplifier drive signal LO3 and their right ends connected to the gate of NMOS transistor Q6. The cathode of diode D10 is connected to the LO3 side, and the anode is connected to the gate side of NMOS transistor Q6.

[0089] It can be understood that NMOS transistors Q1, Q2, and Q3 together constitute the upper arm power group of the three-phase half-bridge. Their drains are connected to the DC positive bus to obtain DC power, and their sources are connected to the drains of the lower arm of each phase respectively. They are used to control the on / off state of the three-phase upper arm. Power NMOS transistors, high-voltage N-channel MOSFETs, and other types can be used.

[0090] NMOS transistors Q4, Q5, and Q6 together form the lower arm power group of a three-phase half-bridge. Their sources are connected to the common node potential PV of the lower arm, and their drains are connected to the sources of the upper arm of each phase, respectively. They are used to control the on / off state of the three-phase lower arm. Power NMOS transistors, high-voltage N-channel MOSFETs, and other types can be used.

[0091] The three sets of anti-parallel two-terminal passive networks, namely resistor R7 and diode D5, resistor R8 and diode D6, and resistor R9 and diode D7, correspond to the gate drive branches of the three-phase upper bridge arms. One end of each set is connected to the corresponding phase's front-stage upper bridge arm drive signal, and the other end is connected to the gate of the corresponding phase's upper bridge arm NMOS transistor. This is used to limit the gate charging current when the power transistor is turned on and to accelerate the gate charge discharge speed when it is turned off. Combinations such as surface-mount metal film resistors with fast recovery diodes and through-hole carbon film resistors with Schottky diodes can be used.

[0092] The three sets of anti-parallel two-terminal passive networks, namely resistor R10 and diode D8, resistor R11 and diode D9, and resistor R12 and diode D10, correspond to the gate drive branches of the three-phase lower bridge arms. One end of each set is connected to the corresponding phase's front-end lower bridge arm drive signal, and the other end is connected to the gate of the corresponding phase's lower bridge arm NMOS transistor. This is used to limit the gate charging current when the power transistor is turned on and to accelerate the gate charge discharge speed when it is turned off. Combinations such as surface-mount metal film resistors with fast recovery diodes and through-hole carbon film resistors with Schottky diodes can be used.

[0093] With this configuration, the gate of each NMOS transistor is connected to a set of anti-parallel resistors and diodes, which receive the corresponding drive signals from the preceding stage. Under the control of the drive signals, the upper and lower bridge arms of each phase alternately switch on and off, converting the DC power input from the DC positive bus into a three-phase AC voltage output to the external load. At the same time, the midpoint potential of each phase half-bridge and the common node potential of the lower bridge arm are output to the reference ground interface module. The two-terminal passive network at the gate helps to adjust the switching speed of the power transistors, which helps to reduce switching losses and interference risks, and together realizes the power conversion and signal transmission functions of the three-phase half-bridge inverter drive module.

[0094] In some embodiments, please refer to Figure 2 The reference ground interface module includes resistors R13, R14, R15, and R16.

[0095] One end of resistor R13 is connected to the midpoint of the U-phase half-bridge, and the other end is connected to the VS1 interface of the gate drive circuit 10.

[0096] One end of resistor R14 is connected to the midpoint of the V-phase half-bridge, and the other end is connected to the VS2 interface of the gate drive circuit 10.

[0097] One end of resistor R15 is connected to the midpoint of the W-phase half-bridge, and the other end is connected to the VS3 interface of the gate drive circuit 10.

[0098] One end of resistor R16 is connected to the common node potential PV of the lower bridge arm, and the other end is connected to the COM interface of the gate drive circuit 10.

[0099] It is understandable that resistors R13, R14, and R15 are used to transmit the real-time potential of the midpoint of each phase half-bridge, while the damping characteristics of the series resistors are used to reduce spike interference. Surface mount metal film resistors, through-hole carbon film resistors, and other types can be used.

[0100] Resistor R16 is used to transmit the common ground potential of the lower bridge arm and limit the ingress of interference current. It can be a surface-mount metal film resistor or a through-hole carbon film resistor.

[0101] With this configuration, the four resistors are connected in series between the corresponding nodes of the power loop and the corresponding interfaces of the gate drive circuit 10. While transmitting the potential of each node as a reference for the gate drive circuit 10, they also dampen and suppress voltage spikes and interference on the power side, assisting the gate drive circuit 10 in obtaining the corresponding potential reference, which helps to improve the stability of the drive circuit operation.

[0102] In some embodiments, please refer to Figure 2 The detection and conditioning module includes a current detection chip U2, resistors R19, R20, and R21, capacitor C9, and a Schottky diode pair D11.

[0103] The DC negative bus is connected to the primary current input terminal of the current detection chip U2, and the primary current output terminal of the current detection chip U2 is connected to the power ground PGND.

[0104] The 5VD auxiliary power supply is connected to pin 8 of the current detection chip U2, and pin 5 of the current detection chip U2 is connected to digital ground DGND.

[0105] Pins 7 and 6 of the current detection chip U2 are shorted. The shorted point is connected to one end of resistor R19, one end of resistor R21, and one end of capacitor C9. The other end of resistor R21 is connected to digital ground DGND, and the other end of capacitor C9 is connected to digital ground DGND. The other end of resistor R19 leads out to the idcadc sampling port and is simultaneously connected to one end of resistor R20 and the common terminal of Schottky diode D11. The other end of resistor R20 is connected to digital ground DGND.

[0106] The Schottky diode pair D11 is connected between the idcadc sampling port and the 3.3V auxiliary power supply and digital ground DGND.

[0107] It is understandable that the current detection chip U2 is used to sense the magnitude of the bus current and output the corresponding analog voltage signal. It can be of the type such as Hall effect current detection chip or magnetic balance current detection chip.

[0108] Resistor R21 and capacitor C9 are connected in parallel between the output of current detection chip U2 and digital ground DGND. Resistor R21 serves as the load resistor for the chip output, and capacitor C9, together with the filter pin of current detection chip U2, sets the detection bandwidth and filters out high-frequency interference components in the output signal. Combinations such as metal film resistor with multilayer ceramic capacitor or carbon film resistor with monolithic ceramic capacitor can be used.

[0109] Resistors R19 and R20 work together to form a resistor voltage divider network, which is used to adjust the voltage amplitude of the sampled signal to match the input range of the sampling port of the subsequent stage. Surface mount precision resistors, through-hole metal film resistors, and other types can be used.

[0110] Schottky diode pair D11 is a port protection device used to provide voltage clamping protection for the sampled output signal, reducing the probability of overvoltage surges damaging subsequent circuits. It can be used in the form of integrated dual Schottky diode pairs or discrete Schottky diode pair combinations.

[0111] With this setup, the DC negative bus is connected to the primary current input terminal of the current detection chip U2, and the primary current output terminal of the current detection chip U2 is connected to power ground PGND, sensing the bus current information through the primary circuit; the 5VD auxiliary power supply is connected to pin 8 of the current detection chip U2 to provide operating power for the chip, and pin 5 of the current detection chip U2 is connected to digital ground DGND to form a secondary power supply reference circuit; pins 7 and 6 of the current detection chip U2 are shorted to output an analog signal, which is simultaneously connected to one end of resistor R19, one end of resistor R21, and one end of capacitor C9; the other end of resistor R21 and the other end of capacitor C9 are connected together to... Digital ground (DGND) forms the output load and filter network. The other end of resistor R19 leads to the idcadc sampling output port and is connected to one end of resistor R20. The other end of resistor R20 is connected to digital ground (DGND) to complete the voltage division adjustment of the sampling signal. Schottky diode pair D11 is connected between the idcadc sampling port and the 3.3V auxiliary power supply and digital ground (DGND) to provide voltage clamping protection for the output sampling signal. All components work together to complete the detection, signal conditioning, and port protection of the bus current, providing the control circuit with an analog sampling signal proportional to the bus current.

[0112] In some embodiments, please refer to Figure 2 The input protection module includes terminal block P1 and fuse F1.

[0113] The positive terminal of the external DC power supply is connected to pin 2 of terminal P1. Pin 2 of terminal P1 is connected in series to one end of fuse F1. The other end of fuse F1 outputs a DC positive bus and is connected to the three-phase half-bridge inverter drive module.

[0114] The negative terminal of the external DC power supply is connected to pin 1 of terminal P1. The negative DC bus is led out from pin 1 of terminal P1 and connected to the primary current input terminal of the current detection chip U2 in the detection and conditioning module. The total current of the bus flows into the power ground PGND after passing through the primary winding of the current detection chip U2, forming a complete current loop with the positive terminal of the power supply.

[0115] It is understandable that terminal P1 is responsible for the physical connection of positive and negative power supplies. Its positive pin is connected in series with fuse F1 to form a positive overcurrent protection branch, which is used to realize overcurrent protection on the positive side. The terminal can be a screw terminal, a plug-in terminal, etc., and the fuse can be a surface mount fuse, a glass tube fuse, etc.

[0116] With this configuration, terminal P1 and fuse F1 together form an input protection module. While enabling physical access to external DC power, it can also cut off the positive power supply path by blowing the fuse when an overcurrent fault occurs in the circuit, which helps to reduce the impact of the fault on the front-end power supply and subsequent circuits.

[0117] Please see Figure 3 This application also provides a fuel pump control method, applied to the fuel pump intelligent controller 100 of any of the above claims. The fuel pump control method includes: Obtain the actual speed and bus current of the fuel pump motor at the current moment; the bus current is used to reflect the load state of the fuel pump motor, and the degree of deviation of the bus current from the steady-state load current base value reflects the deviation of the fuel pump motor output torque.

[0118] If the rate of change of the actual speed is greater than the first threshold and the bus current is less than the base value of the steady-state load current corresponding to the current speed, it is determined that there is a risk of speed overshoot and a first negative correction amount is generated; wherein, the base value of the steady-state load current is the reference load current value required for the fuel pump motor to run in steady state at the current speed, and the first negative correction amount is the correction value used to reduce the target current command.

[0119] The first negative correction is superimposed on the target current command output by the magnetic field orientation control (FOC) speed loop to obtain the first target current command.

[0120] Based on the first target current command, control the output torque of the fuel pump motor.

[0121] As can be seen from the above, the fuel pump control method provided in this application obtains the actual speed and bus current of the fuel pump motor to reflect the load state, and identifies the fluid pressure build-up hysteresis characteristics when the rate of change of the actual speed is greater than a first threshold and the bus current is less than the steady-state load current base value corresponding to the current speed. Then, it generates a first negative correction amount for reducing the target current command. By superimposing the first negative correction amount onto the target current command output by the field-oriented control (FOC) speed loop, a first target current command is obtained. Based on the first target current command, the output torque of the fuel pump motor is controlled to reduce the drive torque in advance before speed overshoot and oil pressure overcharge occur. This reduces the speed overshoot amplitude caused by the time constant mismatch between the fast electrical response of the motor and the slow fluid pressure build-up response under the transient condition of rapid acceleration and pressure build-up, and suppresses the pressure impact peak of the water hammer effect during oil pressure overcharge.

[0122] To better understand the fuel pump control method provided in the embodiments of this application, the specific implementation process of the fuel pump control method provided in the embodiments of this application will be described by way of example below.

[0123] Figure 3 A schematic flowchart of a fuel pump control method provided in an embodiment of this application is shown. The fuel pump control method includes: S100, obtain the actual speed and bus current of the fuel pump motor at the current moment; wherein, the bus current is used to reflect the load state of the fuel pump motor, and the degree of deviation of the bus current from the steady-state load current base value reflects the deviation of the fuel pump motor output torque.

[0124] It can be understood that the current moment refers to the time point at which the fuel pump motor control system performs data sampling within each control cycle. For example, the current moment can be the instant when the control circuit triggers ADC sampling and Hall signal capture every 100 microseconds.

[0125] A fuel pump motor is a brushless DC motor or permanent magnet synchronous motor installed in the vehicle's fuel tank to deliver fuel to the engine. For example, a fuel pump motor can be a BLDC motor with three-phase star windings or a PMSM motor with surface-mounted permanent magnets.

[0126] Actual rotational speed refers to the rotational speed of the fuel pump motor rotor at the current moment, expressed in revolutions per minute (rpm).

[0127] Bus current refers to the total current flowing through the DC input bus of the three-phase inverter power circuit. The magnitude of the bus current is related to the current electromagnetic load of the motor. For example, the bus current is about 2.5 amps when running at a steady state of 3000 rpm, and may be only 1.2 amps at the beginning of rapid acceleration because the fluid resistance has not yet been established.

[0128] Load condition refers to the level of external resistance that the fuel pump motor experiences under current operating conditions, including fuel line flow resistance, fuel rail back pressure, and the mechanical torque that the impeller needs to overcome to pump fluid. For example, load condition can refer to a stable high load condition during smooth cruising, an instantaneous high impact load condition during the initial stage of rapid acceleration and pressure build-up, and a low load condition during idling or deceleration.

[0129] The steady-state load current base value refers to the basic current value required to drive the fuel pump motor when it has reached steady-state operation at the current speed and all fluid resistance has been established. For example, the steady-state load current base value corresponding to the 3000rpm speed range is 2.5 amperes, and the steady-state load current base value corresponding to the 4000rpm speed range is 3.2 amperes, etc.

[0130] The degree of deviation refers to the difference between the actual value of the bus current and the base value of the steady-state load current. The larger the difference, the more the current load deviates from the steady-state level.

[0131] Output torque refers to the rotational torque output by the fuel pump motor rotor through electromagnetic action. Its magnitude is proportional to the q-axis current flowing through the motor windings, i.e., the torque current. For example, in FOC control, when the target q-axis current command is 5 amperes, the corresponding output torque is approximately 0.15 Nm.

[0132] Deviation refers to the difference between the actual controlled quantity and the reference quantity. It refers to the deviation of the bus current relative to the steady-state load current base value. For example, when the bus current is 0.5 amperes lower than the steady-state load current base value, there is a positive deviation.

[0133] The actual speed of the fuel pump motor at the current moment can be obtained by capturing the actual speed through the edge of the Hall position sensor, or by estimating the actual speed through a sensorless back EMF sliding mode observer.

[0134] The current bus current of the fuel pump motor can be obtained by sampling the bus current through a Hall current sensor, or by using a sampling resistor in conjunction with a differential operational amplifier.

[0135] By obtaining the actual speed and bus current of the fuel pump motor at the current moment to reflect the load state of the fuel pump motor, and by reflecting the deviation of the output torque of the fuel pump motor by the degree of deviation of the bus current from the steady-state load current base value, input basis can be provided for subsequent judgment of speed overshoot risk and generation of the first negative correction amount.

[0136] S200: If the rate of change of the actual speed is greater than the first threshold and the bus current is less than the base value of the steady-state load current corresponding to the current speed, it is determined that there is a risk of speed overshoot and a first negative correction amount is generated; wherein, the base value of the steady-state load current is the reference load current value required for the fuel pump motor to run in steady state at the current speed, and the first negative correction amount is the correction value used to reduce the target current command.

[0137] It is understandable that the rate of change of actual speed refers to the amount of change in the actual speed of the fuel pump motor per unit time, reflecting the acceleration or deceleration trend of the motor speed. For example, when the current speed is 3100 rpm, the speed in the previous control cycle is 3050 rpm, and the control cycle is 100 microseconds, the rate of change is 500 rpm per millisecond.

[0138] The first threshold refers to a preset threshold value for the rate of change of speed used to determine whether the fuel pump motor is in a state of rapid acceleration. When the rate of change of speed exceeds this threshold value, it indicates that the motor is operating at a high acceleration that may cause overshoot. The first threshold can be determined based on the rated parameters of the fuel pump motor, through bench rapid acceleration step response tests. Under rated voltage and standard fuel line conditions, the fuel pump motor is accelerated from idle speed to rated speed in a step, and the change in speed overshoot is monitored at different candidate thresholds. The inflection point of the rate of change of speed before the overshoot begins to increase significantly is taken as the first threshold. The value range of the first threshold can be from 30 rpm per millisecond to 80 rpm per millisecond, etc.

[0139] The current speed refers to the actual operating speed of the fuel pump motor at the current moment. The current speed is the speed value corresponding to the steady-state load current base value. For example, when the current speed is 3500 rpm, the steady-state load current base value found from the base value table is 2.8 amps.

[0140] Speed ​​overshoot risk refers to the risk that the fuel pump motor speed may continue to rise and exceed the target value after approaching the target speed, based on the current operating parameters. Speed ​​overshoot risk is determined by the simultaneous satisfaction of two conditions: the actual speed change rate is greater than the first threshold and the bus current is less than the steady-state load current base value.

[0141] The first negative correction amount refers to the correction value generated when it is determined that there is a risk of speed overshoot, which is used to deduct from the target current command. The function of the first negative correction amount is to reduce the motor drive torque in advance before the overshoot actually occurs.

[0142] The target current command refers to the current reference value output by the proportional-integral regulator of the field-oriented control (FOC) speed loop after calculating the deviation between the target speed and the actual speed. This command is used for closed-loop regulation of the current loop.

[0143] The first negative correction can be generated by combining a lookup table based on the speed with the addition of static and dynamic components of the speed change rate gain. For example, the control circuit can look up a pre-stored speed-to-base correction mapping table based on the current actual speed of 3500 rpm. In this mapping table, the base correction for the speed range of 3000 to 3500 rpm is 0.5 amps. The preset gain coefficient is calibrated on the bench to be 0.012 amps multiplied by milliseconds per rpm. The current actual speed change rate is 80 rpm per millisecond. According to the formula that the first negative correction equals the sum of the base correction and the preset gain coefficient multiplied by the speed change rate, the first negative correction = 0.5 + 0.012 * 80 = 1.46 amps. Alternatively, it can be calculated in real time based on proportional control, directly multiplying the speed change rate by a fixed gain coefficient to obtain the first negative correction.

[0144] The first threshold can be determined by calibrating the fuel pump motor based on its rated parameters through bench acceleration tests. For example, the fuel pump motor can be accelerated from idle speed of 800 rpm to the target speed of 3000 rpm under rated voltage and standard fuel line conditions. The range of the first threshold value can be gradually adjusted, and the magnitude of the speed overshoot under different first threshold values ​​can be recorded. The inflection point of the rate of change of speed before the overshoot begins to increase significantly can be selected as the first threshold. For example, the calibration results show that when the first threshold is set to 50 rpm per millisecond, the overshoot remains within 1.5%, while when it exceeds 50 rpm per millisecond, the overshoot increases significantly. The first threshold of the motor is set to 50 rpm per millisecond; alternatively, a segmented variable threshold can be set according to the actual speed range. For example, the operating speed range of the fuel pump motor can be divided into multiple ranges. In the low speed range, such as below 1500 rpm, a smaller first threshold such as 30 rpm per millisecond is set; in the medium speed range, such as 1500 to 3500 rpm, a medium first threshold such as 50 rpm per millisecond is set; and in the high speed range, such as above 3500 rpm, a larger first threshold such as 70 rpm per millisecond is set to adapt to the differences in motor rotational inertia and fluid load characteristics in different speed ranges.

[0145] By determining that there is a risk of speed overshoot when the rate of change of the actual speed is greater than the first threshold and the bus current is less than the base value of the steady-state load current corresponding to the current speed, a first negative correction amount is generated to reduce the target current command. In this way, the fluid pressure build-up hysteresis characteristics of high speed and low load are identified in advance before the fluid resistance is fully established, and a correction basis is provided for subsequent suppression of overshoot.

[0146] In one possible implementation, in S200, generating the first negative correction amount includes: S210, based on the actual speed, query the pre-stored speed-to-basic correction mapping table to obtain the basic correction amount corresponding to the actual speed; wherein, the speed-to-basic correction mapping table is a data table calibrated through bench tests to record the basic reduction amount corresponding to different speed ranges.

[0147] It can be understood that the speed-to-base correction mapping table is a lookup table data structure that uses speed or speed range as index and the required current reduction under the corresponding operating condition as the base correction. It can be stored in the memory of the control circuit for quick lookup during operation by array subscript or range matching.

[0148] The base correction amount refers to the static baseline reduction amount obtained from the mapping table based on the current speed in a rapid acceleration overshoot risk scenario. It reflects the base current value that needs to be deducted from the target current command to suppress overshoot at the current speed level.

[0149] Bench testing refers to an offline testing method in which the fuel pump motor control system is installed on a dedicated test bench in a laboratory to simulate the transient conditions of rapid acceleration and pressure build-up of a vehicle and to collect performance data such as speed overshoot and oil pressure overcharge peak for parameter calibration.

[0150] The basic reduction amount refers to the current value deducted from the target current command corresponding to the basic correction amount. It is positively correlated with the speed, that is, the higher the speed, the greater the basic reduction amount.

[0151] One way to establish a mapping table between engine speed and baseline correction is to determine the baseline correction interval by interval through bench step acceleration calibration. For example, the fuel pump motor is mounted on a test bench, and within each speed interval to be calibrated (e.g., 3000 to 3500 rpm), a target speed step is applied from the lower boundary of the interval (3000 rpm), and the overshoot is observed. The initial value of the baseline correction is set to zero, and the increment step of the baseline correction is gradually increased (e.g., 0.1 amperes each time), and the speed overshoot of each test is recorded. When the overshoot decreases to the target range (e.g., the overshoot does not exceed 1.5%), the current basic correction is recorded. Assuming that after five increments in the 3000 to 3500 rpm range, the overshoot decreases from 8% to 1.2% when the basic correction reaches 0.5 Amperes, the basic correction for this range is calibrated to be 0.5 Amperes. The same steps are used to complete the calibration of all speed ranges such as 3500 to 4000 rpm and 4000 to 4500 rpm, and a complete speed-to-basic correction mapping table is formed.

[0152] By querying the pre-stored speed-to-base correction mapping table based on the actual speed, the base correction corresponding to the actual speed is obtained. This transforms the static operating condition information of speed into a reference current reduction that matches the current speed level, providing a static reference component for the subsequent synthesis of the first negative correction by combining the dynamic rate of change component.

[0153] S220 determines the dynamic correction component based on the product of the actual speed change rate and the preset gain coefficient; whereby the preset gain coefficient is a fixed value calibrated through bench testing.

[0154] It is understandable that the rate of change of actual speed refers to the amount of change in the actual speed of the fuel pump motor per unit time, reflecting the acceleration or deceleration trend of the motor speed.

[0155] The preset gain coefficient refers to a fixed proportional coefficient that is pre-calibrated through bench testing and stored in the controller's memory. Its function is to convert the dynamic indicator of the speed change rate into an increment of current. For example, the preset gain coefficient can be 0.012 amperes per millisecond per rpm, 0.008 amperes per millisecond per rpm, etc.

[0156] The dynamic correction component refers to the incremental correction part obtained by multiplying the actual speed change rate by the preset gain coefficient and superimposed on the basic correction amount. The dynamic correction component is proportional to the speed change rate, that is, the more violent the acceleration, the larger the component.

[0157] The dynamic correction component can be determined by multiplying the actual speed change rate by the preset gain coefficient. One method is to calculate the dynamic correction component based on the preset gain coefficient multiplied by the actual speed change rate. For example, if the preset gain coefficient is 0.012 amperes per millisecond per rpm (calibrated on a test bench), and the current actual speed is 3050 rpm, and the actual speed 100 microseconds before the last control cycle was 3042 rpm, the calculated actual speed change rate is (3050 - 3042) / 0.1 = 80 rpm per millisecond, and the dynamic correction component is 0.012 * 80 = 0.96 amperes. Alternatively, it can be determined based on a segmented preset gain coefficient according to the range of the actual speed. After selecting the corresponding gain coefficient, multiply it by the rate of change of the actual speed. For example, divide the operating range of the fuel pump motor into multiple intervals and calibrate an independent preset gain coefficient in each interval. For example, the preset gain coefficient for the interval below 1500 rpm is 0.005, the preset gain coefficient for the interval from 1500 to 3500 rpm is 0.012, and the preset gain coefficient for the interval above 3500 rpm is 0.015. Assuming the current actual speed is 4000 rpm, the corresponding preset gain coefficient is 0.015, and the rate of change of the actual speed is 60 rpm per millisecond, then the dynamic correction component = 0.015 * 60 = 0.9 amperes, etc.

[0158] The dynamic correction component is determined by multiplying the actual speed change rate by a preset gain coefficient. This quantifies the overshoot trend prediction information contained in the speed change rate into a current increment that can be superimposed on the basic correction amount, providing a dynamic component for the subsequent summation of the static reference and the dynamic prediction to synthesize the first negative correction amount.

[0159] S230, the sum of the basic correction amount and the dynamic correction component is determined as the first negative correction amount.

[0160] It is understandable that the sum of the basic correction amount and the dynamic correction component is determined as the first negative correction amount. Thus, the basic correction amount obtained from the table based on the static operating condition of the speed and the dynamic correction component obtained from the dynamic prediction of the speed change rate are added together to form a complete correction amount that simultaneously integrates the two dimensions of information: the current speed level and the overshoot trend intensity.

[0161] S300, the first negative correction amount is superimposed on the target current command output by the magnetic field orientation control (FOC) speed loop to obtain the first target current command.

[0162] As can be understood, superposition refers to the mathematical operation of subtracting the target current command from the first negative correction amount as a deduction amount, thereby reducing the reference input value of the current loop in advance before overshoot occurs. For example, subtracting the first negative correction amount of 1.46 amps from the target current command of 5 amps results in the first target current command of 3.54 amps.

[0163] The field-oriented control (FOC) speed loop refers to the outer loop regulator in the dual closed-loop architecture of field-oriented control. Its input is the deviation between the target speed issued by the engine ECU (electronic control unit) and the actual speed fed back by the position sensing circuit. After calculation by the proportional-integral regulator, the target current command (i.e., the q-axis current reference value) is output.

[0164] The first target current command refers to the reduced current command obtained by subtracting the first negative correction from the original target current command output by the FOC speed loop. This command replaces the original target current command and is sent into the current loop as the new q-axis current reference value.

[0165] The first target current command is obtained by superimposing the first negative correction amount onto the target current command output by the field-oriented control (FOC) speed loop. This allows the current command to be directly reduced at the output end of the FOC speed loop instead of the input end, thereby reducing the reference input value of the current loop in advance without modifying the speed loop proportional-integral parameters.

[0166] S400 controls the output torque of the fuel pump motor based on the first target current command.

[0167] It is understandable that the method of controlling the output torque of the fuel pump motor based on the first target current command can be to change the amplitude of the three-phase voltage of the motor by adjusting the pulse width modulation duty cycle. The control circuit sends the first target current command to the proportional-integral regulator of the FOC current loop as the q-axis current reference value. The current loop proportional-integral regulator updates the pulse width modulation duty cycle after calculating the deviation between the reference value and the actual bus current fed back by the current sampling circuit. For example, when the quadrature axis current is 4.2 amps, which is greater than the first target current command of 3.54 amps, the current loop proportional-integral regulator outputs a negative duty cycle adjustment, which reduces the duty cycle of the six pulse width modulation signals. The gate drive circuit amplifies the pulse width modulation signal with the reduced duty cycle and drives the MOSFET of the three-phase inverter power circuit, so that the average voltage applied to the motor windings is reduced from the original level corresponding to 5 amps to the level corresponding to 3.54 amps, thereby reducing the output torque of the motor. Assuming the pulse width modulation frequency is 20 kHz, i.e., the period is 50 microseconds, and the duty cycle of the upper bridge arm before adjustment is 60%, i.e., 30 microseconds of conduction, the duty cycle adjustment ratio is based on the ratio of target current to actual current 3.54 / 4.2≈0.843. After adjustment, the duty cycle is 60%*0.843≈50.6%, corresponding to a conduction time of 50*50.6%≈25.3 microseconds. The time for the fuel pump motor winding to draw energy from the DC bus in each pulse width modulation cycle is shortened, and the electromagnetic torque decreases accordingly.

[0168] By using the first target current command as a new q-axis current reference value for the current loop to input the proportional-integral regulator of the field-oriented control (FOC) current loop to adjust the pulse width modulation duty cycle, and then amplifying the pulse width modulation signal through the gate drive circuit to drive the power MOSFET in the three-phase inverter power circuit to adjust the voltage amplitude and phase applied to the three-phase windings of the fuel pump motor, thereby controlling the output torque of the fuel pump motor.

[0169] In one possible implementation, the fuel pump control method further includes: S500: Obtain the steady-state load current base value corresponding to the current speed of the fuel pump motor.

[0170] It is understandable that the current speed refers to the actual operating speed of the fuel pump motor at the current moment.

[0171] The steady-state load current base value corresponding to the current speed of the fuel pump motor can be obtained by querying the pre-stored steady-state feedforward current base value table according to the current speed by interval matching; or it can be calculated by linear interpolation based on the current speed, etc.

[0172] By obtaining the steady-state load current base value corresponding to the current speed of the fuel pump motor, a steady-state reference value is provided for the subsequent calculation of the current difference between the steady-state load current base value and the bus current, so that the current difference can reflect the degree to which the current actual load current deviates from the steady-state normal level.

[0173] S600 calculates the current difference between the steady-state load current base value and the bus current.

[0174] It can be understood that the current difference refers to the absolute value of the difference between the steady-state load current base value and the current bus current. Its magnitude directly reflects the degree to which the actual driving current of the motor is lower than the steady-state normal level, that is, the severity of the fluid pressure build-up hysteresis. For example, when the steady-state load current base value is 2.5 amps and the bus current is 1.2 amps, the current difference is 1.3 amps.

[0175] By calculating the current difference between the steady-state load current base value and the bus current, the difference between the current actual drive current and the steady-state reference current is quantified into a comparable value, providing a basis for subsequent judgment on whether the difference exceeds the second threshold and whether to generate an additional correction amount.

[0176] S700, when the absolute value of the current difference is greater than the second threshold, the product of the current difference and the preset additional gain coefficient is used as the additional correction amount, which is added to the first negative correction amount to obtain the increased first negative correction amount; wherein, the preset additional gain coefficient is a fixed value calibrated by bench test.

[0177] It is understandable that the second threshold refers to a preset current difference threshold value, which is used to determine whether the fluid pressure build-up hysteresis has become significant enough to trigger the need for additional corrections beyond the basic correction and dynamic correction components. For example, the second threshold could be 0.5, 0.8 amperes, etc.

[0178] The preset additional gain coefficient refers to a fixed proportional coefficient that is pre-calibrated through bench testing and stored in the controller's memory. Its function is to convert the current difference into an additional correction amount proportionally. For example, the preset additional gain coefficient can be 0.8 amperes per ampere, 1.0 amperes per ampere, etc.

[0179] The additional correction amount refers to the correction portion that is superimposed on the first negative correction amount, obtained by multiplying the current difference by the preset additional gain coefficient. It is proportional to the current difference, that is, the more severe the fluid hysteresis, the greater the additional correction amount.

[0180] The second threshold can be determined by calibrating the inflection point of the current difference under rapid acceleration conditions in bench tests. For example, multiple rapid acceleration pressure build-up conditions can be set up in the bench test. In each condition, the fuel pump motor accelerates from idle speed of 800 rpm to different target speeds such as 2000 rpm, 3000 rpm, and 4000 rpm. The peak value of the current difference ΔI and its corresponding speed overshoot are recorded for each condition. When the peak value of ΔI is less than a certain value, the speed overshoot remains within 1.5% and no additional correction is needed. When the peak value of ΔI exceeds this value, the overshoot begins to increase significantly and additional correction needs to be introduced. The critical ΔI value is found from the recorded data to calibrate the second threshold. Alternatively, it can be set proportionally to the steady-state load current base value. The second threshold is a certain percentage of the steady-state load current base value corresponding to the current speed, such as 20% of the steady-state load current base value. Assuming the steady-state load current base value is 2.5 amps when the current speed is 3500 rpm, the second threshold = 2.5 * 20% = 0.5 amps; when the current speed is 4000 rpm, the steady-state load current base value is 3.2 amps, the second threshold = 3.2 * 20% = 0.64 amps, and so on.

[0181] When the absolute value of the current difference is greater than the second threshold, the product of the current difference and the preset additional gain coefficient is used as an additional correction amount, which is added to the first negative correction amount to obtain an increased first negative correction amount. This allows the correction amplitude to be increased proportionally according to the magnitude of the current difference under conditions where the fluid hysteresis is significant, so as to provide stronger overshoot suppression. Under conditions where the fluid hysteresis is slight, the additional correction is not triggered to avoid overcorrection.

[0182] In one possible implementation, the fuel pump control method further includes: S200a, before generating the first negative correction amount, if the initial target current of the fuel pump motor is greater than the third threshold, the initial target current is corrected based on the preset static correction parameters to obtain the target current command output by the FOC speed loop.

[0183] It is understandable that the initial target current is an intermediate value that has been calculated and output by the speed loop proportional-integral regulator but has not yet been processed by the static correction stage. Only the corrected value is used as the final target current command output by the speed loop.

[0184] The third threshold refers to the preset safe upper limit of the initial target current, which is used to determine whether the initial target current has exceeded the safe current level allowed at the current speed and needs to trigger static correction. It can be calibrated by bench testing. For example, the third threshold can be 90% of the maximum output current of FOC at the current speed, 6 amps, etc., calibrated by bench testing.

[0185] Preset static correction parameters refer to the correction ratio or correction value that is pre-calibrated through bench testing and stored in the controller memory to reduce the initial target current that exceeds the third threshold. For example, preset static correction parameters can be reduced to 80% of the original value, or a fixed reduction of 2 amperes.

[0186] Based on preset static correction parameters, the initial target current is corrected to obtain the target current command output by the FOC speed loop. One method is to multiply the initial target current by a preset proportional reduction coefficient. For example, if the preset static correction parameter is a proportional coefficient of 0.8, reducing the current to 80% of its original value, and the third threshold is calibrated to 9 amps, the control circuit reads the initial target current output by the FOC speed loop proportional-integral regulator in the current control cycle as 9.5 amps. Since 9.5 amps is greater than the third threshold of 9 amps, static correction is triggered. The corrected target current command = 9.5 * 0.8 = 7.6 amps, and this 7.6 amps is used as the final target current command output by the FOC speed loop and sent to the subsequent dynamic correction stage. Alternatively, the initial target current can be subtracted by a preset fixed reduction amount. For example, if the preset static correction parameter is a fixed reduction amount of 2 amps, and the third threshold is 9 amps, the initial target current is 9.5 amps, which is greater than 9 amps. The corrected target current command = 9.5 - 2 = 7.5 amps, but it must be ensured that the reduced target current command is not lower than the minimum sustaining current value at the current speed.

[0187] By correcting the initial target current based on preset static correction parameters before generating the first negative correction amount and when the initial target current is greater than the third threshold, the target current command output by the field-oriented control (FOC) speed loop is obtained. Thus, before the dynamic prediction correction takes effect, the initial target current is limited to a safe range through static protection correction, thereby reducing the risk of overshoot from the source and providing a reasonable target current reference for subsequent dynamic correction.

[0188] In one possible implementation, the fuel pump control method further includes: S200b, if the rate of change of the actual rotational speed is less than or equal to the first threshold and / or the rate of change of the bus current is less than the fourth threshold, and this state is maintained for more than the preset steady-state determination time threshold, the fuel pump motor is determined to have entered steady-state operation.

[0189] It is understandable that the rate of change of bus current refers to the amount of change of bus current per unit time, reflecting the degree of fluctuation of motor load current. The smaller the value, the more stable the current tends to be.

[0190] The fourth threshold refers to a preset threshold value for the rate of change of bus current, which is used to determine whether the fluctuation of bus current has become small enough to be considered as the system has entered a state of load stability. For example, the fourth threshold can be calibrated on the bench to 0.1 amperes per millisecond, 0.2 amperes per millisecond, etc.

[0191] The preset steady-state determination time threshold refers to the shortest time that the preset steady-state condition needs to be continuously met. It is used to avoid misjudgment of the state due to instantaneous fulfillment of the condition. For example, the steady-state determination time threshold can be set to 100 milliseconds, 50 milliseconds, etc.

[0192] The fuel pump motor is determined to enter steady-state operation when the rate of change of actual speed is less than or equal to the first threshold and / or the rate of change of bus current is less than the fourth threshold and the state is maintained for more than the preset steady-state determination time threshold. This allows for a comprehensive determination of steady-state signals from both the speed and current dimensions, and the addition of a time duration condition to avoid frequent misjudgments of the state under critical operating conditions. This provides a reliable admission criterion for triggering the steady-state self-learning update channel in the future.

[0193] S200c obtains the steady-state compensation current value output by the proportional-integral (PI) regulator of the FOC speed loop under steady-state conditions; the steady-state compensation current value is the non-zero compensation current value output by the PI regulator to maintain the target speed during steady-state operation.

[0194] It can be understood that a proportional-integral (PI) controller is a linear controller consisting of a proportional element and an integral element connected in parallel. The proportional element outputs a control quantity proportional to the current speed deviation instantaneously, while the integral element accumulates historical deviations to eliminate steady-state residual error. The two are then superimposed to form the target current command. For example, when the speed deviation is 200 rpm, a proportional element with a proportional coefficient of 0.05 amps per rpm outputs 10 amps, and the integral element accumulates an output of 2 amps, the total output of the PI controller is 12 amps, etc.

[0195] The steady-state compensation current value refers to the non-zero current value output by the proportional-integral (PI) regulator during steady-state operation to maintain the target speed. This compensation amount reflects the deviation between the system feedforward base value and the actual steady-state load.

[0196] The steady-state compensation current value of the proportional-integral (PI) regulator in the FOC speed loop can be obtained by the control circuit directly accessing the integral output variable of the PI regulator in the current control cycle, which is calculated and stored in RAM (random access memory), after determining that the fuel pump motor has entered steady-state operation. Alternatively, the difference can be obtained by subtracting the feedforward current command from the total output of the PI regulator. For example, if the base value of the feedforward current is 2.5 amps when the current speed is 3500 rpm and the total output of the PI regulator is 2.8 amps, then the steady-state compensation current value is 2.8 - 2.5 = 0.3 amps.

[0197] By acquiring the steady-state compensation current value output by the proportional-integral (PI) regulator of the field-oriented control (FOC) speed loop in steady state, the non-zero compensation amount output by the PI regulator to maintain the target speed during steady-state operation is used as an error indication signal reflecting the accuracy of the feedforward base value. This provides a driving input for subsequent judgment on whether the absolute value of the compensation amount exceeds the fifth threshold and, based on this, whether to trigger the self-learning update of the steady-state feedforward current base value table.

[0198] S200d, when the absolute value of the steady-state compensation current is greater than the fifth threshold and continues to exceed the preset duration threshold, the pre-stored steady-state feedforward current base value table is updated based on the steady-state compensation current value to obtain the updated steady-state feedforward current base value table, and the updated base value is written to the non-volatile memory when the preset persistence condition is met; wherein, the steady-state feedforward current base value table is a pre-calibrated data table recording the reference feedforward current values ​​under different speed ranges, and the preset persistence condition includes at least one of the following: the controller is powered off, the cumulative running time exceeds the writing interval threshold, or a vehicle shutdown signal is detected.

[0199] It is understandable that the fifth threshold refers to the preset absolute value threshold of the steady-state compensation current, which is used to determine whether the deviation of the feedforward base value has become so large that it is necessary to trigger self-learning update. For example, the fifth threshold can be calibrated on the bench to be 0.05 amperes, 0.1 amperes, etc.

[0200] The preset duration threshold refers to the shortest time that the state where the absolute value of the steady-state compensation current is greater than the fifth threshold needs to be continuously satisfied. This is used to avoid frequent triggering of self-learning updates due to transient fluctuations. For example, the duration threshold can be set to 500 milliseconds, 200 milliseconds, etc.

[0201] The steady-state feedforward current base value table is a lookup table data structure that is pre-calibrated through bench tests, indexed by speed range, and records the reference feedforward current values ​​required for steady-state operation of the fuel pump motor at each speed. It can be stored in the RAM of the control circuit for querying during FOC speed loop operation, and written to non-volatile memory when preset persistence conditions are met.

[0202] The reference feedforward current value refers to the reference value of the feedforward current for each speed range in the steady-state feedforward current base value table, representing the open-loop feedforward current required to maintain that speed under ideal conditions.

[0203] The updated base value refers to the new reference feedforward current value obtained by adding the self-learning correction increment to the original base value, which is closer to the actual steady-state load requirements of the current fuel pump motor.

[0204] Preset persistence conditions refer to the set of preset conditions that trigger the writing of the temporary base value table in RAM to non-volatile memory. These conditions are used to control the frequency of erasing and writing non-volatile memory within its lifespan while ensuring that the self-learning results are not lost when power is off.

[0205] Non-volatile memory refers to a storage medium that can retain stored data even after the controller is powered off. For example, non-volatile memory can be EEPROM, Flash memory, etc.

[0206] Controller power-off refers to the event that the control circuit is about to enter a power-off state after receiving an ignition switch off signal or a sleep command from the communication bus.

[0207] Cumulative running time refers to the total steady-state running time accumulated since the controller is powered on, and is used to determine whether the preset write interval threshold has been reached to trigger a non-volatile memory persistent write.

[0208] The write interval threshold refers to the minimum cumulative runtime required between two consecutive persistent writes to non-volatile memory. For example, the write interval threshold can be set to 100 hours, 50 hours, etc.

[0209] The vehicle shutdown signal refers to the indication signal sent by the engine control unit to the fuel pump controller via the LIN bus or hardwired, indicating that the engine has been shut down.

[0210] The steady-state feedforward current base value table can be updated by processing the steady-state compensation current value through a preset self-learning proportional-integral (PI) regulator to obtain the self-learning correction increment, and then adding it to the current base value. For example, the control circuit inputs a steady-state compensation current value of 0.3 amps into a preset self-learning PI regulator. This regulator is independent of the proportional-integral (PI) regulator of the FOC speed loop, with a proportional coefficient of 0.01 and an integral coefficient of 0.001. In each self-learning update cycle (e.g., once every 1 second), the proportional loop output = 0.01 * 0.3 = 0.003 amps, and the integral loop accumulates the integral of all historical steady-state compensation current values. During the first update, the integral loop output = 0.001 * 0.3 = 0.003 amps. = 0.0003 Amperes, self-learning correction increment = 0.003 + 0.0003 = 0.0033 Amperes. Assuming the current speed is 3500 rpm, the current base value in the steady-state feedforward current base value table corresponding to this speed range is 2.5 Amperes. Then the updated base value = 2.5 + 0.0033 = 2.5033 Amperes. This is written into the temporary base value table in RAM, and the above process is repeated in each subsequent self-learning update cycle. The steady-state compensation current value gradually decreases as the base value approaches the actual load, and the self-learning correction increment also decreases accordingly until the steady-state compensation current value converges to within the fifth threshold. Alternatively, the steady-state compensation current value can be directly converted according to a fixed ratio and added to the current base value, etc.

[0211] When the absolute value of the steady-state compensation current exceeds the fifth threshold and continues to exceed the preset duration threshold, the pre-stored steady-state feedforward current base value table is updated based on the steady-state compensation current value. The updated steady-state feedforward current base value table is obtained, and the updated base value is written to non-volatile memory when the preset persistence condition is met. This allows the feedforward base value table to be progressively self-learned and corrected during steady-state operation based on the compensation residual error of the PI regulator, so as to automatically adapt to the changes in mechanical characteristics caused by long-term operation. The erase and write frequency of the non-volatile memory is controlled within the automotive-grade lifespan range through a secondary storage mechanism, ensuring long-term reliability.

[0212] In one possible implementation, S200d updates the pre-stored steady-state feedforward current base value table based on the steady-state compensation current value, including: S200d1 processes the steady-state compensation current value through a preset self-learning proportional-integral (PI) regulator to obtain a self-learning correction increment. The self-learning PI regulator is independent of the PI regulator of the FOC speed loop. The self-learning correction increment is a correction value used to update the base value table after smoothing the steady-state compensation current value through the self-learning PI regulator.

[0213] It is understandable that the preset self-learning proportional-integral (PI) controller refers to a second PI controller, independent of the field-oriented control (FOC) speed loop PI controller, specifically used for the self-learning update channel. Its input is the steady-state compensation current value, and its output is the self-learning correction increment. The PI parameters of the two controllers are not coupled to avoid mutual interference between the parameters of the two controllers, thereby reducing the impact of mutual interference between the parameters of the two controllers on the stability of the control system. For example, the proportional coefficient of the self-learning PI controller can be 0.01 (less than the proportional coefficient of the FOC speed loop PI controller) to reduce the update sensitivity, and the integral coefficient of the self-learning PI controller can be 0.001 (less than the integral coefficient of the FOC speed loop PI controller 0.002) to ensure that the correction increment is smooth and gradual.

[0214] The self-learning correction increment refers to the incremental correction value output by the preset self-learning proportional-integral regulator after processing the steady-state compensation current value. This value is used to update the current base value of the corresponding speed range in the steady-state feedforward current base value table. Its amplitude changes slowly after being smoothed by the PI regulator, gradually increasing or decreasing. For example, the self-learning correction increment can be 0.0033 Amperes, 0.0011 Amperes, etc.

[0215] Smoothing refers to a filtering method that limits the rate of change of the self-learning correction increment through proportional-integral operations to avoid sudden shocks in the correction value. The proportional element provides an instantaneous response proportional to the current steady-state compensation current value, and the integral element accumulates historical steady-state compensation current values ​​so that the correction increment gradually approaches the correction amount corresponding to the steady-state compensation current value approaching zero.

[0216] The self-learning correction increment can be obtained by using an independent self-learning PI controller to synchronously perform proportional and integral operations. The control circuit reads the current steady-state compensation current value in each self-learning update cycle and uses it as the input to the self-learning PI controller. The proportional coefficient of the self-learning PI controller is 0.01, the integral coefficient is 0.001, the self-learning update cycle is 1 second, and the proportional loop output is 0.01 * 0.3 = 0.003 amps. After the first update, the cumulative value of the integral loop is 0 + 0.3 * 1 = 0.3 amps multiplied by the second, and the integral loop output is 0.001 * 0.3 = 0.0003 amps. The self-learning correction increment is 0.003 + 0.0003 = 0.0033 amps. In the second update, assuming the steady-state compensation current remains at 0.3 amps, the cumulative value of the integral element is 0.3 + 0.3 * 1 = 0.6 amps multiplied by seconds, the integral element output is 0.001 * 0.6 = 0.0006 amps, the proportional element output remains at 0.003 amps, and the self-learning correction increment is 0.003 + 0.0006 = 0.0036 amps. As the base value table gradually approaches the actual load demand, the steady-state compensation current will gradually decrease, and the growth rates of both the proportional element output and the integral element cumulative value will decrease accordingly. Finally, when the steady-state compensation current converges below the fifth threshold, the self-learning correction increment... The steady-state compensation current value can be approximated to zero. Alternatively, a step-by-step approximation with a fixed step size can be used. When the absolute value of the steady-state compensation current is greater than the fifth threshold, the control circuit directly uses a preset fixed step size (e.g., 0.005 amperes) as the self-learning correction increment. The direction of the increment is determined by the sign of the steady-state compensation current value. When the steady-state compensation current value is positive (i.e., the feedforward base value is too low), the self-learning correction increment is +0.005 amperes, and the current base value is added to this increment. When the steady-state compensation current value is negative (i.e., the feedforward base value is too high), the self-learning correction increment is -0.005 amperes, and the current base value is subtracted from this increment, and so on.

[0217] By inputting the steady-state compensation current value into a preset self-learning proportional-integral regulator for proportional-integral operation to obtain the self-learning correction increment, the smooth and gradual characteristics of the self-learning channel are ensured through independent PI regulator parameter decoupling while using the PI regulator to compensate for the residual error to drive the self-learning update of the base value table, and the normal control dynamics of the FOC speed loop are not affected.

[0218] S200d2 adds the self-learning correction increment to the current base value corresponding to the current speed range in the steady-state feedforward current base value table to obtain the updated base value.

[0219] It can be understood that the current base value refers to the reference feedforward current value corresponding to the current speed range in the steady-state feedforward current base value table. It is the old base value before the self-learning update. For example, if the control circuit reads the current base value of 2.5 Amperes in the 3000 to 3500 rpm speed range corresponding to the current speed of 3500 rpm in the steady-state feedforward current base value table, and the self-learning correction increment calculated by the self-learning proportional-integral regulator in this update cycle is 0.0033 Amperes, then the updated base value = 2.5 + 0.0033 = 2.5033 Amperes.

[0220] By adding the self-learning correction increment to the current base value corresponding to the current speed range in the steady-state feedforward current base value table, the updated base value is obtained. This allows the correction increment, after being smoothed by the self-learning proportional-integral regulator, to be gradually superimposed on the original base value according to the preset update step size. This gradually approximates the actual steady-state load requirements of the current fuel pump motor and avoids control disturbances caused by sudden changes in the base value.

[0221] S200e, in subsequent steady-state operation, outputs feedforward current commands based on the updated steady-state feedforward current base value table.

[0222] It is understandable that subsequent steady-state operation refers to the period after the fuel pump motor re-enters steady-state operation after the steady-state feedforward current base value table has completed one or more self-learning updates. For example, the vehicle re-enters steady-state operation during the subsequent constant speed cruising phase of the current driving cycle, or the vehicle enters steady-state operation after warming up and entering constant speed driving after the next start.

[0223] The feedforward current command refers to the reference feedforward current value corresponding to the current speed obtained from the steady-state feedforward current base value table. It is directly superimposed on the output of the proportional-integral (PI) regulator of the field-oriented control (FOC) speed loop as an open-loop compensation quantity to reduce the correction burden of the PI regulator.

[0224] The method for determining whether the fuel pump motor has entered subsequent steady-state operation can be the same three-condition joint determination as the first entry into steady-state operation. For example, the conditions for determining entry into steady-state operation are the same as those for the first time, that is, the fuel pump motor is determined to enter subsequent steady-state operation when the rate of change of the actual speed is less than or equal to the first threshold and / or the rate of change of the bus current is less than the fourth threshold and the state is maintained for more than the preset steady-state determination time threshold.

[0225] By outputting feedforward current commands based on the updated steady-state feedforward current base value table during subsequent steady-state operation, the old base value is replaced by a base value that is closer to the actual steady-state load requirement after self-learning correction, thereby reducing the steady-state compensation burden of the FOC speed loop proportional-integral PI regulator and reducing the additional power loss and steady-state pressure deviation caused by feedforward base value deviation.

[0226] In one possible implementation, the fuel pump control method further includes: After updating the steady-state feedforward current base value table, when the S200e1 re-enters steady-state operation within the same speed range, the FOC speed loop directly reads the corresponding base value from the updated steady-state feedforward current base value table as the feedforward current output.

[0227] It is understandable that after updating the steady-state feedforward current base value table, when the system re-enters the same speed range for steady-state operation, the field-oriented control (FOC) speed loop directly reads the corresponding base value from the updated steady-state feedforward current base value table as the feedforward current output. This allows the new base value, which is closer to the actual steady-state load requirements of the fuel pump motor after self-learning correction, to replace the old base value calibrated on the test bench for speed loop feedforward compensation. This reduces the steady-state compensation current value that the proportional-integral (PI) regulator needs to output and reduces the additional energy consumption caused by feedforward deviation.

[0228] S200e2, when the steady-state compensation current value of the PI regulator converges to less than or equal to the fifth threshold, the self-learning update of the steady-state feedforward current base value table is stopped.

[0229] It is understandable that when the steady-state compensation current value of the PI controller converges to less than or equal to the fifth threshold, the feedforward base value has been corrected to a level that reduces the compensation requirement of the PI controller to an acceptable range. Continuing to update may cause the following problems: First, current sampling noise (such as ADC quantization noise or sensor background noise of about ±0.02 amps) will cause the steady-state compensation current value to fluctuate randomly around zero. If self-learning updates continue, the base value will drift up and down irregularly with the noise, causing the control parameters to oscillate back and forth around the convergence point instead of converging stably. Second, excessive self-learning correction may cause the base value to exceed the optimal value and enter an under-compensated or over-compensated state, which will increase the compensation burden of the PI controller.

[0230] By stopping the self-learning update of the steady-state feedforward current base value table when the steady-state compensation current value of the proportional-integral PI regulator converges to less than or equal to the fifth threshold, the update is terminated after the self-learning correction makes the feedforward base value sufficiently close to the actual steady-state load requirement. This avoids control oscillations caused by noise or residual small deviations causing the base value to be repeatedly adjusted back and forth near the convergence point, and achieves coordinated steady-state balance between the feedforward channel and the feedback channel.

[0231] The control circuit of the intelligent fuel pump controller provided in this application embodiment can serve as the execution subject to implement the steps in any of the above-described fuel pump control method embodiments. Those skilled in the art will understand that the above embodiments are merely examples of control circuits and do not constitute a limitation on the control circuit. It may include more or fewer components, or combine certain components, or different components, such as input / output devices, network access devices, buses, etc.

[0232] The above description is merely a preferred embodiment of this application and is not intended to limit this application. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of this application should be included within the protection scope of this application.

Claims

1. A fuel pump intelligent controller, characterized in that, It includes a control circuit, which is connected to the engine ECU, a Hall position sensor installed inside the fuel pump motor, and a Hall current sensor connected in series with the DC bus. The control circuit is used to receive the target speed or target pressure command output by the engine ECU, the rotor position signal output by the Hall position sensor, and the bus current sampling analog voltage output by the Hall current sensor. The intelligent fuel pump controller also includes: A gate drive circuit, connected to the control circuit, is used to receive six PWM signals output by the control circuit, amplify them according to a specified duty cycle and timing, and output corresponding gate drive signals. as well as The three-phase inverter power circuit is connected to the vehicle 12-volt battery and the gate drive circuit. It is used to receive the gate drive signal output by the gate drive circuit, and to invert the DC voltage input from the vehicle 12-volt battery through the DC bus into a three-phase AC voltage with adjustable amplitude and frequency, and then output it to the three-phase winding of the fuel pump motor to drive the motor rotor to rotate and drive the impeller to pump fuel.

2. The intelligent fuel pump controller as described in claim 1, characterized in that, The gate driving circuit includes: The gate driver chip U1 has pins 2, 3, 4, 5, 6, and 7 connected to the control circuit to receive the six PWM signals. Pins 14, 15, 16, 19, 23, and 27 of the gate driver chip U1 are connected to the three-phase inverter power circuit to output the gate drive signal. Pins 13, 18, 22, and 26 of the gate driver chip U1 are reference ground pins connected to the corresponding potential nodes of the three-phase inverter power circuit. The power filtering module has its input end connected to a 15V operating power supply and its output end connected to pin 1 of the gate driver chip U1, and is used to provide a stable 15V operating power supply for the gate driver chip U1 and its surrounding circuits. The bootstrap boost module has its input terminal connected to the 15V working power supply and its output terminal connected to the gate driver chip U1, and is used to provide floating power for the U / V / W three-phase upper bridge arm drive respectively. The overcurrent feedback module has its input terminal connected to the bus current sampling voltage input point and its output terminal connected to pin 9 of the gate driver chip U1.

3. The intelligent fuel pump controller as described in claim 2, characterized in that, The bootstrap boost module includes: resistor R1, diode D1, diode D2, diode D3, capacitor C1, capacitor C2, and capacitor C3; One end of the resistor R1 is connected to the 15V operating power supply and pin 1 of the gate driver chip U1, and the other end is simultaneously connected to the anodes of diodes D1, D2, and D3. The cathode of the diode D3 is connected to pin 28 of the gate driver chip U1 and the anode of the capacitor C1, and the cathode of the capacitor C1 is connected to pin 26 of the gate driver chip U1 and the three-phase inverter power circuit. The cathode of diode D2 is connected to pin 24 of the gate driver chip U1 and the anode of capacitor C2, and the cathode of capacitor C2 is connected to pin 22 of the gate driver chip U1 and the three-phase inverter power circuit. The cathode of diode D1 is connected to pin 20 of the gate driver chip U1 and the anode of capacitor C3, and the cathode of capacitor C3 is connected to pin 18 of the gate driver chip U1 and the three-phase inverter power circuit.

4. The intelligent fuel pump controller as described in claim 2, characterized in that, The overcurrent feedback module includes: resistor R5, resistor R6, and capacitor C8; One end of resistor R5 is connected to the bus current sampling voltage input point, and the other end is connected to one end of resistor R6. The other end of resistor R6 is connected to power ground PGND. Pin 9 of the gate driver chip U1 is connected to the common node between resistor R5 and resistor R6; One end of the capacitor C8 is connected to the common node between the resistor R6 and the power ground PGND, and the other end is connected to the common node between the resistor R5 and the resistor R6.

5. The intelligent fuel pump controller as described in claim 1, characterized in that, The three-phase inverter power circuit includes: The input protection module has its input terminal connected to an external DC power supply and is used to output the DC bus voltage after fuse protection. The DC bus voltage includes a positive DC bus and a negative DC bus. The three-phase half-bridge inverter drive module has its input terminal connected to the DC positive bus and the gate drive circuit. It is used to receive the gate drive signal to output the corresponding three-phase AC voltage to the external load, and to output the lower bridge arm common node potential PV and the three-phase half-bridge midpoint potential. The reference ground interface module has its input terminal connected to the midpoint potential of the three-phase half-bridge and the common node potential PV of the lower bridge arm, and its output terminal connected to the gate drive circuit. The detection and conditioning module has its input terminal connected to the DC negative bus, the 5VD auxiliary power supply, and the 3.3VD auxiliary power supply, and its output terminal connected to the ADC sampling port of the control circuit, for outputting an analog sampling signal idcadc that is proportional to the bus current.

6. The intelligent fuel pump controller as described in claim 5, characterized in that, The detection and conditioning module includes: a current detection chip U2, resistors R19, R20, and R21, a capacitor C9, and a Schottky diode pair D11; The DC negative bus is connected to the primary current input terminal of the current detection chip U2, and the primary current output terminal of the current detection chip U2 is connected to power ground PGND. The 5VD auxiliary power supply is connected to pin 8 of the current detection chip U2, and pin 5 of the current detection chip U2 is connected to digital ground DGND. The current detection chip U2 has pins 7 and 6 shorted. The shorting point is connected to one end of resistor R19, one end of resistor R21, and one end of capacitor C9. The other end of resistor R21 is connected to digital ground DGND, and the other end of capacitor C9 is connected to digital ground DGND. The other end of resistor R19 leads out to the idcadc sampling port and is simultaneously connected to one end of resistor R20 and the common terminal of the Schottky diode and D11. The other end of resistor R20 is connected to digital ground DGND. The Schottky diode pair D11 is connected between the idcadc sampling port and the 3.3VD auxiliary power supply and the digital ground DGND.

7. A fuel pump control method, characterized in that, The method, applied to the fuel pump intelligent controller as described in any one of claims 1 to 6, comprises: Obtain the actual speed and bus current of the fuel pump motor at the current moment; wherein, the bus current is used to reflect the load state of the fuel pump motor, and the degree of deviation of the bus current from the steady-state load current base value reflects the deviation of the output torque of the fuel pump motor; If the rate of change of the actual rotational speed is greater than a first threshold and the bus current is less than the base value of the steady-state load current corresponding to the current rotational speed, it is determined that there is a risk of speed overshoot and a first negative correction amount is generated; wherein, the base value of the steady-state load current is the reference load current value required for the fuel pump motor to operate in steady state at the current rotational speed, and the first negative correction amount is a correction value used to reduce the target current command. The first negative correction amount is superimposed on the target current command output by the magnetic field orientation control (FOC) speed loop to obtain the first target current command; Based on the first target current command, the output torque of the fuel pump motor is controlled.

8. The fuel pump control method as described in claim 7, characterized in that, The generation of the first negative correction amount includes: Based on the actual rotational speed, a pre-stored rotational speed and basic correction amount mapping table is queried to obtain the basic correction amount corresponding to the actual rotational speed; wherein, the rotational speed and basic correction amount mapping table is a data table calibrated by bench testing to record the basic reduction amount corresponding to different rotational speed ranges. The dynamic correction component is determined based on the product of the actual rotational speed change rate and the preset gain coefficient; wherein the preset gain coefficient is a fixed value calibrated through bench testing. The sum of the basic correction amount and the dynamic correction component is determined as the first negative correction amount.

9. The fuel pump control method as described in claim 8, characterized in that, The fuel pump control method further includes: Obtain the steady-state load current base value corresponding to the current speed of the fuel pump motor; Calculate the current difference between the steady-state load current base value and the bus current; When the absolute value of the current difference is greater than the second threshold, the product of the current difference and the preset additional gain coefficient is used as an additional correction amount, which is added to the first negative correction amount to obtain the increased first negative correction amount; wherein, the preset additional gain coefficient is a fixed value calibrated by bench test.

10. The fuel pump control method as described in claim 7, characterized in that, The fuel pump control method further includes: Before generating the first negative correction amount, if the initial target current of the fuel pump motor is greater than the third threshold, the initial target current is corrected based on the preset static correction parameters to obtain the target current command output by the FOC speed loop.

Images