LLC synchronous rectification control method, controller, energy storage system and storage medium

Abstract

The application belongs to the technical field of LLC synchronous rectification control, and particularly relates to an LLC synchronous rectification control method, a controller, an energy storage system and a storage medium. The LLC synchronous rectification control method comprises the following steps: determining whether an LLC resonant converter is in an over-resonance state or an under-resonance state based on a current switching frequency and a first resonant frequency of the LLC resonant converter; calculating a flow interruption duration and a first conduction duration in a first working mode; if the flow interruption duration is greater than zero and less than half of a switching period, determining a rectification drive signal of a secondary rectification tube based on the flow interruption duration and the first conduction duration; otherwise, if the LLC resonant converter is in the over-resonance state, calculating an outward moving phase angle in a second working mode and determining the rectification drive signal based on the outward moving phase angle; if the LLC resonant converter is in the under-resonance state, calculating a second conduction duration in a third working mode and determining the rectification drive signal based on the second conduction duration. The method of the application realizes accurate synchronization of a synchronous rectification tube drive signal and a current waveform in a full working condition range.

Description

Technical Field

[0001] This application belongs to the field of LLC synchronous rectification control technology, specifically relating to an LLC synchronous rectification control method, controller, energy storage system and storage medium. Background Technology

[0002] LLC resonant converters have become the mainstream topology in medium and high power switching power supplies due to their ability to achieve zero-voltage turn-on (ZVS) of the primary-side switches over a wide load range. To further improve efficiency, especially in low-voltage, high-current output applications, secondary-side synchronous rectification (SR) technology is widely used.

[0003] Existing methods for obtaining synchronous rectification drive timing mainly include hardware detection and software lookup table methods. Hardware detection involves sampling the secondary voltage or current of the transformer at high frequency and generating the SR signal using a hardware comparator. This method demands extremely high accuracy and anti-interference capabilities from the sampling circuit and increases hardware cost and layout complexity. Software lookup table methods rely on offline simulation or experimental test data, determining the SR timing through fitting or table lookup. However, this method is greatly affected by hardware parameters, and the large sample size of SR data under different operating conditions can negatively impact SR accuracy if the test samples are insufficient.

[0004] Therefore, there is an urgent need for a synchronous rectification solution that requires no additional hardware testing, no extensive experimental calibration, can adapt to various operating conditions, and provides precise control. Summary of the Invention

[0005] The purpose of this application is to provide an LLC synchronous rectification control method, controller, energy storage system and storage medium, which aims to solve the problems of high hardware dependence or poor adaptability to all operating conditions in the existing synchronous rectification drive timing acquisition methods.

[0006] In a first aspect, embodiments of this application provide an LLC synchronous rectification control method, including: Based on the current switching frequency and the first resonant frequency of the LLC resonant converter, determine whether the LLC resonant converter is in an over-resonance state or an under-resonance state. Calculate the current-off duration and the first conduction duration of the LLC resonant converter in the first operating mode. The first operating mode is the operating mode in which the secondary rectifier is in intermittent conduction mode for half a switching cycle. The current-off duration is the duration from the start time to the first time. The start time is the rising edge of the primary drive signal of the primary switch. The first time is the moment when the primary side begins to transfer power to the secondary side. The first conduction duration is the duration during which the secondary rectifier has current in the switching cycle in the first operating mode. If the interruption duration meets the first condition, the rectification drive signal of the secondary rectifier is determined based on the interruption duration and the first conduction duration. If the interruption duration satisfies the second condition and is in the over-resonance state, calculate the outward phase shift angle of the rectifier drive signal relative to the primary drive signal in the second operating mode of the LLC resonant converter. The second operating mode is that the secondary rectifier is in continuous conduction mode within half a switching cycle. If the interruption duration satisfies the second condition and is in the under-resonance state, calculate the second conduction duration of the LLC resonant converter in the third operating mode, and determine the rectifier drive signal based on the second conduction duration. The third operating mode is an operating mode in which the secondary rectifier is in an intermittent conduction mode for half a switching cycle and the primary side transmits power to the secondary side from the starting time. The second conduction duration is the duration for which the secondary rectifier has current in the switching cycle under the third operating mode. The first condition is that the interruption duration is greater than zero and less than half a switching cycle, and the second condition is that the interruption duration is less than or equal to zero, or that the interruption duration is greater than or equal to half a switching cycle.

[0007] Optionally, based on the current interruption duration and the first conduction duration, the rectification drive signal of the secondary rectifier is determined, including: The first percentage value is determined based on the product of the interruption duration and the current switching frequency; The second percentage value is determined by multiplying the sum of the interruption duration and the first conduction duration by the current switching frequency; The rising edge time of the rectification drive signal of the secondary rectifier tube within the switching cycle is determined based on the first percentage value, and the falling edge time of the rectification drive signal of the secondary rectifier tube within the switching cycle is determined based on the second percentage value.

[0008] Optionally, determining the rectification drive signal of the secondary rectifier based on the second conduction duration includes: The third percentage value is determined based on the product of the second conduction duration and the current switching frequency; The rising edge of the primary-side drive signal within the switching cycle is taken as the rising edge of the secondary-side rectifier drive signal within the switching cycle, and the falling edge of the secondary-side rectifier drive signal within the switching cycle is determined based on the third proportion value.

[0009] Optionally, calculating the disconnection duration and the first conduction duration of the LLC resonant converter in the first operating mode includes: Based on the first time-domain circuit equation of the LLC resonant converter in the first operating mode, the current interruption duration and the first conduction duration are calculated. The first time-domain circuit equation includes a first relation and a second relation obtained by performing second-order circuit time-domain analysis on the first equivalent circuit. The first relation is the relationship between the interruption duration and the switching frequency, the first resonant frequency, the second resonant frequency, the output voltage gain, the quality factor, and the parameters of the resonant cavity. The second relation is the relationship between the first conduction duration and the switching frequency, the first resonant frequency, the second resonant frequency, the output voltage gain, the quality factor, and the parameters of the resonant cavity. Wherein, the first equivalent circuit is the equivalent circuit of the LLC resonant converter in the first operating mode, the quality factor is the quality factor of the LLC resonant converter, and the parameters of the resonant cavity include the inductance value of the resonant inductor and the inductance value of the magnetizing inductor of the LLC resonant converter.

[0010] Optionally, the method for establishing the first time-domain circuit equation includes: Based on the second-order circuit time-domain analysis of the first equivalent circuit, a set of multiple second-order circuit time-domain equations concerning the inductor and capacitor at three time points are obtained. Based on the multiple sets of time-domain equations of the second-order circuit and the Mittag-Leffler approximation, the first relation and the second relation are obtained; Wherein, the inductor is the resonant inductor of the resonant cavity of the LLC resonant converter, the capacitor is the resonant capacitor of the resonant cavity, the three times are the start time, the first time and the second time, and the second time is the time when the primary side stops transmitting power to the secondary side.

[0011] Optional, The first relation is:

[0012] The second relation is:

[0013] in, , , t0 is the interruption duration, t1 is the first conduction duration, Lr is the inductance value of the resonant inductor, Lm is the inductance value of the magnetizing inductor, and fn is the normalized frequency. fs is the switching frequency, and fr is the first resonant frequency. fm is the second resonant frequency. Q is the quality factor. Ntr is the transformer turns ratio, Cr is the resonant capacitor capacitance, Ro is the load resistance, and Gv is the output voltage gain. Vi is the input voltage, and Vo is the output voltage.

[0014] Optionally, calculating the outward phase shift angle of the rectified drive signal relative to the primary-side drive signal in the second operating mode of the LLC resonant converter includes: The external phase shift angle is determined based on the phase shift angle relationship of the LLC resonant converter in the second operating mode. The phase shift angle relationship is the relationship between the external phase shift angle and the parameters of the resonant cavity, and the parameters of the resonant cavity include the inductance value of the resonant inductor and the inductance value of the magnetizing inductor of the LLC resonant converter.

[0015] Optionally, the method for establishing the phase shift angle relationship includes: Based on the phasor-frequency domain model, the current frequency domain equations for the resonant inductor, resonant capacitor, and secondary current are obtained. Based on the fact that the secondary current value is zero at a preset time in the second working mode, a frequency domain equation for the outer phase shift angle is obtained; Perform an inverse Fourier series transform on the frequency domain equation to obtain the phase shift angle relationship; Wherein, the preset time is the falling edge time of the rectifier drive signal, and the preset time is (0.5×θ) / fs, where θ is the outward phase shift angle and fs is the switching frequency.

[0016] Optionally, calculating the second on-time of the LLC resonant converter in the third operating mode includes: The second conduction duration is determined based on the second time-domain circuit equation of the LLC resonant converter in the third operating mode. The second time-domain circuit equation includes a third relation obtained by performing second-order circuit time-domain analysis on the second equivalent circuit. The third relation is the relationship between the second conduction duration and the switching frequency, the first resonant frequency, the output voltage gain, the quality factor, and the parameters of the resonant cavity. Wherein, the second equivalent circuit is the equivalent circuit of the LLC resonant converter in the third operating mode, the quality factor is the quality factor of the LLC resonant converter, and the parameters of the resonant cavity include the inductance value of the resonant inductor and the inductance value of the magnetizing inductor of the LLC resonant converter.

[0017] Optionally, the method for establishing the second time-domain circuit equation includes: Based on the second equivalent circuit time-domain analysis, a set of multiple second-order circuit time-domain equations concerning the inductor and capacitor at two time points are obtained. Based on the multiple sets of time-domain equations of the second-order circuit and the Mittag-Leffler approximation, the third relation is obtained; Wherein, the inductor is the resonant inductor of the resonant cavity of the LLC resonant converter, the capacitor is the resonant capacitor of the resonant cavity, the two moments are the start moment and the second moment, and the second moment is the moment when the primary side stops transmitting power to the secondary side.

[0018] Optionally, the third relation is:

[0019] in, t1 is the second conduction duration, Lr is the inductance value of the resonant inductor, Lm is the inductance value of the magnetizing inductor, and fn is the normalized frequency. fs is the switching frequency, and fr is the first resonant frequency. Q is the quality factor. Ntr is the transformer turns ratio, Cr is the resonant capacitor capacitance, Ro is the load resistance, and Gv is the output voltage gain. Vi is the input voltage, and Vo is the output voltage.

[0020] Secondly, embodiments of this application provide a controller, the controller including at least one processor and a memory communicatively connected to the at least one processor, the memory storing instructions executable by the at least one processor, the instructions being executed by the at least one processor to enable the at least one processor to perform the method described above.

[0021] Thirdly, embodiments of this application provide an energy storage system, including: An LLC resonant converter includes a primary-side bridge circuit, a resonant cavity, a transformer, and a secondary-side bridge circuit connected in sequence. The primary-side bridge circuit includes a first bridge arm and a second bridge arm connected in parallel. The secondary-side bridge circuit includes a third bridge arm and a fourth bridge arm connected in parallel. The first bridge arm and the second bridge arm each include two primary-side switching transistors connected in series. The third bridge arm and the fourth bridge arm each include two secondary-side rectifier transistors connected in series. In the controller described above, each of the primary-side switching transistors and each of the secondary-side rectifier transistors are connected to the controller.

[0022] Fourthly, embodiments of this application provide a computer storage medium storing instructions or programs that, when executed by at least one processor, cause the at least one processor to perform the method described above.

[0023] This application provides an LLC synchronous rectification control method with the following advantages: The resonant state is determined by comparing the current switching frequency with the first resonant frequency, and the interruption duration and first conduction duration of the secondary rectifier diode in the LLC resonant converter in the first operating mode are calculated. Based on the resonant state and interruption duration, the operation of the LLC resonant converter is divided into four operating conditions, and different synchronous rectification control methods are applied to each condition. This ultimately achieves precise synchronization of the rectifier drive signal and current waveform across the entire operating range, significantly reducing rectification losses and improving the conversion efficiency and dynamic performance of the LLC resonant converter. Furthermore, this method eliminates the need for additional current / voltage sampling circuits, comparators, or other hardware, and avoids the need to traverse the entire frequency and load range through experiments or simulations to obtain data tables. This not only reduces costs but also simplifies engineering implementation and increases reliability. Attached Figure Description

[0024] Figure 1 This is a schematic diagram of the circuit topology of the LLC resonant converter provided in the embodiments of this application; Figure 2 This is a schematic diagram of the controller structure provided in the embodiments of this application; Figure 3 This is a flowchart of the LLC synchronous rectification control method provided in the embodiments of this application; Figure 4a This is the equivalent circuit diagram of the LLC resonant converter under over-resonance heavy load condition provided in the embodiments of this application; Figure 4b This is a schematic diagram of the working waveform of the LLC resonant converter under the equivalent circuit diagram corresponding to the over-resonance heavy load condition provided in the embodiments of this application; Figure 5a This is the first equivalent circuit diagram of the LLC resonant converter provided in this application embodiment under under-resonance light load condition or over-resonance light load condition; Figure 5b This is a schematic diagram of the current waveform of the secondary rectifier diode under the first equivalent circuit diagram provided in the embodiments of this application; Figure 6a This is the second equivalent circuit diagram of the LLC resonant converter provided in the embodiments of this application under the under-resonance heavy load condition; Figure 6b The current waveform diagram of the secondary rectifier diode under the second equivalent circuit diagram provided in the embodiments of this application; Figure 7 This is a schematic diagram of the polynomial approximation effect of the cotangent function cot(x) provided in the embodiments of this application; Figure 8 This is a schematic diagram of the simulation waveform of the over-resonance condition provided in the embodiments of this application; Figure 9This is a schematic diagram of the simulation waveform of the underresonant light load condition provided in the embodiments of this application; Figure 10 This is a schematic diagram of the simulation waveform of the underresonant heavy load condition provided in the embodiment of this application. Detailed Implementation

[0025] To make the objectives, technical solutions, and advantages of the embodiments of this application clearer, the technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of this application, not all embodiments. Based on the embodiments of this application, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of this application.

[0026] Please refer to Figure 1 , Figure 1 This is a schematic diagram of the circuit topology of the LLC resonant converter provided in the embodiments of this application. Figure 1 As shown, the LLC resonant converter includes a primary bridge circuit, a resonant cavity, a transformer, and a secondary bridge circuit connected in sequence.

[0027] In this circuit, the input terminal of the primary-side bridge circuit is connected to the input power supply that provides the input voltage Vi, and the output terminal of the primary-side bridge circuit is connected to the input terminal of the resonant cavity. Specifically, the primary-side bridge circuit is a full-bridge circuit, including primary-side switches Q1, Q2, Q3, and Q4. Primary-side switches Q1 and Q2 are connected in series to form the left primary-side bridge arm, and primary-side switches Q4 and Q3 are connected in series to form the right primary-side bridge arm. The left and right primary-side bridge arms are connected in parallel and then bridging the two ends of the input power supply.

[0028] The resonant cavity is composed of a resonant inductor Lr and a resonant capacitor Cr connected in series. The first end of the resonant capacitor Cr is connected to the midpoint A of the left primary bridge arm, and the first end of the resonant inductor Lr is connected to the midpoint B of the right primary bridge arm. The midpoint A of the left primary bridge arm is the connection point of the primary switching transistors Q1 and Q2, and the midpoint B of the right primary bridge arm is the connection point of the primary switching transistors Q3 and Q4.

[0029] Furthermore, the output terminal of the resonant cavity is connected to the primary winding of the transformer, and the secondary winding of the transformer is connected to the input terminal of the secondary bridge circuit. The output terminal of the secondary bridge circuit is connected in parallel with the filter capacitor and the load resistor Ro, and the output voltage is labeled Vo. Specifically, the secondary bridge circuit includes secondary rectifier diodes Q5, Q6, Q7, and Q8. Among them, secondary rectifier diodes Q5 and Q6 are connected in series to form the left arm of the secondary bridge, and secondary rectifier diodes Q7 and Q8 are connected in series to form the right arm of the secondary bridge. The left and right arms of the secondary bridge are connected in parallel. The second terminal of the resonant inductor Lr and the second terminal of the resonant capacitor Cr are respectively connected to the two ends of the primary winding of the transformer. The two ends of the secondary winding of the transformer are respectively connected to the midpoint C of the left arm of the secondary bridge and the midpoint D of the right arm of the secondary bridge. The midpoint C of the left arm is the connection point of the secondary rectifier diodes Q5 and Q6, and the midpoint D of the right arm is the connection point of the secondary rectifier diodes Q7 and Q8.

[0030] like Figure 1 As shown, in the primary-side bridge circuit, the driving signal g1 is a pulse signal that controls the switching of primary-side switching transistors Q1 and Q3, and the driving signal g2 is a pulse signal that controls the switching of primary-side switching transistors Q2 and Q4. The driving signals g1 and g2 are complementary; and the driving signals g1 and g2 can be referred to as primary-side driving signals.

[0031] like Figure 1 As shown, in the secondary bridge circuit, the rectifier drive signal g3 is the drive signal that controls the switching of the secondary rectifier diodes Q5 and Q7; the rectifier drive signal g4 is the drive signal that controls the switching of the secondary rectifier diodes Q6 and Q8. The rectifier drive signals g3 and g4 are complementary.

[0032] As an optional implementation, the LLC resonant converter further includes a controller, which is connected to the primary-side switches Q1, Q2, Q3, Q4 and the secondary-side rectifiers Q5, Q6, Q7, Q8 respectively, and is used to execute the LLC synchronous rectification control method described in any of the following embodiments to control the on and off of the secondary-side rectifiers.

[0033] Please refer to Figure 2 , Figure 2 This is a schematic diagram of the controller provided in an embodiment of this application. Figure 2 As shown, the controller 20 includes at least one processor 201 and a memory 202. The memory 202 can be built into the controller 20 or external to the controller 20. The memory 202 can also be a remotely configured memory connected to the controller 20 via a network.

[0034] Memory 202, as a non-volatile computer-readable storage medium, can be used to store non-volatile software programs, non-volatile computer-executable programs, and modules. Memory 202 may include a program storage area and a data storage area, wherein the program storage area may store the operating system and application programs required for at least one function; the data storage area may store data created based on the use of the terminal, etc. Furthermore, memory 202 may include high-speed random access memory and may also include non-volatile memory, such as at least one disk storage device, flash memory device, or other non-volatile solid-state storage device. In some embodiments, memory 202 may optionally include memory remotely located relative to processor 201, and these remote memories can be connected to the terminal via a network. Examples of such networks include, but are not limited to, the Internet, corporate intranets, local area networks, mobile communication networks, and combinations thereof.

[0035] The processor 201 performs various functions of the terminal and processes data by running or executing software programs and / or modules stored in the memory 202 and calling data stored in the memory 202, thereby performing overall monitoring of the terminal, such as implementing the LLC synchronous rectification control method described in any embodiment of this application.

[0036] Processor 201 can be one or more. Figure 2 The example provided uses a processor 201. The processor 201 and memory 202 can be connected via a bus or other means. The processor 201 may include a central processing unit (CPU), a digital signal processor (DSP), an application-specific integrated circuit (ASIC), a controller, a field-programmable gate array (FPGA) device, etc. The processor 201 can also be implemented as a combination of computing devices, such as a combination of a DSP and a microprocessor, multiple microprocessors, one or more microprocessors combined with a DSP core, or any other such configuration.

[0037] Through the deep coupling of the above hardware architecture and control strategy, the LLC resonant converter provided in this embodiment not only constructs a complete energy transmission path at the physical level, but also migrates the complex synchronous rectification timing generation task from the analog domain to the digital domain at the logical level. This fundamentally solves the technical pain points of high hardware dependence, high debugging difficulty and poor adaptability to all operating conditions in traditional solutions, and achieves accurate synchronization of the secondary rectifier tube drive signal and current waveform across the entire operating range.

[0038] Understandably, for Figure 1When the LLC resonant converter shown is subjected to synchronous rectification control, the rectification drive signals of secondary rectifier diodes Q5 and Q7 are the same, while the rectification drive signals of secondary rectifier diodes Q6 and Q8 are complementary to those of secondary rectifier diodes Q5 and Q7. Therefore, based on the rectification drive signal of any one of the secondary rectifier diodes, the rectification drive signals of the other three secondary rectifier diodes can be determined. For ease of understanding, this application describes the secondary rectifier diode as Q5 as an example when describing the LLC synchronous rectification control method.

[0039] Please refer to the reference. Figures 3 to 7 , Figure 3 This is a flowchart of the LLC synchronous rectification control method provided in the embodiments of this application, which can be applied to... Figure 1 The LLC resonant converter shown, and in Figure 2 Executed in the controller shown. Figure 4a This is the equivalent circuit diagram of the LLC resonant converter provided in this application embodiment under over-resonance heavy load condition. Figure 4b This is a schematic diagram of the working waveform of the LLC resonant converter under the equivalent circuit diagram corresponding to the over-resonance heavy load condition provided in the embodiments of this application. Figure 5a This is the first equivalent circuit diagram of the LLC resonant converter provided in this application embodiment, corresponding to the under-resonance light load condition and / or the over-resonance light load condition; Figure 5b This is a schematic diagram of the secondary rectifier and its current waveform under the first equivalent circuit diagram provided in the embodiments of this application; Figure 6a This is the second equivalent circuit diagram of the LLC resonant converter provided in the embodiments of this application under the under-resonance heavy load condition; Figure 6b The embodiments of this application provide a schematic diagram of the secondary rectifier diode and its current waveform under the second equivalent circuit diagram; Figure 7 This is a schematic diagram of the polynomial approximation effect of the cotangent function cot(x) provided in the embodiments of this application.

[0040] In this embodiment, the LLC synchronous rectification control method is illustrated using two primary-side switches on the same arm of a primary-side bridge circuit and two secondary-side rectifiers on the same arm of a secondary-side bridge circuit as examples; specifically, the LLC synchronous rectification control method is explained using the primary-side drive signal g1 of primary-side switch Q1, the primary-side drive signal g2 of primary-side switch Q2, the rectification drive signal of secondary-side rectifier Q5, and the rectification drive signal of secondary-side rectifier Q6. Figure 4b , Figure 5b and Figure 6b In this diagram, D1 and D2 represent secondary rectifier diodes Q5 and Q6, respectively. I(D1) represents the current flowing through secondary rectifier diode Q5, and I(D2) represents the current flowing through secondary rectifier diode Q6. This LLC synchronous rectification control method includes the following steps S301 and S305.

[0041] Step S301: Based on the current switching frequency and the first resonant frequency of the LLC resonant converter, determine whether the LLC resonant converter is in an over-resonance state or an under-resonance state.

[0042] In this embodiment, the current switching frequency fs is determined in real time by the controller based on closed-loop feedback, and the first resonant frequency fr ( The value of fs is determined by the inherent parameters of the resonant inductance Lr and resonant capacitance Cr in the resonant cavity. Specifically, when fs ≥ fr, the LLC resonant converter is in an over-resonant state; when fs < fr, the LLC resonant converter is in an under-resonant state.

[0043] Step S302: Calculate the interruption time and the first conduction time of the LLC resonant converter in the first operating mode. The first operating mode is the operating mode in which the secondary rectifier is in the intermittent conduction mode for half a switching cycle.

[0044] Wherein, the start time is the rising edge of the primary-side drive signal of the primary-side switch transistor (i.e., Figure 5b At time 0), the interruption duration is the time from the start time to the first time, where the first time is the moment when the primary side begins to transmit power to the secondary side (i.e., ...). Figure 5b At time t0), the first conduction duration is the duration during which current exists in the secondary rectifier diode during the switching cycle in the first operating mode (i.e., Figure 5b The duration t1 is the time from time t0 to time t0+t1. The specific calculation method will be described in detail in the following embodiments.

[0045] In the synchronous rectification control of LLC, the operation of the LLC resonant converter can be divided into four operating conditions based on the resonant state and load: over-resonance light load condition, over-resonance heavy load condition, under-resonance light load condition, and under-resonance heavy load condition. The resonant state can be determined by the magnitude of the current switching frequency relative to the first resonant frequency. However, the load is dynamic, and different load conditions will cause the secondary current flowing through the secondary rectifier diodes to exhibit different characteristics. Specifically, when the load is light, the secondary current exhibits a discontinuous conduction mode (DCM), and the primary side does not begin transferring power to the secondary side from the initial moment. Figure 5b As shown, the current-cutting phase includes the period from time 0 to time t0, the conduction phase from time t0 to time (t0+t1), and the cutoff phase from time (t0+t1) to time 1 / (2fs). When the load is heavily loaded and in an over-resonance state, the secondary current exhibits a continuous conduction mode (CCM) pattern, as shown below. Figure 4bAs shown, the secondary current conducts continuously during the switching cycle. When the load is heavily loaded and in an underresonant state, the secondary current also exhibits an intermittent conduction mode, and power is transferred from the primary side to the secondary side from the initial moment, as shown... Figure 6b As shown, the circuit includes the conduction phase from time t0 to time t1 and the cutoff phase from time t1 to time 1 / (2fs). Compared to the light load state, it lacks the current interruption phase from time 0 to time t0. Therefore, the current interruption duration can be used to identify whether the LLC resonant converter is currently in a light load state or a heavy load state.

[0046] Step S303: If the interruption duration meets the first condition, determine the rectification drive signal of the secondary rectifier tube based on the interruption duration and the first conduction duration.

[0047] The first condition is that the interruption duration is greater than zero and less than half a switching cycle. Based on the foregoing analysis, when the interruption duration is greater than zero and less than half a switching cycle, it indicates that the LLC resonant converter is in a light-load state. Under light-load conditions, regardless of whether the LLC resonant converter is in an under-resonance or over-resonance state, its corresponding equivalent circuit and the current waveform flowing through the secondary rectifier Q5 are the same. Therefore, for both light-load conditions, this application determines the rectification drive signal of the secondary rectifier based on the interruption duration and the first conduction duration calculated in the first operating mode.

[0048] Step S304: If the interruption duration meets the second condition and is in an over-resonance state, calculate the outward phase shift angle of the rectifier drive signal relative to the primary drive signal in the second operating mode of the LLC resonant converter. The second operating mode is the operating mode in which the secondary rectifier tube is in continuous conduction mode within half a switching cycle.

[0049] The second condition is that the interruption duration is less than or equal to zero, or the interruption duration is greater than or equal to half a switching cycle. Understandably, the first and second conditions are opposites of each other.

[0050] When the interruption duration is less than or equal to zero, or the interruption duration is greater than or equal to half a switching cycle, the LLC resonant converter is in a heavy-load state. Since the equivalent circuit and the current waveform flowing through the secondary rectifier are different when the LLC resonant converter is in an over-resonance state and an under-resonance state under heavy load, this application uses different methods to determine the rectification drive signal for the secondary rectifier for these two operating conditions under heavy load.

[0051] In the over-resonance heavy-load condition, the output voltage gain is less than 1, the secondary rectifier diodes are continuously conducting, and the magnetizing inductor is clamped by the output voltage. The LLC resonant converter circuit is equivalent to a series resonant dual active bridge circuit with frequency conversion and external phase shift control. Based on this equivalent circuit, the external phase shift angle of the rectifier drive signal relative to the primary drive signal in the second operating mode of the LLC resonant converter can be calculated, and then the rectifier drive signal of the secondary rectifier diodes can be determined based on the external phase shift angle.

[0052] Step S305: If the interruption duration meets the second condition and the circuit is in an under-resonance state, calculate the second conduction duration of the LLC resonant converter in the third operating mode, and determine the rectifier drive signal based on the second conduction duration. The third operating mode is the operating mode in which the secondary rectifier is in an intermittent conduction mode for half a switching cycle and the primary side transmits power to the secondary side from the start time. The second conduction duration is the duration of current in the secondary rectifier during the switching cycle in the third operating mode.

[0053] Under under-resonant heavy load conditions, the current flowing through the secondary rectifier is still in discontinuous conduction mode. Furthermore, under these conditions, power is transferred from the primary side to the secondary side from the start, lacking a current-cutting phase compared to over-resonant or under-resonant light load conditions. That is, the turn-on time of the secondary rectifier in the switching cycle is the same as the turn-on time of the primary drive signal. Therefore, by determining the turn-off time of the secondary rectifier in the switching cycle, the rectification drive signal for the secondary rectifier can be determined. And by calculating the second conduction duration of the LLC resonant converter in the third operating mode, the turn-off time of the secondary rectifier in the switching cycle can be determined.

[0054] It is worth noting that, in order for the LLC synchronous rectification control method to function properly, the interruption or conduction time of the secondary rectifier tube needs to be less than half a switching cycle, and the calculation time must be greater than 0. In this embodiment, steps S302 and S303 assume that the LLC resonant converter is under a preset operating condition of over-resonance light load or under-resonance light load. The secondary rectifier tubes under the over-resonance light load or under-resonance light load conditions correspond to the first operating mode. Under the preset operating condition, the interruption time and the first conduction time are calculated first. If the interruption time meets the first condition (interruption time greater than zero and less than half a switching cycle), it indicates that the preset operating condition is the current operating condition of the LLC resonant converter.

[0055] In this embodiment, if the interruption duration calculated in step S303 does not meet the first condition, i.e., the interruption duration meets the second condition (interruption duration is less than or equal to zero, or interruption duration is greater than or equal to half a switching cycle), however, if the interruption duration is less than or equal to zero, it is illogical. Furthermore, the interruption duration of the secondary rectifier diode cannot remain continuously closed or continuously open for half a cycle, as this would prevent the rectification process from proceeding normally. Therefore, when the interruption duration meets the second condition, it indicates that the current load on the LLC resonant converter is not lightly loaded, but heavily loaded. When implementing the LLC synchronous rectification control method, two situations arise: First, according to step S301, it is determined that the LLC resonant converter is currently in an over-resonance state, that is, the current operating condition of the LLC resonant converter is an over-resonance heavy-load condition. Under the over-resonance heavy-load condition, the secondary rectifier tube is in continuous conduction mode. At this time, the outward phase shift angle of the rectifier drive signal relative to the primary drive signal in the second operating mode of the LLC resonant converter is calculated. Once the outward phase shift angle is determined, the rectifier drive signal can be obtained. The rectifier drive signal and the primary drive signal differ by an outward phase shift angle value. Second, according to step S301, it is determined that the LLC resonant converter is currently in an under-resonance state, that is, the current operating condition of the LLC resonant converter is an under-resonance heavy-load condition. Under the over-resonance heavy-load condition, the secondary rectifier tube is in the third operating mode. At this time, the second conduction duration of the LLC resonant converter in the third operating mode is calculated, and the rectifier drive signal is determined based on the second conduction duration.

[0056] In this LLC synchronous rectification control method, the resonance state is determined by comparing the current switching frequency with the first resonant frequency, and the interruption time and first conduction time of the secondary rectifier tube of the LLC resonant converter in the first operating mode are calculated. Then, based on the resonance state and the interruption time, the operation of the LLC resonant converter is divided into four operating conditions, and different synchronous rectification control methods are adapted for each operating condition. Finally, accurate synchronization of the rectification drive signal and current waveform is achieved in the entire operating range, which significantly reduces rectification loss and improves the conversion efficiency and dynamic performance of the LLC resonant converter.

[0057] In a specific embodiment, for step S304, if the interruption duration calculated in step S302 satisfies the second condition, it indicates that the load of the LLC resonant converter is heavy at this time. Furthermore, if it is determined according to S301 that the LLC resonant converter is in an over-resonance state, then the LLC resonant converter is in an over-resonance heavy-load condition. The over-resonance heavy-load condition corresponds to a continuous conduction mode. The specific process for calculating the outward phase shift angle of the rectified drive signal of the LLC resonant converter in the continuous conduction mode relative to the primary side drive signal is as follows.

[0058] As an optional implementation, calculating the external phase shift angle of the rectified drive signal relative to the primary drive signal in the second operating mode of the LLC resonant converter includes: determining the external phase shift angle based on the phase shift angle relationship of the LLC resonant converter in the second operating mode; the phase shift angle relationship is the relationship between the external phase shift angle and the parameters of the resonant cavity, and the parameters of the resonant cavity include the inductance value of the resonant inductor and the inductance value of the magnetizing inductor of the LLC resonant converter.

[0059] The method for establishing the phase shift angle relationship includes: obtaining the current frequency domain equations of the resonant inductor, resonant capacitor, and secondary current based on the phasor-frequency domain model; obtaining the frequency domain equation regarding the outer phase shift angle based on the fact that the secondary current value is zero at a preset time in the second operating mode; and obtaining the phase shift angle relationship by performing an inverse Fourier series transform on the frequency domain equation. Here, the preset time is the falling edge time of the rectifier drive signal, and the preset time is (0.5×θ) / fs, where θ is the outer phase shift angle and fs is the switching frequency.

[0060] like Figure 4a and Figure 4b As shown, Figure 4a This is the equivalent circuit diagram of an LLC resonant converter under over-resonance heavy load conditions. Figure 4b This is a schematic diagram of the operating waveforms of an LLC resonant converter under the equivalent circuit diagram corresponding to the over-resonance heavy load condition. Under this condition, the output gain is less than 1, and the operating state of the LLC resonant converter is equivalent to that of a series resonant dual active bridge circuit using frequency conversion and external phase shift control. Figure 4a and Figure 4b In this diagram, g1 and g2 are complementary drive signals for the primary-side switches, I(D1) is the current flowing through the secondary-side rectifier Q5 or Q7, and I(D2) is the current flowing through the secondary-side rectifier Q6 or Q8. At this time, the voltage Vp at the midpoint of the primary-side bridge arm (i.e., the voltage between the midpoints of the left and right primary-side bridge arms) and the voltage Vs at the midpoint of the secondary-side bridge arm (i.e., the voltage between the midpoint of the left and right secondary-side bridge arms) are both square wave signals, and the secondary-side currents I(D1) and I(D2) exhibit an approximately sinusoidal shape. Based on this physical characteristic, this embodiment uses frequency domain analysis to accurately describe the phase relationship between the primary and secondary-side voltages and currents, thereby calculating the precise synchronous rectification drive timing.

[0061] Specifically, let the outward phase shift angle be θ. Based on the phasor-frequency domain model, we first perform a Fourier expansion on Vp and Vs, which gives us: (1.1) (1.2) Then, the expanded harmonics are represented using phasor sums: (1.3) (1.4) Where Vi is the input voltage, Vo is the output voltage, Ntr is the transformer turns ratio, and fs is the switching frequency.

[0062] Furthermore, based on Ohm's theorem in the frequency domain, the frequency domain equations for the currents in the resonant inductor Lr, the magnetizing inductor Lm, and the current flowing through the secondary switch (such as Q5) at time t are: (1.5) (1.6) (1.7) Where ILr is the current in the resonant inductor Lr, ILm is the current in the magnetizing inductor Lm, iL(t) is the current flowing through the secondary rectifier diode at time t, and w is the harmonic angular frequency. Cr is the capacitance of the filter capacitor, Lr is the inductance of the filter inductor, Lm is the inductance of the magnetizing inductor, and Im[.] is the imaginary part.

[0063] like Figure 4b As shown, when t is t1, the secondary current I(D1) is zero (i.e., iL=0). Therefore, let t1=0.5×θ / fs, at which time iL=0, then we have the following frequency domain equation regarding the outward phase shift angle θ: (1.8) Performing an inverse Fourier series transform on this frequency domain equation yields: (1.9) Based on equation (1.9), the following phase shift angle relationship is obtained: (1.10) Where Gv is the output voltage gain. Ntr is the transformer turns ratio, Vi is the input voltage, Vo is the output voltage, and fn is the normalized frequency. fs is the switching frequency, fr is the first resonant frequency, Lr is the inductance of the resonant inductor, Lm is the inductance of the magnetizing inductor, and Cr is the capacitance of the resonant capacitor.

[0064] As can be seen from equation (1.10), the calculation expression for the external phase shift angle θ depends only on the inherent circuit parameters (Lr, Lm, Ntr) and the current output voltage gain and normalized frequency, and is completely decoupled from the load resistance value Ro. This means that under over-resonance heavy load conditions, the controller can directly obtain the optimal synchronous rectification phase shift angle through analytical calculation without acquiring the secondary current signal. It should be noted that the external phase shift angle θ in this embodiment is per unit relative to half a switching cycle (i.e., when θ=1, it corresponds to a 180° phase shift). After obtaining the external phase shift angle θ, the delay time t1 (i.e., t1=0.5×θ / fs) can be calculated based on the switching frequency fs and the external phase shift angle θ. The rectification drive signal of the secondary rectifier tube can be determined based on the delay time. Specifically: the waveform of the primary drive signal g1 is delayed by t1 to obtain the rectification drive signal g3, and the waveform of the primary drive signal g2 is delayed by t1 to obtain the rectification drive signal g4.

[0065] In a specific embodiment, for step S302, when calculating the interruption duration and the first conduction duration, based on whether the LLC resonant converter is in an over-resonance light load condition or an over-resonance heavy load condition, the specific process of calculating the interruption duration and the first conduction duration in the first operating mode is as follows.

[0066] Under over-resonance and under-resonance light load conditions, the current in the secondary rectifier diode (D1) exhibits an intermittent conduction mode. As an optional implementation, calculating the interruption duration and the first conduction duration in the first operating mode includes: calculating the interruption duration and the first conduction duration based on the first time-domain circuit equation of the LLC resonant converter in the first operating mode.

[0067] The first time-domain circuit equation includes a first relation and a second relation obtained by performing second-order circuit time-domain analysis on the first equivalent circuit. The first relation is the relationship between the interruption duration and the switching frequency, the first resonant frequency, the second resonant frequency, the output voltage gain, the quality factor, and the parameters of the resonant cavity. The second relation is the relationship between the first conduction duration and the switching frequency, the first resonant frequency, the second resonant frequency, the output voltage gain, the quality factor, and the parameters of the resonant cavity. The first equivalent circuit is the equivalent circuit of the LLC resonant converter in the first operating mode. The quality factor is the quality factor of the LLC resonant converter. The parameters of the resonant cavity include the inductance value of the resonant inductor and the inductance value of the magnetizing inductor of the LLC resonant converter.

[0068] like Figure 5a This is the first equivalent circuit corresponding to the underresonant light load condition and the overresonant light load condition. Figure 5aThis is a schematic diagram of a second-order DC-excited circuit. To simplify the circuit, in this embodiment, all secondary rectifiers Q5 in the equivalent circuit are represented by diodes D1. In this second-order circuit, when the secondary rectifier (D1) is off, the resonant inductor Lr and the magnetizing inductor Lm are connected in series, and the total inductance in the resonant cavity is the sum of the inductance values ​​of the resonant inductor Lr and the magnetizing inductor Lm; when the secondary rectifier is on, the total inductance in the resonant cavity is the inductance value of the resonant inductor Lr. For a second-order DC-excited circuit, if the initial current of the resonant inductor Lr is IL0 and the initial voltage of the resonant capacitor Cr is Uc0, then in the second-order circuit, The time-domain current expression IL(t) of the resonant inductor Lr and the time-domain voltage expression Uc(t) of the resonant capacitor Cr are shown below, where t represents any time.

[0069] (2.0) (2.1) Where t0 is the initial time, Lr is the inductance of the resonant inductor, and Cr is the capacitance of the resonant capacitor.

[0070] Figure 5b This diagram illustrates the current waveforms of the secondary rectifier diodes Q5 (I(D1)) and Q6 (I(D2)) in an LLC resonant converter under over-resonance light load or under-resonance light load conditions. Under both conditions, the current waveforms of the secondary rectifier diodes exhibit discontinuous conduction, which cannot be analyzed using linear circuits. Therefore, a first time-domain circuit equation based on the first operating mode of the LLC resonant converter is needed to describe the circuit behavior under these conditions.

[0071] As an optional implementation, the method for establishing the first time-domain circuit equation includes: obtaining multiple sets of second-order circuit time-domain equations concerning inductance and capacitance at three time points based on second-order circuit time-domain analysis of the first equivalent circuit; obtaining a first relation and a second relation based on the multiple sets of second-order circuit time-domain equations and the Mitag-Leffler approximation; wherein the inductance is the resonant inductance of the resonant cavity of the LLC resonant converter, the capacitance is the resonant capacitance of the resonant cavity, and the three time points are the start time, the first time point, and the end time of the first conduction duration.

[0072] Reference Figure 5a and Figure 5b Under under-resonant light load and over-resonant light load conditions, the LLC resonant converter operates in its first mode. In this first mode, the current of the secondary rectifier Q5 is in discontinuous conduction mode, and the primary side of the LLC resonant converter does not transfer power to the secondary side from the initial moment. A time-domain analysis of the first equivalent circuit is performed based on the three stages experienced by the current of the secondary rectifier Q5: (1) First stage: The flow interruption stage from time 0 (starting time) to time t0. In this first stage, the secondary rectifier Q5 is cut off, the magnetizing inductor Lm participates in resonance, and the circuit is equivalent to an Lr-Lm-Cr series resonant network, with the current I(D1) of the secondary rectifier Q5 being zero. Here, time 0 is the starting time, which is the rising edge of the primary-side drive signal of the primary-side switch, and t0 is the current cut-off duration. Time t0 corresponds to the first moment, i.e., the moment when the primary side of the transformer begins to transfer power to the secondary side. Combining the general solution expression of the time-domain equation of the second-order circuit, when t is time t0, the formulas for calculating the current IL1 flowing through the resonant inductor and the voltage Uc1 of the resonant capacitor are: (2.2) (2.3) Where IL0 is the current flowing through the resonant inductor at time 0, Uc0 is the voltage across the resonant capacitor at time 0, IL1 is the current flowing through the resonant inductor at time t0, Uc1 is the voltage across the resonant capacitor at time t0, and fm is the second resonant frequency. .

[0073] Based on the voltage divider principle of the magnetizing inductance Lm and the resonant inductance Lr, we have: (2.4) (2) Second stage: the conduction stage from time t0 to time t2, where t2 = t0 + t1 In the second stage, the secondary rectifier diode is turned on, and the circuit is an Lr-Cr series resonant network. During this stage, energy is transferred from the primary side of the transformer to the secondary side. Here, t1 is the first conduction duration, and t2 (t0+t1) is the second moment, i.e., the moment when the primary side of the transformer stops transferring power to the secondary side. Combining the general solution expression of the time-domain equation for a second-order circuit, the formulas for calculating the current IL2 flowing through the resonant inductor and the voltage Uc2 across the resonant capacitor at t=(t0+t1) are: (2.5) (2.6) In the second stage, energy is transferred to the secondary side of the circuit, the secondary rectifier diodes are turned on, and a constant output voltage is applied to Lm, as follows: (2.7) Furthermore, by integrating the current in this stage, averaging the result, and then multiplying it by the output load resistance value Ro, we can obtain the expression for the output voltage: (2.8) (3) The third stage: the cutoff stage from time t2 (t0+t1) to time 1 / (2fs). In the third stage, the secondary rectifier Q5 is cut off again, and the circuit is equivalent to an Lr-Lm-Cr series resonant network. The time 1 / (2fs) is the end of half a switching cycle. Due to circuit symmetry, when t is 1 / (2fs), the current flowing through the resonant inductor is -IL0, and the voltage across the resonant capacitor is -Uc0, therefore: (2.9) (2.10) Equations (2.2) to (2.10) above are a set of nine second-order circuit time-domain equations, which are difficult to solve directly. Simplifying equations (2.4) to (2.8) yields: (2.11) Equation (2.11) is a transcendental equation containing the cotangent function cot(x), which cannot be directly solved analytically in a digital controller. Solving it using traditional numerical iterative algorithms such as the Newton-Raphson method is computationally intensive and time-consuming, and carries the risk of convergence uncertainty, making it difficult to meet the real-time control requirements of LLC resonant converters at switching frequencies of tens to hundreds of kHz. As an alternative implementation, when solving this transcendental equation, a polynomial approximation of the cotangent function is performed based on the Mitag-Leffler partial fraction, reducing the transcendental equation to a four-arithmetic equation.

[0074] Specifically, the Mitag-Leffler partial fraction expansion of the cotangent function is: (2.12) By performing a linear fit on the infinitely accumulating term in equation (2.12), we obtain the fitted equation: (2.13) Where x = π × fr × t1.

[0075] Combining equations (2.11) and (2.13), the second relational expression corresponding to the first conduction duration can be obtained: (2.14) in: (2.15) Substituting t1 into equations (2.2) to (2.4), (2.9), and (2.10), we can solve for the first relational expression corresponding to the interruption duration t0: (2.16) in: (2.17) (2.18) Where fn is the normalized frequency. fs is the switching frequency, and fr is the first resonant frequency. fm is the second resonant frequency. Q is the quality factor. Ntr is the transformer turns ratio, and Gv is the output voltage gain. Vi is the input voltage, and Vo is the output voltage.

[0076] Based on equations (2.14) to (2.18), the interruption duration t0 and the first conduction duration t1 can be calculated.

[0077] In one optional implementation, determining the rectification drive signal of the secondary rectifier based on the interruption duration and the first conduction duration includes: determining a first proportion value, i.e., the first proportion value P0 = t0 × fs, based on the product of the interruption duration and the current switching frequency; determining a second proportion value, i.e., the second proportion value P1 = (t0 + t1) × fs, based on the product of the sum of the interruption duration and the first conduction duration and the current switching frequency; determining the rising edge time of the rectification drive signal of the secondary rectifier within the switching cycle based on the first proportion value P0; and determining the falling edge time of the rectification drive signal of the secondary rectifier within the switching cycle based on the second proportion value P1.

[0078] It is worth noting that, based on the current interruption duration t0 and the first conduction duration t1, the rising edge of the rectification drive signal for the secondary rectifier diode Q5 can be determined as time t0, and the falling edge of the rectification drive signal for the secondary rectifier diode Q5 can be determined as time t2, which is t0 + t1. Thus, the rectification drive signal for the secondary rectifier diode is determined. The second percentage value P1 can be understood as the duty cycle of the pulse signal (rectifier drive signal) driving the switch of the secondary rectifier diode Q5.

[0079] like Figure 7 As shown, Figure 7 This diagram illustrates the polynomial approximation effect of the cotangent function cot(x) provided in this embodiment. Practical testing has verified that the relative error between the above-mentioned fitting formula and the true cotangent function is consistently less than 0.013% across the entire operating range of the LLC resonant converter, i.e., when x varies within the interval (0, π). This level of accuracy is far higher than the system errors caused by power device parameter tolerances and sampling circuit noise, meaning that the timing deviation introduced by mathematical approximation is negligible and will not lead to observable phase misalignment or efficiency loss in the synchronous rectification drive signal. Therefore, this fitting formula significantly reduces the processor's computational burden while ensuring engineering accuracy and avoids the risk of control cycle overflow caused by complex trigonometric function calculations.

[0080] In a specific embodiment, for step S305, if the interruption duration calculated in step S302 satisfies the second condition, it indicates that the load of the LLC resonant converter is heavy load at this time. Furthermore, if it is determined from S301 that the LLC resonant converter is in an under-resonance state, then the LLC resonant converter is in an under-resonance heavy load condition. The under-resonance heavy load condition corresponds to the third operating mode. The specific process of calculating the second conduction duration of the LLC resonant converter in the third operating mode is as follows.

[0081] Based on the foregoing embodiments, this embodiment further details the synchronous rectification control method for underresonant heavy-load conditions.

[0082] Under under-resonance heavy load conditions, the LLC resonant converter is in the third operating mode. The current I (D1) of the secondary rectifier Q5 is still in the intermittent conduction mode. Compared with the first operating mode, in the third operating mode, the current of the secondary rectifier Q5 lacks the interruption stage corresponding to time 0 to time t0. That is, in the third operating mode, the secondary rectifier Q5 is in the conduction mode from the beginning.

[0083] As an optional implementation, calculating the second conduction duration of the LLC resonant converter in the third operating mode includes: determining the second conduction duration based on the second time-domain circuit equation of the LLC resonant converter in the third operating mode.

[0084] The second time-domain circuit equation includes a third relation obtained by performing second-order circuit time-domain analysis on the second equivalent circuit. The third relation is the relationship between the second conduction duration and the switching frequency, the first resonant frequency, the output voltage gain, the quality factor, and the parameters of the resonant cavity. The second equivalent circuit is the equivalent circuit of the LLC resonant converter in the third operating mode. The quality factor is the quality factor of the LLC resonant converter. The parameters of the resonant cavity include the inductance value of the resonant inductor and the inductance value of the magnetizing inductor of the LLC resonant converter.

[0085] Figure 6a This is the second equivalent circuit for an LLC resonant converter under under-resonant heavy load conditions. Figure 6b This diagram illustrates the current waveforms of the secondary rectifier diode Q5 (I(D1)) and the secondary rectifier diode Q6 (I(D2)) within one switching cycle; where, since both the first and third operating modes correspond to intermittent conduction modes, therefore... Figure 5a and Figure 6aThe equivalent circuits are the same; however, the conduction and cutoff states experienced by the secondary rectifier Q5 differ between the first and third operating modes, resulting in different current waveforms for the secondary rectifier Q5. This is because, due to the heavy output load, the second equivalent circuit transfers power from the primary side of the transformer to the secondary side from time 0, thus causing a difference in the current waveform of the secondary rectifier. Figure 5b The interruption phase from time 0 to time t0.

[0086] As an optional implementation, a second time-domain circuit equation is established using a method similar to that used to establish the first time-domain circuit equation. The method for establishing the second time-domain circuit equation includes: obtaining a set of multiple second-order circuit time-domain equations concerning inductance and capacitance at two time points based on second-order circuit time-domain analysis of the second equivalent circuit; and obtaining a third relation based on the multiple second-order circuit time-domain equations and the Mitag-Leffler approximation. Here, the inductance is the resonant inductance of the resonant cavity of the LLC resonant converter, the capacitance is the resonant capacitance of the resonant cavity, and the two time points are the start time and the second time point, where the second time point is the moment when the primary side stops transmitting power to the secondary side. It can be understood that under under-resonant heavy-load conditions, the primary side of the transformer begins transmitting power to the secondary side of the transformer from the start time.

[0087] refer to Figure 6a and Figure 6b During half a switching cycle, under under-resonant heavy load conditions, the LLC resonant converter is in its third operating mode. In this mode, the secondary rectifier Q5 is in discontinuous conduction mode, and the primary side of the LLC resonant converter begins to transfer power to the secondary side from the initial moment. In the second equivalent circuit, the current of the secondary rectifier Q5 undergoes the following two stages: (1) First stage: Conduction stage from time 0 (starting time) to time t1 In the first stage, the secondary rectifier Q5 is in the conducting state, and the circuit is an Lr-Cr series resonant network. Power transfer from the primary side of the transformer to the secondary side begins at time 0. t1 is the second conduction duration, and time t1 is the second moment, i.e., the moment when the primary side stops transferring power to the secondary side. Combining the general solution expression of the time-domain equation of a second-order circuit, the formulas for calculating the current IL1 flowing through the resonant inductor and the voltage Uc1 of the resonant capacitor at time t1 are as follows: (3.1) (3.2) In this stage, the circuit transfers energy to the secondary side, the secondary rectifier diodes conduct, and a constant output voltage acts on Lm, as follows: (3.3) Furthermore, by integrating the current in this stage, averaging the result, and then multiplying it by the output load resistance value Ro, we can obtain the expression for the output voltage: (3.4) (2) Second stage: the cutoff stage from time t1 to time 1 / (2fs). During this stage, the secondary rectifier is cut off, and the circuit is equivalent to an Lr-Lm-Cr series resonant network. The time 1 / (2fs) is the end of half a switching cycle. Due to circuit symmetry, when t is 1 / (2fs), the current flowing through the resonant inductor is -IL0, and the voltage across the resonant capacitor is -Uc0, therefore: (3.5) (3.6) Combining equations (3.1) to (3.6), we have: (3.7) Equation (3.7) is also a transcendental equation containing the cotangent function cot(x). After applying the same partial fraction approximation and fitting to cot(x) as described above, the third relation corresponding to the second conduction duration t1 is obtained: (3.8) in: (3.9) The second conduction duration t1 can be calculated based on equations (3.8) and (3.9). Based on the second conduction duration t1, the falling edge time of the rectifier drive signal of the secondary rectifier tube (i.e., time t1) can be determined, while the rising edge time of the rectifier drive signal of the secondary rectifier tube is time 0.

[0088] In one optional implementation, for ease of control, the second conduction duration t1 can be converted into a duty cycle. Specifically, a third percentage value (i.e., P2 = t1 × fs) is determined based on the product of the second conduction duration and the current switching frequency. The rising edge of the primary-side drive signal within the switching cycle is taken as the rising edge of the secondary-side rectifier drive signal within the switching cycle, and the falling edge of the secondary-side rectifier drive signal within the switching cycle is determined based on the third percentage value. Here, the third percentage value is the duty cycle of the secondary-side rectifier drive signal under underresonant heavy-load conditions. Under this condition, the turn-on time of the secondary-side rectifier is synchronized with the turn-on time of the primary-side switch; therefore, only the third percentage value needs to be calculated to determine the secondary-side rectifier drive signal.

[0089] It should be noted that this application defines the conduction time under the under-resonant light load / over-resonant light load conditions as the "first conduction time" and the conduction time under the under-resonant heavy load conditions as the "second conduction time" in order to distinguish between different operating conditions. In fact, both are the duration of current in the secondary rectifier tube during the switching cycle, and their physical essence is the same. Therefore, the same naming identifier t1 is used in the calculation.

[0090] The entire calculation process described above is based entirely on real-time analytical derivation of the circuit topology and parameters, without relying on any offline experimental data tables or empirical fitting curves. This means that regardless of the input voltage, output voltage, or load combination under which the LLC resonant converter operates, as long as the circuit parameters are known, the controller can calculate the theoretically optimal synchronous rectification timing in real time using the same mathematical model. This purely analytical control architecture fundamentally solves the problem of the traditional lookup table method's accuracy dropping sharply under atypical operating conditions due to incomplete sample coverage. It also avoids the excessive sensitivity of hardware detection methods to the accuracy of the sampling circuit, providing solid technical support for achieving high-efficiency and high-reliability synchronous rectification across the entire operating range.

[0091] Based on the calculation of the external phase shift angle or various percentage values ​​in the aforementioned embodiments, this embodiment further illustrates how to map theoretical timing parameters to safe actual hardware drive signals. As an optional implementation, when generating the rectifier drive signal, a dead time is inserted between the falling edge and rising edge times of the two complementary rectifier drive signals of the same bridge arm of the secondary rectifier.

[0092] Specifically, since the secondary rectifier diodes Q5-Q8 are typically power semiconductor devices such as MOSFETs, their physical switching process is not instantaneous but involves a certain turn-off delay and current tailing effect. If the drive signals of the two switching diodes on the same bridge arm are theoretically completely complementary and switch without interval, a short-circuit can easily form in the bridge arm during the brief overlapping window when one diode has not yet completely turned off while the other has already started conducting, leading to device damage or even system failure. Therefore, a dead time period in which both are at a low level must be forcibly inserted after the falling edge of the complementary rectifier drive signal and before the rising edge to ensure that the first rectifier diode has fully recovered its blocking capability before the other rectifier diode starts conducting. This mechanism is a necessary prerequisite for ensuring the safe and reliable operation of the power stage circuit. In an optional implementation, the external phase shift angle θ, the first proportion value P0, the second proportion value P1, or the third proportion value P2 calculated in the aforementioned embodiments are all theoretically optimal values ​​based on an ideal circuit model and do not contain any non-ideal factors. The controller writes these theoretical values ​​directly into the PWM generator's comparison register, and the hardware PWM module or the underlying driver automatically adds a preset fixed dead time when generating the final gate drive signal.

[0093] To further verify the effectiveness and accuracy of the LLC synchronous rectification control method in the aforementioned embodiments, this embodiment conducted multi-condition simulation tests and analyses based on a unified circuit parameter platform. The specific parameter settings of this platform are as follows: resonant inductance Lr is 50μH, resonant capacitor Cr is 50nF, transformer magnetizing inductance Lm is 320μH, transformer turns ratio Ntr is 8:1, and input voltage Vi is 400V. The calculated first resonant frequency fr is approximately 100kHz. Based on these parameters, verification was conducted for three typical operating scenarios: over-resonance heavy load, over-resonance / under-resonance light load, and under-resonance heavy load, to visually demonstrate the synchronization accuracy of the technical solution in this application across the entire operating range and its effectiveness in addressing the pain points of the background technology.

[0094] In the over-resonance heavy load verification scenario, the load resistance Ro was set to 1.67Ω, and the switching frequency fs was adjusted to 150kHz. At this time, the normalized frequency fn=1.5, and the circuit operates in the over-resonance region. According to the phase shift angle relationship described in the previous embodiment, the controller calculates the external phase shift angle θ to be 0.072 in real time. After applying this calculated value to the drive signal generation of the secondary rectifier diodes Q5-Q8, as follows... Figure 8 The simulation waveforms shown indicate that, under this operating condition, the rising and falling edges of the rectifier drive signals (g3, g4) are highly aligned with the zero-crossing points of the secondary currents I(D1) and I(D2), meaning that the rectifier drive signals are completely synchronized with the secondary currents I(D1) and I(D2), achieving high-precision synchronous rectification.

[0095] In the over-resonance / under-resonance light load verification scenario, the load resistance Ro is adjusted to 5Ω, and the switching frequency is maintained below the first resonant frequency. At this time, the circuit enters a discontinuous current conduction mode, with a significant current interruption gap on the secondary side. Based on the first and second relationships, the interruption duration and the first conduction duration can be calculated respectively. The controller then calculates the first proportion value P0 as 0.021 and the second proportion value P1 as 0.436. The first proportion value accurately represents the turn-on time of the secondary rectifier diode; the second proportion value precisely locks the turn-off time of the secondary rectifier diode. For example... Figure 9 The simulation waveforms shown indicate that, under this operating condition, the rising and falling edges of the rectifier drive signals (g3, g4) are highly aligned with the zero-crossing points of the secondary currents I(D1) and I(D2), meaning that the rectifier drive signals are completely synchronized with the secondary currents I(D1) and I(D2), achieving high-precision synchronous rectification.

[0096] In the underresonant heavy load verification scenario, the load resistance Ro was further reduced to 1.25Ω. As the load increased, the circuit automatically transitioned to continuous current conduction mode. Based on the third relationship, the second conduction time can be calculated, and the controller then calculates the third proportion P2 as 0.4. Compared to the light load scenario, the turn-on delay in the underresonant heavy load condition disappeared, and the secondary rectifier turned on at the beginning of the switching cycle. Figure 10 The simulation waveforms shown indicate that the rising and falling edges of the rectifier drive signals (g3, g4) are highly aligned with the zero-crossing times of the secondary currents I(D1) and I(D2), meaning that the rectifier drive signals are completely synchronized with the secondary currents I(D1) and I(D2), achieving high-precision synchronous rectification.

[0097] In summary, through simulation verification and parameter sensitivity analysis of the three typical operating conditions described above, it is fully demonstrated that the LLC synchronous rectification control method provided in this application can consistently achieve precise synchronization between the secondary rectifier drive signal and the current waveform across the entire load range and under conditions of switching frequency fluctuation. This scheme completely eliminates the dependence on hardware current sampling circuits and overcomes the shortcomings of offline lookup table methods in terms of insufficient accuracy under atypical operating conditions, providing solid technical support for the high-efficiency and high-reliability operation of LLC resonant converters.

[0098] This application also provides a non-volatile computer-readable storage medium storing computer-executable instructions that are executed by one or more processors, for example, to perform the steps of the LLC synchronous rectification control method described above.

[0099] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of this application, and not to limit them; under the concept of this application, the technical features of the above embodiments or different embodiments can also be combined, the steps can be implemented in any order, and there are many other variations of different aspects of this application as described above, which are not provided in detail for the sake of brevity; although this application has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that they can still modify the technical solutions described in the foregoing embodiments, or make equivalent substitutions for some of the technical features; and these modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the scope of the technical solutions of the embodiments of this application.

Claims

1. A method for controlling LLC synchronous rectification, characterized in that, include: Based on the current switching frequency and the first resonant frequency of the LLC resonant converter, determine whether the LLC resonant converter is in an over-resonance state or an under-resonance state. Calculate the current-off duration and the first conduction duration of the LLC resonant converter in the first operating mode. The first operating mode is the operating mode in which the secondary rectifier is in intermittent conduction mode for half a switching cycle. The current-off duration is the duration from the start time to the first time. The start time is the rising edge of the primary drive signal of the primary switch. The first time is the moment when the primary side begins to transfer power to the secondary side. The first conduction duration is the duration during which the secondary rectifier has current in the switching cycle in the first operating mode. If the interruption duration meets the first condition, the rectification drive signal of the secondary rectifier is determined based on the interruption duration and the first conduction duration. If the interruption duration satisfies the second condition and is in the over-resonance state, calculate the outward phase shift angle of the rectifier drive signal relative to the primary drive signal in the second operating mode of the LLC resonant converter. The second operating mode is that the secondary rectifier is in continuous conduction mode within half a switching cycle. If the interruption duration satisfies the second condition and is in the under-resonance state, calculate the second conduction duration of the LLC resonant converter in the third operating mode, and determine the rectifier drive signal based on the second conduction duration. The third operating mode is the operating mode in which the secondary rectifier is in intermittent conduction mode for half a switching cycle and the primary side transmits power to the secondary side from the starting time. The second conduction duration is the duration for which the secondary rectifier has current in the switching cycle under the third operating mode. The first condition is that the interruption duration is greater than zero and less than half a switching cycle, and the second condition is that the interruption duration is less than or equal to zero, or that the interruption duration is greater than or equal to half a switching cycle.

2. The method according to claim 1, characterized in that, Based on the interruption duration and the first conduction duration, the rectification drive signal of the secondary rectifier is determined, including: The first percentage value is determined based on the product of the interruption duration and the current switching frequency; The second percentage value is determined by multiplying the sum of the interruption duration and the first conduction duration by the current switching frequency; The rising edge time of the rectification drive signal of the secondary rectifier tube within the switching cycle is determined based on the first percentage value, and the falling edge time of the rectification drive signal of the secondary rectifier tube within the switching cycle is determined based on the second percentage value.

3. The method according to claim 1, characterized in that, The rectification drive signal of the secondary rectifier is determined based on the second conduction duration, including: The third percentage value is determined based on the product of the second conduction duration and the current switching frequency; The rising edge of the primary-side drive signal within the switching cycle is taken as the rising edge of the secondary-side rectifier drive signal within the switching cycle, and the falling edge of the secondary-side rectifier drive signal within the switching cycle is determined based on the third proportion value.

4. The method according to claim 1, characterized in that, The calculation of the interruption time and the first conduction time of the LLC resonant converter in the first operating mode includes: Based on the first time-domain circuit equation of the LLC resonant converter in the first operating mode, the current interruption duration and the first conduction duration are calculated. The first time-domain circuit equation includes a first relation and a second relation obtained by performing second-order circuit time-domain analysis on the first equivalent circuit. The first relation is the relationship between the interruption duration and the switching frequency, the first resonant frequency, the second resonant frequency, the output voltage gain, the quality factor, and the parameters of the resonant cavity. The second relation is the relationship between the first conduction duration and the switching frequency, the first resonant frequency, the second resonant frequency, the output voltage gain, the quality factor, and the parameters of the resonant cavity. Wherein, the first equivalent circuit is the equivalent circuit of the LLC resonant converter in the first operating mode, the quality factor is the quality factor of the LLC resonant converter, and the parameters of the resonant cavity include the inductance value of the resonant inductor and the inductance value of the magnetizing inductor of the LLC resonant converter.

5. The method according to claim 4, characterized in that, The method for establishing the first time-domain circuit equation includes: Based on the second-order circuit time-domain analysis of the first equivalent circuit, a set of multiple second-order circuit time-domain equations concerning the inductor and capacitor at three time points are obtained. Based on the multiple sets of time-domain equations of the second-order circuit and the Mittag-Leffler approximation, the first relation and the second relation are obtained; Wherein, the inductor is the resonant inductor of the resonant cavity of the LLC resonant converter, the capacitor is the resonant capacitor of the resonant cavity, the three times are the start time, the first time and the second time, and the second time is the time when the primary side stops transmitting power to the secondary side.

6. The method according to claim 5, characterized in that, The first relation is: The second relation is: in, , , t0 is the interruption duration, t1 is the first conduction duration, Lr is the inductance value of the resonant inductor, Lm is the inductance value of the magnetizing inductor, and fn is the normalized frequency. fs is the switching frequency, and fr is the first resonant frequency. fm is the second resonant frequency. Q is the quality factor. Ntr is the transformer turns ratio, Cr is the resonant capacitor capacitance, Ro is the load resistance, and Gv is the output voltage gain. Vi is the input voltage, and Vo is the output voltage.

7. The method according to claim 1, characterized in that, The calculation of the outward phase shift angle of the rectified drive signal relative to the primary-side drive signal of the LLC resonant converter in the second operating mode includes: The external phase shift angle is determined based on the phase shift angle relationship of the LLC resonant converter in the second operating mode. The phase shift angle relationship is the relationship between the external phase shift angle and the parameters of the resonant cavity, and the parameters of the resonant cavity include the inductance value of the resonant inductor and the inductance value of the magnetizing inductor of the LLC resonant converter.

8. The method according to claim 7, characterized in that, The method for establishing the phase shift angle relationship includes: Based on the phasor-frequency domain model, the current frequency domain equations for the resonant inductor, resonant capacitor, and secondary current are obtained. Based on the fact that the secondary current value is zero at a preset time in the second working mode, a frequency domain equation for the outer phase shift angle is obtained; Perform an inverse Fourier series transform on the frequency domain equation to obtain the phase shift angle relationship; Wherein, the preset time is the falling edge time of the rectifier drive signal, and the preset time is (0.5×θ) / fs, where θ is the outward phase shift angle and fs is the switching frequency.

9. The method according to claim 1, characterized in that, The calculation of the second on-time of the LLC resonant converter in the third operating mode includes: The second conduction duration is determined based on the second time-domain circuit equation of the LLC resonant converter in the third operating mode. The second time-domain circuit equation includes a third relation obtained by performing second-order circuit time-domain analysis on the second equivalent circuit. The third relation is the relationship between the second conduction duration and the switching frequency, the first resonant frequency, the output voltage gain, the quality factor, and the parameters of the resonant cavity. Wherein, the second equivalent circuit is the equivalent circuit of the LLC resonant converter in the third operating mode, the quality factor is the quality factor of the LLC resonant converter, and the parameters of the resonant cavity include the inductance value of the resonant inductor and the inductance value of the magnetizing inductor of the LLC resonant converter.

10. The method according to claim 9, characterized in that, The method for establishing the second time-domain circuit equation includes: Based on the second equivalent circuit time-domain analysis, a set of multiple second-order circuit time-domain equations concerning the inductor and capacitor at two time points are obtained. Based on the multiple sets of time-domain equations of the second-order circuit and the Mittag-Leffler approximation, the third relation is obtained; Wherein, the inductor is the resonant inductor of the resonant cavity of the LLC resonant converter, the capacitor is the resonant capacitor of the resonant cavity, the two moments are the start moment and the second moment, and the second moment is the moment when the primary side stops transmitting power to the secondary side.

11. The method according to claim 10, characterized in that, The third relation is: in, t1 is the second conduction duration, Lr is the inductance value of the resonant inductor, Lm is the inductance value of the magnetizing inductor, and fn is the normalized frequency. fs is the switching frequency, and fr is the first resonant frequency. Q is the quality factor. Ntr is the transformer turns ratio, Cr is the resonant capacitor capacitance, Ro is the load resistance, and Gv is the output voltage gain. Vi is the input voltage, and Vo is the output voltage.

12. A controller, characterized in that, The controller includes at least one processor and a memory communicatively connected to the at least one processor, the memory storing instructions executable by the at least one processor to enable the at least one processor to perform the method according to any one of claims 1 to 11.

13. An energy storage system, characterized in that, include: An LLC resonant converter includes a primary-side bridge circuit, a resonant cavity, a transformer, and a secondary-side bridge circuit connected in sequence. The primary-side bridge circuit includes a first bridge arm and a second bridge arm connected in parallel. The secondary-side bridge circuit includes a third bridge arm and a fourth bridge arm connected in parallel. The first bridge arm and the second bridge arm each include two primary-side switching transistors connected in series. The third bridge arm and the fourth bridge arm each include two secondary-side rectifier transistors connected in series. The controller as described in claim 12, wherein each of the primary-side switching transistors and each of the secondary-side rectifier transistors are connected to the controller.

14. A computer storage medium, characterized in that, The computer storage medium stores instructions or programs that, when executed by at least one processor, cause the at least one processor to perform the method as described in any one of claims 1 to 11.

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