Method for enhancing spatial degrees of freedom of a radio channel of a base station antenna

Abstract

The embodiment of the present application provides a kind of method for enhancing the spatial freedom of base station antenna wireless channel.The method comprises: determining the wave number range of the propagation wave corresponding to transmitter according to the structure parameter of local resonant metamaterial;The current distribution corresponding to each antenna feed port is respectively mapped to wave number domain, and a plurality of radiation spectra are obtained;According to the wave number range of the propagation wave and the plurality of radiation spectra, the structure parameter of the local resonant metamaterial is adjusted to enhance the spatial freedom of the base station antenna wireless channel.Through the embodiment of the present application, the problem that the spatial freedom of base station antenna wireless channel is limited under the condition that holographic MIMO port surface is limited in the related art is solved, and the effect of improving the communication performance of HMIMO system is achieved.

Description

Technical Field

[0001] This invention relates to the field of communications, and more specifically, to a method for enhancing the spatial degrees of freedom of a base station antenna wireless channel. Background Technology

[0002] Holographic MIMO (HMIMO) technology, as one of the core technologies of future 6G wireless communication, establishes multiple orthogonal and independent channels by arranging ultra-dense antenna arrays in a limited space and simulating orthogonal current modes on the output plane using dense antenna arrays, thus achieving theoretically extremely high degrees of freedom (DoF).

[0003] Existing HMIMO systems improve communication performance by increasing the number of antenna elements. However, in reality, when the element density is too high, some current modes exist in the form of evanescent waves and cannot be effectively radiated. The spatial degrees of freedom do not increase with the increase of element density. The increase of antenna element density does not break through the upper limit of spatial degrees of freedom set by the electromagnetic wave propagation law and the Nyquist sampling theorem. The upper limit of spatial degrees of freedom is determined by the aperture size of the holographic MIMO. Summary of the Invention

[0004] This invention provides a method for enhancing the spatial degrees of freedom of a base station antenna wireless channel, thereby at least solving the problem of limited spatial degrees of freedom of the base station antenna wireless channel under holographic MIMO aperture constraints in related technologies.

[0005] According to an embodiment of the present invention, a method for enhancing the spatial degrees of freedom of a base station antenna wireless channel is provided, comprising: determining the wavenumber range of the propagating wave corresponding to the transmitter based on the structural parameters of a local resonant metamaterial; mapping the current distribution corresponding to each antenna feed port to the wavenumber domain to obtain multiple radiation spectra; and adjusting the structural parameters of the local resonant metamaterial according to the wavenumber range of the propagating wave and the multiple radiation spectra to enhance the spatial degrees of freedom of the base station antenna wireless channel.

[0006] According to another embodiment of the present invention, a holographic MIMO communication system is provided, comprising: a transmitter and a locally resonant metamaterial disposed in a holographic MIMO near-field region of the transmitter, the locally resonant metamaterial surrounding an antenna array in the transmitter.

[0007] According to yet another embodiment of the present invention, a computer-readable storage medium is also provided, wherein a computer program is stored therein, wherein the computer program is configured to perform the steps in any of the above method embodiments when executed.

[0008] According to yet another embodiment of the present invention, an electronic device is also provided, including a memory and a processor, wherein the memory stores a computer program and the processor is configured to run the computer program to perform the steps in any of the above method embodiments.

[0009] According to yet another embodiment of the present invention, a computer program product is also provided, comprising a computer program that, when executed by a processor, implements the steps in any of the above method embodiments.

[0010] Based on the above embodiments of the present invention, by adjusting the structural parameters of the local resonant metamaterial, the wavenumber range of the propagation wave corresponding to the transmitter can be expanded, thereby enhancing the spatial degree of freedom of the wireless channel. Therefore, the problem of limited spatial degree of freedom of the base station antenna wireless channel under the condition of limited holographic MIMO aperture in related technologies can be solved, thereby improving the communication performance of the HMIMO system. Attached Figure Description

[0011] Figure 1 This is a hardware structure block diagram of the computer terminal used in the embodiments of the method of the present invention;

[0012] Figure 2 This is a schematic diagram of the network architecture of an HMIMO system according to an embodiment of the present invention;

[0013] Figure 3 This is a schematic diagram of a locally resonant metamaterial according to an embodiment of the present invention;

[0014] Figure 4 This is a flowchart of a method for enhancing the spatial degrees of freedom of a base station antenna wireless channel according to an embodiment of the present invention;

[0015] Figure 5 This is a schematic diagram of implementing a current mode of a continuous aperture using a discrete dense antenna according to an embodiment of the present invention;

[0016] Figure 6 This is a flowchart of a method for testing transmitter spatial multiplexing degrees of freedom according to an embodiment of the present invention;

[0017] Figure 7 This is a schematic diagram of the process of constructing a high-dimensional channel for spatial multiplexing using a locally resonant metamaterial according to an embodiment of the present invention;

[0018] Figure 8 This is a schematic diagram illustrating the principle of far-field radiation of evanescent modes using a locally resonant metamaterial according to an embodiment of the present invention.

[0019] Figure 9 This is a flowchart illustrating a method for optimizing the structural parameters of a locally resonant metamaterial according to an embodiment of the present invention.

[0020] Figure 10 This is a schematic diagram of the ultra-dense scatterer resonator structure and its electromagnetic field distribution according to an embodiment of the present invention;

[0021] Figure 11 This is a schematic diagram of the electromagnetic distribution in a locally resonant metamaterial structure according to an embodiment of the present invention;

[0022] Figure 12 This is a schematic diagram of the simulation verification settings for determining the resonant frequency point according to an embodiment of the present invention. Detailed Implementation

[0023] The embodiments of the present invention will be described in detail below with reference to the accompanying drawings and examples.

[0024] It should be noted that the terms "first," "second," etc., in the specification, claims, and drawings of this invention are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence.

[0025] In existing holographic MIMO systems, the spatial degrees of freedom of the system are determined by the emission aperture S. t The number of orthogonal current modes that can be effectively transmitted to the far field is determined by this. According to the theory of electromagnetic wave propagation, the conversion from the near field to the far field is actually a Fourier transform process. The minimum resolution (Δk) in one direction of the wavenumber domain and the minimum resolution (ΔL) in one direction of the spatial domain are Δk = 2π / L and ΔL = λ0 / 2, respectively, where L is the aperture size in the transverse direction of electromagnetic wave propagation and λ0 is the wavelength of the electromagnetic wave in free space.

[0026] For propagating waves, the upper limit of the number of independent channels is given by the ratio of the wavenumber domain to the minimum wavenumber resolution. Where k0 is the wave number of the electromagnetic wave in free space, and the spatial degree of freedom is limited by the aperture size of the holographic MIMO.

[0027] The reason for the low degree of freedom in holographic MIMO systems is that when the element density is too high, some current modes exist in the form of evanescent waves and cannot be effectively radiated. For example, when using N antennas to implement holographic MIMO, N orthogonal bases can be constructed, providing N spatial multiplexing degrees of freedom. However, in reality, not all N orthogonal bases can radiate to the far field.

[0028] Existing holographic MIMO systems are characterized by ultra-dense array element arrangement. As analyzed above, this leads to antenna redundancy in practical applications. Due to the strong coupling effect between antenna elements, the spatial degrees of freedom of the antenna array are no longer linearly related to the number of antennas. When the antenna array density increases, although the ability of holographic MIMO to simulate aperture current is further improved, and higher spatial frequency current modes can be excited, these spatially high-frequency current modes exist as near-field evanescent waves, which attenuate rapidly with increasing propagation distance and cannot effectively radiate to the far field. This results in limited spatial degrees of freedom, making it difficult to directly utilize them for far-field communication.

[0029] This invention provides a method for enhancing the spatial (multiplexing) degrees of freedom of holographic MIMO. The method involves designing a local resonant metamaterial (LRM) outside the transmitter of a holographic MIMO system. By using the LRM to change the dispersion relation, the holographic MIMO system can support near-field evanescent wave radiation to the far field, which can effectively increase the number of orthogonal modes of the holographic MIMO antenna radiating to the far field region, thereby enhancing the spatial degrees of freedom of the holographic MIMO as a transmitter.

[0030] The methods and embodiments provided in this application can be executed on a computer terminal or similar computing device. Taking running on a computer terminal as an example, Figure 1 This is a hardware structure block diagram of the computer terminal used in the embodiments of the method of the present invention. For example... Figure 1 As shown, a computer terminal may include one or more ( Figure 1 Only one is shown in the diagram. A processor 102 (which may include, but is not limited to, a microprocessor MCU or a programmable logic device FPGA, etc.) and a memory 104 for storing data are also shown. The computer terminal may further include a transmission device 106 for communication functions and an input / output device 108. Those skilled in the art will understand that... Figure 1 The structure shown is for illustrative purposes only and does not limit the structure of the computer terminal described above. For example, the computer terminal may also include components that are more complex than those described above. Figure 1 The more or fewer components shown, or having the same Figure 1 The different configurations shown.

[0031] The memory 104 can be used to store computer programs, such as application software programs and modules, like the computer program corresponding to the method for enhancing the spatial degrees of freedom of the base station antenna wireless channel in this embodiment of the invention. The processor 102 executes various functional applications and data processing by running the computer program stored in the memory 104, thereby implementing the above-described method. The memory 104 may include high-speed random access memory, and may also include non-volatile memory, such as one or more magnetic storage devices, flash memory, or other non-volatile solid-state memory. In some instances, the memory 104 may further include memory remotely located relative to the processor 102, and these remote memories can be connected to a computer terminal via a network. Examples of such networks include, but are not limited to, the Internet, corporate intranets, local area networks, mobile communication networks, and combinations thereof.

[0032] The transmission device 106 is used to receive or send data via a network. Specific examples of the network described above may include a wireless network provided by a communication provider for the computer terminal. In one example, the transmission device 106 includes a Network Interface Controller (NIC), which can connect to other network devices via a base station to communicate with the Internet. In another example, the transmission device 106 may be a Radio Frequency (RF) module used for wireless communication with the Internet.

[0033] The embodiments of the present invention can be run in Figure 2 The network architecture of the HMIMO system shown is as follows: Figure 2 As shown, the network architecture includes a transmitter 21 and a locally resonant metamaterial 22 disposed in the near-field region of a holographic multiple-input multiple-output (MIMO) transmitter, the locally resonant metamaterial 22 surrounding an antenna array in the transmitter 21.

[0034] The holographic MIMO system in this embodiment of the invention considers a communication scenario involving two continuous apertures of arbitrary shape and position in a homogeneous medium. The current density at any source point within the transmitting aperture will generate an electric field at any location within the receiving aperture; the relationship between the two is described by a Green's function. It is assumed that the continuous aperture sources or fields at the transmitting and receiving ends use orthogonal modes {Φ... m (s); m = 1, ..., M} and {Ψ n Expand the expansion of (r); n = 1, ..., N, with expansion coefficients x. m and jkZ0y n Thus, the channel model is constructed as shown in formula (1):

[0035]

[0036] Among them, the channel matrix element Hnm Defined as M and N are the dimensions of the input and output signal spaces, respectively. This orthogonal pattern expansion allows for a precise description of the channel characteristics.

[0037] In the holographic MIMO system of this invention embodiment, the holographic MIMO part is composed of ultra-dense dipole antennas. Each dipole antenna is independently fed, and a certain topological structure of subwavelength passive radiation source (local resonant metamaterial) is arranged around the active radiation source (i.e. the dipole antenna constituting the holographic MIMO). The active radiation source and the passive radiation source form a strong coupling, which can generate a complex electromagnetic distribution in space, thereby enabling the holographic MIMO system to create an ultra-high-dimensional spatial multiplexing channel within a confined aperture plane.

[0038] The adjustable parameters of the holographic MIMO system in this embodiment of the invention are: the excitation signal design and structural parameters such as spacing of the holographic MIMO ultra-dense antenna array (active radiation source); the topological parameters such as the element spacing of the local resonant metamaterial; the element configuration (e.g., different structures such as metal wires with loads in the middle, short-circuited metal wires, open-circuit metal wires, split-transformer resonators, etc.); whether the lumped impedance is loaded on each element; and the loaded load impedance value.

[0039] In practical system implementation, each element of a holographic MIMO, i.e. the active radiation source of the holographic MIMO system, can be controlled by a computer's general software-defined radio peripheral to generate various excitation signals. The radiation characteristics of the excitation signals can be evaluated by spatial-wavenumber domain Fourier transform. Local resonant metamaterials can change the spatial dispersion characteristics, transforming the linear relationship of frequency-wavenumber dispersion characteristics in free space into a nonlinear relationship. Thus, the same radiation frequency can carry higher spatial frequencies (corresponding to in the dispersion curve), thereby carrying more refined spatial oscillation information and radiating the higher-order current excitation modes of the holographic MIMO to the far field.

[0040] Figure 3 This is a schematic diagram of a locally resonant metamaterial according to an embodiment of the present invention, as shown below. Figure 3 As shown, the locally resonant metamaterial consists of a densely distributed array of scatterers with adjustable unit parameters. This array of scatterers can form multimode resonances and alter spatial dispersion characteristics. The visible region can be determined by the resonant region and dispersion curves, thereby assessing the number of effective radiation modes of the holographic MIMO.

[0041] Local resonant metamaterials can be realized in various ways, including but not limited to: periodic structures such as periodic split-ring resonators, periodic metal or dielectric sphere scatterers, periodically arranged electromagnetic dipoles, non-periodic structures such as ring scatterer arrays, and continuous structures made of electromagnetic media. Based on the above multiple realization methods, adjustable impedances can also be added to the scatterer units to achieve system optimization effects.

[0042] The dispersive properties of locally resonant metamaterials can be controlled by parameters such as the array element topology, array element configuration, and lumped impedance on the array elements.

[0043] This embodiment provides a method for enhancing the spatial degrees of freedom of a base station antenna wireless channel operating on the aforementioned computer terminal or network architecture. Figure 4 This is a flowchart of a method for enhancing the spatial degrees of freedom of a base station antenna wireless channel according to an embodiment of the present invention, such as... Figure 4 As shown, the process includes the following steps:

[0044] Step S402: Determine the wavenumber range of the propagating wave corresponding to the transmitter based on the structural parameters of the local resonant metamaterial;

[0045] Step S404: Map the current distribution corresponding to each antenna feed port to the wavenumber domain to obtain multiple radiation spectra;

[0046] Due to physical limitations, it is not possible to place an infinite number of antennas on a continuous aperture surface. To address this, a discrete but ultra-dense antenna array can be used, employing mathematical methods (such as interpolation) to simulate the current distribution on the continuous aperture surface. Figure 5 This is a schematic diagram illustrating the implementation of a continuous aperture current mode using a discrete dense antenna according to an embodiment of the present invention, as shown below. Figure 5 As shown, this demonstrates how the current distribution on a continuous aperture is interpolated using the current distribution on a discrete dense antenna at the transmitter antenna location. In other words, at the transmitter antenna location, the current distribution on a continuous aperture is simulated using interpolation techniques based on the current distribution information on a densely distributed discrete antenna array (i.e., an ultra-dense antenna array).

[0047] Step S406: Adjust the structural parameters of the local resonant metamaterial according to the wavenumber range of the propagating wave and the multiple radiation spectra to enhance the spatial degree of freedom of the base station antenna wireless channel.

[0048] In this embodiment of the invention, evanescent wave propagation wave conversion can be achieved by configuring tunable local resonant metamaterials in the near field, which significantly improves the spatial degree of freedom of the holographic MIMO system without increasing the actual aperture of the antenna array.

[0049] In one exemplary embodiment, a locally resonant metamaterial is configured in the near-field region of the transmitter's holographic MIMO and surrounded by an ultra-dense antenna array to allow as many current modes as possible to radiate outward.

[0050] Each antenna in the ultra-dense array transmits a broadband sinusoidal modulated Gaussian pulse, which is obtained by upconverting a baseband sinusoidal modulated Gaussian pulse to the carrier frequency.

[0051] For example: the transmitted signal has a center frequency of f0 and a normalized bandwidth of f0. (bandwidth is) A sinusoidal modulated Gaussian pulse is used as the baseband signal and up-converted to I / Q channels with a frequency of f. c On the carrier wave.

[0052] Using the resonant frequency f c A +f0 dipole antenna is used as the receiving antenna to receive the signal. After down-converting the received signal, the receiving antenna in the far field detects the baseband signal and obtains its time-domain waveform and spectrum. The time-delayed spread signal received by the time-reversal mirror array is time-reversed, and the received signal at the original transmit array port is observed, allowing for mode recovery. Effective mode recovery proves that the transmitter successfully propagated the corresponding mode to the far field.

[0053] The near-field region can be defined as the area where the antenna array size is significantly smaller than the square root of the product of the wavelength and the maximum array size. In this region, the distribution of the electromagnetic field is directly affected by the position and spacing of the antenna elements. The electromagnetic waves exhibit a complex alternating distribution of electric and magnetic fields and contain evanescent wave components. These components decay rapidly with increasing distance and cannot effectively radiate to the far field.

[0054] The far-field region is typically related to the Rayleigh distance, which is the relationship between the maximum size of the antenna array and the operating wavelength. When the distance between the receiving antenna and the antenna array exceeds the Rayleigh distance, the behavior of the electromagnetic waves can be considered to conform to the characteristics of the far-field region. In a holographic MIMO system, the far-field region can refer to the area where the distance between the antenna array and the receiving point is large enough that the propagation of electromagnetic waves can be considered as radiation waves in free space.

[0055] In one embodiment, the structural parameters of the locally resonant metamaterial include array unit structural parameters, topological structural parameters, and lumped circuit parameters.

[0056] In one embodiment, the array unit structural parameters include at least one of the following: array unit shape, length of a single array unit; the topology parameters include at least one of the following: spacing between locally resonant metamaterial array units; and the lumped circuit parameters include at least one of the following: load impedance value applied to each array unit.

[0057] Prior to step S404 in this embodiment, the method further includes: exciting / loading a pre-designed orthogonal current pattern through each antenna feed port in the antenna array independently controlled in the transmitter; wherein each array element is assigned a corresponding current amplitude and phase to achieve a predetermined orthogonal current distribution.

[0058] In one embodiment, before exciting / loading a pre-designed orthogonal current mode through each antenna feed port in an independently controlled antenna array in the transmitter, the method further includes: designing specific current distributions that can be generated at different locations of the aperture by superimposing orthogonal functions to construct multiple orthogonal current modes; wherein the orthogonal functions include, but are not limited to: oblate spheroidal wave function (PSWF) and sine / cosine functions.

[0059] In one exemplary embodiment, cosine and sine functions may be selected to construct orthogonal current modes for use at the aperture D. x ×D y The current distribution on an M×N antenna array with orthogonal current modes is described by the following formula (2):

[0060]

[0061] Where p and q correspond to the row and column indices of the antenna array, respectively, to represent the dipole antennas at different positions in the array; D x This represents the dimension of the antenna array or virtual aperture along the x-axis, i.e., the width of the array; D y This indicates the dimension of the antenna array or virtual aperture in the y-axis direction, i.e., the length of the array; M represents the number of antenna elements in the antenna array in the x-axis direction; N represents the number of antenna elements in the antenna array in the y-axis direction; A pq It describes the amplitude of the current of the (p,q)th dipole, and is a sample of the orthogonal mode of the sine and cosine functions. The current amplitude of the (p,q)th dipole in mode (m,n) is given by the following formula (3):

[0062]

[0063] Where m represents the row number of the orthogonal current mode and n represents the column number of the orthogonal current mode.

[0064] J d To describe the current distribution on each dipole antenna, the boundary condition that the current is zero at both ends needs to be satisfied. This is represented by a sinusoidal distribution, specifically, it can be described by formula (4).

[0065]

[0066] Where J0 represents the initial or reference current intensity, Represents the unit vector in the y-direction;

[0067] This represents the change in current amplitude along the y-axis.

[0068] This represents the current distribution along the y-direction;

[0069] δ(x) represents the current distribution along the x-direction;

[0070] L is the length of each dipole antenna.

[0071] A continuous function is derived from the current amplitude value A on a discrete antenna. pq This is achieved through interpolation. According to the sampling theorem, the current amplitude extracted from M×N discrete antenna elements can be used to perfectly reconstruct the superposition of M×N mutually orthogonal mode currents on a continuous aperture.

[0072] The various orthogonal current modes constructed in the above embodiments do not interfere with each other and can provide high-dimensional spatial freedom.

[0073] When orthogonal current modes are excited, the antennas in the antenna array (dipole antennas or other types of antennas) will generate electromagnetic fields according to the current distribution on them. According to Maxwell's equations, changing currents produce changing electromagnetic fields, which are further converted into electromagnetic waves and diffused into the far field via radiation. The structure and layout of the antenna array affect the mode and directivity of the electromagnetic waves, causing them to propagate according to the designed orthogonal current modes.

[0074] In one embodiment, the method further includes propagating the quadrature current pattern in the form of electromagnetic waves via an antenna array in the transmitter.

[0075] In step S404 of this embodiment, the current distribution corresponding to each antenna feed port is mapped to the wavenumber domain to obtain multiple radiation spectra. This includes performing Fourier transform on the current distribution of the current mode loaded on each antenna feed port to map each current distribution to a radiation spectrum in the wavenumber domain, thereby obtaining the multiple radiation spectra.

[0076] Before step S406 in this embodiment, the method further includes: performing time-domain inversion processing on a first signal transmitted by a far-field receiver to generate an inverted signal; wherein the first signal is transmitted by the far-field receiver based on a first orthogonal current mode signal of the transmitter, and the transmission characteristics of the first orthogonal current mode signal are affected by the local resonant metamaterial; mapping the first orthogonal current mode signal and the second signal transmitted by the far-field receiver based on the inverted signal onto an orthogonal basis, and determining the expansion coefficient vectors of the first orthogonal current mode signal and the second signal on the orthogonal basis respectively; determining a first channel matrix based on the expansion coefficient vectors of the first orthogonal current mode signal and the second signal on the orthogonal basis; and determining that the spatial degrees of freedom of the base station antenna wireless channel needs to be enhanced based on the eigenvalues ​​of the first channel matrix and a first preset feature threshold.

[0077] The spatial degrees of freedom (DoF) of a holographic MIMO system are determined by the aperture S t The number of orthogonal current modes M, such as Prolate Spheroidal Wave Functions (PSWF) and Sine Cosine Functions, that can be effectively transmitted to the far field is determined by the number of these modes.

[0078] This invention also provides a method for testing the spatial multiplexing degree of freedom of a transmitter, which evaluates the spatial multiplexing degree of freedom by determining the number of orthogonal modes that a holographic MIMO transmitter can effectively transmit in the far field.

[0079] This method involves the following devices or tools:

[0080] Time-reversal mirror array: used to receive and time-reverse signals from the original transmitter, and send the time-reversed signals back to the original transmitter as transmission signals to test the effective radiation of orthogonal modes.

[0081] Holographic MIMO transmitter: Contains an ultra-dense antenna array, capable of emitting electromagnetic waves in multiple orthogonal modes.

[0082] Signal analysis software: processes signals and calculates the distribution of eigenvalues.

[0083] In this embodiment, the transmitter is placed within a time-reversal mirror array to ensure that the far-field receiving port can ideally receive the electromagnetic fields radiated from all the transmitting ports.

[0084] Figure 6 This is a flowchart of a test method for transmitter spatial multiplexing degrees of freedom according to an embodiment of the present invention, as shown below. Figure 6 As shown, the method includes the following steps:

[0085] In step S601, the pre-designed orthogonal current pattern is transmitted in the form of electromagnetic wave signals through the antenna array of the holographic MIMO transmitter.

[0086] The orthogonal current modes can be described by the formulas in the above embodiments, and each orthogonal current mode represents a potential independent spatial information channel. This electromagnetic wave signal is functionally equivalent to the first signal in the above embodiments.

[0087] In step S602, the far-field time-reversal mirror array captures the electromagnetic wave signal from the holographic MIMO transmitter, performs time-reversal processing on the received electromagnetic wave signal, and obtains the inverted signal.

[0088] In step S603, the holographic MIMO transmitter receives the second signal transmitted from the far field. This second signal is sent by the far-field receiver based on the inverted signal.

[0089] Step S604: Determine the first channel matrix based on the expansion coefficient vector of the first signal on the orthogonal basis and the expansion coefficient vector of the second signal on the orthogonal basis.

[0090] In an exemplary embodiment, the holographic MIMO transmitter sends a transmission signal x(s,t) to the far field in the form of an electromagnetic wave y(r,t), where x(s,t) is projected onto an orthogonal basis as x(s,t); the electromagnetic wave y(r,t) received in the far field is time-reversed to y TR (r,t) = y(r,-t) and propagates in the back direction to the holographic MIMO transmitter; the holographic MIMO transmitter receives the electromagnetic field x at the emission aperture. TR (s) can be projected onto an orthogonal basis to obtain the expansion coefficient vector x. TR :

[0091] x TR =(HH) H ) T x * Formula (5)

[0092] Where H represents the channel matrix of the MIMO channel or holographic MIMO system; H H x represents the conjugate transpose of matrix H; * It represents the complex conjugate of vector x.

[0093] Step S605: Based on the eigenvalues ​​of the first channel matrix and the first preset eigenvalue threshold, determine that the spatial degree of freedom of the base station antenna wireless channel needs to be enhanced.

[0094] When the receiving surface in the far field ideally receives the radiation field from all the transmitting surfaces, the matrix HH constructed by time reversal in the far field... H The rank and eigenvalue distribution describe the ability of orthogonal current modes to effectively radiate into the far field. Each time the transmitter transmits an orthogonal mode (eigenmode), under ideal time inversion in the far field, if the orthogonal mode can be recovered by the electromagnetic inversion of the time-reversal mirror, then the orthogonal mode can effectively radiate into the far field. If the orthogonal mode cannot be recovered by the electromagnetic inversion of the time-reversal mirror, then the orthogonal mode cannot effectively radiate into the far field, and the system cannot obtain the degrees of freedom provided by that orthogonal mode. Each orthogonal mode corresponds to one spatially multiplexed channel, and the attenuation coefficient of each orthogonal mode in the system reflects the eigenvalue distribution of the channel.

[0095] If, after eigenvalue decomposition, multiple eigenvalues ​​exceed a pre-set threshold, the corresponding orthogonal current modes are effectively received and inverted in the far field, thus providing independent spatial multiplexing channels for the holographic MIMO system. The effective number of orthogonal modes is called the spatial degrees of freedom (DoF), representing the number of independent data streams that the transmitter can transmit simultaneously on a limited aperture.

[0096] In step S406 of this embodiment, adjusting the structural parameters of the local resonant metamaterial according to the wavenumber range of the propagating wave and multiple radiation spectra includes: determining the radiation spectrum corresponding to the evanescent wave according to the multiple radiation spectra and the wavenumber range of the propagating wave, wherein the radiation spectrum corresponding to the evanescent wave is the radiation spectrum that does not fall within the beam range of the propagating wave; and expanding the wavenumber range of the propagating wave by adjusting the structural parameters of the local resonant material so that the radiation spectrum corresponding to the evanescent wave falls within the expanded wavenumber range of the propagating wave.

[0097] In one embodiment, adjusting the structural parameters of the local resonant material to expand the wavenumber range of the propagating wave includes: determining a dispersion relation expression, wherein the dispersion relation expression includes at least the following parameters: electromagnetic wave frequency, wavenumber, array unit structural parameters, topology parameters, and lumped circuit parameters; determining array unit structural parameter values, topology parameters, and lumped circuit parameters to optimize the wavenumber range of the propagating wave based on the dispersion relation expression and the electromagnetic wave frequency corresponding to the signal currently emitted by the transmitter; and adjusting the structural parameters of the local resonant metamaterial to expand the wavenumber range of the propagating wave based on the array unit structural parameter values, array unit structural parameters, topology parameters, and lumped circuit parameters to optimize the wavenumber range of the propagating wave.

[0098] Following step S406 in this embodiment, the method further includes: transmitting a second orthogonal current mode signal to a far-field receiver based on a locally resonant metamaterial with adjusted structural parameters; wherein the frequency of the second orthogonal current mode signal covers the expanded wavenumber range of the propagating wave; receiving a third signal transmitted by the far-field receiver, the third signal being a signal obtained by time-reversal processing of the second orthogonal current mode signal; determining a second channel matrix based on the second orthogonal current mode signal and the third signal; and determining, based on the eigenvalues ​​of the second channel matrix and a second preset characteristic threshold, that the spatial degrees of freedom of the base station antenna wireless channel needs to be further enhanced.

[0099] In one embodiment, determining the second channel matrix based on the second orthogonal current mode signal and the third signal includes: determining the second channel matrix based on the expansion coefficient vector of the second orthogonal current mode signal on the orthogonal basis and the expansion coefficient vector of the third signal on the orthogonal basis; and determining the second channel matrix based on the expansion coefficient vector of the second orthogonal current mode signal on the orthogonal basis and the expansion coefficient vector of the third signal on the orthogonal basis.

[0100] By adjusting the structural parameters of the local resonant metamaterial through the above steps, the wavenumber range of the propagation wave corresponding to the transmitter can be expanded, thereby enhancing the spatial degree of freedom of the wireless channel. Therefore, the problem of limited spatial degree of freedom of the base station antenna wireless channel under the condition of limited antenna aperture in related technologies can be solved, thereby improving the communication performance of the HMIMO system.

[0101] This invention also provides a method for constructing a high-dimensional channel for spatial multiplexing using a locally resonant metamaterial. The system loaded with the locally resonant metamaterial can increase the transmitter's spatial multiplexing stream number, where the transmitter's spatial multiplexing stream number is the number of independent data streams (or channels) that the transmitter can transmit simultaneously.

[0102] This method includes: adjusting the visible region of the locally resonant metamaterial based on its dispersion and resonance characteristics, so that the spectrum of more orthogonal current modes in the wavenumber domain is located in the visible region. The visible region is defined as the frequency of the highest spatial oscillating electromagnetic field that can radiate to the far field; in the above embodiment, the frequency range corresponding to the beam range of the propagating wave is located within the visible region.

[0103] The parameters of locally resonant metamaterials can be categorized into three types: element structure parameters, topological parameters, and lumped circuit parameters. Element structure parameters include the shape of the array elements (e.g., different structures such as short-circuited metal lines, open-circuit metal lines, and split-switch resonators) and the length of a single array element. Topological parameters include the spacing between the array elements of the locally resonant metamaterial. Lumped circuit parameters include whether a lumped impedance is applied to each array element and the value of the applied load impedance.

[0104] Figure 7 This is a schematic diagram illustrating the process of constructing a high-dimensional channel for spatial multiplexing using a locally resonant metamaterial according to an embodiment of the present invention, as shown below. Figure 7 As shown, the process includes the following steps:

[0105] Step S701: Determine the dispersion characteristics of the local resonant metamaterial based on the unit structure parameters and topology parameters, and through analytical models, simulation analysis, or experimental testing.

[0106] In this embodiment, when designing a locally resonant metamaterial, it is necessary to ensure that the kf dispersion curve bends as much as possible toward the k-axis so that the same frequency can carry greater wave number information; where k in the kf dispersion curve is the wave number, that is, the spatial distribution of the periodic characteristics of the wave; and f is the frequency, representing the number of vibrations of the wave per unit time.

[0107] Step S702: Determine the resonance characteristics of the local resonant metamaterial based on the topology parameters and lumped circuit parameters, and through analytical models, simulation analysis, or experimental testing.

[0108] Locally resonant metamaterials possess resonant properties, and their resonant frequencies determine which frequencies of electromagnetic waves can effectively interact with the locally resonant metamaterial. Consequently, these waves may be radiated to distant locations through the conversion of evanescent waves to propagating waves. Therefore, it is necessary to determine the frequencies of electromagnetic waves that can be radiated outward in order to selectively adjust the relevant structural parameters of the locally resonant metamaterial.

[0109] During the design process, the resonant frequency needs to be determined based on the bandwidth range of the transmitted signal from the source antenna, and the resonant frequency must be within the bandwidth of the transmitted signal from the source antenna.

[0110] Step S703: Determine the visible region based on the dispersion and resonance characteristics of the local resonant metamaterial.

[0111] The resonant frequency range gives the frequency distribution of outwardly radiated electromagnetic waves. The dispersion relation curve of the locally resonant metamaterial gives the correspondence between frequency and spatial wavenumber. Therefore, the frequency distribution range of outwardly radiated electromagnetic waves can be mapped to the wavenumber domain range through the dispersion relation curve. This range is called the visible region.

[0112] Without a locally resonant metamaterial, because the far-field time-reversal array surrounds the transmitting antenna, electromagnetic waves radiated from all directions can be received, and the visible region is [missing information]. When a locally resonant metamaterial exists, the size of the visible region needs to be determined by measuring the frequency band of the resonant peak of the radiation field. As the visible region expands, the high-frequency components of space that should propagate as evanescent waves can radiate to the far field as propagating waves, further increasing the propagation modes and thus increasing the spatial multiplexing degree of freedom (DoF).

[0113] By adjusting the structural parameters of locally resonant metamaterials, the dispersion relation and resonant frequency can be controlled and optimized, thereby optimizing the control of the visible region.

[0114] Step S704: Based on the holographic MIMO parameters, determine the current distribution of the orthogonal bases on the holographic MIMO and the radiation spectrum characteristics of the beams corresponding to the orthogonal bases on the holographic MIMO in the wavenumber domain.

[0115] The holographic MIMO parameters include, but are not limited to, at least one of the following: the geometric parameters of the antenna array, the characteristics of the antenna elements, and the operating frequency and bandwidth range of the holographic MIMO system.

[0116] The geometric parameters of an antenna array include, but are not limited to: the size, shape, and spacing of the antenna elements, the overall layout or shape of the antenna array, and the subwavelength spacing of the antenna elements.

[0117] The characteristics of an antenna element include, but are not limited to: antenna type, antenna gain, directivity, polarization characteristics, impedance matching, etc.

[0118] Step S705: Determine the transmitter space division multiplexing current number based on the visible area and radiation spectrum characteristics.

[0119] Step S706: Perform the flow rate verification using the time inversion method.

[0120] Figure 8 This is a schematic diagram illustrating the principle of far-field radiation of evanescent modes using a locally resonant metamaterial according to an embodiment of the present invention. Figure 8 As shown, each port feed is excited by the USRP, loading different orthogonal current modes to simulate the current distribution of a certain orthogonal mode on the continuous port surface.

[0121] The following explanation uses the orthogonal current mode of sine and cosine functions as an example:

[0122] In the embodiment where sine and cosine functions are used as the basis for orthogonal current modes, the field distribution under each current excitation mode is given by the Fourier transform of the electric field in the spatial domain. Using the equivalence principle, when the field on one side of the aperture is set to zero, the current on the aperture can be considered as being generated by... Given the magnetic field, the directional characteristics of the radiation field can be quantitatively described using the Fourier transform of the current. Wherein, J n This represents the current density distribution on the surface of the hole; Unit vector, indicating the direction perpendicular to the antenna array; H n This indicates the distribution of magnetic field strength.

[0123] Performing a Fourier transform on the current distribution of the M×N holographic MIMO yields the Fourier transform form of mode (m, n):

[0124]

[0125] Among them, F d (k x ,k y (m, n) is the Fourier transform form of the dipole current distribution. It can be seen that the spectrum of the mode (m, n) radiation field in the wavenumber domain is a superposition of shifted sinc functions.

[0126] In k x -k y On the wavenumber plane, a circle The inner part is the visible region. Inside the circle, the wavenumber in the direction of propagation... It is a real number that can radiate to the far field. Outside the circle, the wavenumber in the direction of propagation... It is an imaginary number, meaning that it will be in k z Exponential decay in direction, can only occur along k x -k y It propagates in a planar direction and is an evanescent wave that helps store energy.

[0127] Therefore, far-field radiation depends only on the portion of the source current that lies within the visible region after a spatial Fourier transform. Based on the definition of the visible region, the characteristic modes on an antenna array can be divided into two categories: radiating modes and evanescent wave modes. The main peak of the Sinc function constituting the wavenumber spectrum of the radiating modes lies within the visible region, and these modes radiate a large amount of energy. Although the main peaks of these modes are within the visible region, the infinite periodicity of the Fourier transform also results in peaks outside the visible region; these peaks correspond to the evanescent waves that inevitably store energy in the near field. The number of radiating modes of the transmitter corresponds to its spatial degrees of freedom.

[0128] The main peak of the Sinc function on the wavenumber spectrum constituting the evanescent wave mode does not appear in the visible region, therefore it lacks a radiative main lobe. All main lobes will generate evanescent waves, resulting in a larger energy storage. It should also be noted that the evanescent wave mode of an array with finite boundaries will still radiate a small amount of energy because all Sinc functions constituting the wavenumber spectrum will have side lobes in the visible region, thus causing radiation. Theoretically, the surface waves of an infinitely large array will not radiate energy because it will have a Sinc function with a width close to zero, which is equivalent to a delta function with no side lobes to radiate.

[0129] Taking a 1×5 electric dipole array as an example, as the aperture gradually decreases, and the antenna array changes from a normal half-wavelength pitch antenna array to an aperture-constrained holographic MIMO, the changes in spatial degrees of freedom are shown in Table 1:

[0130] Table 1

[0131] Array element spacing λ / 2 λ / 10 λ / 4 λ / 2 Radiation mode / / Mode 1-2 Modes 1-5 Fleeting Mode Modes 1-5 Modes 1-5 Modes 3-5 / DoF 1 1 2 5

[0132] Table 1 classifies the modes of 1×5 dipole arrays with different element spacings, showing the relationship between the mode classification and the degrees of freedom of 1×5 dipole arrays with different element spacings.

[0133] For a typical array with an element spacing of λ / 2, the period of the Sinc function translation is equal to the diameter of the circle defining the visible region. This means it does not include any surface wave modes, maximizing the DoF gained from the number of antennas. For ultra-dense holographic MIMOs with extremely dense element spacing, the spatial DoF is extremely limited under aperture-constrained conditions.

[0134] The size of the visible region indicates the difference in spatial degrees of freedom of the transmitting antenna when there is no local resonant cavity versus when there is one. A time-reversal mirror antenna positioned in the far field, after performing time reversal on the received signal using USRP software (defined as wireless) and related MATLAB programs, and then retransmitting, will, according to the laws of electromagnetic wave inversion, allow modes that successfully propagate to the far field to be focused back to their original position after electromagnetic inversion, resulting in mode recovery. Modes that did not radiate to the far field, however, cannot be recovered.

[0135] It is evident that, compared to the free space scenario, the near-field LRM effectively enhances the spatial degree of freedom of the radiation field of HMIMO composed of ultra-dense linear array antennas, thereby enabling multi-stream transmission of HMIMO in confined spaces.

[0136] In one exemplary embodiment, a method for optimizing the structural parameters of a locally resonant metamaterial composed of finite-length metal wires is provided.

[0137] A locally resonant metamaterial is constructed from multiple periodically arranged metal wires of length L. A 31×31 array of linear ultra-dense scatterers is formed, with an element spacing of 5 mm and an element length of 75 mm. This locally resonant metamaterial array surrounds an ultra-dense antenna array. Each wire in the holographic MIMO ultra-dense array transmits a wideband sinusoidal modulated Gaussian pulse (obtained by up-converting a baseband sinusoidal modulated Gaussian pulse to the carrier frequency). The transmitted signal has a center frequency f0 = 100 MHz and a normalized bandwidth of [missing information]. (bandwidth is) A sinusoidal modulated Gaussian pulse is used as the baseband signal and up-converted to both I and Q carrier waves, with a carrier frequency of f. c =800MHz.

[0138] The transmitting antenna consists of several ultra-dense, subwavelength-spaced antennas. In the prototype provided in this invention, the ultra-dense holographic MIMO antenna array has an element spacing of 10mm and a length of 75mm, with the element spacing being the operating wavelength λ. c 1 / 30th of that. Each antenna has a feed port, and different current modes can be applied to make the aperture of the ultra-dense antenna array present different current modes. The current modes of the continuous aperture of the antenna can be constructed by interpolation.

[0139] Figure 9 This is a flowchart illustrating a method for optimizing the structural parameters of a locally resonant metamaterial according to an embodiment of the present invention, as shown below. Figure 9 As shown, the process includes the following steps:

[0140] Step S901: Determine the analytical expression of the dispersion relation for the structural parameter length L of the locally resonant metamaterial in order to optimize the structural parameters of the locally resonant metamaterial.

[0141] By Floquet's theorem, periodicity in the transverse plane gives E ⊥ (x ⊥ E z (x ⊥ H ⊥ (x ⊥ H z (x ⊥ It has the form of a Bloch wave solution: Where, k ⊥ It is the Bloch wavenumber, U ⊥ (x ⊥ ) represents the electric or magnetic field after vertical and horizontal decomposition.

[0142] Figure 10 This is a schematic diagram of the ultra-dense scatterer resonator structure and its electromagnetic field distribution according to an embodiment of the present invention, as shown below. Figure 10As shown, using Maxwell's equations and the boundary conditions of the periodic structure, the dispersion relation of the TEM modes in the resonant cavity can be obtained as follows: Where, k z ω is the longitudinal wave number, c is the angular frequency, and c is the wave velocity.

[0143] Therefore, the transverse wavenumber k of the TEM mode propagating along the z-direction can be seen. ⊥ It can be any value, thus allowing the evanescent wave component of high-frequency spatial oscillation in the transverse plane to propagate in the metal wire medium, which acts as a waveguide to support the propagation of high-frequency spatial components.

[0144] Since the actual metal wire is not infinitely long (not infinitely long in the z-axis direction), when a TEM mode wave propagates along the z-axis in the metal, it encounters the interface between the metal wire medium and air (or vacuum). Reflection and refraction occur at this interface. and The two reflecting surfaces reflect back and forth, forming an equivalent Fabry-Perot cavity, thus generating Fabry-Perot resonance for each propagation mode. Since the modes penetrate into the air (or vacuum), the length of the Fabry-Perot cavity is not simply the length L of the metal wire medium, but rather the equivalent length l of the Fabry-Perot cavity needs to be calculated using the energy entering the air (or vacuum). eff :

[0145]

[0146] Where, k ⊥ It represents the Bloch wavenumber; k represents the wavenumber along the propagation direction (such as the z-axis).

[0147] Where δ is the energy density at the interface when total internal reflection occurs. and Skin depth decreases exponentially:

[0148]

[0149] Conditions arising from Fabry–Perot resonance:

[0150]

[0151] Where n is a positive integer representing the order or mode number of the resonant mode; λ represents the wavelength of the electromagnetic wave propagating in the cavity.

[0152] From the conditions for generating a first-order Fabry-Perot resonance, the dispersion relation is derived as follows:

[0153]

[0154] in, The initial wavenumber for the first-order Fabry-Perot resonance. The initial frequency of the first-order Fabry-Perot resonance.

[0155] Dispersion relation curves of locally resonant metamaterials with thin metal wires of different lengths L were plotted. Among them, f c and k c These are the center frequency and corresponding wavenumber of the source broadband pulse signal, respectively. Since the resonant frequency will be designed at f... c Therefore, the dispersion curve needs to cover f. c The range, at the same time, is desired so that the dispersion curve corresponds to a larger wavenumber at the same frequency; therefore, the optimal length for a fine metal wire locally resonant metamaterial is... Where, λ c The wavelength corresponds to the center frequency of the source broadband pulse signal.

[0156] Step S902 involves optimizing the element spacing and adjustable load parameters in the local resonant metamaterial to obtain a suitable resonance range that is within the bandwidth coverage of the source antenna's transmitted signal.

[0157] In locally resonant metamaterials composed of metal wires, due to the Fabry-Perot resonance effect at certain frequencies, electromagnetic waves of high-frequency components in space radiate to the far field in the form of propagating waves. Comparing the changes in dispersion relation brought about by the local resonant cavity with those in free space, the bending of the dispersion curve means that, under the same radiation frequency band, the scenario with the local resonant cavity can obtain a larger visible region in the wavenumber domain compared to free space.

[0158] The optimization of the resonance range of a locally resonant metamaterial needs to be determined based on the center frequency and bandwidth of the source antenna's transmitted signal, and it is necessary to ensure that the resonance range is within the bandwidth of the source antenna's transmitted signal.

[0159] In one exemplary embodiment, a full-wave simulation and analytical modeling method for the resonant frequency of a locally resonant metamaterial structure with metal wires is provided:

[0160] Figure 11 This is a schematic diagram of the electromagnetic distribution in a locally resonant metamaterial structure according to an embodiment of the present invention, as shown below. Figure 11As shown, for a locally resonant metamaterial composed of multiple periodically arranged metal wires of length L, all transverse and longitudinal electromagnetic components are solved. Near the metal wire medium along the z-axis, the electric field exhibits strong z-directional polarization, which immediately decays exponentially in an evanescent wave mode. Therefore, energy cannot leave the structure from the top or bottom interface; it can only radiate from the transverse interface. The z-component of the electric field can be modeled using small dipoles located at z = ±L / 2. The spatial distribution of these small dipoles is given by the z-component of the TEM mode at the interface, radiating energy outwards. However, due to the finite nature of the periodic structure, the electromagnetic waves that can radiate outwards are discrete in the frequency domain. This reflects the resonant characteristics of the locally resonant metamaterial, and the resonant frequency can be determined through full-wave simulation or experimental measurement. Figure 12 This is a schematic diagram of the simulation verification settings for determining the resonant frequency point according to an embodiment of the present invention, as shown below. Figure 12 As shown, a full-wave simulation scenario for the aforementioned materials is presented.

[0161] In an exemplary embodiment, the resonant frequency of the above structure can also be determined analytically, considering the above-mentioned N S One active source and N L The system consists of passive sources (scatterer elements). Both active and passive sources are metal lines of length L. A load impedance may exist at the midpoint of the passive source. Consider the following description of the array element specifications: i,j = 1,2,…,N S p,q=1,2,…,N L N S and N L These represent the number of active source array elements and the number of passive source array elements, respectively.

[0162] Define the mutual impedance of array cell i due to array cell j as:

[0163]

[0164] Among them, V ij,oc I is the voltage generated by the near-field produced by the current on element j when element i is open-circuited. j This is the input current on element j. The mutual impedances satisfy reciprocity, i.e., Z... ij =Z ji .

[0165] If element i is open-circuited, and the z-component induced in element i by the electric field generated by element j is E ij (z), then the induced open-circuit voltage will be:

[0166]

[0167] Where h i =l i / 2,I i (z) and I i =I i (0) represents the current of array element i and the input current (current at the origin), respectively. According to the definition of mutual impedance, we can derive:

[0168]

[0169] Assuming the current is sinusoidal, that is:

[0170]

[0171] When calculating the mutual coupling effect between closely packed array elements, it is necessary to know the field generated by the elements at close range, and all field components can be derived from the z-component A of the magnetic vector potential. z In (z,ρ), we obtain:

[0172]

[0173] Where h is the half-length of the array element, h = l / 2. In the above formula (17), the approximate thin-line kernel Green's function is used for integration because it is applicable at distance ρ > a (usually, when...). The difference between the time and the precise core is very small, where a is the characteristic size of the antenna element, such as half the length of a dipole antenna, the diameter of the element, etc.

[0174] Subsequently, the z-component A of the magnetic vector potential z (z,ρ) Derivation of E z :

[0175]

[0176] Where j is the imaginary unit, ω is the angular frequency of the electromagnetic wave, μ is the permeability of the medium, z is the longitudinal coordinate in the cylindrical coordinate system, representing the position along the vertical direction of the antenna array, and ρ is the radial coordinate in the cylindrical coordinate system, that is, the radial distance of the observation point or measurement point relative to the z-axis on the plane perpendicular to the z-axis, used to describe the distribution of the electromagnetic field in the horizontal direction.

[0177] Make G(zz) ′ ,ρ)=e -jkR / R, and use Replaced The differential identity is given by equation (19), where k is the beam and R is the distance between the source point (the position of the transmitting antenna element z') and the observation point (position (z,ρ)):

[0178]

[0179] Due to the assumed form, I(z) ′It satisfies the Helmholtz equation. Therefore, the integrand becomes the complete derivative:

[0180]

[0181] Using the endpoint condition I(h) = I(-h) = 0, integrating the first term yields:

[0182]

[0183] Where h is the half-height of the electromagnetic structure, which represents the upper limit of the integral in formula (21), and G is the Green's function.

[0184] Second item The integral is discontinuous at z = 0. By dividing the integration range, we obtain:

[0185]

[0186] Wherein, G(zh,ρ), G(z+h,ρ), and G(z,ρ) are expressed in the following forms:

[0187]

[0188] Where R0, R1, and R2 are considered as the distances from the center and ends of the array element to the observation point. Therefore, the electric field along array element i can be expressed as formula (26):

[0189]

[0190] Where -h i ≤z≤h i R j ,R i R0 can be represented as:

[0191]

[0192] Substituting into the definition of mutual impedance, we can obtain the mutual impedance Z. ij The final expression:

[0193]

[0194] Among them, h i and h j These represent the height or half-height of array element i and array element j, respectively. In jη, j is the imaginary unit, η is the characteristic impedance of the dielectric, and k is the wave number.

[0195]

[0196] For active source element i, the excitation voltage on it satisfies:

[0197]

[0198] Where s is a source antenna element in the antenna array, and N S N is the number of source antenna array elements. L It is the number of elements in the locally resonant metamaterial (LRM).

[0199] If we consider a passive radiation source with load impedance, the port voltage on the p-element of the passive radiation source satisfies the following formula (33):

[0200]

[0201] Among them, V oc,p I is the voltage when element p is open-circuited. p (0) is the input current on element p, Z L,p Z is the load impedance on element p. A,p It is the self-impedance of array element p.

[0202] Due to V oc,p satisfy:

[0203]

[0204] Therefore, the following is given:

[0205]

[0206] In summary, the voltage and current relationship of the system can be written in the form of a linear equation:

[0207]

[0208] Among them, subarray The element expressions are as follows:

[0209]

[0210] In the known impedance matrix Under these conditions, the resonant frequency of the system can be determined. The resonant frequency corresponds to the frequency at which the system can oscillate spontaneously without external excitation. To address this, consider the case where the system has no external voltage excitation. In this case, to obtain a non-zero current solution, the impedance matrix must satisfy... It is singular at this frequency, that is:

[0211] Where ω is the angular frequency of the system. Solving the above characteristic equation yields the values ​​of all resonant frequencies of the system. This is while ensuring the finite length of the metal wire is λ. cWith the element spacing and adjustable load remaining constant, parameters such as / 2 can be used as optimization parameters. After adjusting the parameters, the dispersive resonant frequency will change accordingly. The optimal parameters can be found by using optimization methods (such as reinforcement learning algorithms) to solve the optimization problem, with the goal of maximizing the visible area.

[0212] After measuring the resonant frequency of electromagnetic waves radiated into the far field by a locally resonant metamaterial composed of multiple periodically arranged metal wires of length L, its visible region range is given as [-91.3m] using dispersion relations. -1 91.3m -1 As the visible region expands, high-frequency spatial components that should propagate as evanescent waves can radiate to the far field as propagating waves, further increasing the propagation modes and thus increasing the spatial multiplexing degrees of freedom (DoF).

[0213] In one exemplary embodiment, the size of the visible region indicates that the spatial degrees of freedom of the transmitting antenna are 1 and 5 when there is no local resonant cavity, respectively. For a holographic MIMO with 5 orthogonal bases, we need to consider a channel matrix HH of size 5. H The eigenvalue distribution is used to give the spatial DoF of the holographic MIMO transmitter. The symbol of the transmitted signal is s. i =e i (i=1, 2, 3, 4, 5) when (e i (The symbol is a unit vector), and the symbol recovered by time inversion is the projection of the received signal onto each set of orthogonal bases, from which the channel matrix HH can be calculated. H .

[0214] Matrix HH H The rank and eigenvalue distribution describe the ability of orthogonal current modes to effectively radiate into the far field. For the case of a near-field locally resonant metamaterial, the eigenvalue distribution obtained after eigenvalue decomposition is σ1 = 1.252, σ2 = 0.804, σ3 = 0.397, σ4 = 0.130, σ5 = 0.049 (×10⁻⁶). -6 For the free space case, the eigenvalue distribution obtained after eigenvalue decomposition is σ1 = 0.4489, σ2 = 0.0008, σ3 = 0.0004, σ4 = 0, σ5 = 0. It can be seen that the rank of the channel matrix is ​​5 when near-field LRM exists, and the eigenvalue distribution is relatively uniform. If channels with eigenvalues ​​greater than 10% of the maximum eigenvalue are considered effective channels, then the number of effective independent orthogonal channels in the above system is 4, while the number of effective channels is 1 when near-field LRM does not exist. LRM greatly improves the DoF of aperture-constrained HMIMO and realizes high-dimensional multi-stream.

[0215] The above embodiments of the present invention are feasible solutions for antenna miniaturization. When the aperture of a holographic MIMO is less than half the wavelength corresponding to the center frequency, the aperture size of the holographic MIMO antenna array can be effectively reduced while ensuring a very high degree of freedom in spatial multiplexing, thereby achieving antenna miniaturization.

[0216] The above embodiments of the present invention can also provide a feasible solution for super-resolution multi-stream line-of-sight. In the transmitter of the embodiments of the present invention, the upper limit of spatial degrees of freedom given by the Nyquist theorem of spatial signals can be broken, super-resolution multi-stream transmission can be obtained, and the number of streams in spatial multiplexing can be greatly increased.

[0217] The above embodiments of the present invention can also solve the spatial division multiple access problem for ultra-dense users, and can deploy more and denser antenna terminals in a limited area, thereby realizing ultra-dense multi-user multiple access.

[0218] Through the above description of the embodiments, those skilled in the art can clearly understand that the methods according to the above embodiments can be implemented by means of software plus necessary general-purpose hardware platforms. Of course, they can also be implemented by hardware, but in many cases the former is a better implementation method. Based on this understanding, the technical solution of the present invention, in essence, or the part that contributes to the prior art, can be embodied in the form of a software product. This computer software product is stored in a storage medium (such as ROM / RAM, magnetic disk, optical disk) and includes several instructions to cause a terminal device (which may be a mobile phone, computer, server, or network device, etc.) to execute the methods described in the various embodiments of the present invention.

[0219] Embodiments of the present invention also provide a computer-readable storage medium storing a computer program, wherein the computer program is configured to perform the steps in any of the above method embodiments when executed.

[0220] In one exemplary embodiment, the aforementioned computer-readable storage medium may include, but is not limited to, various media capable of storing computer programs, such as a USB flash drive, read-only memory (ROM), random access memory (RAM), portable hard disk, magnetic disk, or optical disk.

[0221] Embodiments of the present invention also provide an electronic device including a memory and a processor, the memory storing a computer program and the processor being configured to run the computer program to perform the steps in any of the above method embodiments.

[0222] In one exemplary embodiment, the electronic device may further include a transmission device and an input / output device, wherein the transmission device is connected to the processor and the input / output device is connected to the processor.

[0223] Specific examples in this embodiment can be found in the examples described in the above embodiments and exemplary implementations, and will not be repeated here.

[0224] It is obvious to those skilled in the art that the modules or steps of the present invention described above can be implemented using general-purpose computing devices. They can be centralized on a single computing device or distributed across a network of multiple computing devices. They can be implemented using computer-executable program code, and thus can be stored in a storage device for execution by a computing device. In some cases, the steps shown or described can be performed in a different order than those described herein, or they can be fabricated as separate integrated circuit modules, or multiple modules or steps can be fabricated as a single integrated circuit module. Thus, the present invention is not limited to any particular combination of hardware and software.

[0225] The above description is merely a preferred embodiment of the present invention and is not intended to limit the invention. Various modifications and variations can be made to the present invention by those skilled in the art. Any modifications, equivalent substitutions, or improvements made within the principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A method for enhancing the spatial degrees of freedom of a base station antenna wireless channel, characterized in that, include: The wave number range of the propagating wave corresponding to the transmitter is determined based on the structural parameters of the locally resonant metamaterial. The current distribution corresponding to each antenna feed port is mapped to the wavenumber domain to obtain multiple radiation spectra; The structural parameters of the locally resonant metamaterial are adjusted according to the wavenumber range of the propagating wave and the multiple radiation spectra to enhance the spatial degree of freedom of the base station antenna wireless channel.

2. The method according to claim 1, characterized in that, in, The structural parameters of the locally resonant metamaterial include array unit structural parameters, topological structural parameters, and lumped circuit parameters.

3. The method according to claim 2, characterized in that, in, The array unit structural parameters include at least one of the following: array unit shape, length of a single array unit; the topology parameters include at least one of the following: spacing between locally resonant metamaterial array units; the lumped circuit parameters include at least one of the following: load impedance value applied to each array unit.

4. The method according to claim 1, characterized in that, Mapping the current distribution at each antenna feed port to the wavenumber domain yields multiple radiation spectra, including: Fourier transform is performed on the current distribution of the current mode applied to each antenna feed port to map each current distribution into a radiation spectrum in the wavenumber domain, thus obtaining the multiple radiation spectra.

5. The method according to claim 1, characterized in that, Before adjusting the structural parameters of the locally resonant metamaterial according to the wavenumber range and multiple radiation spectra of the propagating wave, the method further includes: The first signal transmitted by the far-field receiver is subjected to time-domain inversion processing to generate an inverted signal; wherein, the first signal is transmitted by the far-field receiver according to the first orthogonal current mode signal of the transmitter, and the transmission characteristics of the first orthogonal current mode signal are affected by the local resonant metamaterial. The first orthogonal current mode signal and the second signal transmitted by the far-field receiver according to the inverted signal are respectively mapped onto an orthogonal basis, and the expansion coefficient vectors of the first orthogonal current mode signal and the second signal on the orthogonal basis are respectively determined. The first channel matrix is ​​determined based on the expansion coefficient vector of the first signal on the orthogonal basis and the expansion coefficient vector of the second signal on the orthogonal basis; Based on the eigenvalues ​​of the first channel matrix and the first preset eigenvalue threshold, it is determined that the spatial degree of freedom of the base station antenna wireless channel needs to be enhanced.

6. The method according to claim 3, characterized in that, Adjusting the structural parameters of the locally resonant metamaterial based on the wavenumber range and multiple radiation spectra of the propagating wave includes: Based on the plurality of radiation spectra and the wavenumber range of the propagating wave, the radiation spectrum corresponding to the evanescent wave is determined, wherein the radiation spectrum corresponding to the evanescent wave is the radiation spectrum that does not fall within the beam range of the propagating wave. By adjusting the structural parameters of the local resonant material, the wavenumber range of the propagating wave is expanded, so that the radiation spectrum corresponding to the evanescent wave falls within the expanded wavenumber range of the propagating wave.

7. The method according to claim 6, characterized in that, By adjusting the structural parameters of the local resonant material to expand the wavenumber range of the propagating wave, including: Determine the dispersion relation expression, wherein the dispersion relation expression includes at least the following parameters: electromagnetic wave frequency, wave number, array unit structure parameters, topology structure parameters, and lumped circuit parameters; Based on the dispersion relation expression and the electromagnetic wave frequency corresponding to the signal emitted by the transmitter, determine the array unit structure parameter values, topology parameters, and lumped circuit parameters to optimize the wavenumber range of the propagating wave. Based on the array unit structural parameter values, array unit structural parameters, topology parameters, and lumped circuit parameters for optimizing the wavenumber range of the propagating wave, the structural parameters of the locally resonant metamaterial are adjusted to expand the wavenumber range of the propagating wave.

8. The method according to claim 1, characterized in that, After adjusting the structural parameters of the locally resonant metamaterial according to the wavenumber range and multiple radiation spectra of the propagating wave, the method further includes: A third signal is transmitted to a far-field receiver based on a locally resonant metamaterial with adjusted structural parameters; wherein the frequency of the third signal covers the expanded wavenumber range of the propagating wave. Receive a fourth signal transmitted by the far-field receiver, wherein the fourth signal is the third signal after time reversal processing; The second channel matrix is ​​determined based on the third and fourth signals; Based on the eigenvalues ​​of the second channel matrix and the second preset eigenvalue threshold, it is determined that the spatial degrees of freedom of the base station antenna wireless channel need to be further enhanced.

9. The method according to claim 8, characterized in that, Determining the second channel matrix based on the third and fourth signals includes: Based on the expansion coefficient vector of the third signal on the orthogonal basis and the expansion coefficient vector of the fourth signal on the orthogonal basis; The second channel matrix is ​​determined based on the expansion coefficient vector of the third signal on the orthogonal basis and the expansion coefficient vector of the fourth signal on the orthogonal basis.

10. A holographic MIMO communication system, characterized in that, The device includes a transmitter and a locally resonant metamaterial disposed in the near-field region of a holographic multiple-input multiple-output (MIMO) system in the transmitter, the locally resonant metamaterial surrounding an antenna array in the transmitter.

11. The holographic MIMO communication system according to claim 10, characterized in that, in, The local resonant metamaterial consists of a densely distributed array of scatterers with adjustable unit parameters.

12. A computer-readable storage medium, characterized in that, The computer-readable storage medium stores a computer program, wherein the computer program, when executed by a processor, implements the steps of the method described in any one of claims 1 to 9.

13. An electronic device comprising a memory, a processor, and a computer program stored in the memory and executable on the processor, characterized in that, When the processor executes the computer program, it implements the steps of the method described in any one of claims 1 to 9.

14. A computer program product, characterized in that, Includes a computer program, which, when executed by a processor, implements the steps of the method described in any one of claims 1 to 9.

Images