Electromagnetic self-shielded inverter transformer

Abstract

This invention relates to the field of transformer self-shielding technology, specifically to an electromagnetic self-shielded inverter transformer. The invention involves sequentially mounting and fixing the secondary winding, the electrical shielding layer cylinder, and the primary winding on each phase core column of the core from the inside out. Lead-out terminals are installed on the electrical shielding layer to obtain high-frequency resonant current and zero-sequence current, analyze the three-phase imbalance, and determine the connection method of the lead-out terminals to the ground potential network. The overcurrent coefficient is determined based on the actual impact current of the primary and secondary windings, and the required total air gap length is calculated by combining the total impedance design value. The number of unsaturated air gap groups is determined according to the core structure. Low magnetic reluctance channel material is attached to both sides of each unsaturated air gap group to form a leakage flux shunt. The total inductive reactance and total capacitive reactance of the transformer are measured, and the impedance matching deviation rate is analyzed to determine if it is qualified. If not, the cross-sectional area of ​​the low magnetic reluctance channel is adjusted to improve the long-term operational stability of the inverter converter system.

Description

Technical Field

[0001] This invention relates to the field of transformer self-shielding technology, and specifically to an electromagnetic self-shielding inverter transformer. Background Technology

[0002] With the rapid development of society and economy, the advancement of science and technology, and the widespread adoption of power electronic equipment such as energy storage converters, rectifier-inverter power conversion systems have been widely used. The core power circuits of these systems extensively utilize semiconductor nonlinear switching devices such as IGBTs. The high-frequency switching process of these devices injects high-amplitude high-frequency harmonic currents into the power grid and transformer circuits. Simultaneously, load imbalance and differences in three-phase power distribution can easily lead to severe three-phase current imbalance, generating two harmful current components: high-frequency resonant current and zero-sequence current. Therefore, it is desirable to invent an electromagnetically self-shielded inverter transformer.

[0003] However, the existing technology has the following problems: 1. Existing inverter transformers rely solely on external filter capacitors and reactors to suppress harmonics, which can attenuate circuit harmonics in a simple way. However, the transformer body does not integrate a layered electric shielding cylinder structure, and the high-frequency resonant current and zero-sequence current are different under different operating conditions. It is impossible to suppress the 3rd, 5th, and 7th harmonics and zero-sequence current, and it is difficult to solve the problems of overvoltage discharge, insulation aging, and electrochemical corrosion.

[0004] 2. Existing technologies weaken DC bias by using a single iron core with a fixed-size air gap. This can slightly alleviate the iron core saturation problem under normal steady-state conditions. However, it does not take into account the objective reality that there are discrete deviations between the system's transient inrush current, the field compensation capacitor, and the stray capacitor. The iron core is prone to saturation and burnout under the action of impact and bias, and it is difficult to achieve impedance matching between inductance and capacitance, which leads to the potential for long-term resonance. Summary of the Invention

[0005] This invention aims to address the shortcomings of existing technologies by providing an electromagnetic self-shielded inverter transformer. High-frequency resonant current, zero-sequence current, and three-phase imbalance are detected through terminals led out from the electrical shielding layer, and the grounding method is selected accordingly. The air gap is designed based on the inrush current and total impedance, and a low magnetic reluctance channel is incorporated. Furthermore, the cross-sectional area is adjusted based on the deviation between inductive and capacitive reactance to suppress harmonics, resist impacts, and prevent saturation.

[0006] To achieve the above objectives, the present invention adopts the following technical solution: The present invention provides an electromagnetic self-shielded inverter transformer, comprising: sequentially mounting and fixing a secondary winding, an electric shielding layer cylinder and a primary winding on each phase core column of the iron core from the inside out, and setting lead-out terminals on the electric shielding layer to obtain high-frequency resonant current and zero-sequence current and analyze the three-phase unbalance.

[0007] The method of connecting the lead-out terminals of the electrical shielding layer to the ground potential network is determined based on the high-frequency resonant current, zero-sequence current, and three-phase unbalance.

[0008] Measure the actual impact current of the primary and secondary windings, determine the overcurrent coefficient based on the actual impact current, and inversely deduce the required total air gap length by combining the total impedance design value, and determine the number of unsaturated air gap groups according to the core structure.

[0009] Low magnetic resistance channel material is attached to both sides of each group of unsaturated air gaps, so that it spans between the iron core end faces on both sides of the air gap, forming a bypass for leakage flux.

[0010] The total inductive reactance introduced by the self-shielding structure of the magnetic circuit and the total capacitive reactance formed by the compensation capacitor and stray capacitor in the electrical system are measured. Based on the total inductive reactance and total capacitive reactance, the impedance matching deviation rate is analyzed to determine whether it is qualified. If it is not qualified, the cross-sectional area of ​​the low magnetic reluctance channel is adjusted.

[0011] Compared with the prior art, the present invention has the following beneficial effects: (1) The present invention, by sequentially mounting and fixing the secondary winding, the electric shielding layer cylinder and the primary winding on each phase core column of the iron core from the inside out, and setting the lead-out terminals on the electric shielding layer, obtains the high-frequency resonant current and zero-sequence current and analyzes the three-phase imbalance, can effectively attenuate the 3rd, 5th, 7th and above high-frequency harmonic components, weaken the superposition of common-mode interference voltage caused by IGBT switching, and realize the accurate quantification of the harmonic imbalance state of the converter system.

[0012] (2) The present invention determines the way the lead-out terminals of the electrical shielding layer are connected to the ground potential network based on the high-frequency resonant current, zero-sequence current and three-phase unbalance, avoids the mutual coupling of the shielding layers to generate new interference, reduces the probability of local overvoltage discharge, delays insulation aging, inhibits metal electrochemical corrosion, extends the service life of the transformer, and effectively suppresses high-frequency harmonics and zero-sequence current.

[0013] (3) This invention measures the actual impact current of the primary and secondary windings, determines the overcurrent coefficient based on the actual impact current, calculates the required total air gap length by combining the total impedance design value, and determines the number of unsaturated air gap groups according to the core structure. This effectively reduces the core magnetic flux density under DC bias and transient impact, prevents the core from entering the saturation region, solves the problem of core overheating and burnout, and improves the fault tolerance of the inverter system.

[0014] (4) The present invention attaches low magnetic resistance channel material to both sides of each unsaturated air gap, so that it is connected between the iron core end faces on both sides of the air gap, forming a bypass for leakage flux, reducing stray flux passing through the main magnetic circuit of the iron core, further reducing the risk of iron core saturation, suppressing eddy current and hysteresis loss, controlling the temperature rise of the iron core, balancing the magnetic potential difference on both sides of the air gap, reducing the magnetic flux distortion at the edge of the air gap, reducing the local temperature rise of the winding, and improving the uniformity of the heat distribution of the transformer.

[0015] (5) This invention measures the total inductive reactance introduced by the magnetic circuit self-shielding structure and the total capacitive reactance formed by the compensation capacitor and stray capacitor in the electrical system. Based on the total inductive reactance and total capacitive reactance, it analyzes whether the impedance matching deviation rate is qualified. If it is not qualified, it adjusts the cross-sectional area of ​​the low magnetic reluctance channel to prevent the resonance amplification of harmonics and the generation of destructive overvoltage, thereby reducing the risk of insulation breakdown and equipment damage caused by system harmonic amplification. Attached Figure Description

[0016] To more clearly illustrate the technical solutions of the embodiments of the present invention, the accompanying drawings used in the description of the embodiments will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0017] Figure 1 This is a schematic diagram of the method steps of the present invention.

[0018] Figure 2 This is a schematic diagram illustrating the specific steps of the method for determining the number of unsaturated air gap groups in this invention.

[0019] Figure 3 This is a schematic diagram of the steps for adjusting the cross-sectional area of ​​the low magnetic reluctance channel in this invention. Detailed Implementation

[0020] Various exemplary embodiments of the present invention will now be described in detail with reference to the accompanying drawings. It should be noted that, unless otherwise specifically stated, the relative arrangement, numerical expressions, and values ​​of the components and steps set forth in these embodiments do not limit the scope of the invention. Furthermore, it should be understood that, for ease of description, the dimensions of the various parts shown in the drawings are not drawn to actual scale.

[0021] The following description of at least one exemplary embodiment is merely illustrative and is in no way intended to limit the invention or its application or use. Techniques, methods, and apparatus known to those skilled in the art may not be discussed in detail, but where appropriate, such techniques, methods, and apparatus should be considered part of the specification.

[0022] In all examples shown and discussed herein, any specific values ​​should be interpreted as merely exemplary and not as limitations. Therefore, other examples of exemplary embodiments may have different values.

[0023] Please see Figure 1 As shown, the present invention provides an electromagnetic self-shielded inverter transformer, comprising: S1, a secondary winding with fixed phases, an electric shielding layer cylinder and a primary winding, and analyzes the three-phase unbalance.

[0024] Considering that the high-frequency switching of IGBTs in the inverter system continuously generates high-frequency common-mode voltage, superimposed with the zero-sequence component caused by the three-phase load imbalance, conventional transformers without layered shielding structures cannot isolate the high-frequency stray electric fields between windings and between windings and ground. High-frequency harmonics and zero-sequence currents will continuously erode the main insulation. At the same time, the harmonic amplitude and the degree of three-phase imbalance vary greatly under different operating conditions. A uniform grounding method will cause circulating current coupling and local overvoltage in the shielding layer. Therefore, this invention sets an electrical shielding layer between the primary and secondary windings of the inverter transformer and collects current components through the shielded lead-out terminals to quantify the harmonics and the degree of imbalance, providing data basis for differentiated grounding schemes.

[0025] Based on this, the specific implementation steps of S1 include: S11, sequentially mounting and fixing the secondary winding, the electric shielding layer cylinder and the primary winding on each phase core column of the iron core from the inside out.

[0026] S12, and set out terminals on the electric shielding layer to obtain high-frequency resonant current and zero-sequence current and analyze the three-phase unbalance.

[0027] The method for obtaining the high-frequency resonant current and zero-sequence current includes: S121, acquiring the instantaneous current signal of the secondary winding of each phase, transforming it to obtain a current frequency domain signal, extracting all frequency points with frequencies greater than or equal to a preset cutoff frequency, obtaining the root of the square of the amplitude of each frequency point, and recording it as the high-frequency resonant current of the corresponding phase. The current frequency domain signal is obtained by Fourier transforming the instantaneous current signal. The Fourier transform is existing technology and will not be described in detail here.

[0028] Furthermore, in a specific embodiment of the present invention, the preset cutoff frequency is set to 2kHz. When the frequency of a certain point is greater than 2kHz, that point is considered a high-frequency point. The square root of the sum of squares among all high-frequency points is recorded as the high-frequency resonant current. In this embodiment, the square root of the sum of squares is a comprehensive representation of the amplitude of all high-frequency points within the same phase, equivalent to the total effective value of the high-frequency harmonic components of that phase. The implementer may also set the preset cutoff frequency to other specific values.

[0029] S122. Extract the fundamental component from the current frequency domain signal corresponding to each phase winding, and vector synthesize the fundamental components of the three-phase winding, then use its amplitude as the zero-sequence current. The method for obtaining the fundamental components of the three-phase winding is to obtain the complex amplitude at a frequency equal to the system fundamental frequency, such as 50Hz. Specifically, vector synthesis involves summing the fundamental components of the three-phase winding and then multiplying the sum by 1 / 3.

[0030] In addition, it should be noted that the analysis method for the three-phase unbalance includes: W1, extracting the effective value of the fundamental current from the frequency domain signal of the current corresponding to each phase winding, and calculating the average value of the effective value of the fundamental current of each phase.

[0031] W2. Calculate the absolute deviation between the effective value and the average value of the current in each phase, and record it as the fundamental current deviation of the corresponding phase.

[0032] W3. Obtain the maximum value of the fundamental current deviation of each phase, and record the ratio of the maximum value to the average value of the effective value of the fundamental current of each phase as the three-phase unbalance.

[0033] This invention obtains high-frequency resonant current and zero-sequence current and analyzes three-phase imbalance by sequentially mounting and fixing secondary windings, electrical shielding cylinders and primary windings on each phase core column of the iron core from the inside out, and setting lead terminals on the electrical shielding layer. It can effectively attenuate 3rd, 5th, 7th and above high-frequency harmonic components, weaken the superposition of common-mode interference voltage caused by IGBT switching, and achieve accurate quantification of the harmonic imbalance state of the converter system.

[0034] S2. Determine the method of connecting the lead-out terminals of the electrical shielding layer to the ground potential network based on the high-frequency resonant current, zero-sequence current and three-phase unbalance.

[0035] Considering that the requirements for the flow path of the induced interference current in the shielding layer are completely different under different operating conditions: under severe harmonic zero-sequence three-phase imbalance, star grounding can discharge the unbalanced zero-sequence component through the neutral point resistor; when there is only slight three-phase imbalance and no high-frequency large harmonics, the open delta structure can close the loop to cancel most of the third zero-sequence harmonics; under steady-state balance, independent grounding can avoid mutual coupling between shielding layers to avoid generating new electromagnetic interference. Therefore, it is necessary to match the corresponding grounding topology in three levels according to the three types of current index thresholds.

[0036] Based on this, the specific implementation steps of S2 include: S21, when the high-frequency resonant current of a certain phase is greater than or equal to a set high-frequency resonant current threshold, or the zero-sequence current is greater than or equal to a set zero-sequence current threshold, and the three-phase imbalance is greater than or equal to a set imbalance threshold, then the leads of each phase are first connected in a star configuration, and the neutral point is connected to the ground potential network through a resistor. The high-frequency resonant current and zero-sequence current absorbed by the self-shielding network of the three-phase star circuit are first absorbed through the three-phase star (Y) circuit. N After the vectors connected by the (shape) cancel each other out, the unbalanced current is absorbed by the resistor or low impedance element and then flows directly into the ground potential network, where it is directly absorbed by the ground potential network itself.

[0037] S22. If the high-frequency resonant current of a certain phase is greater than or equal to the set high-frequency resonant current threshold, or the zero-sequence current is greater than or equal to the set zero-sequence current threshold, and the three-phase imbalance is less than the set imbalance threshold, then the leads of each phase are first connected into an open delta configuration. A resistor is then connected at the opening of the open delta before connecting to the ground potential network. The high-frequency resonant current and zero-sequence current absorbed by the three-phase open delta (Z-shaped) self-shielding network are first vector-canceled through the three-phase open delta (Z-shaped) connection. The unbalanced current is then absorbed by a resistor or low-impedance element and flows directly into the ground potential network, where it is directly absorbed by the ground potential network.

[0038] S23. When the high-frequency resonant current of each phase is less than the set high-frequency resonant current threshold, the zero-sequence current is less than the set zero-sequence current threshold, and the three-phase imbalance is less than the set imbalance threshold, then the leads of the three independent circuits are connected to the ground potential network separately. The high-frequency resonant current and zero-sequence current absorbed by the electric shielding network of the three independent circuits of the three phases flow directly into the ground potential network and are uniformly absorbed by the ground potential network.

[0039] It should be noted that the above-mentioned high-frequency resonant current threshold and zero-sequence current threshold can be determined based on the rated current of the inverter transformer and the system's allowable limit on harmonic content, typically taking 5%~10% and 2%~5% of the rated current, respectively; the unbalance threshold can be determined based on the allowable unbalance of national standards, typically taking 10%~15%; the above thresholds are technical parameters that can be conventionally determined by those skilled in the art based on specific application scenarios, and the specific values ​​should not be construed as limitations on the present invention.

[0040] This invention determines the connection method of the lead-out terminals of the electrical shielding layer to the ground potential network based on high-frequency resonant current, zero-sequence current and three-phase unbalance. This avoids mutual coupling of the shielding layers and the generation of new interference, reduces the probability of local overvoltage discharge, delays insulation aging, inhibits metal electrochemical corrosion, extends the service life of the transformer, and effectively suppresses high-frequency harmonics and zero-sequence current.

[0041] S3. Determine the overcurrent coefficient and the number of unsaturated air gap groups based on the actual impact current of the primary and secondary windings.

[0042] Considering that inverter transformers will frequently withstand instantaneous surge currents caused by startup, short circuits, and sudden load changes, fixed-size air gaps can only be adapted to rated steady-state conditions. Under impact, the magnetic flux is very likely to exceed the core's saturation flux density, causing the core to overheat and burn out. At the same time, the total length of the air gap is constrained by both the anti-saturation lower limit and the system impedance matching target. A single wide air gap will exacerbate the distortion of the magnetic circuit edge and increase local eddy current losses. The mechanical strength of the core also limits the maximum width of a single air gap. Therefore, it is necessary to determine the overcurrent coefficient in conjunction with the surge current, calculate the total length of the air gap under the dual constraints, and then split it into multiple narrow unsaturated air gaps to balance the magnetic circuit performance and mechanical reliability.

[0043] Based on this, the specific implementation steps of S3 are as follows: S31, measure the actual impact current of the primary and secondary windings, and determine the overcurrent coefficient based on the actual impact current. The method for determining the overcurrent coefficient is as follows: S31, collect the peak current of the primary and secondary windings under the set operating conditions, record the ratio of each peak current to the corresponding rated current as the corresponding impact current multiple, and take the maximum value as the overcurrent coefficient.

[0044] S32. Based on the total impedance design value, deduce the required total air gap length, and determine the number of unsaturated air gap groups according to the core structure. For example... Figure 2 As shown, the specific implementation steps include: S321, obtaining the number of turns of the primary winding, overcurrent coefficient, rated current and saturation magnetic flux density of the core material of each phase of the inverter transformer, and comprehensively calculating the lower limit of the required total air gap length.

[0045] The formula for calculating the lower limit of the required total air gap length is as follows:

[0046] ;

[0047] in This represents the lower limit of the required total air gap length. Representing the vacuum permeability, in this embodiment it is taken as 4π × 10⁻⁶. - 7 H / m, Represents the number of turns in the primary winding. Represents the current coefficient. Represents the rated current. Representing the saturation magnetic flux density, this formula is based on the design principle that under an impact of K times the rated current, the magnetic flux density of the core must never exceed its saturation magnetic flux density, and the formula for calculating the magnetic flux density of the core is: ;in The magnetic flux density representing the magnetic core, according to the inequality ,roll out Therefore, the formula for the lower limit of the required total air gap length is obtained. The formula for calculating the magnetic flux density of the magnetic core is existing technology and will not be described in detail in this invention. The effective cross-sectional area of ​​the magnetic core and the number of turns of the primary winding are determined according to the rated voltage, rated capacity and selected core model of the inverter transformer, according to the conventional transformer electromagnetic design formula. These are well-known design input parameters in the field and will not be described in detail in this invention.

[0048] S322. Obtain the total impedance design value. Based on the total impedance design value, the effective cross-sectional area of ​​the magnetic core and the number of turns of the primary winding, determine the target value of the required total air gap length for the corresponding phase. Take the maximum value between the target value and the lower limit of the required total air gap length as the required total air gap length.

[0049] The formula for calculating the target value of the required total air gap length is as follows:

[0050] ;

[0051] in This represents the target value for the total required air gap length. Represents the total impedance design value. Represents the effective cross-sectional area of ​​the magnetic core. Represents the system's rated angular frequency, where This represents the fundamental frequency of the system, determined by the applicable power system standard, and is typically taken as 50Hz. This is because the magnetizing inductance of the air-gap core and the total length of the air gap satisfy the following inverse relationship: And because of the magnetizing inductance Therefore, the expression for the required total air gap length target value is derived, where This represents the magnetizing inductor with an air gap core.

[0052] S323. Determine the first group number based on the number of phases. Obtain the maximum allowable gap width of a single air gap from the core mechanical strength requirements. Record the ratio of the required total air gap length to the maximum allowable gap width as the second group number. The method for determining the first group number is to use twice the number of phases as the first group number, ensuring that each phase has at least one unsaturated air gap at its first and second ends, thereby guaranteeing the symmetry and mechanical balance of the magnetic circuit structure of each phase. In this embodiment, the three-phase transformer has 3 phases, therefore the first group number is 6. The required total air gap length and the maximum allowable gap width are both dimensions of the air gap along the core magnetic circuit direction. The required total air gap length is the sum of the gap widths of all air gaps.

[0053] S324. If the number in the first group is greater than or equal to the number in the second group, then the number in the first group is taken as the number of unsaturated air gap groups; otherwise, the number in the second group is calculated by comparing the number in the second group with the number of phases and rounding up to obtain the number of unsaturated air gap groups.

[0054] This invention measures the actual impact current of the primary and secondary windings, determines the overcurrent coefficient based on the actual impact current, and back-calculates the required total air gap length by combining the total impedance design value. It also determines the number of unsaturated air gap groups based on the core structure. This effectively reduces the core magnetic flux density under DC bias and transient impact, prevents the core from entering the saturation region, solves the problems of core overheating and burnout, and improves the fault tolerance of the inverter system.

[0055] S4. Attach low magnetic resistance channel material to both sides of each group of unsaturated air gaps, so that it spans between the iron core end faces on both sides of the air gap, forming a bypass for leakage flux.

[0056] Considering that the magnetic reluctance of the air at the unsaturated air gap is much greater than that of the silicon steel core, a large amount of leakage flux will diffuse outward from the edge of the air gap, resulting in severe edge flux distortion, increasing winding eddy currents, hysteresis losses, and local temperature rise. The magnetic potential difference on both sides of the air gap will also exacerbate stray flux passing through the main magnetic circuit, further increasing the risk of core saturation. Simply increasing the number of air gap groups can only slightly alleviate this problem and cannot divert leakage flux across the air gap. Therefore, a high-permeability, low-resistance silicon steel bypass is connected across the end face of each air gap group to provide a low-resistance diversion channel for leakage flux, weaken the edge effect, and balance the magnetic potential and temperature rise.

[0057] Based on this, the preferred material for the low magnetic reluctance channel is a high-permeability cold-rolled silicon steel sheet, which is bound together with semi-dry glass adhesive tape and then dried and cured. This material is made into a sheet and is tightly adhered to and spans the core end faces on both sides of the unsaturated air gap through adhesive bonding, forming good magnetic contact. Its function is to provide a parallel bypass with a magnetic reluctance much lower than that of air for the leakage flux that would otherwise need to cross the air gap. According to the principle of parallel magnetic circuit current diversion, most of the leakage flux will be diverted to this low magnetic reluctance channel, thereby effectively suppressing the adverse effects of the air gap edge effect on the surrounding assembly and reducing the resulting eddy current losses.

[0058] This invention attaches a low magnetic reluctance channel material to both sides of each unsaturated air gap, bridging the core end faces on both sides of the air gap to form a bypass for leakage flux. This reduces stray flux passing through the main magnetic circuit of the core, further reducing the risk of core saturation, suppressing eddy current and hysteresis losses, controlling core temperature rise, balancing the magnetic potential difference on both sides of the air gap, reducing magnetic flux distortion at the air gap edge, reducing local temperature rise in the windings, and improving the uniformity of transformer heat distribution.

[0059] S5. Obtain the total inductive reactance and total capacitive reactance values, and analyze whether the impedance matching deviation rate is qualified.

[0060] Considering that the switching of on-site compensation capacitors and changes in line length can cause discrete fluctuations in the total capacitive reactance of the system, the cross-sectional area of ​​the low reluctance channel directly changes the equivalent excitation inductance of the magnetic circuit, thereby changing the total inductive reactance of the system. If the deviation between the total inductive reactance and the total capacitive reactance is too large, parallel resonance can easily form, and harmonics will be amplified by resonance to generate destructive overvoltages and break down the transformer insulation. The fixed cross-sectional area channel cannot adapt to changes in capacitive reactance on-site. Therefore, the total inductive reactance is measured by short-circuit test, the total capacitive reactance is obtained by capacitance measurement, the matching deviation rate is calculated, and the cross-sectional area of ​​the low reluctance channel is increased or decreased according to the magnitude of the deviation to dynamically correct the magnetic circuit inductive reactance and achieve impedance matching throughout the entire process.

[0061] Based on this, the specific implementation steps of S5 include: S51, measuring the total inductive reactance introduced by the magnetic circuit self-shielding structure and the total capacitive reactance formed by the compensation capacitor and stray capacitor in the electrical system. In a specific embodiment of the present invention, the method for obtaining the total inductive reactance includes: S511, applying a low voltage of rated frequency to the primary winding under a short-circuit condition on the secondary winding of the inverter transformer, adjusting the voltage to bring the primary current to its rated value, and measuring the effective value of the primary input voltage, the effective value of the input current, and the input active power.

[0062] S512. Obtain the ratio of the effective value of the input voltage to the effective value of the input current, and record it as the short-circuit impedance. Calculate the short-circuit resistance based on the effective value of the input active power and the input current, and calculate the short-circuit reactance in combination with the short-circuit impedance.

[0063] The short-circuit resistance is the ratio of the input active power to the square of the effective value of the input current under short-circuit conditions. The square root of the difference between the squares of the short-circuit resistance and the short-circuit impedance is denoted as the short-circuit reactance.

[0064] S513. Based on the phase voltages of the primary and secondary windings of the inverter transformer, determine the actual turns ratio of the primary and secondary windings. Based on the actual turns ratio, calculate the short-circuit reactance to the secondary side to obtain the total inductive reactance value. The specific steps for calculating the short-circuit reactance to the secondary side are as follows: In the short-circuit test, due to the low applied voltage, the core flux density is far below the saturation point, and the excitation current accounts for a very small proportion of the rated current. Its shunt effect is negligible within the allowable range of engineering errors. In this embodiment, the short-circuit reactance is approximately equivalent to the total inductive reactance. Therefore, the ratio of the short-circuit reactance to the square of the actual turns ratio is recorded as the total inductive reactance value.

[0065] In addition, the method for obtaining the total capacitive reactance value includes: T1, using a capacitance meter to measure the capacitance value of each centralized compensation capacitor in the power system, and summing them to obtain the total compensation capacitance.

[0066] T2. Disconnect the line or equipment under test from the system, use a capacitance meter to directly measure its stray capacitance to ground and sum the stray capacitances to obtain the total stray capacitance.

[0067] T3. Calculate the sum of the compensation capacitance and the stray capacitance, and record it as the total system capacitance. Combine this with the system fundamental frequency to obtain the total capacitive reactance of the system.

[0068] The formula for calculating the total capacitive reactance of the system is as follows:

[0069] ;

[0070] in, Represents the total capacitive reactance of the system. This represents the total capacitance of the system. This formula is existing technology and will not be described in detail here.

[0071] S52. Analyze whether the impedance matching deviation rate is qualified based on the total inductive reactance and total capacitive reactance values. If it is not qualified, adjust the cross-sectional area of ​​the low magnetic reluctance channel.

[0072] In a specific embodiment of the present invention, the method for determining whether the impedance matching deviation rate is qualified includes: S521, calculating the absolute difference between the total inductive reactance value and the total capacitive reactance value, and recording the ratio of the difference to the total capacitive reactance value as the impedance matching deviation rate.

[0073] S522. If the impedance matching deviation rate is less than or equal to the set deviation threshold, the deviation is deemed acceptable, and the cross-sectional area of ​​the low magnetic reluctance channel is not adjusted. In this embodiment, the deviation threshold is set to 5%. The implementer may also set other specific values, but it should not be too small, as this would result in excessively high requirements for the air gap processing accuracy and the adjustment accuracy of the low magnetic reluctance channel cross-sectional area, significantly increasing manufacturing costs and scrap rate. It should also not be too high, as this would result in insufficient matching between inductive and capacitive reactance, reducing the suppression effect on target frequency harmonics.

[0074] S523. If the impedance matching deviation rate is greater than the set deviation rate threshold, the deviation rate is deemed unqualified, and the cross-sectional area of ​​the low magnetic reluctance channel is adjusted based on the total inductive reactance and total capacitive reactance values. For example... Figure 3As shown, the specific implementation steps include: S5231, obtaining the change in inductive reactance corresponding to the change in unit area from the transformer system's backend database. The change in inductive reactance corresponding to the change in unit area refers to the change in the total inductive reactance introduced by the magnetic circuit self-shielding structure when the cross-sectional area of ​​the low magnetic reluctance channel changes by a unit area (e.g., 1 cm²), with units of Ω / cm². Multiple sample transformers of the same model with different low magnetic reluctance channel cross-sectional areas can be manufactured according to a set interval. The total inductive reactance value on the secondary side of each sample transformer is measured using the short-circuit test method in step S511. The change in total inductive reactance value corresponding to adjacent low magnetic reluctance channel cross-sectional areas is obtained. The ratio of this change to the set interval is recorded as the inductive reactance change coefficient. The average of all inductive reactance change coefficients corresponding to the same model transformer is calculated to obtain the inductive reactance change corresponding to the change in unit area for the corresponding model, and this value is stored in the transformer system's backend database.

[0075] S5232. Obtain the absolute difference between the total inductive reactance value and the total capacitive reactance value, and record the ratio of this difference to the change in inductive reactance corresponding to the change in unit area as the cross-sectional area adjustment amount of the low magnetic reluctance channel.

[0076] S5233. If the total inductive reactance is greater than the total capacitive reactance, reduce the cross-sectional area of ​​the low magnetic reluctance channel according to the adjustment amount of the low magnetic reluctance channel.

[0077] S5234. If the total inductive reactance is less than the total capacitive reactance, increase the cross-sectional area of ​​the low-resistance channel by the adjustment amount. According to Ohm's law for magnetic circuits, when the cross-sectional area of ​​the low-resistance channel increases, the magnetic reluctance decreases, the inductance increases, and the inductive reactance increases accordingly; conversely, when the cross-sectional area decreases, the magnetic reluctance increases, the inductance decreases, and the inductive reactance decreases accordingly. Therefore, when the total inductive reactance is greater than the total capacitive reactance, the total inductive reactance needs to be reduced, so the cross-sectional area of ​​the low-resistance channel is reduced by the aforementioned adjustment amount. Similarly, when the total inductive reactance is less than the total capacitive reactance, the total inductive reactance needs to be increased, so the cross-sectional area of ​​the reluctance channel is increased by the aforementioned adjustment amount.

[0078] This invention measures the total inductive reactance introduced by the self-shielding structure of the magnetic circuit and the total capacitive reactance formed by the compensation capacitor and stray capacitor in the electrical system. Based on the total inductive reactance and total capacitive reactance values, it analyzes whether the impedance matching deviation rate is qualified. If it is not qualified, it adjusts the cross-sectional area of ​​the low magnetic reluctance channel to prevent the resonance amplification of harmonics and the generation of destructive overvoltages, thereby reducing the risk of insulation breakdown and equipment damage caused by system harmonic amplification.

[0079] The above embodiments can be implemented, in whole or in part, by software, hardware, firmware, or any other combination thereof. When implemented using software, the above embodiments can be implemented, in whole or in part, in the form of a computer program product.

[0080] Those skilled in the art will recognize that the modules and algorithm steps of the various examples described in conjunction with the embodiments disclosed herein can be implemented in electronic hardware, or a combination of computer software and electronic hardware. Whether these functions are implemented in hardware or software depends on the specific application and design constraints of the technical solution. Those skilled in the art can use different methods to implement the described functions for each specific application, but such implementation should not be considered beyond the scope of this application.

[0081] In addition, the functional modules in the various embodiments of this application can be integrated into one processing module, or each module can exist physically separately, or two or more modules can be integrated into one module.

[0082] The above description is merely a specific embodiment of this application, but the scope of protection of this application is not limited thereto. Any variations or substitutions that can be easily conceived by those skilled in the art within the scope of the technology disclosed in this application should be included within the scope of protection of this application. Therefore, the scope of protection of this application should be determined by the scope of the claims.

[0083] Finally, the above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the protection scope of the present invention.

Claims

1. An electromagnetic self-shielded inverter transformer, characterized in that, include: The secondary winding, the electric shielding layer cylinder and the primary winding are sequentially mounted and fixed on each phase core column of the iron core from the inside to the outside. Lead-out terminals are set on the electric shielding layer to obtain high-frequency resonant current and zero-sequence current and analyze the three-phase unbalance. The method of connecting the lead-out terminals of the electrical shielding layer to the ground potential network is determined based on the high-frequency resonant current, zero-sequence current, and three-phase unbalance. Measure the actual impact current of the primary and secondary windings, determine the overcurrent coefficient based on the actual impact current, and back-calculate the required total air gap length by combining the total impedance design value, and determine the number of unsaturated air gap groups according to the core structure. Low magnetic resistance channel material is attached to both sides of each group of unsaturated air gaps, so that it is connected between the iron core end faces on both sides of the air gap to form a bypass for leakage flux. The total inductive reactance introduced by the self-shielding structure of the magnetic circuit and the total capacitive reactance formed by the compensation capacitor and stray capacitor in the electrical system are measured. Based on the total inductive reactance and total capacitive reactance, the impedance matching deviation rate is analyzed to determine whether it is qualified. If it is not qualified, the cross-sectional area of ​​the low magnetic reluctance channel is adjusted.

2. The electromagnetic self-shielded inverter transformer according to claim 1, characterized in that, The methods for obtaining the high-frequency resonant current and zero-sequence current include: The instantaneous current signal of the secondary winding of each phase is collected, and it is transformed to obtain the current frequency domain signal. All frequency points with frequencies greater than or equal to the preset cutoff frequency are extracted from it. The square root of the amplitude of each frequency point is obtained and recorded as the high-frequency resonant current of the corresponding phase. The fundamental component is extracted from the current frequency domain signal corresponding to each phase winding. After vector synthesis of the fundamental components of the three-phase winding, its amplitude is used as the zero-sequence current.

3. The electromagnetic self-shielded inverter transformer according to claim 2, characterized in that, The analysis method for the three-phase imbalance includes: Extract the effective value of the fundamental current from the frequency domain signal corresponding to each phase winding, and calculate the average value of the effective value of the fundamental current of each phase. Calculate the absolute deviation between the effective value and the average value of the current in each phase, and record it as the fundamental current deviation of the corresponding phase. The maximum value of the fundamental current deviation of each phase is obtained, and the ratio of this value to the average value of the effective value of the fundamental current of each phase is recorded as the three-phase unbalance.

4. The electromagnetic self-shielded inverter transformer according to claim 3, characterized in that, The method for determining the connection method of the ground potential network includes: When the high-frequency resonant current of a certain phase is greater than or equal to the set high-frequency resonant current threshold, or the zero-sequence current is greater than or equal to the set zero-sequence current threshold, and the three-phase unbalance is greater than or equal to the set unbalance threshold, the lead terminals of each phase are first connected in a star configuration, and the neutral point is connected to the ground potential network through a resistor. If the high-frequency resonant current of a certain phase is greater than or equal to the set high-frequency resonant current threshold, or the zero-sequence current is greater than or equal to the set zero-sequence current threshold, and the three-phase unbalance is less than the set unbalance threshold, then the lead terminals of each phase are first connected into an open delta, and a resistor is connected at the opening of the open delta before connecting to the ground potential network. When the high-frequency resonant current of each phase is less than the set high-frequency resonant current threshold, the zero-sequence current is less than the set zero-sequence current threshold, and the three-phase imbalance is less than the set imbalance threshold, the leads of the three independent circuits are connected to the ground potential network separately.

5. The electromagnetic self-shielded inverter transformer according to claim 1, characterized in that, The method for determining the overcurrent coefficient includes: The peak current of the primary and secondary windings under the set operating conditions is collected separately, and the ratio of the peak current to the corresponding rated current is recorded as the corresponding impact current multiple. The maximum value is taken as the overcurrent coefficient.

6. An electromagnetic self-shielded inverter transformer according to claim 5, characterized in that, The method for determining the number of unsaturated air gap groups includes: Obtain the number of turns, overcurrent coefficient, rated current, and saturation magnetic flux density of the core material of each phase of the inverter transformer, and comprehensively calculate the lower limit of the required total air gap length. Obtain the total impedance design value, and determine the target value of the required total air gap length for the corresponding phase based on the total impedance design value, the effective cross-sectional area of ​​the magnetic core and the number of turns of the primary winding. The maximum value between this target value and the lower limit of the required total air gap length is taken as the required total air gap length. The first group number is determined based on the number of phases. The maximum allowable gap width of a single air gap is obtained from the mechanical strength requirements of the iron core. The ratio of the required total air gap length to the maximum allowable gap width is recorded as the second group number. If the number in the first group is greater than or equal to the number in the second group, then the number in the first group is taken as the number of unsaturated air gap groups; otherwise, the number in the second group is calculated by comparing the number in the second group with the number of phases and rounding up to obtain the number of unsaturated air gap groups.

7. The electromagnetic self-shielded inverter transformer according to claim 1, characterized in that, The method for obtaining the total inductively coupled plasma (TIP) value includes: With the secondary winding of the inverter transformer short-circuited, a low voltage of rated frequency is applied to the primary winding, the voltage is adjusted to make the primary current reach the rated value, and the effective value of the primary input voltage, the effective value of the input current, and the input active power are measured. Obtain the ratio of the effective value of the input voltage to the effective value of the input current, and record it as the short-circuit impedance. Calculate the short-circuit resistance based on the effective value of the input active power and the input current, and calculate the short-circuit reactance in combination with the short-circuit impedance. The actual turns ratio of the primary and secondary windings of the inverter transformer is determined based on the phase voltages of the primary and secondary windings. The short-circuit reactance is then transferred to the secondary side based on the actual turns ratio to obtain the total inductive reactance value.

8. The electromagnetic self-shielded inverter transformer according to claim 1, characterized in that, The method for obtaining the total capacitive reactance value includes: The capacitance values ​​of each centralized compensation capacitor in the power system are measured using a capacitance meter, and the total compensation capacitance is obtained by summing the values. Disconnect the line or equipment under test from the system, use a capacitance meter to directly measure its stray capacitance to ground and sum them to obtain the total stray capacitance; Calculate the sum of the compensation capacitance and the stray capacitance, and denote it as the total system capacitance. Combine this with the system fundamental frequency to obtain the total capacitive reactance of the system.

9. An electromagnetic self-shielded inverter transformer according to claim 1, characterized in that, The methods for determining whether the impedance matching deviation rate is qualified include: Calculate the absolute difference between the total inductive reactance and the total capacitive reactance, and record the ratio of this difference to the total capacitive reactance as the impedance matching deviation rate. If the impedance matching deviation rate is less than or equal to the set deviation threshold, the deviation is deemed acceptable, and the cross-sectional area of ​​the low magnetoresistance channel is not adjusted. If the impedance matching deviation rate is greater than the set deviation rate threshold, the deviation rate is deemed unqualified, and the cross-sectional area of ​​the low magnetic reluctance channel is adjusted based on the total inductive reactance and total capacitive reactance values.

10. An electromagnetic self-shielded inverter transformer according to claim 9, characterized in that, The method for adjusting the cross-sectional area of ​​the low magnetic reluctance channel includes: Obtain the change in inductive reactance corresponding to the change in unit area from the transformer system's backend database; Obtain the absolute difference between the total inductive reactance and the total capacitive reactance, and record the ratio of this difference to the change in inductive reactance corresponding to the change in unit area as the cross-sectional area adjustment amount of the low magnetic reluctance channel. If the total inductive reactance is greater than the total capacitive reactance, reduce the cross-sectional area of ​​the low reluctance channel by adjusting the cross-sectional area of ​​the low reluctance channel. If the total inductive reactance is less than the total capacitive reactance, increase the cross-sectional area of ​​the low magnetic reluctance channel according to the adjustment amount of the low magnetic reluctance channel.

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