Dynamic regulation device and method for exciting inductance of LLC resonant converter

Abstract

This invention discloses a dynamic adjustment device and method for the excitation inductance of an LLC resonant converter. The magnetically controlled variable inductor includes a magnetic core, a main power winding, and a bias winding. The bias winding is wound on both sides of the magnetic core and connected to an AC bias current sampling circuit for injecting power frequency AC bias current. The main power winding is wound in the middle of the magnetic core and connected to the LLC resonant converter. The voltage gain and instantaneous transmission power required by the LLC resonant converter at different power frequency phases are calculated, and the inductance ratio K of the excitation inductance to the resonant inductance and the characteristic impedance Z are obtained. r Based on different inductance ratios K and characteristic impedance Z r The combined data yields K-Z values ​​that satisfy the current power frequency phase conditions. r Feasible solution domain, based on K-Z r The characteristic curves of the target excitation inductance and bias current are obtained from the feasible solution domain. The target bias current is then calculated based on the characteristic curves. By dynamically adjusting the parameters, the bias winding obtains an AC bias current corresponding to the target bias current.

Description

Technical Field

[0001] This invention relates to the field of power electronic converters and magnetically controlled variable inductors, and particularly to a device and method for dynamically adjusting the excitation inductance of an LLC resonant converter. Background Technology

[0002] LLC resonant converters are widely used in AC-DC power supplies and on-board chargers due to their advantages such as soft-switching characteristics, high efficiency, and high power density. Their topology is as follows: Figure 1 As shown. For a single-stage AC-DC LLC converter without a large-capacity DC bus capacitor or using a small-value bus capacitor, the rectified input voltage exhibits a double-power-frequency pulsation characteristic, causing the required voltage gain and instantaneous power transfer of the converter to continuously change within one power-frequency cycle. The required voltage gain G of the converter is... req and instantaneous power P in The variation curve within the power frequency cycle is as follows: Figure 2 As shown.

[0003] The resonant parameters of a traditional single-stage AC-DC LLC resonant converter are typically designed based on the worst-case operating conditions. For example, the ratio of the magnetizing inductance to the resonant inductance, K=L, is determined under the conditions of highest voltage gain, maximum output power, and lowest input voltage. m / L r and characteristic impedance Z r While this method can ensure that the converter meets voltage gain and power transfer requirements under all operating conditions, it severely limits the design of resonant parameters under extreme conditions. In non-extreme conditions, such as the non-peak range of AC input voltage, light load range, or range with lower gain requirements, traditional fixed resonant parameters cannot fully utilize the available parameter space, easily leading to an excessively large effective value of the resonant current, increased circulating current loss, increased conduction loss and magnetic component loss, thereby reducing the overall efficiency of the converter.

[0004] Existing variable resonant element solutions mainly include switched capacitors, switched inductors, and magnetically controlled variable inductors. Among them, capacitor or inductor solutions based on switching regulation usually suffer from problems such as a large number of components, high cost, discontinuous adjustment, and complex control. Existing DC bias magnetically controlled inductors are usually used to set a fixed inductance value or for slower operating condition switching. Their bias current usually does not change continuously with the power frequency phase, making it difficult to adjust the excitation inductance in real time according to the instantaneous input voltage, instantaneous power, and required gain of a single-stage AC-DC LLC converter within the power frequency cycle.

[0005] Therefore, there is an urgent need to propose a new dynamic adjustment scheme for resonant parameters, so that the LLC resonant converter can dynamically adjust the equivalent inductance parameters according to the input voltage, output power and required gain within the power frequency cycle, thereby reducing the effective value of the resonant current during the power frequency cycle and improving the efficiency under all operating conditions. Summary of the Invention

[0006] In view of the above-mentioned shortcomings in the current power electronic converter and magnetically controlled variable inductor technology, the present invention provides a device and method for dynamically adjusting the excitation inductance of an LLC resonant converter. By injecting an AC bias current that varies with the power frequency cycle into the bias winding of the magnetically controlled variable excitation inductor, the excitation inductance of the LLC resonant converter is dynamically changed within the power frequency cycle, thereby reducing the effective value of the resonant current, reducing circulating current loss and improving converter efficiency.

[0007] To achieve the above objectives, the embodiments of the present invention adopt the following technical solutions:

[0008] A dynamic adjustment device for the excitation inductance of an LLC resonant converter, implemented based on the LLC resonant converter, includes an inductor module connected to the LLC resonant converter. The inductor module includes an AC bias current sampling circuit and a magnetically controlled variable inductor. The magnetically controlled variable inductor includes a magnetic core, a main power winding, and a bias winding. The bias winding is wound on both sides of the magnetic core and connected to the AC bias current sampling circuit for injecting power frequency AC bias current. The main power winding is wound in the middle of the magnetic core and connected to the LLC resonant converter. The device calculates the voltage gain and instantaneous transmission power required by the LLC resonant converter at different power frequency phases, obtaining the inductance ratio K of the excitation inductance to the resonant inductance and the characteristic impedance Z. r Based on different inductance ratios K and characteristic impedance Z r The combined data yields KZ that satisfies the current power frequency phase condition. r Feasible solution domain, based on KZ r The characteristic curves of the target excitation inductance and bias current are obtained from the feasible solution domain. The target bias current is then calculated based on the characteristic curves. By dynamically adjusting the parameters, the bias winding obtains an AC bias current corresponding to the target bias current.

[0009] According to one aspect of the present invention, the AC bias current sampling circuit includes a current transformer and a compensation capacitor. The primary winding of the current transformer is connected in series with the input current of an LLC resonant converter. One end of the secondary winding of the current transformer is connected to a magnetically controlled variable inductor through the compensation capacitor, and the other end is connected to a magnetically controlled variable inductor.

[0010] According to one aspect of the invention, the compensation capacitor is selected based on the equivalent inductance of the bias winding and the target power frequency angular frequency, thereby reducing the equivalent impedance of the bias injection circuit at the power frequency.

[0011] According to one aspect of the present invention, the AC bias current sampling circuit includes a clamping circuit for preventing core saturation. The clamping circuit is configured as a bidirectional Zener diode, one end of which is connected to one end of the secondary winding and a compensation capacitor, and the other end of which is connected to the other end of the secondary winding.

[0012] According to one aspect of the invention, the inductance ratio K of the magnetizing inductor and the resonant inductor and the characteristic impedance Z are obtained. r This includes: establishing a time-domain model of the LLC resonant converter based on voltage gain and instantaneous transmitted power, and calculating the inductance ratio K and characteristic impedance Z for different inductance ratios. r Voltage gain, soft-switching conditions, effective value of resonant current, and device voltage stress under the combined conditions.

[0013] According to one aspect of the invention, the KZ-based r The feasible solution domain yields the characteristic curves of the target excitation inductance versus bias current, including: in the KZ... r Within the feasible solution domain, with the minimum effective value of the resonant current as the objective, the ratio of the target inductance is selected, and the target magnetizing inductance is calculated.

[0014] According to one aspect of the present invention, the dynamic adjustment parameters include: current transformer sampling injection, direct injection of AC input current, or active current source injection.

[0015] According to one aspect of the present invention, the magnetic core includes a central column and two side columns next to the central column, wherein a main power winding is wound on the central column and a bias winding is wound on the side columns.

[0016] According to one aspect of the present invention, the main power winding includes a transformer primary winding and a transformer secondary winding, wherein the two ends of the transformer primary winding and the transformer secondary winding serve as the output terminals of a magnetically controlled variable inductor and are respectively connected to the primary power circuit and the secondary power circuit of an LLC resonant converter.

[0017] A method for dynamically adjusting the excitation inductance of an LLC resonant converter includes an inductor module connected to the LLC resonant converter. The inductor module includes an AC bias current sampling circuit and a magnetically controlled variable inductor. The magnetically controlled variable inductor includes a magnetic core, a main power winding, and a bias winding. The bias winding is wound on both sides of the magnetic core and connected to the AC bias current sampling circuit for injecting power frequency AC bias current. The main power winding is wound on the center column of the magnetic core and connected to the LLC resonant converter. The method further includes the following steps:

[0018] Calculate the voltage gain and instantaneous power transfer required by the LLC resonant converter at different power frequency phases;

[0019] Calculate the inductance ratio K and characteristic impedance Z of the magnetizing inductor and resonant inductor based on voltage gain and instantaneous transmitted power.r ;

[0020] Based on different inductance ratios K and characteristic impedances Z r The combined data yields KZ that satisfies the current power frequency phase condition. r Feasible solution domain;

[0021] Based on KZ r The characteristic curves of the target excitation inductance and bias current are obtained from the feasible solution domain, and the target bias current is calculated inversely based on the characteristic curves.

[0022] By dynamically adjusting the parameters, the bias winding can obtain an AC bias current corresponding to the target bias current.

[0023] The advantages of this invention are: by injecting an AC bias current that varies with the power frequency cycle into the bias winding of the magnetically controlled variable excitation inductor, the excitation inductance of the LLC resonant converter changes dynamically within the power frequency cycle, thereby reducing the effective value of the resonant current, reducing circulating current loss, and improving converter efficiency. Attached Figure Description

[0024] To more clearly illustrate the technical solutions in the embodiments of the present invention, the accompanying drawings used in the embodiments will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0025] Figure 1 This is a topology diagram of a conventional unipolar AC-DC LLC resonant converter in the existing technology;

[0026] Figure 2 This is a graph showing the changes in input / output voltage gain and instantaneous power over half a power frequency cycle in the existing technology;

[0027] Figure 3 This is a graph showing the change in inductance of a conventional magnetically controlled variable inductor as a function of bias current in the prior art.

[0028] Figure 4 This is a structural diagram of the inductor module of the excitation inductor dynamic adjustment device for an LLC resonant converter according to the present invention;

[0029] Figure 5 This is a graph showing the change of the inductance of the magnetically controlled excitation inductor with AC bias current in a dynamic adjustment device and method for the excitation inductor of an LLC resonant converter according to the present invention.

[0030] Figure 6 KZ is the excitation inductance dynamic adjustment device and method for an LLC resonant converter described in this invention. r Feasible solution domain graph;

[0031] Figure 7 This is a structural diagram of the LLC resonant converter in Example 2;

[0032] Figure 8 This is a structural diagram of the LLC resonant converter in Example 3;

[0033] Figure 9 A schematic diagram comparing the effective values ​​of the resonant current before and after using a variable excitation inductor under constant current charging conditions;

[0034] Figure 10 A schematic diagram comparing the effective values ​​of the resonant current before and after using a variable excitation inductor under constant voltage charging conditions;

[0035] Figure 11 This is a flowchart illustrating a method for dynamically adjusting the excitation inductance of an LLC resonant converter according to the present invention. Detailed Implementation

[0036] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0037] Example 1

[0038] like Figure 4 As shown, a dynamic adjustment device for the excitation inductance of an LLC resonant converter is implemented based on a single-stage AC-DC LLC resonant converter. It includes an inductor module connected to the LLC resonant converter. The inductor module includes an AC bias current sampling circuit and a magnetically controlled variable inductor.

[0039] The AC bias current sampling circuit includes a current transformer (CT), a bidirectional Zener diode (DZ), and a compensation capacitor (C). p The primary winding of the current transformer CT is connected in series with the input current of the LLC resonant converter, and one end of the secondary winding of the current transformer is connected to one end of the bidirectional Zener diode DZ and the compensation capacitor C. P One end of the current transformer's secondary winding is connected to the other end of the bidirectional Zener diode DZ, and serves as the output of the AC bias sampling circuit. The compensation capacitor C... p The other end serves as the output of the AC bias current sampling circuit.

[0040] The magnetically controlled variable inductor includes a magnetic core, a main power winding, and a bias winding. The selected magnetic core can be of type EE, EI, PQ, or other multi-circuit core structures. This embodiment uses an EE-type magnetic core as an example, which includes a central column and two side columns beside it. The main power winding is wound on the central column, and the bias winding is wound on the side columns. The bias winding is wound on both sides of the magnetic core and connected to an AC bias current sampling circuit, serving as the input terminal of the magnetically controlled variable inductor for injecting power frequency AC bias current. The main power winding is wound in the middle of the magnetic core and connected to an LLC resonant converter. The main power winding includes a transformer primary winding and a transformer secondary winding. The two ends of the transformer primary and secondary windings serve as the output terminals of the magnetically controlled variable inductor and are respectively connected to the primary and secondary power circuits of the LLC resonant converter.

[0041] This invention utilizes a compensation capacitor C p The equivalent inductance L formed by the bias winding and the bias winding bias Series resonance occurs, thereby significantly reducing the equivalent impedance of the load connected to the secondary side of the CT. Compensation capacitor C p The equivalent inductance L of the bias winding can be used as a reference. bias and target power frequency ω g Choose one that satisfies or approximately satisfies: ω g L bias ≈1 / ω g C p This reduces the equivalent impedance of the bias injection circuit at power frequency and improves the bias current injection capability. In practical applications, the influence of actual capacitance tolerance must also be considered, which can be compensated for by appropriately increasing the equivalent inductance of the CT secondary side. Furthermore, to reduce the phase error during CT detection, the excitation inductance in the CT also needs to be increased to reduce sampling error. Therefore, the CT core designed in this invention can be made of amorphous material with high permeability at low frequencies. In addition, since the CT secondary side voltage is prone to core saturation due to abnormal operating conditions, a clamping circuit composed of a bidirectional Zener diode DZ is added to the output side of the CT secondary side.

[0042] Figure 5 This illustrates the effect of injecting a power frequency AC bias current i into the inductor module, which varies with the power frequency. bias Post-equivalent magnetizing inductance L m The variation pattern within the power frequency cycle.

[0043] When an AC bias current is applied to the bias winding, the permeability of the side column magnetic circuit changes, thereby altering the equivalent reluctance and inductance of the main power winding. By rationally designing the air gap location, the number of turns in the bias winding, the number of turns in the main winding, and the core cross-sectional area, the desired inductance adjustment range can be obtained while avoiding core saturation. Unlike traditional magnetically controlled variable inductors based on steady-state DC injection, this invention uses the power frequency AC bias current as the control variable for the magnetically controlled variable inductor, achieving dynamic inductance adjustment within the power frequency cycle. Since the frequency of the AC bias current change is the power frequency, and the switching frequency of the LLC resonant converter is much higher than the power frequency cycle frequency, the amplitude of the AC bias current can be approximated as constant within one switching cycle. Under this condition, the main power winding of the magnetically controlled variable inductor can be equivalent to a fixed inductor in each switching cycle, while its inductance changes slowly with the AC bias current amplitude within the power frequency cycle scale. This allows for dynamic variation of the equivalent inductance of the main power winding with the bias current and the power frequency phase angle.

[0044] This invention focuses on the mechanism of minimizing resonant current during the power frequency cycle. It employs a time-domain method to analyze the relationship between resonant parameters and the effective value of the resonant current, and solves the feasible region. This provides a theoretical basis for reducing the effective value of the resonant current, optimizing the design of converter resonant parameters, and improving converter efficiency. According to the variation law of the effective value of the resonant current, increasing the inductance ratio K of the excitation inductance to the resonant inductance of the LLC resonant converter can effectively reduce the effective value of the resonant current. However, considering constraints such as switching frequency limitations, resonant capacitor withstand voltage, and soft-switching conditions, when solving for different combinations of resonant parameters satisfying different voltage gains and output power within the power frequency cycle based on a time-domain model, it is found that the inductance ratio K and characteristic impedance Z... r The feasible solution domain, such as Figure 6 As shown.

[0045] Analysis of the feasible domain variation reveals that, under the same voltage gain, a smaller output power results in a larger feasible domain; conversely, under the same output power, a smaller required voltage gain results in a larger feasible domain. In other words, to minimize the effective value of the resonant current, KZ needs to be selected. r Choose appropriate parameters near the right boundary of the feasible region. However, according to the variation law of the right boundary, K and Z... r They are approximately inversely proportional and difficult to increase simultaneously. Clearly, for a traditional single-stage LLC topology AC-DC converter, parameter design only needs to consider the worst-case conditions for voltage gain and output power to meet all operating conditions. Of course, in actual design, switching frequency, inductor / transformer size, and resonant capacitor voltage rating also need to be considered. To ensure sufficient margin in the design parameters, a proper distance must be left between the design parameters and the boundary conditions.

[0046] Based on the above theoretical analysis, the design of resonant parameters for traditional single-stage LLC topologies of AC-DC converters faces significant constraints. While parameter design considering extreme operating conditions of voltage gain and output power can ensure the gain requirements across all operating conditions, it also leads to limitations in KZ. r The available options are strictly limited to a finite feasible region, resulting in low efficiency under non-peak power frequency and light load conditions. Under non-extreme conditions, theoretical analysis has shown that KZ... r The feasible solution boundary of the parameters has an extensible space, which means that the alternative KZ under non-extreme conditions is possible. r The parameters can be much larger than the resonant parameters selected under traditional extreme operating conditions, thereby greatly optimizing the effective value of the power frequency periodic resonant current.

[0047] Based on the AC input voltage, rectified input voltage, output voltage, output power, and power frequency phase angle, calculate the required voltage gain G of the LLC resonant converter at different power frequency phases. req (θ) and instantaneous transmission power P in (θ). The formula for calculating the voltage gain Greq(θ) is: V ac (θ) is the instantaneous expression for the AC input voltage; θ is the phase of the power frequency cycle; V in V represents the effective value of the AC input voltage. o The output voltage is n; n is the turns ratio of the primary and secondary sides of the transformer, and the required voltage gain within the power frequency cycle is as follows: Figure 6 The solid line represents the required voltage gain G when the instantaneous input voltage approaches zero, i.e., θ = 0 or π. req It will tend towards infinity. On the other hand, according to the instantaneous AC input power P... in The expression for (θ) is: I in P represents the effective value of the input current. in This represents the input power. The instantaneous power change curve is shown below. Figure 6 As shown by the dashed line, the voltage gain G req and instantaneous input power P in The trends are opposite within the power frequency cycle. Based on the voltage gain Greq(θ) and instantaneous transmitted power pin(θ), a time-domain model of the LLC resonant converter is established, and the inductance ratio K and characteristic impedance Z of different excitation inductors and resonant inductors are calculated. r The voltage gain, soft-switching conditions, RMS resonant current, and device voltage stress are considered under combined conditions. Within the constraints of switching frequency range, soft-switching conditions, resonant capacitor voltage stress, switching device voltage stress, and maximum current stress, the K-Z coefficient satisfying the current power frequency phase condition is obtained. r Feasible solution domain. Within the feasible solution domain, with the goal of minimizing the effective value of the resonant current, the target inductance ratio K(θ) is selected, and the target magnetizing inductance L is calculated.m (θ)=K(θ)L r According to the L of the magnetically controlled variable excitation inductor m -i bias Characteristic curves, inversely calculate the target bias current i bias (θ). By using current transformer sampling injection, direct injection of AC input current, or active current source injection, the bias winding obtains an AC bias current corresponding to the target bias current, thereby realizing the dynamic adjustment of the excitation inductance within the power frequency cycle.

[0048] The advantages of this invention are: by injecting an AC bias current that varies with the power frequency cycle into the bias winding of the magnetically controlled variable excitation inductor, the excitation inductance of the LLC resonant converter changes dynamically within the power frequency cycle, thereby reducing the effective value of the resonant current, reducing circulating current loss, and improving converter efficiency.

[0049] Example 2

[0050] like Figure 7 As shown, a dynamic adjustment device for the excitation inductance of an LLC resonant converter is implemented based on a single-stage AC-DC LLC resonant converter. It includes an inductor module connected to the LLC resonant converter. The inductor module includes an AC bias current sampling circuit and a magnetically controlled variable inductor.

[0051] The AC bias current sampling circuit includes a compensation capacitor C. p Compensation capacitor C P One end is connected to an LLC resonant converter, and the compensation capacitor C p The other end serves as the output of the AC bias current sampling circuit.

[0052] The magnetically controlled variable inductor includes a magnetic core, a main power winding, and a bias winding. The selected magnetic core can be of type EE, EI, PQ, or other multi-circuit core structures. This embodiment uses an EE type magnetic core as an example, which includes a central column and two side columns beside the central column. The main power winding is wound on the central column, and the bias winding is wound on the side columns. The bias winding is wound on both sides of the magnetic core and connected to the compensation capacitor C of the AC bias current sampling circuit. p Connected to the input terminal of the magnetically controlled variable inductor, it is used to inject AC bias current at the power frequency. The main power winding is wound in the middle of the magnetic core and connected to the LLC resonant converter. The main power winding includes the primary winding and the secondary winding of the transformer. The two ends of the primary and secondary windings serve as the output terminals of the magnetically controlled variable inductor and are respectively connected to the primary and secondary power circuits of the LLC resonant converter.

[0053] This invention utilizes a compensation capacitor C p The equivalent inductance L formed by the bias winding and the bias winding biasSeries resonance occurs, thereby significantly reducing the equivalent impedance of the load connected to the secondary side of the CT. Compensation capacitor C p The equivalent inductance L of the bias winding can be used as a reference. bias and target power frequency ω g Choose one that satisfies or approximately satisfies: ω g L bias ≈1 / ω g C p This reduces the equivalent impedance of the bias injection circuit at power frequency and improves the bias current injection capability.

[0054] The secondary power circuit of the LLC resonant converter includes a first diode D1, a second diode D2, a third diode D3, a fourth diode D4, and an output capacitor C. o and load resistance R o One end of the secondary winding of the transformer is connected to the input terminal of the first diode D1 and the output terminal of the second diode D2. The output terminal of the first diode D1 is connected to the output terminal of the third diode D3 and the output capacitor C. o one end and load resistor R o One end of the second diode D2 is connected to the input of the fourth diode D4 and the output capacitor C. o The other end and the load resistor R o At the other end, the output terminal of the fourth diode D4 is connected to the input terminal of the third diode D3 and the other end of the secondary winding of the transformer.

[0055] The primary power circuit of the LLC resonant converter includes a resonant inductor L r Resonant capacitor C r Input capacitor C in Fifth diode D b1 The sixth diode D b2 7th diode D b3 The eighth diode D b4 First switch Q1, second switch Q2, third switch Q3, fourth switch Q4, AC power supply V ac and filters. AC power supply V ac Connected to the filter, one output terminal of the filter is connected to the fifth diode D. b1 The input terminal and the sixth diode D b2 The output terminal of the fifth diode D b1 The output terminals are respectively connected to the seventh diode D. b3 Output terminal, input capacitor C in One end of the diode, the drain of the first switching transistor Q1, and the drain of the third switching transistor Q3. The sixth diode D... b2 The input terminals are respectively connected to the eighth diode D b4 The input terminal and the input capacitor Cin The other end, the source of the second switch Q2, and the source of the fourth switch Q4 are grounded. The eighth diode D... b4 The output terminal is connected to the seventh diode D. b3 Input terminal and compensation capacitor C p One end, compensation capacitor C p The other end is connected to one end of the bias winding, and the other end of the bias winding is directly connected to the filter. The drain of the second switch Q2 is connected to the source of the first switch Q1 and the resonant capacitor C, respectively. r One end of the fourth switch Q4 is connected to the drain of the third switch Q3 and one end of the primary winding of the transformer. The resonant capacitor C... r The other end is connected to the other end of the transformer primary winding via a resonant inductor Lr.

[0056] Figure 5 This illustrates the effect of injecting a power frequency AC bias current i into the inductor module, which varies with the power frequency. bias Post-equivalent magnetizing inductance L m The variation pattern within the power frequency cycle.

[0057] When an AC bias current is applied to the bias winding, the permeability of the side column magnetic circuit changes, thereby altering the equivalent reluctance and inductance of the main power winding. By rationally designing the air gap location, the number of turns in the bias winding, the number of turns in the main winding, and the core cross-sectional area, the desired inductance adjustment range can be obtained while avoiding core saturation. Unlike traditional magnetically controlled variable inductors based on steady-state DC injection, this invention uses the power frequency AC bias current as the control variable for the magnetically controlled variable inductor, achieving dynamic inductance adjustment within the power frequency cycle. Since the frequency of the AC bias current change is the power frequency, and the switching frequency of the LLC resonant converter is much higher than the power frequency cycle frequency, the amplitude of the AC bias current can be approximated as constant within one switching cycle. Under this condition, the main power winding of the magnetically controlled variable inductor can be equivalent to a fixed inductor in each switching cycle, while its inductance changes slowly with the AC bias current amplitude within the power frequency cycle scale. This allows for dynamic variation of the equivalent inductance of the main power winding with the bias current and the power frequency phase angle.

[0058] This invention focuses on the mechanism of minimizing resonant current during the power frequency cycle. It employs a time-domain method to analyze the relationship between resonant parameters and the effective value of the resonant current, and solves the feasible region. This provides a theoretical basis for reducing the effective value of the resonant current, optimizing the design of converter resonant parameters, and improving converter efficiency. According to the variation law of the effective value of the resonant current, increasing the inductance ratio K of the excitation inductance to the resonant inductance of the LLC resonant converter can effectively reduce the effective value of the resonant current. However, considering constraints such as switching frequency limitations, resonant capacitor withstand voltage, and soft-switching conditions, when solving for different combinations of resonant parameters satisfying different voltage gains and output power within the power frequency cycle based on a time-domain model, it is found that the inductance ratio K and characteristic impedance Z... r The feasible solution domain, such as Figure 6 As shown.

[0059] Analysis of the feasible domain variation reveals that, under the same voltage gain, a smaller output power results in a larger feasible domain; conversely, under the same output power, a smaller required voltage gain results in a larger feasible domain. In other words, to minimize the effective value of the resonant current, KZ needs to be selected. r Choose appropriate parameters near the right boundary of the feasible region. However, according to the variation law of the right boundary, K and Z... r They are approximately inversely proportional and difficult to increase simultaneously. Clearly, for a traditional single-stage LLC topology AC-DC converter, parameter design only needs to consider the worst-case conditions for voltage gain and output power to meet all operating conditions. Of course, in actual design, switching frequency, inductor / transformer size, and resonant capacitor voltage rating also need to be considered. To ensure sufficient margin in the design parameters, a proper distance must be left between the design parameters and the boundary conditions.

[0060] Based on the above theoretical analysis, the design of resonant parameters for traditional single-stage LLC topologies of AC-DC converters faces significant constraints. While parameter design considering extreme operating conditions of voltage gain and output power can ensure the gain requirements across all operating conditions, it also leads to limitations in KZ. r The available options are strictly limited to a finite feasible region, resulting in low efficiency under non-peak power frequency and light load conditions. Under non-extreme conditions, theoretical analysis has shown that KZ... r The feasible solution boundary of the parameters has an extensible space, which means that the alternative KZ under non-extreme conditions is possible. r The parameters can be much larger than the resonant parameters selected under traditional extreme operating conditions, thereby greatly optimizing the effective value of the power frequency periodic resonant current.

[0061] Based on the AC input voltage, rectified input voltage, output voltage, output power, and power frequency phase angle, calculate the required voltage gain G of the LLC resonant converter at different power frequency phases. req (θ) and instantaneous transmission power P in(θ). Wherein, voltage gain G req The formula for calculating (θ) is: V ac (θ) is the instantaneous expression for the AC input voltage; θ is the phase of the power frequency cycle; V in V represents the effective value of the AC input voltage. o The output voltage is n; n is the turns ratio of the primary and secondary sides of the transformer, and the required voltage gain within the power frequency cycle is as follows: Figure 6 The solid line represents the required voltage gain G when the instantaneous input voltage approaches zero, i.e., θ = 0 or π. req It will tend towards infinity. On the other hand, according to the instantaneous AC input power P... in The expression for (θ) is: I in P represents the effective value of the input current. in This represents the input power. The instantaneous power change curve is shown below. Figure 6 As shown by the dashed line, the voltage gain G req and instantaneous input power P in The trends are opposite within the power frequency cycle. Based on voltage gain G req (θ) and instantaneous transmission power P in (θ), establish the time-domain model of the LLC resonant converter, and calculate the inductance ratio K and characteristic impedance Z of different excitation inductors and resonant inductors. r The voltage gain, soft-switching conditions, RMS resonant current, and device voltage stress are considered under combined conditions. Within the constraints of switching frequency range, soft-switching conditions, resonant capacitor voltage stress, switching device voltage stress, and maximum current stress, the K-Z coefficient satisfying the current power frequency phase condition is obtained. r Feasible solution domain. Within the feasible solution domain, with the goal of minimizing the effective value of the resonant current, the target inductance ratio K(θ) is selected, and the target magnetizing inductance L is calculated. m (θ)=K(θ)L r According to the L of the magnetically controlled variable excitation inductor m -i bias Characteristic curves, inversely calculate the target bias current i bias (θ). By using current transformer sampling injection, direct injection of AC input current, or active current source injection, the bias winding obtains an AC bias current corresponding to the target bias current, thereby realizing the dynamic adjustment of the excitation inductance within the power frequency cycle.

[0062] The advantages of this invention are as follows: By injecting an AC bias current that varies with the power frequency cycle into the bias winding of the magnetically controlled variable excitation inductor, the excitation inductance of the LLC resonant converter changes dynamically within the power frequency cycle, thereby reducing the effective value of the resonant current, reducing circulating current loss and improving converter efficiency. At the same time, it greatly reduces the number of components, lowers circuit cost and size, and the bias winding can be further reused as an input filter inductor, which is beneficial to further reduce system cost.

[0063] Example 3

[0064] like Figure 8 As shown, a dynamic adjustment device for the excitation inductance of an LLC resonant converter is implemented based on a single-stage AC-DC LLC resonant converter. It includes an inductor module connected to the LLC resonant converter. The inductor module includes an AC bias current sampling circuit and a magnetically controlled variable inductor.

[0065] The AC bias current sampling circuit includes a current transformer (CT), a bidirectional Zener diode (DZ), and a compensation capacitor (C). p The primary winding of the current transformer CT is connected in series with the input current of the LLC resonant converter, and one end of the secondary winding of the current transformer is connected to one end of the bidirectional Zener diode DZ and the compensation capacitor C. P One end of the current transformer's secondary winding is connected to the other end of the bidirectional Zener diode DZ, and serves as the output of the AC bias sampling circuit. The compensation capacitor C... p The other end serves as the output of the AC bias current sampling circuit.

[0066] The magnetically controlled variable inductor includes a magnetic core, a main power winding, and a bias winding. The selected magnetic core can be of type EE, EI, PQ, or other multi-circuit core structures. This embodiment uses an EE-type magnetic core as an example, which includes a central column and two side columns beside it. The main power winding is wound on the central column, and the bias winding is wound on the side columns. The bias winding is wound on both sides of the magnetic core and connected to an AC bias current sampling circuit, serving as the input terminal of the magnetically controlled variable inductor for injecting power frequency AC bias current. The main power winding is wound in the middle of the magnetic core and connected to an LLC resonant converter. The main power winding includes a transformer primary winding and a transformer secondary winding. The two ends of the transformer primary and secondary windings serve as the output terminals of the magnetically controlled variable inductor and are respectively connected to the primary and secondary power circuits of the LLC resonant converter.

[0067] This invention utilizes a compensation capacitor C p The equivalent inductance L formed by the bias winding and the bias winding bias Series resonance occurs, thereby significantly reducing the equivalent impedance of the load connected to the secondary side of the CT. Compensation capacitor C p The equivalent inductance L of the bias winding can be used as a reference. biasand target power frequency ω g Choose one that satisfies or approximately satisfies: ω g L bias ≈1 / ω g C p This reduces the equivalent impedance of the bias injection circuit at power frequency and improves the bias current injection capability. In practical applications, the influence of actual capacitance tolerance must also be considered, which can be compensated for by appropriately increasing the equivalent inductance of the CT secondary side. Furthermore, to reduce the phase error during CT detection, the excitation inductance in the CT also needs to be increased to reduce sampling error. Therefore, the CT core designed in this invention can be made of amorphous material with high permeability at low frequencies. In addition, since the CT secondary side voltage is prone to core saturation due to abnormal operating conditions, a clamping circuit composed of a bidirectional Zener diode DZ is added to the output side of the CT secondary side.

[0068] The secondary power circuit of the LLC resonant converter includes a first diode D1, a second diode D2, a third diode D3, a fourth diode D4, and an output capacitor C. o and load resistance R o One end of the secondary winding of the transformer is connected to the input terminal of the first diode D1 and the output terminal of the second diode D2. The output terminal of the first diode D1 is connected to the output terminal of the third diode D3 and the output capacitor C. o one end and load resistor R o One end of the second diode D2 is connected to the input of the fourth diode D4 and the output capacitor C. o The other end and the load resistor R o At the other end, the output terminal of the fourth diode D4 is connected to the input terminal of the third diode D3 and the other end of the secondary winding of the transformer.

[0069] The primary power circuit of the LLC resonant converter includes a resonant inductor L r Resonant capacitor C r Input capacitor C in Fifth diode D b1 The sixth diode D b2 7th diode D b3 The eighth diode D b4 First switch Q1, second switch Q2, third switch Q3, fourth switch Q4, AC power supply V ac and filters. AC power supply V ac Connected to the filter, one output terminal of the filter is connected to the fifth diode D. b1 The input terminal and the sixth diode D b2 The output terminal of the fifth diode D b1 The output terminals are respectively connected to the seventh diode D. b3Output terminal, input capacitor C in One end of the diode, the drain of the first switching transistor Q1, and the drain of the third switching transistor Q3. The sixth diode D... b2 The input terminals are respectively connected to the eighth diode D b4 The input terminal and the input capacitor C in The other end, the source of the second switch Q2, and the source of the fourth switch Q4 are grounded. The eighth diode D... b4 The output terminal passes through the current transformer CT and is connected to the filter. The drain of the second switch Q2 is connected to the source of the first switch Q1 and the resonant capacitor C. r One end of the fourth switch Q4 is connected to the drain of the third switch Q3 and one end of the primary winding of the transformer. The resonant capacitor C... r The other end is connected to the other end of the transformer primary winding via a resonant inductor Lr.

[0070] Figure 5 This illustrates the effect of injecting a power frequency AC bias current i into the inductor module, which varies with the power frequency. bias Post-equivalent magnetizing inductance L m The variation pattern within the power frequency cycle.

[0071] When an AC bias current is applied to the bias winding, the permeability of the side column magnetic circuit changes, thereby altering the equivalent reluctance and inductance of the main power winding. By rationally designing the air gap location, the number of turns in the bias winding, the number of turns in the main winding, and the core cross-sectional area, the desired inductance adjustment range can be obtained while avoiding core saturation. Unlike traditional magnetically controlled variable inductors based on steady-state DC injection, this invention uses the power frequency AC bias current as the control variable for the magnetically controlled variable inductor, achieving dynamic inductance adjustment within the power frequency cycle. Since the frequency of the AC bias current change is the power frequency, and the switching frequency of the LLC resonant converter is much higher than the power frequency cycle frequency, the amplitude of the AC bias current can be approximated as constant within one switching cycle. Under this condition, the main power winding of the magnetically controlled variable inductor can be equivalent to a fixed inductor in each switching cycle, while its inductance changes slowly with the AC bias current amplitude within the power frequency cycle scale. This allows for dynamic variation of the equivalent inductance of the main power winding with the bias current and the power frequency phase angle.

[0072] This invention focuses on the mechanism of minimizing resonant current during the power frequency cycle. It employs a time-domain method to analyze the relationship between resonant parameters and the effective value of the resonant current, and solves the feasible region. This provides a theoretical basis for reducing the effective value of the resonant current, optimizing the design of converter resonant parameters, and improving converter efficiency. According to the variation law of the effective value of the resonant current, increasing the inductance ratio K of the excitation inductance to the resonant inductance of the LLC resonant converter can effectively reduce the effective value of the resonant current. However, considering constraints such as switching frequency limitations, resonant capacitor withstand voltage, and soft-switching conditions, when solving for different combinations of resonant parameters satisfying different voltage gains and output power within the power frequency cycle based on a time-domain model, it is found that the inductance ratio K and characteristic impedance Z... r The feasible solution domain, such as Figure 6 As shown.

[0073] Analysis of the feasible domain variation reveals that, under the same voltage gain, a smaller output power results in a larger feasible domain; conversely, under the same output power, a smaller required voltage gain results in a larger feasible domain. In other words, to minimize the effective value of the resonant current, KZ needs to be selected. r Choose appropriate parameters near the right boundary of the feasible region. However, according to the variation law of the right boundary, K and Z... r They are approximately inversely proportional and difficult to increase simultaneously. Clearly, for a traditional single-stage LLC topology AC-DC converter, parameter design only needs to consider the worst-case conditions for voltage gain and output power to meet all operating conditions. Of course, in actual design, switching frequency, inductor / transformer size, and resonant capacitor voltage rating also need to be considered. To ensure sufficient margin in the design parameters, a proper distance must be left between the design parameters and the boundary conditions.

[0074] Based on the above theoretical analysis, the design of resonant parameters for traditional single-stage LLC topologies of AC-DC converters faces significant constraints. While parameter design considering extreme operating conditions of voltage gain and output power can ensure the gain requirements across all operating conditions, it also leads to limitations in KZ. r The available options are strictly limited to a finite feasible region, resulting in low efficiency under non-peak power frequency and light load conditions. Under non-extreme conditions, theoretical analysis has shown that KZ... r The feasible solution boundary of the parameters has an extensible space, which means that the alternative KZ under non-extreme conditions is possible. r The parameters can be much larger than the resonant parameters selected under traditional extreme operating conditions, thereby greatly optimizing the effective value of the power frequency periodic resonant current.

[0075] Based on the AC input voltage, rectified input voltage, output voltage, output power, and power frequency phase angle, calculate the required voltage gain G of the LLC resonant converter at different power frequency phases. req (θ) and instantaneous transmission power P in(θ). Wherein, voltage gain G req The formula for calculating (θ) is: V ac (θ) is the instantaneous expression for the AC input voltage; θ is the phase of the power frequency cycle; V in V represents the effective value of the AC input voltage. o The output voltage is n; n is the turns ratio of the primary and secondary sides of the transformer, and the required voltage gain within the power frequency cycle is as follows: Figure 6 The solid line represents the required voltage gain G when the instantaneous input voltage approaches zero, i.e., θ = 0 or π. req It will tend towards infinity. On the other hand, according to the instantaneous AC input power P... in The expression for (θ) is: I in P represents the effective value of the input current. in This represents the input power. The instantaneous power change curve is shown below. Figure 6 As shown by the dashed line, the voltage gain G req and instantaneous input power P in The trends are opposite within the power frequency cycle. Based on voltage gain G req (θ) and instantaneous transmission power P in (θ), establish the time-domain model of the LLC resonant converter, and calculate the inductance ratio K and characteristic impedance Z of different excitation inductors and resonant inductors. r The voltage gain, soft-switching conditions, RMS resonant current, and device voltage stress are considered under combined conditions. Within the constraints of switching frequency range, soft-switching conditions, resonant capacitor voltage stress, switching device voltage stress, and maximum current stress, the K-Z coefficient satisfying the current power frequency phase condition is obtained. r Feasible solution domain. Within the feasible solution domain, with the goal of minimizing the effective value of the resonant current, the target inductance ratio K(θ) is selected, and the target magnetizing inductance L is calculated. m (θ)=K(θ)L r According to the L of the magnetically controlled variable excitation inductor m -i bias Characteristic curves, inversely calculate the target bias current i bias (θ). By using current transformer sampling injection, direct injection of AC input current, or active current source injection, the bias winding obtains an AC bias current corresponding to the target bias current, thereby realizing the dynamic adjustment of the excitation inductance within the power frequency cycle.

[0076] by Figure 8 Taking the single-stage LLC architecture of a 3.3kW / 250V-420V on-board charger as an example, the traditional parameter design method yields an inductance ratio of K=4, Z... r =10. Construct a variable excitation inductor based on an EE65 magnetic core. When the bias current varies from 0-2A, the measured excitation inductance L... mThe sensitivity varies in the range of 61μH-133μH. Figure 9 and Figure 10 The effective value changes of the resonant current before and after using a variable excitation inductor during the power frequency cycle were compared. Under constant current output conditions, the effective value of the resonant current of the converter based on the variable inductor (VL) decreases significantly near the zero crossing of the AC voltage compared to the fixed inductor (CL) scheme. Figure 9 As shown. Under constant voltage output conditions, under light load, the AC input current decreases accordingly, the bias current decreases, and the equivalent magnetizing inductance increases further, resulting in a significant decrease in the effective value of the resonant current, such as... Figure 10 As shown. Meanwhile, theoretically, further increasing the excitation inductance L... m The variation amplitude of the inductance within the power frequency cycle can be used to further optimize the effective value of the resonant current, thereby improving the converter efficiency.

[0077] The advantages of this invention are as follows: By injecting an AC bias current that varies with the power frequency cycle into the bias winding of the magnetically controlled variable excitation inductor, the excitation inductance of the LLC resonant converter changes dynamically within the power frequency cycle, thereby reducing the effective value of the resonant current, reducing circulating current loss and improving converter efficiency. At the same time, the addition of a current transformer and a bidirectional Zener diode enhances anti-interference capability and improves accuracy.

[0078] Example 4

[0079] like Figure 11 As shown, a method for dynamically adjusting the excitation inductance of an LLC resonant converter is based on Embodiment 1 and... Figure 4 The excitation inductance dynamic adjustment device of the LLC resonant converter includes an inductor module connected to the LLC resonant converter. The inductor module includes an AC bias current sampling circuit and a magnetically controlled variable inductor.

[0080] The AC bias current sampling circuit includes a current transformer (CT), a bidirectional Zener diode (DZ), and a compensation capacitor (C). p The primary winding of the current transformer CT is connected in series with the input current of the LLC resonant converter, and one end of the secondary winding of the current transformer is connected to one end of the bidirectional Zener diode DZ and the compensation capacitor C. P One end of the current transformer's secondary winding is connected to the other end of the bidirectional Zener diode DZ, and serves as the output of the AC bias sampling circuit. The compensation capacitor C... p The other end serves as the output of the AC bias current sampling circuit.

[0081] The magnetically controlled variable inductor includes a magnetic core, a main power winding, and a bias winding. The selected magnetic core can be of type EE, EI, PQ, or other multi-circuit core structures. This embodiment uses an EE-type magnetic core as an example, which includes a central column and two side columns beside it. The main power winding is wound on the central column, and the bias winding is wound on the side columns. The bias winding is wound on both sides of the magnetic core and connected to an AC bias current sampling circuit, serving as the input terminal of the magnetically controlled variable inductor for injecting power frequency AC bias current. The main power winding is wound in the middle of the magnetic core and connected to an LLC resonant converter. The main power winding includes a transformer primary winding and a transformer secondary winding. The two ends of the transformer primary and secondary windings serve as the output terminals of the magnetically controlled variable inductor and are respectively connected to the primary and secondary power circuits of the LLC resonant converter.

[0082] This invention utilizes a compensation capacitor C p The equivalent inductance L formed by the bias winding and the bias winding bias Series resonance occurs, thereby significantly reducing the equivalent impedance of the load connected to the secondary side of the CT. Compensation capacitor C p The equivalent inductance L of the bias winding can be used as a reference. bias and target power frequency ω g Choose one that satisfies or approximately satisfies: ω g L bias ≈1 / ω g C p This reduces the equivalent impedance of the bias injection circuit at power frequency and improves the bias current injection capability. In practical applications, the influence of actual capacitance tolerance must also be considered, which can be compensated for by appropriately increasing the equivalent inductance of the CT secondary side. Furthermore, to reduce the phase error during CT detection, the excitation inductance in the CT also needs to be increased to reduce sampling error. Therefore, the CT core designed in this invention can be made of amorphous material with high permeability at low frequencies. In addition, since the CT secondary side voltage is prone to core saturation due to abnormal operating conditions, a clamping circuit composed of a bidirectional Zener diode DZ is added to the output side of the CT secondary side.

[0083] Figure 5 This illustrates the effect of injecting a power frequency AC bias current i into the inductor module, which varies with the power frequency. bias Post-equivalent magnetizing inductance L m The variation pattern within the power frequency cycle.

[0084] When an AC bias current is applied to the bias winding, the permeability of the side column magnetic circuit changes, thereby altering the equivalent reluctance and inductance of the main power winding. By rationally designing the air gap location, the number of turns in the bias winding, the number of turns in the main winding, and the core cross-sectional area, the desired inductance adjustment range can be obtained while avoiding core saturation. Unlike traditional magnetically controlled variable inductors based on steady-state DC injection, this invention uses the power frequency AC bias current as the control variable for the magnetically controlled variable inductor, achieving dynamic inductance adjustment within the power frequency cycle. Since the frequency of the AC bias current change is the power frequency, and the switching frequency of the LLC resonant converter is much higher than the power frequency cycle frequency, the amplitude of the AC bias current can be approximated as constant within one switching cycle. Under this condition, the main power winding of the magnetically controlled variable inductor can be equivalent to a fixed inductor in each switching cycle, while its inductance changes slowly with the AC bias current amplitude within the power frequency cycle scale. This allows for dynamic variation of the equivalent inductance of the main power winding with the bias current and the power frequency phase angle.

[0085] This invention focuses on the mechanism of minimizing resonant current during the power frequency cycle. It employs a time-domain method to analyze the relationship between resonant parameters and the effective value of the resonant current, and solves the feasible region. This provides a theoretical basis for reducing the effective value of the resonant current, optimizing the design of converter resonant parameters, and improving converter efficiency. According to the variation law of the effective value of the resonant current, increasing the inductance ratio K of the excitation inductance to the resonant inductance of the LLC resonant converter can effectively reduce the effective value of the resonant current. However, considering constraints such as switching frequency limitations, resonant capacitor withstand voltage, and soft-switching conditions, when solving for different combinations of resonant parameters satisfying different voltage gains and output power within the power frequency cycle based on a time-domain model, it is found that the inductance ratio K and characteristic impedance Z... r The feasible solution domain, such as Figure 6 As shown.

[0086] Analysis of the feasible domain variation reveals that, under the same voltage gain, a smaller output power results in a larger feasible domain; conversely, under the same output power, a smaller required voltage gain results in a larger feasible domain. In other words, to minimize the effective value of the resonant current, KZ needs to be selected. r Choose appropriate parameters near the right boundary of the feasible region. However, according to the variation law of the right boundary, K and Z... r They are approximately inversely proportional and difficult to increase simultaneously. Clearly, for a traditional single-stage LLC topology AC-DC converter, parameter design only needs to consider the worst-case conditions for voltage gain and output power to meet all operating conditions. Of course, in actual design, switching frequency, inductor / transformer size, and resonant capacitor voltage rating also need to be considered. To ensure sufficient margin in the design parameters, a proper distance must be left between the design parameters and the boundary conditions.

[0087] Based on the above theoretical analysis, the design of resonant parameters for traditional single-stage LLC topologies of AC-DC converters faces significant constraints. While parameter design considering extreme operating conditions of voltage gain and output power can ensure the gain requirements across all operating conditions, it also leads to limitations in KZ. r The available options are strictly limited to a finite feasible region, resulting in low efficiency under non-peak power frequency and light load conditions. Under non-extreme conditions, theoretical analysis has shown that KZ... r The feasible solution boundary of the parameters has an extensible space, which means that the alternative KZ under non-extreme conditions is possible. r The parameters can be much larger than the resonant parameters selected under traditional extreme operating conditions, thereby greatly optimizing the effective value of the power frequency periodic resonant current.

[0088] It also includes the following steps:

[0089] S1: Calculate the voltage gain and instantaneous transmission power required by the LLC resonant converter at different power frequency phases.

[0090] Based on the AC input voltage, rectified input voltage, output voltage, output power, and power frequency phase angle, calculate the required voltage gain G of the LLC resonant converter at different power frequency phases. req (θ) and instantaneous transmission power P in (θ). The formula for calculating the voltage gain Greq(θ) is: V ac (θ) is the instantaneous expression for the AC input voltage; θ is the phase of the power frequency cycle; V in V represents the effective value of the AC input voltage. o The output voltage is n; n is the turns ratio of the primary and secondary sides of the transformer, and the required voltage gain within the power frequency cycle is as follows: Figure 6 The solid line represents the required voltage gain G when the instantaneous input voltage approaches zero, i.e., θ = 0 or π. req It will tend towards infinity. On the other hand, the instantaneous AC input power P in(θ)的 The expression is: I in P represents the effective value of the input current. in This represents the input power. The instantaneous power change curve is shown below. Figure 6 As shown by the dashed line, the voltage gain G req and instantaneous input power P in The trends are opposite within the power frequency cycle.

[0091] S2: Calculate the inductance ratio K and characteristic impedance Z of the magnetizing inductor and resonant inductor based on voltage gain and instantaneous transmitted power. r .

[0092] Based on voltage gain G req (θ) and instantaneous transmission power P in(θ), establish the time-domain model of the LLC resonant converter, and calculate the inductance ratio K and characteristic impedance Z of different excitation inductors and resonant inductors. r Voltage gain, soft-switching conditions, effective value of resonant current, and device voltage stress under the combined conditions.

[0093] S3: Based on different inductance ratios K and characteristic impedance Z r The combined data yields KZ that satisfies the current power frequency phase condition. r Feasible solution domain.

[0094] Under constraints of switching frequency range, soft-switching conditions, resonant capacitor voltage stress, switching device voltage stress, and maximum current stress, K−Z is obtained to satisfy the current power frequency phase condition. r Feasible solution domain.

[0095] S4: Based on KZ r The characteristic curves of the target excitation inductance and bias current are obtained from the feasible solution domain, and the target bias current is calculated inversely based on the characteristic curves.

[0096] Within the feasible solution domain, with the goal of minimizing the effective value of the resonant current, the target inductance ratio K(θ) is selected, and the target excitation inductance L is calculated. m (θ)=K(θ)L r According to the L of the magnetically controlled variable excitation inductor m -i bias Characteristic curves, inversely calculate the target bias current i bias (θ).

[0097] S5: By dynamically adjusting the parameters, the bias winding obtains an AC bias current corresponding to the target bias current.

[0098] By using current transformer sampling injection, direct injection of AC input current, or active current source injection, the bias winding can obtain an AC bias current corresponding to the target bias current, thereby realizing the dynamic adjustment of the excitation inductance within the power frequency cycle.

[0099] The advantages of this invention are: by injecting an AC bias current that varies with the power frequency cycle into the bias winding of the magnetically controlled variable excitation inductor, the excitation inductance of the LLC resonant converter changes dynamically within the power frequency cycle, thereby reducing the effective value of the resonant current, reducing circulating current loss, and improving converter efficiency.

[0100] The above description is merely a specific embodiment of the present invention, but the scope of protection of the present invention is not limited thereto. Any variations or substitutions that can be easily conceived by those skilled in the art within the scope of the technology disclosed in the present invention should be included within the scope of protection of the present invention. Therefore, the scope of protection of the present invention should be determined by the scope of the claims.

Claims

1. A dynamic adjustment device for the excitation inductance of an LLC resonant converter, implemented based on an LLC resonant converter, characterized in that, The system includes an inductor module connected to the LLC resonant converter. This inductor module comprises an AC bias current sampling circuit and a magnetically controlled variable inductor. The magnetically controlled variable inductor includes a magnetic core, a main power winding, and a bias winding. The bias winding is wound on both sides of the magnetic core and connected to the AC bias current sampling circuit for injecting power frequency AC bias current. The main power winding is wound in the middle of the magnetic core and connected to the LLC resonant converter. The system calculates the voltage gain and instantaneous power transfer required by the LLC resonant converter at different power frequency phases, obtaining the inductance ratio K of the magnetizing inductor and the resonant inductor, and the characteristic impedance Z. r Based on different inductance ratios K and characteristic impedance Z r The combined data yields KZ that satisfies the current power frequency phase condition. r Feasible solution domain, based on KZ r The characteristic curves of the target excitation inductance and bias current are obtained from the feasible solution domain. The target bias current is then calculated based on the characteristic curves. By dynamically adjusting the parameters, the bias winding obtains an AC bias current corresponding to the target bias current.

2. The dynamic adjustment device for the excitation inductance of the LLC resonant converter according to claim 1, characterized in that, The AC bias current sampling circuit includes a current transformer and a compensation capacitor. The primary winding of the current transformer is connected in series with the input current of the LLC resonant converter. One end of the secondary winding of the current transformer is connected to a magnetically controlled variable inductor through the compensation capacitor, and the other end is connected to a magnetically controlled variable inductor.

3. The dynamic adjustment device for the excitation inductance of the LLC resonant converter according to claim 2, characterized in that, The compensation capacitor is selected based on the equivalent inductance formed by the bias winding and the target power frequency angular frequency, thereby reducing the equivalent impedance of the bias injection circuit at the power frequency.

4. The dynamic adjustment device for the excitation inductance of the LLC resonant converter according to claim 2, characterized in that, The AC bias current sampling circuit includes a clamping circuit to prevent core saturation. The clamping circuit is configured as a bidirectional Zener diode, with one end of the bidirectional Zener diode connected to one end of the secondary winding and a compensation capacitor, and the other end of the bidirectional Zener diode connected to the other end of the secondary winding.

5. The dynamic adjustment device for the excitation inductance of the LLC resonant converter according to claim 1, characterized in that, The inductance ratio K of the excitation inductor and the resonant inductor and the characteristic impedance Z are obtained. r This includes: establishing a time-domain model of the LLC resonant converter based on voltage gain and instantaneous transmitted power, and calculating the inductance ratio K and characteristic impedance Z for different inductance ratios. r Voltage gain, soft-switching conditions, effective value of resonant current, and device voltage stress under the combined conditions.

6. The dynamic adjustment device for the excitation inductance of the LLC resonant converter according to claim 1, characterized in that, The KZ-based r The feasible solution domain yields the characteristic curves of the target excitation inductance versus bias current, including: in the KZ... r Within the feasible solution domain, with the minimum effective value of the resonant current as the objective, the ratio of the target inductance is selected, and the target magnetizing inductance is calculated.

7. The dynamic adjustment device for the excitation inductance of the LLC resonant converter according to claim 1, characterized in that, The dynamic adjustment parameters include: current transformer sampling injection, direct injection of AC input current, or active current source injection.

8. The dynamic adjustment device for the excitation inductance of the LLC resonant converter according to any one of claims 1 to 7, characterized in that, The magnetic core includes a central column and two side columns next to the central column. The central column is wound with a main power winding, and the side columns are wound with a bias winding.

9. The dynamic adjustment device for the excitation inductance of the LLC resonant converter according to claim 8, characterized in that, The main power winding includes a transformer primary winding and a transformer secondary winding. The two ends of the transformer primary winding and the transformer secondary winding serve as the output terminals of a magnetically controlled variable inductor, and are respectively connected to the primary power circuit and the secondary power circuit of the LLC resonant converter.

10. A method for dynamically adjusting the excitation inductance of an LLC resonant converter, implemented based on the dynamic adjustment device for the excitation inductance of the LLC resonant converter as described in claim 1, characterized in that, It also includes the following steps: Based on different inductance ratios K and characteristic impedances Z of the magnetizing inductor and resonant inductor... r The combined data yields KZ that satisfies the current power frequency phase condition. r Feasible solution domain; Based on KZ r The characteristic curves of the target excitation inductance and bias current are obtained from the feasible solution domain, and the target bias current is calculated inversely based on the characteristic curves.

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