ISOP-CLLC Traction Transformer Topology and Control Method for Medium-Voltage DC Power Supply Electric Multiple Units

Abstract

This invention discloses a topology and control method for an ISOP-CLLC traction transformer in medium-voltage DC-powered electric multiple units (EMUs), belonging to the field of EMU traction transmission technology. This invention combines the voltage fluctuations of the medium-voltage DC traction network, the voltage stabilization requirements of the on-board DC bus, the discrete parameters of multiple modules, and the bidirectional energy flow operation requirements of EMUs. It constructs a collaborative control method that combines common-mode gain adaptation voltage stabilization with zero-sum and differential-mode balancing. Through a systematic scheme of "ISOP-CLLC on-board DC traction transformer topology—gain adaptation interval determination—common-mode voltage stabilization—zero-sum and differential-mode balancing," the DC traction transformer can simultaneously meet the requirements of DC voltage transformation, electrical isolation, modular power expansion, traction network voltage fluctuation adaptation, module balanced operation, and bidirectional energy transmission for traction / regenerative braking in the actual operation scenario of medium-voltage DC-powered EMUs.

Description

Technical Field

[0001] This invention relates to the field of traction transmission technology for electric multiple units (EMUs), and particularly to a topology and control method for an ISOP-CLLC traction transformer in a medium-voltage DC-powered electric multiple unit (EMU). Background Technology

[0002] With the continuous development of high-speed railways, heavy-haul railways, and the demand for new energy integration, the existing single-phase AC traction power supply system has gradually revealed certain limitations. Traditional 25kV single-phase AC traction power supply systems require phase separation during operation. When trains pass through the phase separation zone, speed losses and electrical shocks occur. Simultaneously, single-phase AC traction loads are prone to power quality issues such as negative sequence, voltage fluctuations, and harmonics. Furthermore, the existing EMU traction transmission systems contain large-volume and heavy-duty traction transformers, which is detrimental to the development of lightweight vehicles, high power density, and high-speed operation.

[0003] In medium-voltage DC traction power supply systems, the traction network is transformed from the traditional power frequency AC system to a DC system, and the EMU (Electric Multiple Unit) draws power directly from the DC traction network. Since the input energy is no longer power frequency AC, the existing front-end structure of "traction transformer—four-quadrant rectifier—intermediate DC link—traction inverter—traction motor" in AC EMUs cannot be directly reused. For DC-powered EMUs, their traction drive system needs to be able to convert the medium-voltage DC traction network voltage to the onboard DC bus voltage, while also meeting requirements for high-voltage electrical isolation, high-power energy transmission, and regenerative braking energy feedback. Therefore, onboard power electronic DC traction transformers suitable for medium-voltage DC power supply systems become key equipment in the traction drive system of this type of EMU.

[0004] On the other hand, the existing EMU's downstream "intermediate DC link - traction inverter - traction motor" structure is already relatively mature. If the entire vehicle's traction drive system were completely reconstructed to adapt to the medium-voltage DC power supply system, it would significantly increase the difficulty of vehicle system modification, equipment adaptation costs, and engineering verification workload. Therefore, a more reasonable technical approach is to replace the original AC-based power frequency traction transformer and four-quadrant rectifier with a power electronic DC traction transformer suitable for medium-voltage DC input, while retaining the existing downstream traction inverter and traction motor system as much as possible. This would enable the transformer to complete voltage level conversion, electrical isolation, and bidirectional energy transmission between the medium-voltage DC traction network and the on-board low-voltage DC bus.

[0005] For onboard DC traction transformers in high-speed trains, the application scenarios differ from those of typical DC-DC converter systems with fixed input and fixed loads. During train operation, factors such as changes in traction power, line voltage drop, power supply section status, load changes of adjacent vehicles, and regenerative energy feedback can all cause voltage fluctuations in the medium-voltage DC traction network. The onboard DC traction transformer not only needs to perform voltage level conversion under rated operating conditions but also needs to maintain stable onboard DC bus voltage under traction network voltage fluctuations to ensure the normal operation of downstream traction inverters and traction motor systems. Furthermore, the high power level of the high-speed train traction system means that a single power electronic conversion module cannot directly handle all medium-voltage and high-power conversion tasks. Therefore, the onboard DC traction transformer also needs to possess modular voltage division, current paralleling, and power expansion capabilities.

[0006] In existing technologies, such as Figure 1 The multi-phase bidirectional Buck-Boost DC-DC converter shown typically consists of multiple Buck-Boost power units operating in parallel or interleaved configurations. It reduces input and output current ripple through multi-phase interleaving control and achieves bidirectional power flow under traction and regenerative braking conditions via bidirectional switching devices. Under traction conditions, this type of converter can step down the medium-voltage DC traction grid voltage to the onboard DC bus voltage; under regenerative braking conditions, it can boost the energy from the onboard DC bus back to the DC traction grid. However, this topology is essentially a non-isolated DC-DC converter, lacking a high-frequency isolation transformer between its input and output terminals, making it difficult to meet the electrical isolation and safety insulation requirements between the medium-voltage traction grid and the onboard low-voltage DC bus of the EMU. For high-ratio transformation scenarios between the medium-voltage DC traction grid and the onboard DC bus, this topology needs to withstand significant voltage level differences, placing high stress on the switching devices, inductors, capacitors, and insulation systems.

[0007] In addition, such as Figure 2The dual active bridge (DAB) DC power electronic traction transformer with input series-output parallel (ISOP) and output parallel (OSP) configuration typically consists of multiple DAB sub-modules. The high-voltage input terminals of each DAB sub-module are connected in series to the medium-voltage DC traction network to achieve input voltage sharing; the low-voltage output terminals are connected in parallel to the on-board DC bus to achieve output current sharing. However, the DAB topology relies primarily on transformer leakage inductance or external series inductance to transfer power. Under conditions of input-output voltage mismatch, light load, or wide power range operation, the DAB converter is prone to generating significant circulating current power and current stress, limiting the soft-switching range and consequently affecting high power density and high-efficiency operation. For high-voltage, high-power, wide-condition, bidirectional energy flow scenarios like EMU traction systems, simply adopting the ISOP-DAB scheme would require significant control and design costs in areas such as power balancing, input voltage equalization, soft-switching maintenance, circulating current suppression, and loss control. In addition, some ISOP-DAB type DC power electronic traction transformers have concentrated capacitors at the input end, which can easily lead to overcurrent risks in the event of a fault. Furthermore, existing ISOP-DAB type power electronic traction transformers focus more on the modular structure of series input and parallel output and the phase-shifting power control itself. They still lack targeted collaborative control methods for the voltage stabilization of the on-board DC bus under voltage fluctuation conditions of the medium-voltage DC traction network of the EMU, as well as the coupling problem between voltage stabilization control and module equalization control during the operation of multiple sub-modules.

[0008] Furthermore, existing CLLC-type bidirectional DC / DC converters, such as the Chinese patent application CN120750191A, primarily address single-unit voltage conversion and bidirectional energy transfer, typically focusing on single-module voltage gain adjustment, soft-switching implementation, and wide-range input / output adaptation. However, in medium-voltage DC-DC train applications, CLLC submodules need to further address the challenges of input voltage division, output current division, multi-module parameter discretization, and on-board DC bus voltage regulation and module balancing coordination under the ISOP structure. Existing single-module CLLC control methods are difficult to apply directly. Therefore, while existing isolated modular DC power electronic transformer solutions can meet the basic requirements of input series, output parallel, and bidirectional isolation conversion, there is still room for improvement in adapting to voltage fluctuations in the medium-voltage DC traction network of EMU trains, enhancing resonant energy transfer characteristics, and achieving multi-module voltage regulation and balancing coordination. Summary of the Invention

[0009] One of the objectives of this invention is to propose a modular, isolated, bidirectional ISOP-CLLC type DC traction transformer topology suitable for medium-voltage DC traction network access scenarios, addressing the on-board traction drive application requirements of medium-voltage DC power supply EMUs, in order to solve the aforementioned problems.

[0010] To achieve the above objectives, the technical solution adopted by the present invention is as follows:

[0011] An ISOP-CLLC traction transformer topology for medium-voltage DC-powered electric multiple units (EMUs) is disclosed. The traction transformer comprises N isolated bidirectional CLLC resonant converter submodules. Each submodule adopts the same topology, and the parameters of each submodule may be the same or different. The high-voltage input terminals of each submodule are connected in series, and the low-voltage output terminals are connected in parallel, forming an input-series-output-parallel structure (ISOP), where N≥2. The isolated bidirectional CLLC resonant converter includes a high-voltage full-bridge converter unit, a primary-side resonant capacitor C1, a primary-side resonant inductor L1, a high-frequency isolation transformer T, and a magnetizing inductor L... m Secondary resonant inductor L2, secondary resonant capacitor C2, low-voltage side full-bridge converter unit, and DC support capacitors on both sides C d1 and C f1 The high-voltage side full bridge consists of four controllable switching devices and their anti-parallel diodes, namely S1-S4. The low-voltage side full bridge also consists of four controllable switching devices and their anti-parallel diodes, namely Q1-Q4. The primary resonant inductor L1 and the primary resonant capacitor C1 form a primary series resonant network, and the secondary resonant inductor L2 and the secondary resonant capacitor C2 form a secondary series resonant network.

[0012] The basic operating process of a single isolated bidirectional CLLC resonant converter is as follows: The active-side full-bridge converter transforms the DC voltage into a high-frequency AC square wave, which acts on the CLLC resonant cavity. The resonant cavity primarily transfers energy to the fundamental component of the square wave voltage and suppresses some higher harmonics, resulting in a resonant current with an approximately sinusoidal waveform. Subsequently, the high-frequency energy is transferred to the other side via a high-frequency isolation transformer, where it is rectified into DC voltage by the full-bridge converter on the other side. Since this structure incorporates resonant networks on both the primary and secondary sides, it essentially adds dual resonant cavities to a dual active bridge structure, transforming power transmission from "direct transmission of hard-switched square waves through leakage inductance" to "transmission of high-frequency energy shaped by the resonant network."

[0013] Under forward traction conditions, the high-voltage side full-bridge of the CLLC resonant converter submodule acts as the active inverter bridge, while the low-voltage side full-bridge acts as the rectifier side. Energy is transmitted from the medium-voltage DC traction network through the high-voltage side full-bridge, the primary resonant network, the high-frequency isolation transformer, and the secondary resonant network to the on-board DC bus. Under regenerative braking conditions, the low-voltage side full-bridge acts as the active inverter bridge, while the high-voltage side full-bridge acts as the rectifier side. Energy is transmitted in reverse from the on-board DC bus to the medium-voltage DC traction network. Because resonant networks are set on both the primary and secondary sides, the submodule has resonant energy transmission characteristics under both forward traction and reverse feedback energy flows.

[0014] Preferably, the traction transformer further includes a common-mode frequency converter control module for determining the common switching cycle based on traction network voltage changes, and a module equalization control module for generating zero and differential mode correction quantities based on the input voltage deviation and output current deviation of each submodule. Details of the control modules are described later.

[0015] The number of submodules is determined based on the input voltage level, output voltage level, and the withstand voltage capability of a single module. Preferably, the number of submodules N=5, and the rated power of a single isolated bidirectional CLLC resonant submodule is... It is 1MW.

[0016] The CLLC resonant parameters are determined based on the target gain range and the soft-switching operating region. Preferably, according to claim 1, for a single submodule, the primary-to-secondary turns ratio of the high-frequency isolation transformer is n=10 / 9, the magnetizing inductance ratio is k=5, and the quality factor Q of the CLLC resonant cavity is 0.3.

[0017] The traction drive system composed of the aforementioned traction transformer includes a medium-voltage DC traction network, a DC traction transformer, a three-phase inverter, and a traction motor arranged in sequence. The high-voltage side input terminals of each sub-module of the DC traction transformer are connected in series to the medium-voltage DC traction network, and the low-voltage side output terminals of each sub-module of the DC traction transformer are connected in parallel to the vehicle-mounted DC bus.

[0018] Compared to existing AC-powered EMUs, this invention eliminates the use of power frequency traction transformers and four-quadrant rectifiers. Instead, it directly transforms the DC voltage and provides electrical isolation from the medium-voltage DC traction network to the onboard DC bus using a modular CLLC-type DC traction transformer. This approach allows the EMU to adapt to medium-voltage DC power supply systems while preserving as much of the existing EMU's main structure as possible, which consists of the onboard DC bus, traction inverter, and traction motor.

[0019] Simultaneously, voltage sharing on the grid side is achieved through series connection of high-voltage side inputs, and current sharing on the low-voltage side is achieved through parallel connection of low-voltage side outputs. For medium-voltage DC-DC train traction drive systems, this structure allows for flexible adjustment of the number of sub-modules according to the traction grid voltage level, the on-board DC bus voltage level, and the overall vehicle traction power requirements, exhibiting good modular expansion capabilities.

[0020] This structure addresses the constraints of onboard, medium-voltage, high-power, and bidirectional energy flow applications in high-speed trains. Through input series connection, the high DC voltage on the medium-voltage side is shared by multiple sub-modules; through output parallel connection, the high current on the onboard DC bus side is shared by multiple sub-modules, thereby reducing the voltage stress, current stress, insulation design pressure, and heat dissipation pressure of individual power modules.

[0021] This invention combines the voltage fluctuations of the medium-voltage DC traction network of the EMU, the voltage stabilization requirements of the on-board DC bus, the discrete parameters of multiple modules, and the bidirectional energy flow operation requirements, and also constructs a collaborative control method that combines common-mode gain adaptation voltage stabilization with zero-sum differential mode balancing.

[0022] The common-mode frequency converter voltage regulation control is as follows: based on the traction network voltage fluctuation range, the target voltage of the on-board DC bus, the number of sub-modules, the high-frequency transformer ratio, and the CLLC gain curve, the frequency converter operating range that meets the requirements of gain adjustment, gain monotonicity, and soft switching operation is determined, and the overall voltage gain of each sub-module is adjusted by the common switch cycle or common switch frequency to keep the on-board DC bus stable under the traction network voltage fluctuation conditions.

[0023] The differential mode equalization control is as follows: a module-level correction amount is generated based on the input voltage and low-voltage side port current deviation of each sub-module, and the differential mode correction amount is processed by zero-sum and superimposed on the common switching cycle command to form an independent switching cycle for each sub-module without changing the common average cycle of the system. Thus, without destroying the voltage regulation effect of the on-board DC bus, the input series module voltage equalization, the output parallel module current equalization, and the power sharing coordination are achieved.

[0024] Specifically, this invention addresses the issue of easy coupling between "onboard DC bus voltage regulation" and "module voltage, current, and power balancing" in ISOP-CLLC type DC traction transformers by proposing a control method that decouples common-mode frequency conversion voltage regulation and differential-mode balancing. The common-mode control quantity characterizes the common switching cycle or frequency command for all CLLC submodules, primarily used to adjust the overall system voltage gain and stabilize the onboard DC bus voltage. The differential-mode control quantity characterizes the correction amount of each submodule relative to the common command, primarily used to adjust the relative power distribution between modules. By decomposing the control quantity into common-mode and differential-mode components, the problem of module balancing adjustment directly interfering with the overall system voltage regulation operating point is avoided. This method is suitable for operating scenarios in EMU onboard DC traction transformers with a large number of modules, high power ratings, unavoidable parameter dispersion, and strict bus voltage regulation requirements.

[0025] This invention addresses the voltage fluctuation problem in medium-voltage DC traction networks caused by factors such as traction power variations, line voltage drop, power supply section status changes, and regenerative braking feedback during EMU operation. It proposes a common-mode gain adaptation method accommodating traction network voltage fluctuations and CLLC soft-switching boundary constraints. Instead of simply performing ordinary voltage closed-loop regulation with a fixed reference period, this method determines the required voltage gain range and corresponding frequency converter operating range for each sub-module based on the traction network voltage fluctuation range, the target voltage of the onboard DC bus, the number of sub-modules, the high-frequency transformer turns ratio, and the CLLC positive gain curve.

[0026] In the EMU application scenario described in this invention, when the traction grid voltage fluctuates within a certain range, the required gain of the CLLC submodule must simultaneously meet the on-board DC bus voltage stabilization requirements, gain adjustment monotonicity, and ZVS soft-switching operation conditions. This invention limits the common switching cycle or common switching frequency to a gain adaptation range jointly determined by the CLLC gain curve and the soft-switching region. This allows each submodule to increase voltage gain when the traction grid voltage decreases and decrease voltage gain when the traction grid voltage increases, thereby achieving stable control of the on-board DC bus under medium-voltage DC traction grid voltage fluctuation conditions.

[0027] This invention addresses the uneven load sharing problem among multiple CLLC sub-modules in an on-board DC traction transformer caused by variations in resonant parameters, high-frequency transformer turns ratio, switching device characteristics, drive delay, losses, and heat dissipation conditions. It proposes a differential-mode equalization method based on input voltage deviation and low-voltage side port current deviation. This method collects the high-voltage side input voltage and low-voltage side port current of each sub-module, calculates their deviations relative to the average value, and generates a module-level differential-mode correction to improve the voltage sharing consistency of the input series modules and the current sharing consistency of the output parallel modules.

[0028] Since the low-voltage sides of each submodule are connected in parallel to the same on-board DC bus, the low-voltage side port voltages are approximately consistent. Therefore, low-voltage side port current balancing can reflect the consistency of power sharing among the modules. This invention, through the coordinated control of input voltage balancing and low-voltage side port current balancing, prevents some modules from being subjected to excessively high voltage, current, or thermal stresses for extended periods due to parameter discrepancies. This improves the reliability, power expansion capability, and engineering applicability of the on-board DC traction transformer in medium-voltage DC train sets.

[0029] Compared with the prior art, the advantages of the present invention are as follows:

[0030] Compared to the ISOP-DAB type DC power electronic traction transformer, this invention, while retaining the modular structure of input series and output parallel, introduces a CLLC resonator module. This transforms power transmission from square wave phase-shifting transmission relying on leakage inductance or external inductance to high-frequency resonant energy transmission shaped by the primary and secondary resonant networks. This structure improves the current waveform, reduces circulating current power and switching losses, and enhances efficiency and power density potential in medium-voltage high-power traction applications. Furthermore, this topology, while replacing the traditional power frequency traction transformer and front-end rectifier stage, retains as much of the existing "on-board DC bus—traction inverter—traction motor" downstream structure as possible, making it more suitable for system retrofitting and engineering applications of medium-voltage DC-powered electric multiple units.

[0031] Furthermore, this invention addresses the unique constraints encountered in actual EMU operation, such as traction network voltage fluctuations, on-board DC bus voltage regulation, inconsistent parameters among multiple submodules, and bidirectional energy flow in traction / regenerative braking. It proposes a collaborative control method combining common-mode gain-adaptive voltage regulation and zero-sum differential-mode equalization. Common-mode frequency conversion voltage regulation determines the frequency conversion operating range based on the traction network voltage fluctuation range, the target voltage of the on-board DC bus, the number of submodules, the transformer ratio, and the CLLC gain curve, ensuring the system maintains on-board DC bus stability even when the traction network voltage changes. Differential-mode equalization control generates module-level corrections based on the input voltage of each submodule and the low-voltage side port current deviation, and avoids altering the system's common average switching cycle through zero-sum processing. This improves the consistency of module voltage division, current division, and power sharing without compromising bus voltage regulation. Therefore, this invention not only enables isolated voltage transformation between the medium-voltage DC traction network and the on-board DC bus but also enhances the adaptability of the DC traction transformer under actual EMU operating conditions. Attached Figure Description

[0032] Figure 1 This is a diagram of the topology of a triple bidirectional buck-boost DC-DC converter in the prior art;

[0033] Figure 2 The diagram shows the topology of the existing 2 ISOP-DAB type DC power electronic transformer.

[0034] Figure 3 This is a general structural diagram of the traction transmission system of the present invention;

[0035] Figure 4 Structure diagram of a bidirectional CLLC resonant converter;

[0036] Figure 5 Voltage gain curves for different Q values;

[0037] Figure 6 The voltage gain curve is shown for Q=0.3.

[0038] Figure 7 This is the waveform of the on-board DC bus voltage.

[0039] Figure 8 The waveforms are those of the on-board DC bus voltage and the common switch frequency.

[0040] Figure 9 This is the high-voltage input voltage waveform of the submodule;

[0041] Figure 10 The output current waveform on the low-voltage side of the submodule;

[0042] Figure 11 Output power waveform for the submodule;

[0043] Figure 12 The waveforms of the final switching cycle, cycle correction amount, and sum of the values ​​for the submodules;

[0044] Figure 13 The waveforms are the high-voltage side input voltage and the vehicle DC voltage.

[0045] Figure 14 The waveforms of the output voltage of a single submodule on the high-voltage side and the output voltage in series.

[0046] Figure 15 This is the zero-current turn-off transient process of DAB;

[0047] Figure 16 The waveforms of the full-bridge output voltage and resonant current of the CLLC submodule are shown. Detailed Implementation

[0048] To explain the technical content, objectives, and effects of the present invention in detail, the following specific embodiments are provided to further illustrate the content of the present invention. However, the content of the present invention is far more than the following examples.

[0049] Example 1

[0050] This embodiment provides a traction drive system for a medium-voltage DC-powered EMU (Electric Multiple Unit). The system connection is as follows: the medium-voltage DC traction network is connected to a modular ISOP-CLLC DC traction transformer via a pantograph. The modular ISOP-CLLC DC traction transformer consists of N isolated bidirectional CLLC resonant converter sub-modules with series inputs and parallel outputs. The output of the traction DC transformer is connected to the onboard DC bus, which is further connected to a traction inverter. The traction inverter outputs three-phase AC power to drive the traction motor. Its structure is as follows: Figure 3 As shown.

[0051] The circuit structure of a single isolated bidirectional CLLC resonant converter is as follows: Figure 4 As shown. It includes a high-voltage side full-bridge converter unit, a primary-side resonant capacitor C1, a primary-side resonant inductor L1, a high-frequency isolation transformer T, and a magnetizing inductor L. m Secondary resonant inductor L2, secondary resonant capacitor C2, low-voltage side full-bridge converter unit, and DC support capacitors on both sides C d1 and C f1 The high-voltage side full bridge consists of four controllable switching devices and their anti-parallel diodes, designated S1-S4. The low-voltage side full bridge also consists of four controllable switching devices and their anti-parallel diodes, designated Q1-Q4. The primary-side resonant inductor L1 and primary-side resonant capacitor C1 form the primary-side resonant network, and the secondary-side resonant inductor L2 and secondary-side resonant capacitor C2 form the secondary-side resonant network. The high-frequency isolation transformer T is used to achieve electrical isolation and voltage transformation between the high-voltage and low-voltage sides. The magnetizing inductor L... mUsed to form the transformer excitation branch and participate in the soft switching process.

[0052] (1) Number of sub-modules, voltage level, power level and high-frequency transformer turns ratio configuration

[0053] The number of sub-modules, voltage level, and transformer ratio of the ISOP-CLLC DC traction transformer should be determined comprehensively based on the traction network voltage, on-board DC bus voltage, vehicle traction power, and the withstand voltage level of individual module components.

[0054] In this embodiment, the rated voltage of the medium-voltage DC traction network is... Set to 20kV, rated voltage of intermediate DC bus The voltage is set to 3600V. This is to ensure that each submodule operates under reduced voltage conditions during traction, while maximizing voltage division capacity and not exceeding the rated voltage that each submodule can withstand. In this embodiment, the number of submodules is set to 20000 / 4000=5.

[0055] This invention employs a symmetrical CLLC resonant parameter design, where the secondary-side resonant parameters, after being converted using a high-frequency transformer, are matched to the primary-side resonant parameters. The primary-to-secondary turns ratio n of the high-frequency isolation transformer is:

[0056] (1),

[0057] Where N1 is the number of turns in the high-voltage side winding of the high-frequency isolation transformer, and N2 is the number of turns in the low-voltage side winding.

[0058] Based on the above design, the rated power of this DC-powered EMU is... The rated power of a single submodule is 5MW. It is 1MW.

[0059] (2) CLLC resonant parameter configuration and soft-switching operation constraints

[0060] Primary-side series resonant frequency of a single submodule Represented as:

[0061] (2),

[0062] Secondary-side series resonant frequency It can be represented as:

[0063] (3),

[0064] To ensure a more consistent resonance characteristic between the primary and secondary resonant networks, a symmetrical CLLC resonant parameter design is adopted. The secondary resonant parameters are then designed according to the following relationship:

[0065] (4),

[0066] This allows the secondary resonant inductor to be matched with L1 after being referred to the primary side, and the secondary resonant capacitor to be matched with C1 after being referred to the primary side, thus giving the primary and secondary resonant networks more consistent resonance characteristics, which is beneficial to improving the consistency of forward and reverse operation characteristics.

[0067] The magnetizing inductance ratio k represents the magnetizing inductance. With resonant inductor The ratio between them:

[0068] (5),

[0069] The value of k affects the magnitude of the excitation current, the resonant current stress, and the voltage gain adjustment capability. When k is too small, the excitation inductance is small, and the effective values ​​of the excitation current and resonant current may increase; when k is too large, the resonant cavity gain adjustment capability may weaken. Therefore, k can be comprehensively selected based on the input voltage fluctuation range, the required gain range, the device current stress, and efficiency requirements. In this invention, considering the input voltage fluctuation range, gain adjustment capability, and current stress, k=5 is set.

[0070] The quality factor Q of a CLLC resonant cavity can be used to describe the relationship between load conditions and the impedance of the resonant network. Let the equivalent load of the on-board DC bus be R0, then:

[0071] (6),

[0072] In fundamental frequency equivalent analysis, the load can be converted to the AC side to obtain the equivalent AC load. :

[0073] (7),

[0074] Characteristic impedance of resonant network It can be represented as:

[0075] (8),

[0076] The quality factor Q can be defined as:

[0077] (9),

[0078] Among them, R o,i For the equivalent load of a single submodule, and The equivalent resonant inductance and equivalent resonant capacitance are respectively used to inversely deduce the resonant parameters based on the required quality factor and resonant frequency. In this embodiment, the quality factor is set to Q=0.3 to obtain a gain curve and soft-switching operating margin that meet the voltage fluctuation range of the traction network.

[0079] The above-mentioned resonant cavity configuration has the following technical significance: First, the resonant network can make the current under square wave voltage excitation tend to be sinusoidal, reducing current spikes and high-frequency harmonic content, and reducing current stress on switching devices and high-frequency transformers; Second, the resonant network can form the required impedance characteristics near the switching frequency. By controlling the equivalent input impedance of the resonant cavity to be inductive, the resonant current can charge and discharge the junction capacitance of the switching transistor during the dead time, thereby creating conditions for zero-voltage turn-on of the switching device; Third, the primary and secondary side resonant network enables the submodule to have good symmetry in both forward traction and reverse feedback energy flow directions, which is beneficial to improving bidirectional operating efficiency and power density.

[0080] (3) CLLC voltage gain requirements and gain adaptation range

[0081] During the operation of high-speed trains, the voltage of the medium-voltage DC traction network fluctuates due to factors such as changes in traction power, line voltage drop, power supply section status, and regenerative braking energy feedback. For ISOP-CLLC type on-board DC traction transformers, changes in traction network voltage directly alter the required voltage gain of a single CLLC submodule. Therefore, it is necessary to determine the required gain range of the CLLC submodule based on the traction network voltage fluctuation range, and to determine the switching frequency or switching cycle adjustment range by combining the CLLC gain curve and the soft-switching region.

[0082] Assuming the rated voltage of the medium-voltage DC traction network is V1, the number of submodules is N, the target voltage of the on-board DC bus is V2 (in this embodiment, V1 is 20kV and V2 is 3600V, but this gain adaptation method can be extended to different voltage levels), and the high-frequency transformer turns ratio is n, then the required normalized voltage gain of the submodule is... It can be represented as:

[0083] (10)

[0084] If a voltage fluctuation range for the medium-voltage DC traction network is set, this embodiment sets the voltage fluctuation range to the rated value ±5% to verify the voltage adaptability of the proposed control method, i.e., 5%, with a network voltage fluctuation range of 19-21kV. Based on the aforementioned parameters of the EMU, the normalized gain range that the CLLC submodule needs to cover is approximately:

[0085] (11),

[0086] This embodiment derives a normalized model of the primary and secondary sides of an ISOP-CLLC type DC transformer, obtaining a graph showing the relationship between the voltage gain of the CLLC submodule and the switching frequency and quality factor. Figure 5 As shown.

[0087] pass Figure 5The gain curve of the CLLC submodule shown is used to determine the inverter operating range that meets the above gain requirements. This operating range should simultaneously meet the following conditions:

[0088] (12),

[0089] (13)

[0090] (14)

[0091] Among them, f s The common switching frequency for all submodules, For the positive voltage gain of the CLLC submodule, The first condition represents the imaginary part of the equivalent input impedance of the resonant cavity. The second condition indicates that the gain has a monotonic adjustment characteristic within the selected interval, avoiding control ambiguity caused by the same gain corresponding to multiple frequencies; the third condition indicates that the input impedance of the resonant cavity is inductive, which is beneficial for achieving ZVS soft-switching operation.

[0092] In this embodiment, the quality factor Q is set to 0.3 based on relevant literature and engineering experience, and the common switching period of the CLLC submodule is constrained to:

[0093] (15)

[0094] The corresponding common switch frequency range is:

[0095] (16)

[0096] This range is used to ensure that the submodule has the gain adjustment capability to meet the voltage regulation requirements of the on-board DC bus under input voltage fluctuation conditions. At the same time, the parameter design should also combine the equivalent impedance characteristics of the resonant cavity, the phase of the resonant current and the commutation requirements of the device to verify the soft-switching operation conditions within this period range. The frequency conversion range given by Equation (16) can cover the required voltage gain and can be kept within the preset soft-switching operation range, thus taking into account both input voltage adaptation and high-efficiency operation requirements.

[0097] (4) Common-mode gain adaptive voltage regulation control for traction network voltage fluctuations

[0098] To stabilize the on-board DC bus voltage under traction network voltage fluctuations, this embodiment also proposes a common-mode gain adaptive voltage regulation control method. The common-mode control quantity is a common switching cycle or common switching frequency command used by all CLLC submodules, which is used to adjust the overall voltage gain of the DC traction transformer.

[0099] The normalized gain required by the CLLC submodule under the current operating conditions is calculated. Then, within the gain adaptation range that satisfies the gain monotonicity and ZVS soft-switching conditions, the corresponding basic switching period is obtained through data fitting. :

[0100] (17)

[0101] in, This represents the mapping relationship between the target gain and the switching period within a preset monotonic gain range.

[0102] If the target voltage of the vehicle-mounted DC bus is The actual on-board DC bus voltage is Then voltage error for:

[0103] (18)

[0104] The voltage error is used by the voltage regulator to generate a common switching cycle command. , represented as:

[0105] (19)

[0106] in, The basic switching cycle of the CLLC submodule is set to 10kHz in this embodiment, so the basic switching cycle is 1 / 10k = 100us. and These are the proportional and integral coefficients of the voltage regulator, respectively.

[0107] At the same time, in accordance with the switching cycle constraint, the common switching cycle command needs to be limited to a preset gain adaptation range:

[0108] (20)

[0109] in This is the amplitude limiting function.

[0110] The function of common-mode frequency converter voltage regulation is as follows: when the traction grid voltage decreases, leading to an increase in the required gain, the gain of the submodule is increased by adjusting the common switch cycle or switching frequency; when the traction grid voltage increases, leading to a decrease in the required gain, the gain of the submodule is decreased by adjusting the common switch cycle or switching frequency in the opposite direction. Thus, all submodules coordinately adjust under the same common-mode command, stabilizing the on-board DC bus voltage near the target value.

[0111] (5) Balanced control of differential input voltage and low-voltage side port current for modules with uneven distribution

[0112] While common-mode frequency converter voltage regulation can adjust the overall system output voltage, it is difficult to actively suppress uneven input voltage, output current, and power transmission between sub-modules. For ISOP-CLLC type DC traction transformers with series inputs and parallel outputs, each sub-module's high-voltage side is connected in series to the medium-voltage DC traction network, requiring consistent high-voltage input voltage sharing across all sub-modules. Similarly, each sub-module's low-voltage side is connected in parallel to the vehicle's DC bus, requiring consistent low-voltage port current sharing. Since each sub-module's low-voltage side is connected in parallel to the same vehicle's DC bus, their port voltages are approximately identical; therefore, balanced low-voltage port current also reflects the consistency of power transmission sharing among the sub-modules.

[0113] To accommodate both traction and regenerative braking energy flows, this embodiment defines an energy flow direction variable:

[0114] (twenty one),

[0115] Define the direction-normalized low-voltage side port current as:

[0116] (twenty two),

[0117] in, Let be the low-voltage side port current of the i-th submodule. Through this definition, the port currents for both forward traction and regenerative braking energy flows can be unified under the same module sharing criterion.

[0118] Let the high-voltage side input voltage of the i-th submodule be... The low-voltage side port current after directional normalization is The low-voltage side port current can be defined according to a unified reference direction; under forward traction conditions, it can be represented as the current output from the submodule to the on-board DC bus; under regenerative braking conditions, it can be represented as the current input from the on-board DC bus to the submodule, or, after sign correction based on the power flow direction, it can be used in the equalization calculation. Average input voltage of each submodule. Average output current They are represented as follows:

[0119] (twenty three),

[0120] Based on the input voltage deviation and output current deviation, the original differential mode correction amount of the i-th submodule can be generated. :

[0121] (twenty four),

[0122] in, , These are the voltage equalization and current equalization adjustment coefficients, respectively. The above formulas illustrate that the differential correction can be generated by both voltage and current deviations. In practical applications, one, two, or a combination of these can be selected according to system requirements, and the signs of each variable can be adjusted according to the control direction.

[0123] (6) Zeroing out of differential mode correction, superposition of common mode and differential mode and periodic amplitude limiting

[0124] To prevent differential mode equalization control from altering the overall average switching cycle of the system and thus interfering with the on-board DC bus voltage regulation control, this invention performs zero-sum processing on the original differential mode correction. If the original differential mode correction of the i-th submodule is... Then the modulus correction after zero-summing is:

[0125] (25),

[0126] After this processing, the differential mode correction values ​​of each submodule satisfy the following:

[0127] (26)

[0128] This zero-sum constraint means that the differential mode correction only changes the relative switching period between each sub-module, without changing the average switching period of all sub-modules, thereby avoiding the disruption of common mode frequency converter voltage regulation control by module equalization control.

[0129] Ultimately, the switching cycle of the i-th CLLC submodule is determined by both the common-mode cycle instruction and the differential-mode correction:

[0130] (27)

[0131] To ensure that each CLLC submodule remains within its operating range that meets the requirements for voltage gain and soft-switching operation, the final switching cycle needs to be limited to a preset operating range: (28)

[0132] The switching cycle set for the aforementioned 5% traction network voltage fluctuation is as follows:

[0133] (29)

[0134] It should be noted that within the normal equalization correction range, zero-sum differential correction can ensure that the module equalization control does not change the system's common average period; when individual sub-modules reach the final safety limit boundary, the limit priority ensures that the sub-modules do not deviate from the gain adaptation range and soft-switching operating range.

[0135] (7) Bidirectional operation and inter-module staggered drive mode

[0136] The ISOP-CLLC type DC traction transformer in this embodiment has bidirectional energy transmission capabilities for traction power supply and regenerative braking. Under forward traction conditions, the medium-voltage DC traction network supplies power to the on-board DC bus. The high-voltage side full-bridge acts as an active inverter bridge, and the low-voltage side full-bridge or its anti-parallel diodes act as the rectifier side. The low-voltage side outputs of each submodule are connected in parallel to the on-board DC bus, jointly supplying power to the traction inverter and traction motor. Under regenerative braking conditions, energy is fed back to the medium-voltage DC traction network from the on-board DC bus. The low-voltage side full-bridge can act as an active inverter bridge, and the high-voltage side full-bridge or its anti-parallel diodes act as the rectifier side. The series-connected high-voltage side ports of multiple submodules are superimposed to feed energy back to the medium-voltage DC traction network.

[0137] To reduce input and output voltage and current ripple, multiple CLLC submodules employ a phase-interleaved drive method. When the system contains N submodules, the drive signals of adjacent submodules can be set to have a phase difference of... :

[0138] (30)

[0139] This interleaved drive method enables the high-frequency current ripple of multiple sub-modules to partially cancel each other out at the input and output ends, thereby reducing the DC support capacitor ripple current and bus voltage fluctuations and improving the system's operational stability.

[0140] Example 2

[0141] Simulation verification:

[0142] (1) Simulation model parameter settings

[0143] To verify the feasibility of the ISOP-CLLC type EMU DC traction transformer and its common-mode gain adaptation voltage regulation and differential-mode equalization control method proposed in Example 1, a simulation model of the CLLC type DC traction transformer with five modules connected in series at the input and parallel at the output was built based on MATLAB / Simulink. The simulation model mainly includes a medium-voltage DC traction network, five CLLC resonant oscillator modules, an on-board DC bus, a DC load, a common-mode frequency converter voltage regulator, a differential-mode equalization controller, and a secondary-side synchronous rectification control circuit.

[0144] In the simulation, the rated voltage of the medium-voltage DC traction network is set to 20kV, and the rated voltage of the vehicle-mounted DC bus is set to 3600V. The system consists of five identical CLLC resonant submodules. The high-voltage side ports of each submodule are connected in series to the medium-voltage DC traction network, and the low-voltage side ports are connected in parallel to the vehicle-mounted DC bus. Under rated operating conditions, the high-voltage side input voltage of a single submodule is approximately 4000V, and the rated transmission power of a single module is 1MW. The reference turns ratio of the high-frequency transformer is n=10 / 9, the rated switching period is 100us, and the corresponding switching frequency is 10kHz. The resonant frequency is also set to 10kHz, so that the converter operates at the resonant point. The quality factor Q is set to 0.3, and k=5.

[0145] (2) CLLC gain adaptation range to be verified in the ZVS region

[0146] Based on the above calculations, the switching cycle adjustment range can be set to 92µs-112µs. Figure 6 The voltage gain curves of the submodules are presented when Q=0.3 and k=5. The voltage gain exhibits monotonicity within the switching period range of 92µs-112µs and does not exceed the soft-switching range. This ensures that each submodule remains within the operating range that satisfies the conditions for voltage gain, gain monotonicity, and soft switching.

[0147] (3) Verification of phased control of common-mode voltage regulation and differential-mode equalization under inconsistent module parameters

[0148] To verify the effectiveness of common-mode gain-adaptive voltage regulation and differential-mode equalization control, a simulation was conducted with inconsistent module parameters. In the simulation, the medium-voltage DC traction network voltage was maintained at 20kV, and the high-frequency transformer turns ratios of the first and third CLLC submodules were set to 0.995 and 1.005 times the baseline values, respectively. The control process consisted of three phases: a fixed-cycle operation phase from 0-0.2s; common-mode voltage regulation control at 0.2s; and differential-mode voltage and current equalization control at 0.4s. The simulation was set to the traction operating condition.

[0149] During the fixed-cycle operation phase, each submodule operates according to its rated cycle, and the system can achieve basic voltage conversion functions. However, due to deviations in the module turns ratio, there may be some deviations in the on-board DC bus voltage, the input voltage of each submodule, and the low-voltage side output current. After common-mode voltage regulation control is engaged at 0.2s, the controller adjusts the common switch cycle based on the feedback of the on-board DC bus voltage, gradually stabilizing the low-voltage side DC bus voltage around 3600V. This stage demonstrates that common-mode voltage regulation control can effectively adjust the overall system voltage gain and achieve on-board DC bus voltage regulation. The on-board DC bus voltage waveform is as follows: Figure 7 As shown.

[0150] Figure 8The waveforms of the changes in the on-board DC bus voltage and the common switching frequency are presented. It can be seen that after common-mode voltage regulation is activated at 0.2s, the common switching frequency adjusts according to the output voltage error, and the on-board DC bus voltage gradually recovers from the deviation state to around 3600V and remains stable. This result demonstrates that common-mode voltage regulation can achieve on-board DC bus voltage regulation by adjusting the common switching frequency shared by all CLLC submodules, thereby changing the overall system voltage gain.

[0151] However, when only common-mode control is engaged, the five sub-modules share the same common cycle command. The controller only adjusts the overall average operating point of the system, making it difficult to actively suppress the uneven distribution of input voltage and output current among the modules caused by turns ratio deviation. Therefore, in the 0.2-0.4s stage, although the on-board DC bus voltage can stabilize near the target value, there are still certain differences in the high-voltage side input voltage and low-voltage side output current of each sub-module. Figure 9 and Figure 10 The waveform from 0.2s to 0.4s is shown.

[0152] After further differential-mode equalization control at 0.4s, the controller generates module-level periodic corrections based on the deviations of the high-voltage side input voltage and low-voltage side port current of each submodule from their average values. These differential-mode corrections are then summed and added to the common periodic command, enabling each submodule to adjust its relative power while maintaining the system's average switching cycle constant. Figure 9 and Figure 10 The waveform after 0.4s shows that after the differential mode equalization control is put into operation, the high-voltage side input voltage of each sub-module gradually approaches the average value, and the difference in low-voltage side output current decreases. This indicates that the proposed differential mode equalization control can improve the voltage division consistency of the input series module and the current division consistency of the output parallel module.

[0153] It should be noted that under forward traction conditions, the low-voltage side port current represents the current output by each submodule to the on-board DC bus; under regenerative braking conditions, the energy flow is reversed, and the low-voltage side port current can be defined in the opposite direction or, after sign correction, participate in the equalization control. Therefore, this differential-mode current equalization method is not limited to forward traction conditions.

[0154] Figure 11 The waveforms of the output power of each submodule are given. It can be seen that when the module parameters are inconsistent, the addition of differential mode control ensures that the output power of each submodule can be stabilized at 1MW by balancing the output current of each submodule.

[0155] Figure 12The waveforms showing the changes in the final switching period, period correction, and sum of each submodule after differential-mode equalization control is activated are presented. It can be seen that the final switching period of each submodule varies slightly around the common switching period and remains within the preset gain adaptation range of 92µs-112µs. This result indicates that differential-mode equalization control improves the consistency of module load sharing without causing any submodule to deviate from the effective gain adjustment range of the CLLC resonant converter. Furthermore, after differential-mode control is activated, the period correction of each submodule exhibits a positive and negative distribution, used to adjust the relative operating points between the submodules. The sum of the differential-mode corrections of each submodule remains close to zero, indicating that differential-mode equalization control only changes the relative switching periods between the submodules and does not change the average switching period of the five submodules. This avoids significant interference from differential-mode equalization control to the common-mode voltage regulation control.

[0156] (4) Verification of common-mode gain adaptation and voltage regulation under traction network voltage fluctuation

[0157] To verify the adaptability of the proposed common-mode gain adaptive voltage regulation control to changes in the medium-voltage DC traction network voltage, a further simulation was performed with varying traction network input voltage. In the simulation, the initial medium-voltage DC input voltage was reduced to 19kV, while the on-board DC bus reference voltage remained at 3600V. Since the effectiveness of the differential-mode voltage and current balancing method has been verified in section 6.2, the parameters of each CLLC submodule were restored to consistency.

[0158] Under this operating condition, the high-voltage side input voltage of a single CLLC submodule decreases. If the switching cycle remains unchanged, the on-board DC bus voltage will show a downward trend. After common-mode voltage regulation is activated, the controller adjusts the common switching cycle according to the bus voltage error, thereby increasing the voltage gain of the CLLC submodule and compensating for the impact of the input voltage drop on the output voltage. Simulation results show that after the input voltage drops, if common-mode voltage regulation is activated at 0.2s, the on-board DC bus voltage can still stabilize around 3600V after a brief dynamic process. Figure 13 The waveforms of the high-voltage side input voltage and the vehicle DC voltage under this operating condition are given.

[0159] The results demonstrate that the common-mode voltage regulation control proposed in this invention can adapt to changes in the voltage of the medium-voltage DC traction network and maintain the stability of the on-board DC bus voltage when the input voltage of the traction network decreases.

[0160] (5) Verification of bidirectional energy transfer under regenerative braking conditions

[0161] To verify the bidirectional energy transmission capability of the proposed ISOP-CLLC DC traction transformer, a regenerative braking condition was further established. Under this condition, the onboard 3600V DC bus serves as the energy input, and the low-voltage side full-bridge of each CLLC submodule acts as an active inverter bridge, converting low-voltage DC energy into high-frequency AC energy. This AC energy is then transmitted to the high-voltage side via the secondary resonant network, the high-frequency isolation transformer, and the primary resonant network. The high-voltage side ports of each submodule are connected in series and superimposed to form a medium-voltage DC output, which feeds back energy to the DC traction network.

[0162] Figure 14 This study demonstrates that, under a 3600V on-board DC bus input condition, a single CLLC submodule's high-voltage side output port can output 4000V DC voltage, and five CLLC submodules connected in series on the high-voltage side can output 20kV DC voltage. This proves that the proposed topology can not only achieve forward traction power supply but also reverse transmission of regenerative braking energy from the on-board DC bus to the medium-voltage DC traction network, verifying the bidirectional energy flow capability of this topology.

[0163] It should be noted that the regenerative braking simulation in this section is mainly used to verify the reverse voltage transformation capability and the feasibility of the bidirectional energy channel of the proposed topology. In practical engineering applications, a regenerative braking closed-loop control strategy can be further designed based on the traction network voltage, the on-board DC bus voltage, and the regenerative power requirements.

[0164] (6) Comparison and verification with the ISOP-DAB scheme

[0165] Figure 15This demonstrates the physical process by which the ISOP-DAB transformer, under the same conditions of 20000V on the high-voltage side and 3600V on the low-voltage side, fails to achieve zero-current turn-off on its secondary side using traditional single-phase-shift control. Observing the falling edge of the drive pulse signal in the figure (around 0.53246s), the branch current flowing through the switching transistor does not naturally cross zero, but is at a high amplitude of approximately 400A~500A. Since the DAB topology mainly relies on leakage inductance or external inductance for power transmission, its current waveform is mainly determined by the phase shift relationship. Compared with the CLLC double-sided resonant structure, its current shaping capability is limited, and the current is difficult to naturally fall back, forcing the switching transistor to forcibly cut off the load current. This causes the terminal voltage across the switching transistor to rise sharply in a very short time, momentarily bearing a 3600V DC bus blocking voltage. During this turn-off transient, the high voltage and high current severely overlap, increasing switching losses. In addition, the forced interruption of non-zero current also drastically worsens the reverse recovery stress of the body diode on the other side of the bridge arm. Therefore, compared to CLLC type DC transformers, ISOP-DAB type transformers, under current operating conditions, if single-phase-shift control is used, will experience non-zero current commutation issues in the secondary-side switching devices due to the input / output voltage matching relationship and load conditions. This increases switching losses and thermal stress under wide voltage and power ranges. Although multi-degree-of-freedom control strategies such as extended phase-shifting and triple phase-shifting can further improve the soft-switching range, they increase the complexity of control variables and modulation logic, reducing the difficulty of control system implementation and the ease of engineering application.

[0166] In contrast, the CLLC submodule incorporates resonant networks on both the primary and secondary sides. After the full-bridge square wave voltage passes through the resonant cavity, the resonant current exhibits an approximately sinusoidal variation, shaping the current during high-frequency energy transfer. With appropriate switching frequency and dead-time settings, the resonant current provides conditions for the charging and discharging of the output capacitors of the switching devices, significantly improving soft-switching conditions and reducing switching current surges and high-frequency harmonic content. Therefore, compared to the ISOP-DAB scheme, the ISOP-CLLC scheme has inherent advantages in reducing switching stress, improving soft-switching conditions, and enhancing the potential for high-efficiency operation in medium-voltage, high-power, and wide-condition EMU DC traction transformer applications. Figure 16 The waveforms of the full-bridge output voltage and resonant current of the CLLC submodule are given.

[0167] It should be noted that this section's comparison is primarily based on a mechanistic explanation of typical switching transient waveforms, and does not directly provide a conclusion on full-condition efficiency from a single waveform. A complete efficiency comparison requires further investigation considering device parameters, load range, control strategy, and thermal design conditions. The comparison results in this paper mainly illustrate that, under the same voltage level and similar power conditions, CLLC resonant cavities can improve potential problems with DAB topologies in terms of non-zero current turn-off, circulating current power, and switching stress.

[0168] (7) Conclusion

[0169] Based on the comprehensive simulation results of positive traction, grid voltage change, regenerative braking, and ISOP-DAB comparison, it can be seen that the ISOP-CLLC type EMU DC traction transformer proposed in this invention can realize isolated DC voltage transformation and bidirectional energy transmission between the 20kV medium-voltage DC traction grid and the 3600V on-board DC bus.

[0170] Common-mode gain-adaptive voltage regulation control can adjust the common switching cycle based on the feedback of the on-board DC bus voltage, enabling the system to maintain stable on-board DC bus voltage even when module parameters are inconsistent. When only common-mode control is used, there may still be uneven distribution of high-voltage side input voltage and low-voltage side port current among the sub-modules. After the zero-sum differential-mode equalization control is implemented, the high-voltage side input voltage and low-voltage side port current of each sub-module gradually approach the average value, indicating that the proposed differential-mode voltage and current equalization control can improve the consistency of input voltage division and low-voltage side port current distribution among modules.

[0171] Compared to the ISOP-DAB scheme, the ISOP-CLLC submodule uses a primary-secondary resonant network to resonate and shape the high-frequency current, making the resonant current exhibit an approximately sinusoidal variation. This is beneficial for improving the commutation conditions of the switching devices and reducing switching stress. Therefore, the ISOP-CLLC type DC traction transformer and its common-mode gain adaptation voltage regulation and differential-mode equalization control method proposed in this invention can better meet the requirements of medium-voltage DC power supply EMUs for voltage level matching, electrical isolation, modular voltage division and parallel current, bidirectional energy transmission, and operational adaptability of the on-board DC traction transformer.

[0172] The above description is merely a preferred embodiment of the present invention and is not intended to limit the invention. Various modifications and variations can be made to the present invention by those skilled in the art. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A medium-voltage DC-powered electric multiple unit (EMU) ISOP-CLLC traction transformer topology, characterized in that, The traction transformer consists of N isolated bidirectional CLLC resonant converter submodules. Each submodule adopts the same topology, and the parameters of each submodule may be the same or different. The high-voltage input terminals of each submodule are connected in series, and the low-voltage output terminals are connected in parallel, forming an input-series, output-parallel structure, where N≥2. The isolated bidirectional CLLC resonant converter includes a high-voltage full-bridge converter unit, a primary-side resonant capacitor C1, a primary-side resonant inductor L1, a high-frequency isolation transformer T, and a magnetizing inductor L. m Secondary resonant inductor L2, secondary resonant capacitor C2, low-voltage side full-bridge converter unit, and DC support capacitors on both sides C d1 and C f1 The high-voltage side full bridge consists of four controllable switching devices and their anti-parallel diodes, namely S1-S4. The low-voltage side full bridge also consists of four controllable switching devices and their anti-parallel diodes, namely Q1-Q4. The primary resonant inductor L1 and the primary resonant capacitor C1 form a primary series resonant network, and the secondary resonant inductor L2 and the secondary resonant capacitor C2 form a secondary series resonant network.

2. The ISOP-CLLC traction transformer topology for medium-voltage DC-powered electric multiple units according to claim 1, characterized in that, The traction transformer also includes a common-mode frequency converter control module for determining the common switching cycle based on the traction network voltage change, and a module equalization control module for generating zero and differential mode corrections based on the input voltage deviation and output current deviation of each submodule.

3. The ISOP-CLLC traction transformer topology for medium-voltage DC-powered electric multiple units according to claim 1, characterized in that, The number of submodules is N=5, and the rated power of a single isolated bidirectional CLLC resonant submodule is... It is 1MW.

4. The ISOP-CLLC traction transformer topology for medium-voltage DC-powered electric multiple units according to claim 1, characterized in that, For a single submodule, the primary-to-secondary turns ratio of the high-frequency isolation transformer is n=10 / 9, the magnetizing inductance ratio is k=5, and the quality factor of the CLLC resonant cavity is Q=0.

3.

5. A traction drive system composed of the ISOP-CLLC traction transformer topology of the medium-voltage DC power supply electric multiple unit as described in claim 1, characterized in that, It includes a medium-voltage DC traction network, a DC traction transformer, a three-phase inverter, and a traction motor arranged sequentially, among which, The high-voltage input terminals of each sub-module of the DC traction transformer are connected in series to the medium-voltage DC traction network, and the low-voltage output terminals of each sub-module of the DC traction transformer are connected in parallel to the vehicle-mounted DC bus.

6. The control method for the traction drive system according to claim 5, characterized in that, The control methods include common-mode frequency conversion voltage regulation control and differential-mode equalization control.

7. The control method according to claim 6, characterized in that, The common-mode frequency converter voltage regulation control is as follows: based on the traction network voltage fluctuation range, the target voltage of the on-board DC bus, the number of sub-modules, the high-frequency transformer ratio, and the CLLC gain curve, the frequency converter operating range that meets the requirements of gain adjustment, gain monotonicity, and soft switching operation is determined, and the overall voltage gain of each sub-module is adjusted by the common switch cycle or common switch frequency to keep the on-board DC bus stable under the traction network voltage fluctuation conditions.

8. The control method according to claim 6, characterized in that, The differential mode equalization control is as follows: a module-level correction amount is generated based on the input voltage and low-voltage side port current deviation of each sub-module, and the differential mode correction amount is processed by zero-sum and superimposed on the common switching cycle command to form an independent switching cycle for each sub-module without changing the common average cycle of the system. Thus, without destroying the voltage regulation effect of the on-board DC bus, the input series module voltage equalization, the output parallel module current equalization, and the power sharing coordination are achieved.

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