Improved double active bridge converter double-sided variable duty ratio phase-shift modulation method of full-soft switching

Abstract

An improved dual-side duty cycle phase-shift modulation method for a fully soft-switching dual active bridge converter belongs to the field of converter control technology. This invention addresses the problem that existing converter modulation methods cannot achieve zero-voltage turn-on under light loads, leading to decreased efficiency. It includes: introducing switch parasitic capacitance constraints based on traditional DVDM control; analyzing the resonant commutation process of inductors and parasitic capacitances; ensuring that the inductor provides sufficient energy to complete the charging and discharging of the bridge arm parasitic capacitances before the switch turns on, thereby actively reducing the voltage before the switch turns on and achieving zero-voltage turn-on for each switch; and establishing optimized control laws for different power ranges by minimizing the peak-to-peak inductor current, combined with power transfer conditions, operating mode boundaries, and ZVS constraints, clarifying mode selection and switching boundaries. This invention can extend the ZVS range under light loads while maintaining the original high efficiency characteristics under heavy loads, taking into account low current stress and full-power range soft switching.

Description

Technical Field

[0001] This invention relates to an improved dual active bridge converter with fully soft switching and a bilateral variable duty cycle phase shift modulation method, belonging to the field of converter control technology. Background Technology

[0002] Dual active bridge (DAB) converters have been widely used in DC microgrids, energy storage interfaces, solid-state transformers and multi-port energy management systems due to their advantages such as bidirectional power flow, high-frequency isolation and high power density.

[0003] Traditional symmetrical duty cycle modulation methods, such as single-phase shift, extended-phase shift, dual-phase shift, and triple-phase shift, have simple structures and are relatively easy to implement soft switching under medium to high loads. However, under light loads or voltage mismatch conditions, the inductor commutation energy is insufficient, the zero-voltage turn-on range is significantly reduced, and large circulating currents, reverse power, and current stress are easily generated, leading to increased conduction and switching losses and decreased efficiency under light loads. While some multi-degree-of-freedom phase-shift control can optimize the inductor current waveform and reduce current stress, it requires consideration of factors such as dead time, parasitic capacitance, and nonlinear output capacitance of devices, resulting in complex control models and high requirements for real-time computing power and parameter accuracy, making engineering implementation difficult. Asymmetric duty cycle modulation and dual-sided variable duty cycle phase-shift modulation can expand the soft-switching range and improve efficiency under light loads, but they usually bring problems such as complex mode classification, parameter sensitivity, and unclear mode switching boundaries.

[0004] Existing bilateral variable duty cycle modulation (DVDM) methods primarily establish zero-voltage switching (ZVS) conditions based on the direction of inductor current, without fully considering the energy required for charging and discharging the parasitic capacitance of the switches. This results in some switches still failing to achieve zero-voltage turn-on under light loads, limiting the improvement in soft-switching reliability and efficiency. Therefore, current technologies struggle to simultaneously achieve full-switching ZVS, low current stress, low control complexity, and high efficiency across a wide voltage range and full power range. Summary of the Invention

[0005] To address the problem that existing converter modulation methods cannot achieve zero-voltage turn-on under light load, resulting in decreased efficiency under light load, this invention provides an improved dual-active bridge converter bilateral variable duty cycle phase-shift modulation method with fully soft switching.

[0006] This invention discloses an improved dual-active-bridge converter with fully soft switching and a bilateral variable duty cycle phase-shift modulation method. The dual-active-bridge converter includes a primary-side full-bridge and a secondary-side full-bridge; the primary-side full-bridge includes upper-arm switching transistors. and lower bridge arm switch tube The primary side bridge arm consists of one component and the upper bridge arm switch tube. and lower bridge arm switch tube The primary side bridge arm consists of two sections; the secondary side bridge consists of the upper bridge arm switch tube. and lower bridge arm switch tube The secondary side bridge arm consists of one component and the upper bridge arm switch tube. and lower bridge arm switch tube The second secondary bridge arm is formed by the sum of the transformer leakage inductance and the series auxiliary inductance, which is the inductance. The methods include,

[0007] definition for and The shift between them; Compared to the original edge's internal shift; Compared to the external shift of the primary edge and the secondary edge;

[0008] Two control modes are set:

[0009] Mode 1: ;

[0010] Mode 2: ;

[0011] Based on inductance Based on the volt-second balance principle and reference values ​​of inductor current and transmission power, the normalized peak-to-peak inductor current and normalized transmission power are calculated, and the expressions for the normalized peak-to-peak inductor current and normalized transmission power in mode one and mode two are obtained respectively.

[0012] In the first mode, inequality constraints and ZVS constraints are determined, and KKT conditions for soft turn-on of eight switches are defined. Based on the KKT conditions, a normalized objective function for the inductor peak-to-peak current is established, and the solution is obtained. , and The optimal solution is obtained similarly under mode two. , and The optimal solution;

[0013] In both modes , and Under the optimal solution, for the overlapping region of mode 1 and mode 2, the optimal solution corresponding to the smaller value of the normalized inductor peak-to-peak current is selected as the optimal value of the phase shift ratio, and finally the final value of the phase shift ratio of the bilateral variable duty cycle phase shift modulation is obtained.

[0014] According to the improved dual active bridge converter bilateral variable duty cycle phase shift modulation method of the present invention, the internal phase shift ratio of the primary side is... for and or and Compared to the shifts between the two sides; compared to the external shifts between the primary edge and the secondary edge. for and The shifts between them.

[0015] The improved dual-active-bridge converter bilateral variable duty cycle phase-shift modulation method of the present invention, based on inductance, is a fully soft-switching method. The volt-second balance principle yields:

[0016] ,

[0017] In the formula For inductor current, For inductor voltage, let's try it out. In order Conduction timing, Conduction timing, and Conduction timing, Conduction timing, Conduction timing, Conduction timing, Conduction timing and Conduction timing; Mode 2 In order Conduction timing, Conduction timing, and Conduction timing, Conduction timing, Conduction timing, Conduction timing, Conduction timing and Conduction timing; for Time and The time difference between moments;

[0018] Inductor current reference value and transmission power reference value Defined as:

[0019] ,

[0020] In the formula For transformer turns ratio, The primary side DC voltage, This is the secondary DC voltage. The switching frequency;

[0021] Calculate the normalized peak-to-peak inductor current and normalized transmission power for:

[0022] ,

[0023] In the formula For time, For the switching cycle, This is the voltage at the midpoint of the primary side of the full-bridge.

[0024] Define voltage transfer ratio ,when ,get:

[0025] Let's try it out: ;

[0026] Mode 2 (or below): .

[0027] According to the improved dual active bridge converter bilateral variable duty cycle phase shift modulation method of the present invention with full soft switching, in mode one, the inequality constraints and ZVS constraints are:

[0028] ,

[0029] ,

[0030] ,

[0031] In the formula For the parasitic capacitance of the switching transistor, for or for, This is the critical ZVS current on the primary side. This is the critical ZVS current on the secondary side after being referred to the primary side;

[0032] Based on the inductor current reference value and transmission power reference value ,get and per unit value and :

[0033] ;

[0034] The constraints are then simplified to:

[0035] ,

[0036] In the formula for The inductor current after time-normalization.

[0037] According to the improved dual active bridge converter bilateral variable duty cycle phase shift modulation method of the present invention, in mode one, the KKT conditions for soft turn-on of eight switches are defined. for:

[0038] ,

[0039] make and ;

[0040] Establish the objective function:

[0041] .

[0042] According to the improved dual-active-bridge converter bilateral variable duty cycle phase-shift modulation method of the present invention, the method for solving the objective function is as follows:

[0043] Define the Lagrange function based on the objective function. :

[0044] ,

[0045] In the formula For the equality-bound Lagrange multiplier one, For equality-bound Lagrange multipliers two, For the corresponding Lagrange multipliers constrained by inequality;

[0046] Establish the corresponding KKT conditions to obtain the optimality criterion:

[0047] ;

[0048] The calculation yields the following pattern: , and The optimal solution:

[0049] ,

[0050] In the formula , The normalized upper limit of transmission power for the mode is:

[0051] ;

[0052] This leads to the normalized peak-to-peak inductor current. for:

[0053] .

[0054] According to the improved dual-active-bridge converter bilateral variable duty cycle phase-shift modulation method of the present invention, which is based on the same principle as mode two, the simplified form of the constraint conditions in mode two is as follows:

[0055] .

[0056] According to the improved dual-active bridge converter bilateral variable duty cycle phase shift modulation method of the present invention, in mode two, the KKT conditions for soft turn-on of eight switches are defined. for, :

[0057] ;

[0058] The objective function is defined as follows:

[0059] .

[0060] The beneficial effects of this invention are as follows: This invention improves soft-switching capability under light load conditions without significantly increasing control complexity. The method of this invention introduces switch parasitic capacitance constraints based on traditional DVDM control, analyzes the resonant commutation process of inductors and parasitic capacitances, and ensures that the inductor provides sufficient energy to complete the charging and discharging of the bridge arm parasitic capacitances before the switch is turned on, thereby actively reducing the voltage before the switch is turned on and achieving zero-voltage turn-on for each switch. This invention aims to minimize the peak-to-peak inductor current, and, combined with power transfer conditions, operating mode boundaries, and ZVS constraints, establishes optimized control laws for different power ranges, clarifying mode selection and switching boundaries. This expands the ZVS range under light loads while maintaining the original high efficiency characteristics under heavy loads, taking into account low current stress and full-power range soft switching. Attached Figure Description

[0061] Figure 1 This is a topology diagram of the dual active bridge converter described in this invention;

[0062] Figure 2 This is the steady-state waveform diagram for Mode 1;

[0063] Figure 3 This is the steady-state waveform diagram for mode two;

[0064] Figure 4 This is the steady-state waveform diagram for Mode 3;

[0065] Figure 5 It is the normalized transmission power of Mode 1. With voltage transfer ratio A curve graph;

[0066] Figure 6 It is the normalized transmission power of Mode 2. With voltage transfer ratio A curve graph;

[0067] Figure 7 This is a diagram showing the mode selection of the method of the present invention across the full power range;

[0068] Figure 8 It is the voltage transfer ratio Peak-to-peak inductor current in both modes within the range of 1 to 3, at the same power rating. Comparison chart;

[0069] Figure 9 This is a steady-state waveform diagram of the dual active bridge converter controlled by the method of this invention in the verification experiment;

[0070] Figure 10 yes , The voltage and drive signal waveforms of the dual active bridge converter switches are controlled using the method of this invention.

[0071] Figure 11 yes , The voltage and drive signal waveforms of the dual active bridge converter switches are controlled using the method of this invention. Detailed Implementation

[0072] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0073] Specific Implementation Method 1: Combination Figures 1 to 7 As shown, this invention provides an improved dual-side variable duty cycle phase-shift modulation method for a fully soft-switching dual active bridge converter. The dual active bridge converter includes a primary-side full bridge and a secondary-side full bridge; the primary-side full bridge includes upper-side switching transistors... and lower bridge arm switch tube The primary side bridge arm consists of one component and the upper bridge arm switch tube. and lower bridge arm switch tube The primary side bridge arm consists of two sections; the secondary side bridge consists of the upper bridge arm switch tube. and lower bridge arm switch tube The secondary side bridge arm consists of one component and the upper bridge arm switch tube. and lower bridge arm switch tube The second secondary bridge arm is formed by the sum of the transformer leakage inductance and the series auxiliary inductance, which is the inductance. ;include,

[0074] definition for and The shift between them; Compared to the original edge's internal shift; Compared to the external shift of the primary edge and the secondary edge;

[0075] Two control modes are set:

[0076] Mode 1: ;

[0077] Mode 2: ;

[0078] Based on inductance Based on the volt-second balance principle and reference values ​​of inductor current and transmission power, the normalized peak-to-peak inductor current and normalized transmission power are calculated, and the expressions for the normalized peak-to-peak inductor current and normalized transmission power in mode one and mode two are obtained respectively.

[0079] In the first mode, inequality constraints and ZVS constraints are determined, and KKT conditions for soft turn-on of eight switches are defined. Based on the KKT conditions, a normalized objective function for the inductor peak-to-peak current is established, and the solution is obtained. , and The optimal solution is obtained similarly under mode two. , and The optimal solution;

[0080] In both modes , and Under the optimal solution, for the overlapping region of mode 1 and mode 2, the optimal solution corresponding to the smaller value of the normalized inductor peak-to-peak current is selected as the optimal value of the phase shift ratio, and finally the final value of the phase shift ratio of the bilateral variable duty cycle phase shift modulation is obtained.

[0081] The topology of the DAB converter is as follows: Figure 1 As shown. Figure 1 middle and These are the input and output filter capacitors, respectively; the high-frequency transformer has a turns ratio of n; U ab and U cd These represent the midpoint voltages of the primary and secondary sides of the full-bridge circuit, respectively.

[0082] To ensure that the DAB circuit can operate stably over a wide voltage range under the DVDM scheme, according to , and Regarding the relationship between the two, the control mode of this implementation also includes mode three:

[0083] ;

[0084] Figures 2 to 4 Steady-state waveforms for three operating modes are given.

[0085] Furthermore, the internal shift of the original edge compared to for and or and Compared to the shifts between the two sides; compared to the external shifts between the primary edge and the secondary edge. for and The shifts between them.

[0086] Furthermore, based on inductors The volt-second balance principle yields:

[0087] ,

[0088] In the formula For inductor current, For inductor voltage, let's try it out. In order Conduction timing, Conduction timing, and Conduction timing, Conduction timing, Conduction timing, Conduction timing, Conduction timing and Conduction timing; Mode 2 In order Conduction timing, Conduction timing, and Conduction timing, Conduction timing, Conduction timing, Conduction timing, Conduction timing and Conduction timing; for Time and The time difference between moments;

[0089] For ease of subsequent analysis, the inductor current reference value is... and transmission power reference value Defined as:

[0090] ,

[0091] In the formula For transformer turns ratio, The primary side DC voltage, This is the secondary DC voltage. The switching frequency;

[0092] Calculate the normalized peak-to-peak inductor current and normalized transmission power for:

[0093] ,

[0094] In the formula For time, For the switching cycle, This is the voltage at the midpoint of the primary side of the full-bridge.

[0095] Define voltage transfer ratio This implementation method only considers In this case, we get:

[0096] Let's try it out: ;

[0097] Mode 2 (or below): .

[0098] Mode 3 (or below): .

[0099] Taking Mode 1 as an example, before the corresponding switch is turned on, the parasitic capacitance must be discharged to reduce its voltage to zero. The polarity of the inductor current can be determined based on the direction of the discharge current, thus establishing the ZVS constraint.

[0100] However, in practical DAB converters, achieving ZVS requires not only the correct direction of the inductor current but also a sufficiently large inductor current amplitude. Especially under light load conditions, the ZVS range of traditional modulation methods is extremely narrow. Therefore, to improve light load efficiency, specific ZVS constraints must be considered, namely, requiring the inductor to store enough energy so that the output capacitors on both sides of the same bridge arm can complete charging and discharging within the dead time.

[0101] In this model, the inequality constraints and ZVS constraints are as follows:

[0102] To ensure that the drain-source voltage drops to zero before the switch is turned on, the energy stored in the inductor must satisfy the following:

[0103] ;

[0104] For primary-side switches to The bridge arm commutation voltage is Therefore, the critical current of the primary-side ZVS can be expressed as: ;

[0105] Similarly, for the secondary switch to The secondary arm commutation voltage is The critical ZVS current on the secondary side, after being referred to the primary side, can be expressed as:

[0106] ,

[0107] In the formula For the parasitic capacitance of the switching transistor, for or for, This is the critical ZVS current on the primary side. This is the critical ZVS current on the secondary side after being referred to the primary side;

[0108] Based on the inductor current reference value and transmission power reference value The derivation yields and per unit value and :

[0109] ;

[0110] In practical calculations and applications, especially under light load conditions, the critical current value can be set slightly higher than the theoretical value to expand the soft-switching range.

[0111] However, considering the specific capacitance constraints required for all switching devices to achieve ZVS increases the complexity of analysis and calculation, as well as the complexity of the optimal solution. Therefore, there is a trade-off between optimization complexity and power loss reduction. In this embodiment, for mode one, and All are located on the secondary side of the entire bridge, and in Since both are always on, their ZVS boundary conditions are the same. and Corresponding to and The inductor current must be zero at these two moments: when they are on the secondary side of the full-bridge and when they are on the primary side, respectively, and the required ZVS currents are in opposite directions. Finally, we also need to consider... Time-based ZVS constraints to ensure Soft switching is achieved. Therefore, ZVS for five switching devices can be achieved with only three ZVS constraints. Since... and The conduction times occur at the peak and trough of the inductor current, respectively, while The conduction time is located at the falling edge of the inductor current, so ZVS can be automatically satisfied without additional constraints.

[0112] Therefore, the constraints are simplified to:

[0113] ,

[0114] In the formula for The inductor current after time-normalization.

[0115] Based on the normalized peak-to-peak inductor current and normalized transmission power under the three modes, and control variables , and If a set of relationships can be found , and By minimizing the peak-to-peak inductor current while satisfying power requirements and ZVS constraints, current stress can be reduced while achieving ZVS for all switching devices, thereby improving power transmission efficiency.

[0116] Under light load conditions, the ZVS range of the DAB converter is narrow, while ZVS is more easily achieved under heavy load conditions. Therefore, if only the inductor current direction is considered under light load conditions, it is difficult to ensure that all switches achieve ZVS. Specific ZVS constraints, i.e., simplified constraints, must be considered to actively reduce the switching voltage to 0 before turn-on.

[0117] Taking mode one as an example, Figure 2 The steady-state waveform for mode one is given. Within a certain power range, if... , and If the constraints can be satisfied simultaneously, then eight switches on the primary and secondary sides can be implemented. Therefore, considering both the inequality constraints and ZVS constraints in Mode 1, the KKT conditions for soft turn-on of the eight switches in Mode 1 can be determined. Defined as:

[0118] ,

[0119] make and This ensures that the inequality constraints and ZVS constraints of Mode 1 are satisfied simultaneously.

[0120] Further combining the power condition and the peak-to-peak inductor current, the objective function is established as follows:

[0121] .

[0122] The method for solving the objective function is as follows: the Karush-Kuhn-Tucker (KKT) conditions are a classic tool for solving optimization models that simultaneously contain equality and inequality constraints, and are very suitable for constrained optimization analysis. To construct a solution framework for such problems, the Lagrangian function is usually defined first.

[0123] For the constrained optimization model of Mode 1, the Lagrangian function is defined according to the objective function. :

[0124] ,

[0125] In the formula For the equality-bound Lagrange multiplier one, For equality-bound Lagrange multipliers two, For the corresponding Lagrange multipliers constrained by inequality;

[0126] Based on this, the corresponding KKT conditions are established, and the optimality criteria for the optimization problem are obtained:

[0127] ;

[0128] The calculation yields the following pattern: , and The optimal solution:

[0129] ,

[0130] In the formula , The normalized upper limit of transmission power for the mode is:

[0131] ;

[0132] This leads to the normalized peak-to-peak inductor current. for:

[0133] .

[0134] when When the inequality constraints cannot be satisfied simultaneously, the original optimization problem has no optimal solution. Therefore, Mode 1 only has an optimal solution that satisfies all constraints within the power range.

[0135] Furthermore, similar to the previous model, under Model 2, the simplified form of the constraints is:

[0136] .

[0137] For mode 3, the soft-switching constraints are more complex, and subsequent analysis shows that its feasible power range is extremely narrow. Therefore, the control method for mode 3 is not considered in this implementation.

[0138] In Mode 2, KKT conditions for soft turn-on of eight switching transistors are defined. for, :

[0139] ;

[0140] The objective function is defined as follows:

[0141] .

[0142] In Mode 2, the method for solving the objective function is as follows:

[0143] Define the Lagrange function based on the objective function. :

[0144] ,

[0145] activation and Two constraints, will and If all Lagrange multipliers are set to 0, then the optimal solution that satisfies all constraints can be obtained.

[0146] Solving the three power ranges under mode two , and The optimal solution:

[0147] Set the upper limit of transmission power after normalization in the low power range to be [value]. :

[0148] ,

[0149] low power range , and The optimal solution is:

[0150] ;

[0151] This leads to the normalized peak-to-peak inductor current. for:

[0152] ;

[0153] It can be seen that in the low power range , and The optimal solution completely satisfies all inequality constraints.

[0154] when When this happens, the above optimized solution will not satisfy all constraints, among which... and It will reach the critical value first, therefore the constraints need to be adjusted. , and Incorporate these constraints into the Lagrange equations. Once the above constraints are introduced into the Lagrange equations... It will be automatically satisfied, and other ZVS constraints will also be satisfied.

[0155] The upper limit of the transmission power after normalization of the medium power range is set to be [value]. :

[0156] ;

[0157] medium power range , and The optimal solution is:

[0158] ;

[0159] This leads to the normalized peak-to-peak inductor current. for:

[0160] ;

[0161] After normalization, the upper limit of transmission power in the high-power range is 1, requiring only activation. The high power range can then be obtained. , and The optimal solution:

[0162] ;

[0163] Obtain the normalized peak-to-peak inductor current for:

[0164] ;

[0165] The power range in the low power range is The power range in the medium power range is The power range in the high-power range is .

[0166] when equal , Satisfy the following equation:

[0167] ,

[0168] set up The solution to the equation; if , , and The optimal solution is divided into three power intervals; if Mode 2 only exists in the low power range and the high power range.

[0169] Pick , Therefore, the boundary conditions are as follows: when When it is greater than 1.32, mode 2 has the above three intervals; when When the value is less than 1.32, there is no medium power range in Mode 2.

[0170] For Mode 1, the optimal solution exists only in the interval. ;when When there is no feasible optimal solution, the boundary condition is: by Impact. The segmentation of Mode 1 and Mode 2 is as follows: Figure 5 and Figure 6 As shown. In addition to the above analysis, the selection of the operating mode is also based on the optimized inductor peak-to-peak current. When When it is greater than 1.32:

[0171] 1) If The peak-to-peak inductor current in Mode 1 is lower than that in Mode 2, so the optimized solution in Mode 1 is selected as the control strategy.

[0172] 2) If The peak-to-peak inductor current in Mode 1 is still smaller than that in Mode 2, so the optimal solution of Mode 1 is still selected as the control strategy.

[0173] 3) If The peak-to-peak inductor current in mode 1 will exceed that in mode 2, so the optimal solution of mode 2 is selected as the control strategy.

[0174] 4) If If no feasible optimal solution exists for Mode 1, then Mode 2 must be selected.

[0175] The final phase shift value of the bilateral variable duty cycle phase shift modulation is:

[0176] .

[0177] It should be noted that when the power is at Within this range, the duty cycle of all switching devices is 0.5. In this case, the control scheme is essentially the same as the EPS scheme. Specifically, compared with the original DVDM, the improved DVDM proposed in this invention has better performance in the light load power range. Optimizations are made to achieve soft switching; while in the heavy-load power range Internally, its control strategy is the same as the original DVDM.

[0178] Based on the above analysis, when When the value is between 1 and 1.32, the medium power range of Mode 2 disappears. Therefore, when When selecting mode one; and when Exceed At that time, the system will switch directly to mode two.

[0179] along with Increase, in the medium power range If the peak-to-peak inductor current in Mode 1 exceeds that in Mode 2, then it is necessary to switch to Mode 2 in advance to further reduce current stress.

[0180] Therefore, for those between 1 and 3 value, Figure 7 It shows the selection of each mode across the full power range.

[0181] Figure 8 Comparison Two modes within the range of 1 to 3, at the same power level. The size thus verifies the above analysis.

[0182] Verification Experiment: To verify the above theoretical analysis, a 100 kHz, 10 A experimental prototype was built, with an input voltage of 50 V and an output voltage of 25 V or 41.67 V.

[0183] Since the control strategy under heavy load conditions is consistent with that of DVDM, only the experimental waveforms under light load conditions are given. Figure 9 When , as well as , The steady-state operating waveforms of the DVDM control of this invention are shown under different transmission power conditions. Regarding soft switching, analysis is performed separately. and Soft-switching behavior during operation. Figure 10 Given , The switching voltage and drive signal are controlled by the method of this invention. Figure 11 Given , The switching voltage and drive signal under the control of the method of this invention are as follows: At that time, the duty cycle of all switches is not 50%, achieving asymmetric modulation. Furthermore, regardless of... Whether set to 2 or 1.2, the method of this invention can be used in [the following context]. This enables all eight primary and secondary switches to achieve zero-voltage switching. In contrast, existing DVDM solutions cannot achieve this under light load conditions. and ZVS is achieved. Therefore, the method of this invention outperforms existing methods in terms of ZVS performance. Figure 9 In the text, "time" represents the time period. Figure 10 middle for The drain-source voltage, for The gate-source drive voltage, where Duty is the duty cycle.

[0184] While the invention has been described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the invention. Therefore, it should be understood that many modifications can be made to the exemplary embodiments, and other arrangements can be designed without departing from the spirit and scope of the invention as defined by the appended claims. It should be understood that different dependent claims and features described herein can be combined in ways different from those described in the original claims. It is also understood that features described in conjunction with individual embodiments can be used in other described embodiments.

Claims

1. An improved dual-active bridge converter with fully soft switching and dual-side variable duty cycle phase-shift modulation method, wherein the dual-active bridge converter includes a primary-side full-bridge and a secondary-side full-bridge; the primary-side full-bridge includes upper-arm switching transistors... and lower bridge arm switch tube The primary side bridge arm consists of one component and the upper bridge arm switch tube. and lower bridge arm switch tube The primary side bridge arm consists of two sections; the secondary side bridge consists of the upper bridge arm switch tube. and lower bridge arm switch tube The secondary side bridge arm consists of one component and the upper bridge arm switch tube. and lower bridge arm switch tube The second secondary bridge arm is formed by the sum of the transformer leakage inductance and the series auxiliary inductance, which is the inductance. Its characteristics include, definition for and The shift between them; Compared to the original edge's internal shift; Compared to the external shift of the primary edge and the secondary edge; Two control modes are set: Mode 1: ; Mode 2: ; Based on inductance Based on the volt-second balance principle and reference values ​​of inductor current and transmission power, the normalized peak-to-peak inductor current and normalized transmission power are calculated, and the expressions for the normalized peak-to-peak inductor current and normalized transmission power in mode one and mode two are obtained respectively. In the first mode, inequality constraints and ZVS constraints are determined, and KKT conditions for soft turn-on of eight switches are defined. Based on the KKT conditions, a normalized objective function for the inductor peak-to-peak current is established, and the solution is obtained. , and The optimal solution is obtained similarly under mode two. , and The optimal solution; In both modes , and Under the optimal solution, for the overlapping region of mode 1 and mode 2, the optimal solution corresponding to the smaller value of the normalized inductor peak-to-peak current is selected as the optimal value of the phase shift ratio, and finally the final value of the phase shift ratio of the bilateral variable duty cycle phase shift modulation is obtained.

2. The improved dual active bridge converter bilateral variable duty cycle phase shift modulation method for fully soft switching according to claim 1, characterized in that, Compared to the original edge internal shift for and or and Compared to the shifts between the two sides; compared to the external shifts between the primary edge and the secondary edge. for and The shifts between them.

3. The improved dual active bridge converter bilateral variable duty cycle phase shift modulation method for fully soft switching according to claim 2, characterized in that, Based on inductance The volt-second balance principle yields: ， In the formula For inductor current, For inductor voltage, let's try it out. In order Conduction timing, Conduction timing, and Conduction timing, Conduction timing, Conduction timing, Conduction timing, Conduction timing and Conduction timing; Mode 2 In order Conduction timing, Conduction timing, and Conduction timing, Conduction timing, Conduction timing, Conduction timing, Conduction timing and Conduction timing; for Time and The time difference between moments; Inductor current reference value and transmission power reference value Defined as: ， In the formula For transformer turns ratio, The primary side DC voltage, This is the secondary DC voltage. The switching frequency; Calculate the normalized peak-to-peak inductor current and normalized transmission power for: ， In the formula For time, For the switching cycle, This is the voltage at the midpoint of the primary side of the full-bridge. Define voltage transfer ratio ,when ,get: Let's try it out: ; Mode 2 (or below): .

4. The improved dual active bridge converter bilateral variable duty cycle phase shift modulation method for fully soft switching according to claim 3, characterized in that, In this model, the inequality constraints and ZVS constraints are as follows: ， ， ， In the formula For the parasitic capacitance of the switching transistor, for or for, This is the critical ZVS current on the primary side. This is the critical ZVS current on the secondary side after being referred to the primary side; Based on the inductor current reference value and transmission power reference value ,get and per unit value and : ； The constraints are then simplified to: ， In the formula for The inductor current after time-normalization.

5. The improved dual active bridge converter bilateral variable duty cycle phase shift modulation method for fully soft switching according to claim 4, characterized in that, In the following mode, define the KKT conditions for soft turn-on of eight switching transistors. for: ， make and ; Establish the objective function: 。 6. The improved dual active bridge converter bilateral variable duty cycle phase shift modulation method for fully soft switching according to claim 5, characterized in that, The method for solving the objective function is as follows: Define the Lagrange function based on the objective function. : ， In the formula For the equality-bound Lagrange multiplier one, For equality-bound Lagrange multipliers two, For the corresponding Lagrange multipliers constrained by inequality; Establish the corresponding KKT conditions to obtain the optimality criterion: ； The calculation yields the following pattern: , and The optimal solution: ， In the formula , The normalized upper limit of transmission power for the mode is: ； This leads to the normalized peak-to-peak inductor current. for: 。 7. The improved dual active bridge converter bilateral variable duty cycle phase shift modulation method for fully soft switching according to claim 6, characterized in that, Similar to Mode 2, the simplified form of the constraints is as follows: 。 8. The improved dual active bridge converter bilateral variable duty cycle phase shift modulation method for fully soft switching according to claim 7, characterized in that, In Mode 2, KKT conditions for soft turn-on of eight switching transistors are defined. for, : ； The objective function is defined as follows: 。 9. The improved dual active bridge converter bilateral variable duty cycle phase shift modulation method for fully soft switching according to claim 8, characterized in that, In Mode 2, the method for solving the objective function is as follows: Define the Lagrange function based on the objective function. : ， activation and Two constraints, will and All Lagrange multipliers are set to 0; Solving the three power ranges under mode two , and The optimal solution: Set the upper limit of transmission power after normalization in the low power range to be [value]. : ， low power range , and The optimal solution is: ； This leads to the normalized peak-to-peak inductor current. for: ； The upper limit of the transmission power after normalization of the medium power range is set to be [value]. : ； medium power range , and The optimal solution is: ； This leads to the normalized peak-to-peak inductor current. for: ； The normalized upper limit of transmission power in the high-power range is 1. , and The optimal solution is: ； Obtain the normalized peak-to-peak inductor current for: ； when equal , Satisfy the following equation: ， set up The solution to the equation; if , , and The optimal solution is divided into three power intervals; if It only exists in the low power range and the high power range.

10. The improved dual active bridge converter bilateral variable duty cycle phase shift modulation method for fully soft switching according to claim 9, characterized in that, The final phase shift value of the bilateral variable duty cycle phase shift modulation is: 。

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