A metasurface antenna phase synthesis method for realizing same-frequency co-modulation

Abstract

The application discloses a metasurface antenna phase synthesis method for realizing same-frequency co-modulation, relates to the metasurface antenna modulation technical field, and provides a unified design framework, controls two wave patterns simultaneously through the single physical quantity of reflection phase, and avoids the complicated process of respectively designing surface impedance and reflection array in the surface impedance method; the reflection phase has a fixed and explicit value range, and the optimization parameters have clear physical meanings, so that independent control of the spatial wave and the surface wave in the same-frequency collaborative modulation is realized; the phase synthesis method is proposed by introducing a first weight coefficient and a second weight coefficient, and normalization constraints are applied to the two, so that the peak value of the superimposed synthesized phase is actively compressed to the middle safe area of the physical adjustable interval, and the method has significant advantages in realizing circularly polarized radiation and compact array design.

Description

Technical Field

[0001] This invention relates to the field of metasurface antenna modulation technology, and in particular to a metasurface antenna phase synthesis method for achieving co-modulation at the same frequency and a metasurface antenna with co-modulation at the same frequency. Background Technology

[0002] Metasurface antennas, with their low profile, low loss and flexible wavefront modulation capabilities, have become a key technology for millimeter-wave communication systems. In complex application scenarios such as radar detection and full-duplex communication, it is often necessary to independently control surface waves (SW) and free-space waves (FSW) within the same physical aperture.

[0003] Researchers have proposed an interleaved spatial arrangement scheme, which divides the array elements into two groups: one group is dedicated to controlling surface waves, and the other group is dedicated to controlling free-space waves. However, since the phase distribution of surface waves (concentric elliptical gradient) and the phase distribution of free-space waves (concentric circles or linear gradient) are fundamentally conflicting in mathematical form, when the phase distribution required for surface waves and the phase distribution required for free-space waves are directly added on independent elements, the resulting value will almost inevitably exceed the physical phase adjustment range that the reflecting elements can provide at the array edge or phase singularity because the two phases are superimposed at the same location.

[0004] Therefore, researchers perform folding mapping on phase values ​​that exceed the boundary by adding or subtracting integer multiples of 360° to fold them back into the physical phase adjustment range. However, when the phase distributions of two completely different spatial waves are forcibly superimposed and folded, the folding operation will interfere with the original phase gradient, causing a large distortion of the main radiation beam. Ultimately, it is difficult to stably achieve beamforming of the metasurface antenna with dual-wave co-frequency co-modulation. Summary of the Invention

[0005] This invention provides a method for phase synthesis of metasurface antennas to achieve co-modulation at the same frequency, which can solve the problems existing in the prior art.

[0006] This invention provides a method for phase synthesis of a metasurface antenna for achieving co-modulation at the same frequency. The metasurface antenna comprises multiple reflective elements arranged in an array, and includes the following steps: Obtain the surface wave phase control distribution and free space wave phase control distribution of the target metasurface antenna; The actual adjustable physical phase range that the internal reflection unit of the target metasurface antenna can achieve is obtained. The preset first weighting coefficient and second weighting coefficient are compressed and constrained to form the constrained first weighting coefficient and second weighting coefficient. Using the constrained first weighting coefficient and second weighting coefficient, the surface wave phase control distribution and the free space wave phase control distribution of the target metasurface antenna are superimposed on each reflection phase to form the phase synthesis distribution of the target metasurface antenna, so that each phase value in the phase synthesis distribution is located within the actual adjustable physical phase range. The phase values ​​in the phase synthesis distribution of the target metasurface antenna are mapped to the physical structural parameters of the corresponding reflective element to complete the phase configuration of the target metasurface antenna.

[0007] Preferably, constraining the preset first weighting coefficient and second weighting coefficient includes: The range of reflection phase changes that this type of reflective element structure in the target metasurface antenna can achieve within a preset operating frequency band is extracted by full-wave simulation or actual measurement, forming an actual adjustable physical phase interval applicable to all reflective elements in the target metasurface antenna; wherein, the actual adjustable physical phase interval is defined by the minimum achievable phase value and the maximum achievable phase value of this type of reflective element structure; Based on the span of the actual adjustable physical phase range, the relative ratio of the preset first weight coefficient and the second weight coefficient is adjusted to achieve normalization constraint on the first weight coefficient and the second weight coefficient, so that the sum of the constrained first weight coefficient and the second weight coefficient is 1.

[0008] Preferably, the surface wave phase control distribution is decomposed into a first polarization direction phase distribution and a second polarization direction phase distribution that are orthogonal to each other; The first weighting coefficient is decomposed into a third weighting coefficient and a fourth weighting coefficient, so that the third weighting coefficient is configured on the phase distribution of the first polarization direction and the fourth weighting coefficient is configured on the phase distribution of the second polarization direction; When the third weighting coefficient is equal to the fourth weighting coefficient, and the phase difference between the configured first polarization direction phase distribution and the second polarization direction phase distribution is 90°, the target metasurface antenna radiates a circularly polarized beam under surface wave excitation.

[0009] Preferably, after the first weight coefficient and the second weight coefficient are normalized and constrained, the sum of the first weight coefficient and the second weight coefficient after normalization and constraining is 1. After the first weight coefficient is decomposed into the third weight coefficient and the fourth weight coefficient and normalized, the sum of the normalized third weight coefficient, the fourth weight coefficient and the second weight coefficient is 1.

[0010] Preferably, the values ​​of the third weighting coefficient, the fourth weighting coefficient, and the second weighting coefficient after normalization constraint are proportionally allocated according to the span of the actual adjustable physical phase interval, so that all phase values ​​in the phase synthesis distribution of the target metasurface antenna are limited to the actual adjustable physical phase interval. When the actual adjustable physical phase interval spans 180°, the third, fourth, and second weighting coefficients after normalization constraints are all set to one-third, and all phase values ​​in the phase synthesis distribution of the target metasurface antenna are confined within the actual adjustable physical phase interval.

[0011] Preferably, when it is necessary to prioritize free space wave radiation, under constraints, the second weighting coefficient is configured to be greater than the values ​​of the third and fourth weighting coefficients; When it is necessary to prioritize surface wave polarization radiation, under constraints, the third and fourth weighting coefficients are configured to be greater than the second weighting coefficient. When it is necessary to simultaneously ensure free space wave radiation and surface wave polarization radiation, under constraints, the third weighting coefficient, the fourth weighting coefficient, and the second weighting coefficient are configured to be equal.

[0012] Preferably, mapping each phase value in the phase synthesis distribution of the target metasurface antenna to the physical structural parameters of the corresponding reflective element includes: Based on a preset quantization bit depth, the continuous phase values ​​in the phase synthesis distribution are discretized into a finite number of reflection phase state values; Based on the reflection phase state values, the physical structure parameters of each reflection unit are determined to adjust the equivalent reflection phase of each reflection unit.

[0013] This invention also provides a metasurface antenna fabricated using the phase synthesis method for achieving co-modulation of a metasurface antenna as described above, comprising a dielectric substrate, a metal ground layer disposed on the bottom surface of the dielectric substrate, and co-frequency reflection units arranged in an array disposed on the top surface of the dielectric substrate. The co-modulated metasurface antenna also includes a surface wave excitation port located on a dielectric substrate and a free space wave excitation port in the outer space region surrounding the central region.

[0014] Preferably, when fed through the surface wave excitation port, the energy diffuses radially along the dielectric substrate in the form of surface waves, and after being modulated by each of the same-frequency reflection units according to the phase values ​​in the phase synthesis distribution, it is radiated into free space; When fed through the free space wave excitation port, the energy is irradiated onto the surface of the co-modulated metasurface antenna in the form of a space plane wave, and after being modulated by each co-frequency reflection element, it is reflected to the preset direction. When both the surface wave excitation port and the free space wave excitation port are simultaneously fed, the two beams maintain their radiation directions in space independently and without interference.

[0015] Preferably, the same-frequency reflection unit is a rectangular patch structure, and the equivalent reflection phase is changed by adjusting the gap width between adjacent patches.

[0016] This invention provides a phase synthesis method for metasurface antennas to achieve co-modulation at the same frequency. Compared with the prior art, its advantages are as follows: This invention avoids the traditional method of adding or subtracting 360° integer multiples to the phase values ​​that exceed the boundary. Instead, under the control of the actual adjustable physical phase range, it introduces a first weighting coefficient after compression constraint to linearly superimpose the surface wave phase control distribution and the free space wave phase control distribution on a per-reflection-unit basis. This process, through active linear compression of the two phase contributions, ensures that the resulting composite phase response value falls naturally within the actual adjustable physical phase range without any periodic angular folding. This compression-only operation, without folding, eliminates distortion at the source, ensuring that the reflection phase distributions of surface waves and free space waves on the aperture do not overlap, thus guaranteeing efficient radiation of the two beams. Ultimately, it stably and independently achieves beamforming of the metasurface antenna with dual-wave co-frequency co-modulation.

[0017] Furthermore, the unified design framework of this invention controls two waveforms simultaneously through the single physical quantity of reflection phase, eliminating the cumbersome process of designing surface impedance and reflection array separately in the surface impedance method. Attached Figure Description

[0018] Figure 1 A schematic diagram of the phase distribution of the phase synthesis scheme provided in the embodiment of the present invention; wherein: (a) is the phase combination strategy; (b) is the phase distribution; Figure 2 This is a schematic diagram of phase distribution adjustment provided in an embodiment of the present invention; wherein: (a) is the phase distribution of the phase synthesis method when a=b=1 is set; (b) is the phase distribution of the phase synthesis method adjusted to the range of [-180°, 180°] by adding or subtracting 360° as needed; Figure 3 Gain as a function of parameters provided in embodiments of the present invention a and b A schematic diagram of the changing relationship; where: (a) represents the relationship when... a SFC SW ⊙ ϕ When excited by surface waves, the gain increases with a (b) is the relationship of change; b SFCFW ⊙ φ When driven by free-space waves, the gain increases with... b The relationship of change; (c) is when Φ SFC When excited by surface waves, the gain contributed by surface waves increases with... a and b The relationship of change; (d) is when Φ SFC When driven by free-space waves, the gain of free-space wave radiation increases with... a and b The changing relationship; Figure 4 This is a schematic diagram of the radiation modes of the SFC antenna provided in an embodiment of the present invention; wherein: (a) is the array mode calculated when the SFC antenna is fed with a 30 GHz surface wave; (b) is the array mode calculated when the SFC antenna is fed with a 30 GHz free space wave. Figure 5 This is a schematic diagram of the adjusted phase distribution provided in an embodiment of the present invention; wherein: (a) is in a 1= a 2= b (a) Phase synthesis distribution under the condition of =1; (b) Phase distribution after applying ±360° adjustment; Figure 6 This is a schematic diagram illustrating the staggered arrangement of circular polarizations and its gain calculation according to an embodiment of the present invention; wherein: (a) is the staggered arrangement of circular polarizations; (b) is the circular polarization phase synthesis method; (c) is... a 1= a 2= b =1 / 3 and using a 1-bit configuration, the calculated gain contributed by the free space wave; (d) is the gain contributed by the free space wave under the condition of 1 / 3 and 1-bit configuration; a 1= a 2= b The calculated gain contributed by surface waves under the condition that =1 / 3 and 1-bit configuration is used; Figure 7 The following is a schematic diagram of the components and simulation of the SFC antenna provided in the embodiments of the present invention; wherein: (a) is the geometric structure of the SFC antenna components; (b) is a schematic diagram of different... g Simulation results of reflection phase and amplitude under certain values; Figure 8 Provided for embodiments of the present invention ϕ The phase distribution and simulation diagram; where: (a) is ϕ x , ϕ y and φ (a) is the phase distribution; (b) is ϕ (c) shows the 1-bit phase distribution; (d) shows the simulation S-parameter results; ϕ 1— ϕ Simulated radiation mode of a 1-bit SFC antenna when θ = 90° and operating frequency is 30 GHz; Figure 9 This is a schematic diagram provided for an embodiment of the present invention; wherein: (a) is the simulated shaft ratio (AR) as a function of... ϕ 1 and ϕ (a) The relationship between the phase difference between 0 and 0; (b) The simulation AR follows ϕ The results of 0 taking different values; Figure 10 This is a schematic diagram provided for an embodiment of the present invention; wherein: (a) is φ The 1-phase distribution, in which ϕ = ϕ x + ϕ y (b) is φ Phase distribution; (c) is (d) shows the 1-bit phase distribution; (d) shows the simulation results of AR under different quantization conditions. Figure 11 The following is a schematic diagram provided for an embodiment of the present invention; wherein: (a) is a simulated radiation mode under different quantization conditions when the SFC antenna is fed by surface wave at a frequency of 30 GHz; (b) is a simulated radiation mode under different quantization conditions when the SFC antenna is fed by free space wave at a frequency of 30 GHz. Figure 12 The following diagram illustrates the antenna prototype and measurement process provided for embodiments of the present invention: (a) is a top view of the prototype antenna; (b) is a bottom view of the antenna; and (c) illustrates the measurement process. Figure 13 A schematic diagram of the simulated and measured S-parameters provided for embodiments of the present invention; Figure 14 Simulated and measured gains contributed by surface waves and free space waves, and a schematic diagram of AR value of surface waves provided for embodiments of the present invention. Figure 15 The following is a schematic diagram of the simulated and measured radiation modes at frequencies of 29, 30 and 31 GHz when the SFC antenna is fed by surface wave, provided for embodiments of the present invention. Figure 16 The diagram shows the simulated and measured radiation modes at frequencies of 29, 30, and 31 GHz when the SFC antenna is fed by free space waves, as provided in the embodiments of the present invention. Detailed Implementation

[0019] To make the above-mentioned objects, features, and advantages of the present invention more apparent and understandable, specific embodiments of the present invention will be described in detail below with reference to the accompanying drawings. Many specific details are set forth in the following description to provide a thorough understanding of the present invention. However, the present invention can be practiced in many other ways different from those described herein, and those skilled in the art can make similar modifications without departing from the spirit of the present invention. Therefore, the present invention is not limited to the specific embodiments disclosed below.

[0020] In recent years, metasurface antennas have gradually become an important communication technology. Despite their advantages in beam scanning, most reported structures are custom-designed in terms of design flexibility. The ability to control surface waves or free-space waves separately limits their versatility. When both surface waves and free-space waves occur simultaneously, manipulation of both is required. Existing methods often rely on complex feeding networks.

[0021] Researchers have proposed a phase-modulated metasurface antenna with an interleaved arrangement that can simultaneously control surface waves and free-space waves at the same frequency. This antenna design simplifies the surface wave design process and feeding structure, but its components can only modulate surface waves or free-space waves separately. Therefore, for small arrays (especially in circularly polarized scenarios), this method becomes significantly less efficient as the number of components decreases drastically. The interleaved arrangement requires a large number of components to operate effectively, resulting in a bulky antenna that is difficult to meet the requirements of compact or integrated antenna applications.

[0022] Based on this, this invention proposes a phase synthesis strategy for realizing a metasurface antenna with co-modulation at the same frequency. This antenna can simultaneously process surface waves and free-space waves within a shared aperture. Unlike using a single element to control a single type of wave, this invention integrates the phase distributions of both waves, allowing each element to contribute to both simultaneously. This not only improves aperture utilization but is also particularly suitable for compact arrays and circularly polarized radiation scenarios. A comprehensive analysis of radiation characteristics and polarization response confirms the inherent decoupling relationship between surface waves and free-space waves. Under phase quantization conditions, this invention establishes and optimizes a design method for generating circular polarization from surface waves.

[0023] Specifically: I. Working principle of surface waves and free-space waves co-modulated (SFC) antenna based on phase synthesis.

[0024] Surface impedance has been widely used in controlling surface waves; as a fundamental parameter of metasurfaces, surface impedance is closely related to reflection impedance; Equation (1) provides the equation from the input impedance Z in To surface impedance Z surfThe transformation relationship of Z: surf Z represents the surface impedance of a single metasurface unit, while Z in This is obtained through simulation of normal incident scattering under periodic boundary conditions; ϵ r For thickness is d The relative permittivity of the substrate material, k 0 represents the free space wavenumber. Z 0 represents the intrinsic impedance of free space.

[0025] (1) Due to the correlation between the mappings, the reflection phase can replace the surface impedance to effectively control surface waves, as described in equation (2).

[0026] (2) in: x t Represents a point on the XOY plane. r This is the distance between the element and the feed source. Parameter θ L Indicates the direction of the narrow pencil-shaped beam in the XOZ plane. n The average effective refractive index.

[0027] Furthermore, equation (2) also possesses the potential to control free-space waves. Equation (3) is a well-known expression for manipulating free-space waves, expressed as:

[0028] (3) in: φ ( x t () indicates the reflection phase of the element; The position vector of the element; Indicates phase; Indicates the desired beam direction; This indicates adjustable design parameters.

[0029] Equations (2) and (3) show that the reflected phase can simultaneously modulate surface waves and free space waves.

[0030] Compared to surface impedance-based models, a significant advantage of the reflection phase method is its ability to unify the control of surface waves and free-space waves. By establishing this joint control mechanism in a single metasurface antenna, analytical tools originally developed for free-space applications (such as holographic imaging, focusing, and beam steering) can now be extended to the control of surface waves. This breakthrough completely bridges the methodological gap that previously separated these two distinct metasurface antenna fields.

[0031] Figure 1 (a) illustrates the phase synthesis scheme of the present invention, where black rectangles represent units for controlling surface waves, green rectangles correspond to units for manipulating free-space waves, and orange rectangles realize linear phase synthesis, enabling simultaneous modulation of two wave modes; as Figure 1 As shown in (b), the resulting distribution contains two modes: one (concentric circles) controls free space radiation, and the other (concentric ellipses) dominates surface wave propagation; this arrangement establishes a co-modulation mechanism that allows each unit antenna to modulate both surface waves and free space waves simultaneously.

[0032] (4) (5) (6) a· SFC SW ⊙ ϕ = a 1 · SFC SW ⊙ ϕ x + a 2 · SFC SW ⊙ ϕ y (7) Formulas (4), (5), and (6) describe the construction method of the SFC antenna proposed in this invention, where ⊙ represents the Hadamard product. Formula (7) elucidates the basic principle of circularly polarized surface wave radiation.

[0033] Figure 2 (a) shows Φ SFC The phase distribution pattern, corresponding to a = b The case where =1. In phase synthesis methods, a and b This is a key control parameter that determines the phase range of the proposed SFC antenna. In Φ SFC middle, ϕ and φ The range of values ​​for is limited to the interval [-180°, 180°]. When a = b When =1, the combined phase Φ SFC The reflection phase may exceed 180° or fall below -180°, exceeding the achievable reflection phase range of the component; at this point, no cell can support the required reflection phase, making it difficult to realize a physical object. The phase can be adjusted to the range of [-180°, 180°] by adding or subtracting 360°, but this will cause pattern distortion (e.g., ...). Figure 2 (b) shown), so that ϕ and φ The contributions of these distortions are difficult to distinguish. In practical SFC antenna design, these distortions manifest as beam distortion, increased sidelobe levels, and reduced aperture efficiency, thus limiting overall radiation performance; therefore, it is necessary to... a and b Optimize to ensure Φ SFC The phase range falls within the interval [-180 ◦, 180 ◦].

[0034] Figure 3 It shows the gain as a function of parameters a and b The changing relationship; Figure 3 (a) shows the corresponding phase distribution a SFC SW ⊙ ϕ The gain curve, which is in a= It reaches its maximum value at time 1; when a When the value exceeds 1, the gain contributed by surface waves gradually decreases; free-space waves also exhibit a similar trend: such as Figure 3 As shown in (b), when b When the value is greater than 1, the gain decreases accordingly; Figure 3 (c) shows Φ SFC Surface wave excitation gain varies with parameters a and b The pattern of change: When a =1 and b When =0.1, the phase distribution is similar to that of SFC. SW ⊙ ϕ Approximately, due to the optimal phase distribution of the surface wave control unit, the gain reaches its maximum. With... b Increase and a With the gain fixed at 1, the gain gradually decreases because the increased free-space wave control phase contribution distorts the phase distribution and weakens the coherent interference of the surface wave control unit. When b When the gain exceeds 1, the gain of free-space waves decreases significantly, such as... Figure 3 As shown in (b), this leads to Figure 3 (c) a =1 and a Another high-gain region forms nearby, where surface wave contributions once again dominate. b The free-space excitation gain >1 is further demonstrated in Figure 3 A similar trend is observed in (d), with the same explanation. The phase distributions of surface waves and free-space waves compete to control the radiation aperture. Enhancing the phase distribution of one wave inevitably interferes with the other; therefore, an increase in surface wave gain comes at the cost of a decrease in free-space wave gain, and vice versa. This competition leads to a fundamental trade-off between the two waves in terms of achievable radiation performance.

[0035] In practical applications, phase quantization simplifies the design of control surface wave and free space wave elements and possesses the potential for reconfigurable operation. Equation (8) describes... Figure 3 The quantization method in (a), where a ≠0. Figure 3 (a) Demonstrates 1-bit quantization under surface wave excitation. a SFC SW ⊙ ϕ The array's gain. As expected, its gain is lower than that of a continuous-phase design, but relative to... a The overall trend remains consistent. This illustrates the inherent trade-off between gain and phase quantization resolution, where higher quantization levels can get closer to the ideal phase profile, thus resulting in higher gain.

[0036] The observed attenuation stems from phase quantization errors, which cause the signal to deviate from the expected continuous distribution. Similar phenomena have also been observed under free-space excitation conditions (e.g., Figure 3 (d) shows: using a 1-bit quantization array b SFC FW ⊙ φ At that time, its gain still decreases compared to the continuous phase case.

[0037] The radiation mode of the SFC antenna proposed in this invention is as follows: Figure 4 As shown, where is set a + b =1 and a = b =0.5. Figure 4 (a) Demonstrates the computational array radiation modes of the SFC metasurface antenna under surface wave excitation: ϕ The (full array) configuration achieves the highest gain, serving as a benchmark; when the phase distribution is reduced to... ϕ At / 2, the radiation gain decreases by approximately 1.68 dB, indicating a reduction in constructive interference effects between elements; Φ SFC The gain further decreased by about 3.5 dB under this configuration, indicating that the phase mismatch was more severe within the aperture range; ϕ In the / 2 configuration, no obvious highly directional beam is formed at (-30°, 0°), while significant sidelobes appear at 30°, which is consistent with Φ SFC The configurations behaved consistently; this result indicates that... ϕ / 2 modulation disrupts phase coherence, resulting in a high level of sidelobes when the SFC antenna modulates surface waves.

[0038] For the free-space excitation mode, the calculated array radiation pattern is as follows: Figure 4 As shown in (b). Among the various configurations studied, φ (Full array) achieved the highest gain; while φ The gain of the / 2 configuration is reduced by 1.95 dB; Φ SFC The gain of the configuration further decreased by approximately 2.75 dB. In contrast, when the SFC antenna uses... φ When the / 2 phase distribution is configured, its radiation pattern cannot form a clear directional beam. This fully demonstrates the significant isolation characteristics of surface wave excitation and free space excitation in beam control, allowing the two beams to be independently controlled without causing significant interference between them.

[0039] II. Configuration of Circularly Polarized SFC Antenna.

[0040] Equation (9) describes the proposed circularly polarized (CP) surface wave antenna array, where a 1. a 2 and b This is a constant used to adjust the phase range. ϕ x Indicates the vertical polarization direction of surface wave radiation; in ϕ y Under modulation, surface waves will generate horizontal polarization.

[0041] (9) In the circular polarization scheme a 1. a 2. b , ϕ 0 and ϕ 1 is a key parameter. For example... Figure 3 As shown, when the array is fed by surface waves, a 1 and a 2. The radiation gain of vertical polarization and horizontal polarization are controlled respectively; at the same time, they also determine the phase range of the SFC antenna proposed in this invention.

[0042] Figure 5 (a) showed The phase distribution pattern, in which a 1= a 2= b = 1 is set. ϕ x , ϕ y and φ The range is limited to [−180 ◦ , 180 ◦ ]. When a 1= a 2= b When = 1, combined phase Values ​​greater than 180 ◦ or less than −180 ◦ result in no element being able to support the corresponding reflection phase. By adding or subtracting 360 ◦, the phase can be mapped to the range [−180 ◦, 180 ◦]. However, in this case, it also leads to pattern distortion, such as… Figure 5 As shown in (b), the waveform becomes chaotic and difficult to distinguish. ϕ x , ϕ y and φ Each of their respective contributions, in the proposed SFC antenna implementation scheme, these waveform distortions lead to beam and polarization distortion; through optimization a 1. a 2 and b Parameters can ensure The phase range falls within the interval [-180°, 180°]. a 1= a Under condition 2, it can be adjusted a The 1 / b ratio is used to optimize the gain of surface waves and free-space waves. In practical applications, it is typically set... a 1+ a 2+ b =1 and a 1= a 2= b =1 / 3.

[0043] Under surface wave feeding conditions ϕ 0 and ϕ 1 describes the far-field phase characteristics of vertical and horizontal polarization respectively. For circular polarization, the required phase difference between the orthogonal components is ±90°, corresponding to right-handed and left-handed circular polarization respectively; while for linear polarization, the phase difference is ±180°, which ensures that the combined electric field vector propagates along a straight trajectory, thereby producing linearly polarized radiation of ±45°.

[0044] In staggered or column / row staggered arrangements, only half of the array elements (i.e., half the aperture) are dedicated to controlling free-space waves or surface waves. When the total array size is less than 16×16, the effective number of elements for each wave type is less than 8×8, which is insufficient to support complex beam manipulation. In contrast, the proposed phase synthesis method is particularly suitable for small arrays because each element can simultaneously modulate free-space waves and surface waves. This advantage is especially significant in circularly polarized applications.

[0045] Specifically, Figure 6(a) illustrates the staggered arrangement structure used to generate circularly polarized surface waves. In the 16×16 surface wave array employing this scheme, only a 4×4 subarray is configured to generate vertically polarized surface waves, while another 4×4 subarray is configured to generate horizontally polarized surface waves. The limited number of active elements severely restricts the ability to effectively modulate and radiate surface waves—a limitation fundamentally addressed in the proposed phase synthesis method (e.g., ...). Figure 6 (b) shows that each unit can simultaneously modulate free space waves and surface waves.

[0046] Figure 6 (c) shows the calculated array gain in a 1-bit configuration when the SFC antenna is excited by free-space waves. For array sizes ranging from 10×10 to 60×60, the gain contribution from free-space waves is almost identical in both methods.

[0047] Figure 6 (d) illustrates the array gain in the vertical polarization direction when the SFC antenna is excited by surface waves. In the staggered arrangement structure, the insufficient number of dedicated cells results in a gain significantly lower than that of the phase synthesis method proposed in this invention; while the phase synthesis method of this invention can achieve a higher overall gain for circular polarization synthesis that relies on orthogonal linear components.

[0048] The SFC antenna proposed in this invention is composed of components. The substrate uses Rogers RO5880, whose dielectric constant is... ε r and loss tangent tan δ The values ​​are 2.2 and 0.0009 respectively. This component is a rectangular patch antenna with a period of p = 2 mm (e.g., ...). Figure 7 As shown in (a), it is printed on a substrate with a thickness of h = 0.787 mm. The period is defined as p = a + g, where g represents the spacing between adjacent patches. The simulation results of the rectangular patch are as follows. Figure 7 As shown in (b), the reflection phases of −70° and 110° were used to construct a 1-bit SFC antenna. The corresponding reflection amplitudes are above −0.5 dB in the 28–32 GHz range.

[0049] Figure 8 (a) showed ϕ x , ϕ y The phase distribution of its superposition distribution, wherein the phase distribution is adopted. θ L = -30 ◦and ϕ 0- ϕ The parameter setting is 1=90◦. ϕ x and ϕy The main difference lies in the direction of the phase split line: ϕ x The splitting occurs along the y-direction, causing the phase to reverse on both sides of the x-axis; while ϕ y The splitting is aligned with the x-direction, causing the phase to reverse on both sides of the y-axis. These two orthogonal arrangements produce different field distributions and polarization characteristics. In the superposition distribution, the two patterns intertwine but remain distinguishable. In practical applications, the metasurface array is quantized to a 1-bit phase value: one is -70 ◦ (g=0.2), and the other is 110 ◦ (g=1.2). This 1-bit distribution is as follows: Figure 8 As shown in (b): when the phase value is greater than 20°, it is quantized to 110°; when the phase value is equal to or less than 20°, it is quantized to -70°. Simulation results are as follows: Figure 8 As shown in (c), the reflection coefficient remains below -10 dB throughout the 28–32 GHz band, reaching -13.4 dB at 30 GHz, indicating good impedance matching within the operating frequency band. The corresponding radiation modes are as follows: Figure 8 As shown in (d): Figure 8 The phase distribution in (b) produces left-hand circularly polarized radiation (LHCP). Figure 9 (a) showcased different ϕ 1 and ϕ The axial ratio (AR) corresponding to the phase difference between 0 and 0: When the phase difference is 70°, the AR remains above 3 dB throughout the 28-32 GHz band, indicating that there is no circular polarization phenomenon; when the phase difference is increased to 90°, the AR drops to a lower level.

[0050] Within the 28.8–32 GHz band, the gain is less than 3 dB, achieving circular polarization. As the phase difference further increases, the AR band shifts towards higher frequencies. Figure 9 (b) Demonstrates the performance of simulated AR as... ϕ A curve relationship where 0 changes. ϕ 0 and ϕ 1- ϕ All of these parameters will affect the AR performance of the surface wave antenna; a wider AR bandwidth can be obtained by optimizing these parameters.

[0051] Figure 10 (a) showed ϕ = ϕ x + ϕ y The 1-bit phase distribution generated at that time, this configuration can produce LHCP radiation; Figure 10 (b) presents ϕThe 1-bit phase distribution produces a lateral beam with linear polarization characteristics. The combined 1-bit phase distribution of both is as follows: Figure 10 As shown in (a) and 10(b).

[0052] like Figure 10 As shown in (c), the two patterns are interwoven. They are still clearly distinguishable. 1-bit The simulated AR value is as follows Figure 10 As shown in (d). For Q ( ϕ AR values ​​below 3dB cover the range of 28.8–31.8 GHz, while Q ( The coverage area is 28.4–31.8 GHz. The simulated radiation pattern is shown in the figure. Figure 11 (a) Under surface wave excitation Q ( The distribution pattern and its characteristics are Q ( ϕ The results are highly consistent, with almost identical gain values; furthermore, Q ( The sidelobe level is low at approximately 30°. This indicates that surface wave radiation can effectively resist the phase mismatch problem caused by the free-space wave unit in this configuration. Figure 11 (b) Demonstrates the simulated radiation pattern when a 1-bit SFC antenna is fed by free-space waves: the main beam direction remains unchanged, but the gain decreases by approximately 6 dB. This attenuation stems from the competition between free-space waves and surface waves in the radiation aperture. a 1= a 2= b When the free space radiation rate is 1 / 3, surface waves dominate, suppressing free space radiation; as... a 1 / b Ratio (where a 1= a 2) Decreasing the free-space wave amplitude leads to an increase in its gain, indicating that it occupies a larger share of the radiation aperture. However, this improvement comes at the cost of sacrificing surface wave radiation modes and standing wave ratio. Therefore, careful optimization is necessary. a 1 / b The ratio is used to balance the free-space wave gain and maintain acceptable surface wave performance.

[0053] Specific experiment: Figure 12 The image shows a photograph of the fabricated SFC antenna (including the support bracket). The antenna employs a 60×60 element array structure, with overall dimensions of 126 mm × 126 mm × 0.787 mm (12.6λ0×12.6λ0×0.08λ0) at a 30 GHz operating frequency. In the 1-bit encoding configuration, the parameters are set as follows:a 1= a 2= b =1 / 3; the geometric parameter g takes values ​​of 0.2 and 1.2 in the two encoding states, respectively. Port 1 excites surface waves (such as...) through a 50Ω metasurface waveguide probe welded to the bottom layer. Figure 12 (b) is shown), while port 2 is connected to a horn antenna mounted on a bracket to excite free-space waves. Photos of the measurement process are shown below. Figure 12 (c)

[0054] like Figure 13 As shown, the simulated impedance bandwidths of both port 1 and port 2 cover the range of 28–32 GHz; the measured impedance bandwidths of both remain below -10 dB within the same frequency range. The simulated isolation (>30 dB) is achieved in the 28–31 GHz range, while the measured isolation (>30 dB) covers the 28.2–31.9 GHz range. The slight difference between the two is mainly due to manufacturing tolerances.

[0055] Gain, axial ratio (AR), and radiation pattern can be obtained through far-field measurements. During surface wave measurements, port 1 of the antenna is connected to the test system, and port 2 is connected to a matched load. For example... Figure 14 As shown, simulation results indicate that the maximum gain of the surface wave at 31.8 GHz is 22.4 dBic, and the maximum gain of the free-space wave at 30.2 GHz is 22.9 dBi. Measured results show that the maximum gain of the surface wave at 30.4 GHz is 20.7 dBic, and the maximum gain of the free-space wave at 31.8 GHz is 22.5 dBi. The peak gain of the surface wave exhibits a slight upward frequency shift, which is consistent with the measured trend of the circular polarization axial ratio. The simulated circular polarization axial ratio bandwidth ranges are 28.4–31.6 GHz, while the measured values ​​are 28.8–32.2 GHz. The difference between the two is mainly due to differences in the SMA connector and the measurement device.

[0056] Figure 15 Simulated and measured radiation patterns of the surface wave mode at 29, 30, and 31 GHz are presented, with the main beam deflected to -30° in the xOy plane. Radiation patterns of free-space waves at the same frequencies are shown below. Figure 16 As shown, the simulation results are in high agreement with the measured values; minor differences are still attributed to the SMA connector and the experimental setup.

[0057] The embodiments described above are merely illustrative of several implementations of the present invention, and while the descriptions are relatively specific and detailed, they should not be construed as limiting the scope of the invention patent. It should be noted that those skilled in the art can make various modifications and improvements without departing from the concept of the present invention, and these all fall within the protection scope of the present invention. Therefore, the protection scope of this invention patent should be determined by the appended claims.

Claims

1. A method for phase synthesis of a metasurface antenna for achieving co-modulation at the same frequency, wherein the metasurface antenna comprises a plurality of reflective elements arranged in an array, characterized in that, Includes the following steps: Obtain the surface wave phase control distribution and free space wave phase control distribution of the target metasurface antenna; The actual adjustable physical phase range that the internal reflection unit of the target metasurface antenna can achieve is obtained. The preset first weighting coefficient and second weighting coefficient are compressed and constrained to form the constrained first weighting coefficient and second weighting coefficient. Using the constrained first weighting coefficient and second weighting coefficient, the surface wave phase control distribution and the free space wave phase control distribution of the target metasurface antenna are superimposed on each reflection phase to form the phase synthesis distribution of the target metasurface antenna, so that each phase value in the phase synthesis distribution is located within the actual adjustable physical phase range. The phase values ​​in the phase synthesis distribution of the target metasurface antenna are mapped to the physical structural parameters of the corresponding reflective element to complete the phase configuration of the target metasurface antenna.

2. The method for phase synthesis of a metasurface antenna for achieving co-modulation at the same frequency, as described in claim 1, is characterized in that... The compression constraint on the preset first weight coefficient and second weight coefficient includes: The range of reflection phase changes that this type of reflective element structure in the target metasurface antenna can achieve within a preset operating frequency band is extracted by full-wave simulation or actual measurement, forming an actual adjustable physical phase interval applicable to all reflective elements in the target metasurface antenna; wherein, the actual adjustable physical phase interval is defined by the minimum achievable phase value and the maximum achievable phase value of this type of reflective element structure; Based on the span of the actual adjustable physical phase range, the relative ratio of the preset first weight coefficient and the second weight coefficient is adjusted to achieve normalization constraint on the first weight coefficient and the second weight coefficient, so that the sum of the constrained first weight coefficient and the second weight coefficient is 1.

3. The method for phase synthesis of a metasurface antenna for achieving co-modulation at the same frequency, as described in claim 1, is characterized in that... The surface wave phase control distribution is decomposed into a first polarization direction phase distribution and a second polarization direction phase distribution that are orthogonal to each other; The first weighting coefficient is decomposed into a third weighting coefficient and a fourth weighting coefficient, so that the third weighting coefficient is configured on the phase distribution of the first polarization direction and the fourth weighting coefficient is configured on the phase distribution of the second polarization direction; When the third weighting coefficient is equal to the fourth weighting coefficient, and the phase difference between the configured first polarization direction phase distribution and the second polarization direction phase distribution is 90°, the target metasurface antenna radiates a circularly polarized beam under surface wave excitation.

4. The method for phase synthesis of a metasurface antenna for achieving co-modulation at the same frequency, as described in claim 3, is characterized in that... After the first and second weight coefficients are normalized and constrained, the sum of the first and second weight coefficients after normalization and constraint is 1. After the first weight coefficient is decomposed into the third weight coefficient and the fourth weight coefficient and normalized, the sum of the normalized third weight coefficient, the fourth weight coefficient and the second weight coefficient is 1.

5. The method for phase synthesis of a metasurface antenna for achieving co-modulation at the same frequency, as described in claim 4, is characterized in that... Based on the span of the actual adjustable physical phase interval, the values ​​of the third, fourth, and second weighting coefficients after normalization constraints are proportionally allocated so that all phase values ​​in the phase synthesis distribution of the target metasurface antenna are limited to the actual adjustable physical phase interval. When the actual adjustable physical phase interval spans 180°, the third, fourth, and second weighting coefficients after normalization constraints are all set to one-third, and all phase values ​​in the phase synthesis distribution of the target metasurface antenna are confined within the actual adjustable physical phase interval.

6. The method for phase synthesis of a metasurface antenna for achieving co-modulation at the same frequency, as described in claim 4, is characterized in that... When it is necessary to prioritize free space wave radiation, under constraints, the second weighting coefficient is configured to be greater than the values ​​of the third and fourth weighting coefficients. When it is necessary to prioritize surface wave polarization radiation, under constraints, the third and fourth weighting coefficients are configured to be greater than the second weighting coefficient. When it is necessary to simultaneously ensure free space wave radiation and surface wave polarization radiation, under constraints, the third weighting coefficient, the fourth weighting coefficient, and the second weighting coefficient are configured to be equal.

7. The method for phase synthesis of a metasurface antenna for achieving co-modulation at the same frequency, as described in claim 1, is characterized in that, The step of mapping each phase value in the phase synthesis distribution of the target metasurface antenna to the physical structural parameters of the corresponding reflective element includes: Based on a preset quantization bit depth, the continuous phase values ​​in the phase synthesis distribution are discretized into a finite number of reflection phase state values; Based on the reflection phase state values, the physical structure parameters of each reflection unit are determined to adjust the equivalent reflection phase of each reflection unit.

8. A metasurface antenna fabricated using the phase synthesis method for achieving co-modulation of a metasurface antenna as described in any one of claims 1 to 7, characterized in that, include: A dielectric substrate, a metal grounding layer disposed on the bottom surface of the dielectric substrate, and an array of co-frequency reflecting units disposed on the top surface of the dielectric substrate; The co-modulated metasurface antenna also includes a surface wave excitation port located on a dielectric substrate and a free space wave excitation port in the outer space region surrounding the central region.

9. A co-modulated metasurface antenna according to claim 8, characterized in that, When fed through the surface wave excitation port, the energy diffuses radially along the dielectric substrate in the form of surface waves. After being modulated by each same-frequency reflection unit according to the phase values ​​in the phase synthesis distribution, it is radiated into free space. When fed through the free space wave excitation port, the energy is irradiated onto the surface of the co-modulated metasurface antenna in the form of a space plane wave, and after being modulated by each co-frequency reflection element, it is reflected to the preset direction. When both the surface wave excitation port and the free space wave excitation port are simultaneously fed, the two beams maintain their radiation directions in space independently and without interference.

10. A co-modulated metasurface antenna according to claim 8, characterized in that, The same-frequency reflection unit is a rectangular patch structure, and the equivalent reflection phase is changed by adjusting the gap width between adjacent patches.

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