A hybrid three-level converter for inhibiting reactive circulating current and a control method and device thereof, and an electronic device

Abstract

The application discloses a hybrid three-level converter for inhibiting reactive circulating current and a control method and device thereof and electronic equipment, and belongs to the technical field of high-power high-voltage electric energy conversion. The method comprises the following steps: obtaining an actual input voltage of a primary side and an actual output voltage of a secondary side of the hybrid three-level converter; determining a total phase-shifting demand according to the input and output voltages, a transformer ratio and a reference output voltage; calculating a target inner phase-shifting angle duty cycle and a target outer phase-shifting angle duty cycle according to a zero-voltage turn-on condition; generating a primary side driving pulse signal and a secondary side driving pulse signal according to the target inner and outer phase-shifting angle duty cycles, and controlling the primary side three-level bridge arm and the secondary side full-bridge to act in coordination. Through implementation of the application, the problem that, in the prior art, when the hybrid three-level converter deals with wide-range input and output voltage fluctuation, the phase-shifting instruction is prone to over-limiting due to the lack of dynamic physical boundary constraint on the phase-shifting variable, thereby causing system reactive circulating current and increasing switching loss can be solved.

Description

Technical Field

[0001] This invention relates to the field of high-power high-voltage power conversion technology, specifically to a hybrid three-level converter for suppressing reactive circulating current and its control method, device, and electronic equipment. Background Technology

[0002] In the field of high-power high-voltage power conversion, hybrid three-level converters, which include a three-level bridge arm with a primary-side diode midpoint clamp and a secondary-side full-bridge, have been widely used due to their excellent device withstand voltage reduction capability and high-frequency isolation characteristics. In actual system operation, effectively suppressing reactive circulating currents within the converter's high-frequency links is crucial for reducing device conduction losses, minimizing heat generation, and improving overall energy conversion efficiency.

[0003] However, current hybrid three-level converter control strategies still face technical drawbacks when dealing with wide-range input and output voltage fluctuations, including difficulty in effectively suppressing reactive circulating currents and a tendency to lose zero-voltage turn-on characteristics. The main reason for this problem is that existing multi-degree-of-freedom phase-shift control methods fail to deeply integrate the coupling constraint mechanism between the converter's real-time terminal voltage ratio and the system's zero-voltage turn-on critical condition. In actual operation, because the reasonable operating range of the phase-shift variable is not dynamically derived based on the actual operating voltages at both AC and DC ends, the controller often lacks an upper limit limit on the external phase-shift angle itself, and also lacks a dynamic lower limit constraint on the correlation ratio between the external and internal phase-shift angles when decoupling and allocating the total phase-shift demand. This allocation mechanism, detached from the real-time voltage ratio and physical soft-switching boundaries, makes it easy for phase-shift commands to exceed limits under non-rated operating conditions, causing the primary and secondary bridge arm switching devices to deviate from the soft-switching feasible region, thereby generating reactive circulating currents that oscillate back and forth on both sides of the isolation transformer, increasing the overall system losses. Summary of the Invention

[0004] This invention provides a hybrid three-level converter for suppressing reactive circulating current, and its control method, device, and electronic equipment. It can solve the problem that in the prior art, when hybrid three-level converters deal with wide-range input and output voltage fluctuations, the lack of dynamic physical boundary constraints on the phase shift variable makes it easy for the phase shift command to exceed the limit, thereby causing reactive circulating current in the system and increasing switching losses.

[0005] An embodiment of the present invention provides a control method for a hybrid three-level converter to suppress reactive circulating current. The hybrid three-level converter includes a primary-side diode midpoint clamped three-level bridge arm, an isolation transformer, and a secondary-side full bridge. The control method includes: Obtain the actual primary input voltage and actual secondary output voltage of the hybrid three-level converter; Calculate the terminal voltage ratio based on the actual input voltage of the primary side, the actual output voltage of the secondary side, and the preset isolation transformer turns ratio; calculate the total phase shift requirement based on the actual output voltage of the secondary side and the preset reference output voltage. Based on the preset zero-voltage turn-on critical condition and the terminal voltage ratio, determine the upper limit threshold of the outer phase shift duty cycle and the lower limit of the outer phase shift duty cycle relative to the inner phase shift duty cycle; Based on the total phase shift requirement, upper threshold, and associated lower limit, calculate the target inner phase shift duty cycle and the target outer phase shift duty cycle; wherein, the target inner phase shift duty cycle is the phase shift variable that controls the operation of the internal switch of the three-level bridge arm clamped at the midpoint of the primary diode; the target outer phase shift duty cycle is the phase shift variable that controls the cross-bridge delay operation of the switch of the secondary full bridge relative to the switch of the three-level bridge arm clamped at the midpoint of the primary diode; Based on the duty cycle of the target's inner phase shift angle and the duty cycle of the target's outer phase shift angle, primary-side drive pulse signals and secondary-side drive pulse signals are generated; the primary-side drive pulse signals are used to control the action of the three-level bridge arm clamped at the midpoint of the primary-side diodes, and the secondary-side drive pulse signals are used to control the action of the secondary-side full bridge.

[0006] Furthermore, based on the actual input voltage on the primary side, the actual output voltage on the secondary side, and the preset isolation transformer turns ratio, the terminal voltage ratio is calculated, including: Based on the preset isolation transformer turns ratio, the actual input voltage on the primary side is voltage-reduced to generate an equivalent voltage-reduced voltage. The terminal voltage ratio is determined based on the equivalent converted voltage and the actual output voltage on the secondary side.

[0007] Furthermore, based on the actual secondary output voltage and the preset reference output voltage, the total phase shift requirement is calculated, including: The voltage deviation value is determined based on the preset reference output voltage and the actual output voltage on the secondary side. The voltage deviation value is input to a preset closed-loop voltage regulator for adjustment, generating an adjustment output result; Based on the adjustment output results, determine the total phase shift requirement.

[0008] Furthermore, based on the preset zero-voltage turn-on critical condition and the terminal voltage ratio, the upper limit threshold of the outer phase shift duty cycle and the lower limit of the outer phase shift duty cycle relative to the inner phase shift duty cycle are determined, including: Based on the terminal voltage ratio and the preset zero-voltage turn-on critical condition, a soft-switching boundary constraint model is constructed for the target operating mode. From the soft-switching boundary constraint model, the maximum outward phase shift duty cycle critical value that satisfies the preset power transmission direction and maintains zero voltage turn-on is extracted as the upper limit threshold of the outward phase shift duty cycle; Based on the soft-switching boundary constraint model, the minimum boundary condition is determined to ensure that the duty cycle of the outer phase shift angle dynamically changes with the duty cycle of the inner phase shift angle when the switching device is turned on at zero voltage. The minimum boundary condition is used as the lower limit of the correlation between the duty cycle of the outer phase shift angle and the duty cycle of the inner phase shift angle.

[0009] Furthermore, based on the total phase shift requirement, the upper threshold, and the associated lower limit, the target inner phase shift duty cycle and the target outer phase shift duty cycle are calculated, including: Based on the upper threshold and the associated lower limit, the boundary of the phase-shifting working range is constructed; The total phase shift demand is compared with the phase shift operating range boundary to determine the target phase shift operating range to which the converter currently belongs; Based on the preset mapping relationship between phase shifting working intervals and allocation rules, the target allocation rule for the corresponding target phase shifting working interval is obtained; Based on the target allocation rules, the total phase shift requirement is decoupled and calculated to generate the target inner phase shift duty cycle and the target outer phase shift duty cycle.

[0010] Furthermore, based on the target's inner phase shift duty cycle and the target's outer phase shift duty cycle, primary-side drive pulse signals and secondary-side drive pulse signals are generated, including: Extract a preset global switching periodic carrier as a unified phase reference; Based on the phase reference, the duty cycle of the target inner phase shift angle is converted into the inner phase shift time of the primary side bridge arm. The timing difference between the action of the inner and outer switches in the three-level bridge arm is configured according to the inner phase shift time of the primary side bridge arm, and the primary side drive pulse signal is generated. The target phase shift angle duty cycle is converted into the phase shift time between the primary and secondary bridges. Starting from the emission phase of the primary driving pulse signal, the delay action timing of the secondary full-bridge switch relative to the midpoint of the primary diode is configured according to the phase shift time between the primary and secondary bridges to generate the secondary driving pulse signal.

[0011] Furthermore, the primary-side drive pulse signal is used to control the operation of the three-level bridge arm clamped at the midpoint of the primary-side diode, and the secondary-side drive pulse signal is used to control the operation of the secondary-side full bridge, including: The primary-side drive pulse signal is applied to the control terminals of the outer and inner switches of the three-level bridge arm clamped at the midpoint of the primary-side diode, respectively, to synthesize a primary-side high-frequency AC voltage containing positive, zero, and negative level sequences. The secondary-side drive pulse signal is applied to the control terminal of each switch of the secondary-side full bridge, controlling the secondary-side full bridge to perform turn-on and turn-off actions according to the preset asymmetrical conduction sequence, and synthesizing the corresponding secondary-side high-frequency AC voltage. Active power transmission is driven by the voltage difference generated between the primary and secondary windings of a high-frequency transformer through the primary-side high-frequency AC voltage and the secondary-side high-frequency AC voltage.

[0012] Based on the above method embodiments, the present invention provides corresponding apparatus embodiments.

[0013] One embodiment of the present invention provides a hybrid three-level converter control device for suppressing reactive circulating current. The hybrid three-level converter includes a primary-side diode midpoint clamped three-level bridge arm, an isolation transformer, and a secondary-side full bridge. The control device includes: a data acquisition module, a parameter calculation module, a constraint boundary determination module, a duty cycle decoupling calculation module, and a drive control module; The data acquisition module is used to acquire the actual input voltage of the primary side and the actual output voltage of the secondary side of the hybrid three-level converter; The parameter calculation module is used to calculate the terminal voltage ratio based on the actual input voltage of the primary side, the actual output voltage of the secondary side, and the preset isolation transformer ratio; and to calculate the total phase shift requirement based on the actual output voltage of the secondary side and the preset reference output voltage. The constraint boundary determination module is used to determine the upper limit threshold of the outer phase shift duty cycle and the lower limit of the outer phase shift duty cycle relative to the inner phase shift duty cycle based on the preset zero-voltage turn-on critical condition and the terminal voltage ratio. The duty cycle decoupling calculation module is used to calculate the target inner phase shift angle duty cycle and the target outer phase shift angle duty cycle based on the total phase shift requirement, the upper threshold, and the associated lower limit. The target inner phase shift angle duty cycle is a phase shift variable that controls the operation of the internal switching transistor of the three-level bridge arm clamped at the midpoint of the primary diode. The target outer phase shift angle duty cycle is a phase shift variable that controls the cross-bridge delay operation of the switching transistor of the secondary full-bridge relative to the switching transistor of the three-level bridge arm clamped at the midpoint of the primary diode. The drive control module is used to generate primary-side drive pulse signals and secondary-side drive pulse signals according to the target inner phase shift duty cycle and the target outer phase shift duty cycle; the primary-side drive pulse signals are used to control the action of the three-level bridge arm clamped at the midpoint of the primary-side diodes, and the secondary-side drive pulse signals are used to control the action of the secondary-side full bridge.

[0014] Based on the above method embodiments, the present invention provides corresponding electronic device embodiments.

[0015] An embodiment of the present invention provides an electronic device, including a processor, a memory, and a computer program stored in the memory and configured to be executed by the processor. When the processor executes the computer program, it implements the hybrid three-level converter control method for suppressing reactive circulating current as described in any of the above-described method embodiments.

[0016] One embodiment of the present invention provides a hybrid three-level converter for suppressing reactive circulating current, comprising: a primary-side diode midpoint clamped three-level bridge arm, a high-frequency isolation transformer, and a secondary-side full bridge; The hybrid three-level converter has a primary-side DC input terminal. A first voltage-dividing capacitor and a second voltage-dividing capacitor are connected between the positive and negative terminals of the primary-side DC input terminal. The first end of the first voltage-dividing capacitor is connected to the positive terminal, and the second end of the second voltage-dividing capacitor is connected to the negative terminal. The second end of the first voltage-dividing capacitor and the first end of the second voltage-dividing capacitor are connected to form a first common connection point. The primary-side diode midpoint clamping three-level bridge arm includes the first to eighth switching transistors and the corresponding clamping diodes, wherein the first to fourth switching transistors are connected end to end in sequence to form a leading bridge arm, and the fifth to eighth switching transistors are connected end to end in sequence to form a lagging bridge arm. The first common connection point is not only connected to the clamping midpoint of the three-level bridge arm of the primary diode, but also directly extended to the center tap of the primary winding of the high-frequency isolation transformer. The hybrid three-level converter also includes an equivalent resonant inductor, the first end of which is connected to the midpoint of the three-level bridge arm clamped by the midpoint of the primary diode, and the second end of which is connected to the primary winding of the high-frequency isolation transformer. The secondary-side full bridge includes the ninth to twelfth switching transistors; the hybrid three-level converter also includes a DC blocking capacitor, the first end of which is connected to the secondary winding of the high-frequency isolation transformer, and the second end of which is connected to the input midpoint of the secondary-side full bridge. The hybrid three-level converter has a secondary output terminal. A first output filter capacitor and a second output filter capacitor are connected between the positive and negative terminals of the secondary output terminal. The first end of the first output filter capacitor is connected to the positive terminal of the secondary output terminal, and the second end of the second output filter capacitor is connected to the negative terminal of the secondary output terminal for connecting a load.

[0017] Compared with the prior art, the present invention has the following beneficial effects: This invention provides a hybrid three-level converter for suppressing reactive circulating current, and its control method, device, and electronic equipment. The method obtains the actual input voltage of the primary side and the actual output voltage of the secondary side of the hybrid three-level converter; calculates the terminal voltage ratio based on the actual input voltage of the primary side, the actual output voltage of the secondary side, and the turns ratio of the isolation transformer, and determines the total phase shift requirement in conjunction with the reference output voltage; determines the upper limit threshold of the outer phase shift duty cycle and its lower limit related to the inner phase shift duty cycle based on the zero-voltage turn-on critical condition and the terminal voltage ratio; calculates the target inner phase shift duty cycle and the target outer phase shift duty cycle based on the total phase shift requirement; and generates primary-side drive pulse signals and secondary-side drive pulse signals based on the target inner and outer phase shift duty cycles to control the coordinated operation of the primary three-level bridge arms and the secondary full bridge.

[0018] This invention dynamically determines the upper threshold of the outer phase shift duty cycle and the lower limit of its correlation with the inner phase shift duty cycle by real-time calculation of the terminal voltage ratio and a preset zero-voltage turn-on critical condition. This overcomes the deficiency of existing technologies in lacking dynamic physical boundary constraints under wide voltage conditions. Based on this, the present invention decouples the total phase shift requirement under the constraints of the aforementioned upper threshold and lower limit, ensuring that the final output target inner and outer phase shift duty cycles remain within the feasible soft-switching region. This effectively suppresses reactive power circulating current caused by phase shift command exceeding limits and reduces the hard switching losses of the converter. Attached Figure Description

[0019] Figure 1 This is a flowchart illustrating a hybrid three-level converter control method for suppressing reactive circulating current according to an embodiment of the present invention. Figure 2 This is a schematic diagram of the hardware topology of a hybrid three-level converter provided in an embodiment of the present invention; Figure 3 This is a schematic diagram of the driving pulse and waveform of a converter in the forward small phase shift interval according to an embodiment of the present invention; Figure 4 This is a schematic diagram of the driving pulse and waveform of a converter in the reverse small phase shift interval according to an embodiment of the present invention; Figure 5 This is a schematic diagram of the driving pulse and waveform of a converter in the forward large phase shift range according to an embodiment of the present invention; Figure 6 This is a schematic diagram of the driving pulse and waveform of a converter in the reverse large phase shift interval provided by an embodiment of the present invention; Figure 7 This is a schematic diagram of the operation of a hybrid three-level converter in mode 1 according to an embodiment of the present invention; Figure 8 This is a schematic diagram of the operation of a hybrid three-level converter in mode 2 according to an embodiment of the present invention; Figure 9 This is a schematic diagram of the operation of a hybrid three-level converter in mode 3 according to an embodiment of the present invention; Figure 10 This is a schematic diagram of the operation of a hybrid three-level converter in mode 4 according to an embodiment of the present invention; Figure 11 This is a schematic diagram of the operation of a hybrid three-level converter in mode 5 according to an embodiment of the present invention; Figure 12 This is a schematic diagram of the operation of a hybrid three-level converter in mode 6 according to an embodiment of the present invention; Figure 13 This is a schematic diagram of the operation of a hybrid three-level converter in mode 7 according to an embodiment of the present invention; Figure 14 This is a schematic diagram of the operation of a hybrid three-level converter in mode 8 according to an embodiment of the present invention; Figure 15 This is a schematic diagram of a hybrid three-level converter control device for suppressing reactive circulating current, provided in an embodiment of the present invention. Detailed Implementation

[0020] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0021] like Figure 1 As shown, to address the problem in existing hybrid three-level converters that, when dealing with wide-range input and output voltage fluctuations, suffer from a lack of dynamic physical boundary constraints on the phase shift variable, leading to easily exceeded phase shift limits and consequently reactive circulating current and increased switching losses, an embodiment of this invention provides a control method for suppressing reactive circulating current in a hybrid three-level converter. The hybrid three-level converter includes a primary-side diode-clamped three-level bridge arm, an isolation transformer, and a secondary-side full bridge. The control method includes at least: Step A1: Obtain the actual input voltage on the primary side and the actual output voltage on the secondary side of the hybrid three-level converter; Specifically, in the control process of a hybrid three-level converter for suppressing reactive circulating current, the first action is to comprehensively and accurately perceive the real-time electrical state of the hybrid three-level converter under the current operating conditions. Combined with... Figure 2The schematic diagram of the hardware topology of the hybrid three-level converter is presented. The primary side of the converter has a diode-clamped three-level bridge arm, and the secondary side has a full bridge. The diode-clamped three-level bridge arm on the primary side and the full bridge on the secondary side are connected via a high-frequency isolation transformer to achieve isolated energy exchange. See details. Figure 2 As shown, the primary side DC input of the hybrid three-level converter is connected in parallel with voltage dividing capacitors C1 and C2. The primary side diode-clamped three-level bridge arm consists of switches S1 to S8 and their corresponding clamping diodes. Switches S1 to S4 form the leading bridge arm, and switches S5 to S8 form the lagging bridge arm. The secondary side full bridge consists of switches S9 to S8. 12 The circuit is configured such that output filter capacitors C3 and C4 are connected in parallel at the secondary output terminal to connect to the load. R load An equivalent resonant inductor is connected in series between the primary winding of the high-frequency isolation transformer and the midpoint of the three-level bridge arm. Lr The turns ratio of the primary and secondary windings is configured as 1:1. N .

[0022] Further observation Figure 2 The circuit topology shown has a DC blocking capacitor connected in series between the secondary winding of the high-frequency isolation transformer and the input midpoint of the secondary full-bridge. C s Meanwhile, the voltage divider capacitor at the primary side DC input terminal... C 1 With voltage divider capacitor C 2 The common connection point between them is not only connected to the clamping midpoint of the three-level bridge arm clamped by the midpoint of the primary diode, but also directly extended to the center tap of the primary winding of the high-frequency isolation transformer to construct a complete physical clamping and commutation circuit.

[0023] To address the stringent physical constraints faced by high-power charging and storage emergency power vehicles when connected to medium- and high-voltage DC buses, traditional two-level dual active bridge converters require the switching devices on the high-voltage side to withstand the full DC bus voltage stress at the moment of turn-off, which easily leads to overvoltage breakdown risk. Furthermore, forcing high-voltage withstand devices into series significantly increases hardware costs and the complexity of the drive circuit. Compared to conventional three-level topologies, the upper and lower switches of the main bridge arm in a T-type three-level structure still withstand the full bus voltage during turn-off, failing to fundamentally solve the pain point of difficult high-voltage device selection. While the flying capacitor type three-level structure can broaden the soft-switching range, it introduces a bulky floating capacitor component, severely encroaching on the limited physical space on the vehicle and facing extremely complex pre-charging timing challenges during cold start. The hybrid three-level converter cleverly utilizes the physical conduction characteristics of passive clamping diodes by using diodes to clamp the three-level bridge arms at the midpoint of the primary side, strictly and stably limiting the steady-state voltage stress of all primary-side switching devices to half of the actual input voltage on the primary side. The aforementioned hardware architecture not only allows the use of high-efficiency semiconductor power devices with low withstand voltage and low on-resistance, thereby reducing manufacturing costs and suppressing the accumulation of switching heat from a physical perspective, but also eliminates the need for large floating capacitors. Combined with a compact full-bridge circuit on the secondary side, it maximizes the power density of the entire isolated bidirectional converter, laying a solid hardware foundation for the subsequent implementation of a two-degree-of-freedom phase-shift modulation strategy.

[0024] For bidirectional power conversion scenarios where the converter is connected to a high-voltage DC bus and a low-voltage energy storage load, a voltage detection circuit located on the primary side DC bus is used to collect the physical voltage values ​​at both ends of the DC bus in real time, and the collected voltage values ​​at both ends of the DC bus are defined as the actual input voltage on the primary side; simultaneously, a voltage detection circuit located on the secondary side load is used to collect the physical voltage values ​​at both ends of the load in real time, and the collected voltage values ​​at both ends of the load are defined as the actual output voltage on the secondary side.

[0025] To accurately map the actual voltage reduction mechanism of the primary-side diode midpoint clamped three-level bridge arm during physical operation, based on the previously obtained actual primary-side input voltage, the equivalent excitation amplitude experienced by the primary-side diode midpoint clamped three-level bridge arm during operation is further derived. The diode midpoint clamping structure, through the hardware topology circuit of the passive clamping diode, can strictly limit the steady-state voltage stress of all switching devices to half of the bus voltage. The mathematical derivation expression of the equivalent excitation amplitude is as follows: In the formula, This represents the equivalent excitation amplitude of the three-level bridge arm clamped at the midpoint of the primary-side diode. This represents the actual primary-side input voltage obtained in real time.

[0026] When facing high-power wide-voltage fluctuation conditions or complex operations such as black start of distribution networks using a parallel input and series output composite architecture, the above-mentioned physical acquisition and analysis steps accurately extract the real-time base voltage characteristics at both ends of the primary and secondary sides of the converter. This provides the lowest-level closed-loop feedforward data reference for the main control feedback link, fundamentally ensuring the absolute calculation accuracy of subsequent dynamic optimization of the physical boundary of zero-voltage turn-on and the target phase shift parameters. This effectively avoids command overruns and reactive power surges caused by fuzzy initial voltage boundaries.

[0027] Step A2: Calculate the terminal voltage ratio based on the actual input voltage of the primary side, the actual output voltage of the secondary side, and the preset isolation transformer turns ratio; calculate the total phase shift requirement based on the actual output voltage of the secondary side and the preset reference output voltage. In a preferred embodiment, the terminal voltage ratio is calculated based on the actual primary input voltage, the actual secondary output voltage, and the preset isolation transformer turns ratio, including: Based on the preset isolation transformer turns ratio, the actual input voltage on the primary side is voltage-reduced to generate an equivalent voltage-reduced voltage. The terminal voltage ratio is determined based on the equivalent converted voltage and the actual output voltage on the secondary side.

[0028] In a preferred embodiment, the total phase shift requirement is calculated based on the actual secondary-side output voltage and a preset reference output voltage, including: The voltage deviation value is determined based on the preset reference output voltage and the actual output voltage on the secondary side. The voltage deviation value is input to a preset closed-loop voltage regulator for adjustment, generating an adjustment output result; Based on the adjustment output results, determine the total phase shift requirement.

[0029] Specifically, combined Figure 1 The flowchart presents a control method for a hybrid three-level converter to suppress reactive current circulation. After physically acquiring the voltage signals at both ends, the control method enters the parameter conversion and demand calculation stage. To accurately assess the voltage matching degree between the primary and secondary sides of the high-frequency transformer, the control method performs a low-level voltage reduction operation. Specifically, based on the preset isolation transformer turns ratio, the actual input voltage on the primary side is reduced to generate an equivalent reduced voltage. The equivalent reduced voltage physically represents the equivalent excitation reference mapped from the primary side DC bus potential to the secondary winding of the high-frequency transformer. Subsequently, the terminal voltage ratio is determined based on the equivalent reduced voltage and the actual output voltage on the secondary side. The specific mathematical relationship is expressed as follows: In the formula, Indicates the equivalent voltage. This represents the actual input voltage on the primary side. This indicates the preset isolation transformer turns ratio. Indicates the terminal voltage ratio. This indicates the actual output voltage on the secondary side.

[0030] During the synchronous operation cycle of the calculated voltage ratio, to maintain voltage stability at the secondary load end under wide operating condition fluctuations, the control method constructs a voltage regulation feedback feedforward mechanism. First, based on the preset reference output voltage and the actual secondary output voltage, the voltage deviation value is determined through node subtraction. After obtaining the voltage deviation value, it is input to a preset closed-loop voltage regulator for adjustment. The preset closed-loop voltage regulator integrates a zero steady-state error control algorithm, which can dynamically compensate for and smooth continuously fluctuating voltage tracking errors, thereby generating the regulated output result. Finally, based on the regulated output result, the total phase shift requirement of the hybrid three-level converter in the high-frequency AC link is directly determined. The entire dynamic closed-loop calculation process satisfies the following mathematical logic: In the formula, Indicates the voltage deviation value. This indicates the preset reference output voltage. Indicates the total phase shift requirement. This represents the proportional adjustment coefficient. This represents the integral adjustment coefficient.

[0031] By accurately calculating the terminal voltage ratio and dynamically adjusting the total phase shift requirement through a closed loop, a reliable quantitative data benchmark is established for the subsequent decoupling allocation of internal and external phase shift variables, ensuring that the hybrid three-level converter can respond quickly and accurately to load disturbances and external input voltage fluctuations.

[0032] After obtaining the total phase shift requirement, if only a traditional single-dimensional phase shift control strategy is adopted, the converter is prone to deviating from the optimal voltage matching point when facing wide-ratio operating conditions where the voltage of the emergency power vehicle's energy storage battery terminal fluctuates significantly with the depth of discharge, as it only has the single degree of freedom to control the phase between the primary and secondary bridges. Under non-rated voltage matching conditions, single-dimensional phase shift control cannot actively suppress the surge of reactive power backflow in the high-frequency resonant cavity, resulting in a significant increase in transformer link current stress and causing some switching devices to leave the zero-voltage turn-on range, thus leading to severe hard switching losses. To overcome the physical limitations of single-phase shift control under wide voltage and light load conditions, the control method introduces an extended phase shift dual-degree-of-freedom collaborative optimization mechanism. The extended phase shift dual-degree-of-freedom collaborative optimization mechanism constructs an inner phase shift duty cycle by clamping the three-level bridge arm at the midpoint of the primary diode and deeply decouples the inner phase shift duty cycle from the outer phase shift duty cycle that controls the interaction of power between the primary and secondary bridges. The inner phase shift duty cycle directly determines the duration of the zero level in the primary-side high-frequency AC voltage waveform, while the outer phase shift duty cycle determines the fundamental phase difference between the primary-side and secondary-side high-frequency AC voltage waveforms. Through two-dimensional coordinated adjustment of these two phase shift variables, the control system can flexibly reconstruct the voltage pulse sequence combination on both sides of the high-frequency transformer while meeting the target active power transmission requirements. This enables it to actively reduce reactive circulating current and broaden the global zero-voltage switching operating range.

[0033] Step A3: Based on the preset zero-voltage turn-on critical condition and the terminal voltage ratio, determine the upper limit threshold of the outer phase shift duty cycle and the lower limit of the outer phase shift duty cycle relative to the inner phase shift duty cycle. In a preferred embodiment, based on a preset zero-voltage turn-on critical condition and the terminal voltage ratio, an upper limit threshold for the outer phase shift duty cycle and a lower limit relating the outer phase shift duty cycle to the inner phase shift duty cycle are determined, including: Based on the terminal voltage ratio and the preset zero-voltage turn-on critical condition, a soft-switching boundary constraint model is constructed for the target operating mode. From the soft-switching boundary constraint model, the maximum outward phase shift duty cycle critical value that satisfies the preset power transmission direction and maintains zero voltage turn-on is extracted as the upper limit threshold of the outward phase shift duty cycle; Based on the soft-switching boundary constraint model, the minimum boundary condition is determined to ensure that the duty cycle of the outer phase shift angle dynamically changes with the duty cycle of the inner phase shift angle when the switching device is turned on at zero voltage. The minimum boundary condition is used as the lower limit of the correlation between the duty cycle of the outer phase shift angle and the duty cycle of the inner phase shift angle.

[0034] Specifically, after completing the initial closed-loop calculation of the total phase shift requirement, the control method enters the physical optimization phase of the zero-voltage turn-on operation boundary. Combined with... Figures 3 to 6The provided diagrams illustrate the drive pulses and waveforms of the converter in different operating ranges. To prevent hard switching of the switching devices in the primary-side diode midpoint clamping three-level bridge arm and the secondary-side full-bridge during commutation, a soft-switching boundary constraint model for the target operating mode needs to be constructed based on the terminal voltage ratio and a preset zero-voltage turn-on critical condition. The preset zero-voltage turn-on critical condition strictly requires, at the physical level, that the equivalent leakage inductance current at each switching transistor conduction node must be less than or equal to zero to ensure that the anti-parallel diodes of each switching transistor can preferentially conduct freewheeling current.

[0035] To address diverse active power flow demands, the target operating mode is divided into a forward small-phase-shift interval and a forward large-phase-shift interval. Combined with... Figures 7 to 14 The schematic diagrams of the physical operating modes of the hybrid three-level converter at different commutation stages are shown. By simultaneously solving the inductor current state equations of each commutation node within a single cycle, the mathematical algebraic relationships for maintaining the zero-voltage turn-on state of all switches can be derived. When the converter operates in the forward small-phase-shift interval, the constructed soft-switching boundary constraint model includes the following set of algebraic inequalities: In the formula, Indicates the duty cycle of the outward phase shift. This indicates the duty cycle of the inward phase shift angle.

[0036] When the converter moves into the positive large phase shift region to perform operations, the corresponding soft-switching boundary constraint model switches to the following set of algebraic inequalities: Based on the constructed soft-switching boundary constraint model, the control method further extracts the critical value of the maximum outward phase shift duty cycle that satisfies the preset power transmission direction and maintains zero-voltage operation, and uses this critical value as the upper limit threshold of the outward phase shift duty cycle. Specifically, under the model constraint of the forward small phase shift interval, in order to ensure that the secondary full bridge is always in a zero-voltage operating state, there is an absolute physical upper limit to the outward phase shift duty cycle. The control logic directly locks the real-time value of the terminal voltage ratio as the upper limit threshold of the outward phase shift duty cycle.

[0037] Subsequently, based on the soft-switching boundary constraint model, the minimum boundary condition is determined to ensure that the duty cycle of the outer phase shift angle dynamically changes with the duty cycle of the inner phase shift angle when the switching device is turned on at zero voltage. This minimum boundary condition is then used as the lower limit of the correlation between the duty cycle of the outer phase shift angle and the duty cycle of the inner phase shift angle. In the underlying control algorithm, to maintain the soft-switching characteristics of the inner and outer switches of the three-level bridge arm clamped at the midpoint of the primary diode, the duty cycle of the outer phase shift angle cannot be reduced indefinitely. The duty cycle of the outer phase shift angle must always be greater than or equal to a product term determined by the ratio of the duty cycle of the inner phase shift angle to the terminal voltage. The result of this product operation directly establishes the lower limit of the correlation for dynamic changes.

[0038] By establishing a dual operational boundary constraint mechanism that includes a fixed upper limit threshold and a dynamically correlated lower limit, the solution space for invalid phase shift commands that deviate from the physical soft switching boundary is completely eliminated. This eliminates the risk of exceeding limits for outputting absolutely safe decoupled duty cycle data in subsequent stages, suppresses the backflow oscillation of reactive power on both sides of the high-frequency transformer from the source of the control algorithm, and significantly reduces switching losses across the entire domain.

[0039] Step A4: Calculate the target inner phase shift duty cycle and the target outer phase shift duty cycle based on the total phase shift requirement, upper threshold, and associated lower limit; wherein, the target inner phase shift duty cycle is the phase shift variable that controls the operation of the internal switch of the three-level bridge arm clamped at the midpoint of the primary diode; the target outer phase shift duty cycle is the phase shift variable that controls the cross-bridge delay operation of the switch of the secondary full bridge relative to the switch of the three-level bridge arm clamped at the midpoint of the primary diode; In a preferred embodiment, the target inner phase shift duty cycle and the target outer phase shift duty cycle are calculated based on the total phase shift requirement, the upper threshold, and the associated lower limit, including: Based on the upper threshold and the associated lower limit, the boundary of the phase-shifting working range is constructed; The total phase shift demand is compared with the phase shift operating range boundary to determine the target phase shift operating range to which the converter currently belongs; Based on the preset mapping relationship between phase shifting working intervals and allocation rules, the target allocation rule for the corresponding target phase shifting working interval is obtained; Based on the target allocation rules, the total phase shift requirement is decoupled and calculated to generate the target inner phase shift duty cycle and the target outer phase shift duty cycle.

[0040] Specifically, after establishing the absolutely safe physical operating boundary of soft switching, the control logic enters the pulse allocation and variable calculation stage. Based on the defined upper threshold of the outer phase shift duty cycle and its corresponding lower limit relative to the inner phase shift duty cycle, the control logic performs numerical calculations on the target inner and outer phase shift duty cycles. The target inner phase shift duty cycle serves as the absolute phase shift variable in the underlying architecture, controlling the action sequence of the switching transistors inside the three-level bridge arm clamped at the midpoint of the primary diode. Simultaneously, the target outer phase shift duty cycle is designated as the core phase shift variable controlling the cross-bridge interaction between the three-level bridge arm clamped at the midpoint of the primary diode and the secondary full bridge.

[0041] To achieve accurate analysis of the two-degree-of-freedom variables, the control logic constructs a continuous and closed phase-shifting operating interval boundary in a two-dimensional coordinate domain based on the acquired upper threshold and associated lower limit. Physically, this boundary divides the full-power operating domain of the hybrid three-level converter into multiple discrete steady-state mode spaces, such as a forward small phase-shift interval and a forward large phase-shift interval. Subsequently, the control logic quantitatively compares the total phase-shift demand calculated by the feedforward with the constructed phase-shifting operating interval boundary, and uses a strict threshold determination to lock the current target phase-shifting operating interval of the converter.

[0042] The full-power operating domain of the hybrid three-level converter under extended phase-shift control is finely divided into four core steady-state operating modes. Specifically, when the converter executes a unidirectional forward power transmission command, based on the different algebraic constraints of the inner and outer phase-shift duty cycles, the target phase-shift operating range includes a forward small phase-shift range and a forward large phase-shift range. Within the forward small phase-shift range, the inner and outer phase-shift duty cycles satisfy the constraint that the inner phase-shift duty cycle is greater than or equal to zero and less than or equal to half the difference between a constant and the outer phase-shift duty cycle. Within the forward large phase-shift range, the inner and outer phase-shift duty cycles satisfy the constraint that the inner phase-shift duty cycle is greater than half the difference between a constant and the outer phase-shift duty cycle and less than or equal to a constant of 0.5. When the converter executes a power feedback command, the target phase-shift operating range correspondingly switches to a reverse small phase-shift range and a reverse large phase-shift range. Within the reverse small phase shift interval, the duty cycle of the inner phase shift angle and the duty cycle of the outer phase shift angle satisfy the constraint that the duty cycle of the inner phase shift angle is greater than or equal to half the difference between the duty cycle of the outer phase shift angle and the constant, and is less than or equal to zero. Within the reverse large phase shift interval, the duty cycle of the inner phase shift angle and the duty cycle of the outer phase shift angle satisfy the constraint that the duty cycle of the inner phase shift angle is greater than or equal to -0.5 and is less than half the difference between the duty cycle of the outer phase shift angle and the constant. Under the above four distinct target phase shift operating intervals, the phase combination sequence of the voltages on both sides of the primary and secondary sides of the high-frequency transformer is fundamentally different, directly causing changes in the transient volt-second characteristics at both ends of the equivalent resonant inductor, which in turn determines the piecewise linear evolution trend of the inductor current trajectory and the extreme boundary of the transmitted active power.

[0043] After locking onto a specific target phase-shifting operating interval, the control logic retrieves a preset mapping table between phase-shifting operating intervals and allocation rules, extracting the target allocation rule that perfectly corresponds to the current target phase-shifting operating interval. The target allocation rule deeply integrates an optimization strategy for minimizing the effective value of the inductor current in high-frequency AC links and a closed-form equation for the per-unit transmission power of each operating interval. Specifically, when the comparison determines that the target phase-shifting operating interval belongs to a positive small phase-shifting interval, the transmission power equality constraint condition built into the target allocation rule is satisfied: In the formula, This represents the per-unit total power transmission demand within the first target interval.

[0044] When the target phase-shifting working interval transitions to a positive large phase-shifting interval, the transmission power equality constraint condition built into the target allocation rule changes, specifically satisfying: In the formula, This represents the per-unit total power transmission demand within the second target interval.

[0045] Based on the acquired target allocation rules, the control logic employs a constraint optimization algorithm to perform deep decoupling calculations on the total power transmission demand variable representing the total phase shift requirement. It identifies and locks the duty cycle coordinate combination that satisfies the transmission equality constraint and minimizes the return power in the corresponding operating mode. In steady state, it generates absolutely unique target inner and outer phase shift duty cycles. By deploying nonlinear interval optimization mapping and duty cycle decoupling calculations, the single-dimensional total phase shift requirement is successfully reconstructed into a refined two-degree-of-freedom control command. This not only completely eliminates the potential for transient abrupt changes at the boundaries of each phase shift interval but also reduces the reactive power return of the converter to its physical limit under full load conditions, significantly improving the overall energy conversion efficiency of the unit under wide voltage ratio conditions.

[0046] Step A5: Based on the target inner phase shift duty cycle and the target outer phase shift duty cycle, generate primary-side drive pulse signal and secondary-side drive pulse signal; use the primary-side drive pulse signal to control the action of the three-level bridge arm clamped at the midpoint of the primary-side diode, and use the secondary-side drive pulse signal to control the action of the secondary-side full bridge.

[0047] In a preferred embodiment, primary-side drive pulse signals and secondary-side drive pulse signals are generated based on the target inner phase shift duty cycle and the target outer phase shift duty cycle, including: Extract a preset global switching periodic carrier as a unified phase reference; Based on the phase reference, the duty cycle of the target inner phase shift angle is converted into the inner phase shift time of the primary side bridge arm. The timing difference between the action of the inner and outer switches in the three-level bridge arm is configured according to the inner phase shift time of the primary side bridge arm, and the primary side drive pulse signal is generated. The target phase shift angle duty cycle is converted into the phase shift time between the primary and secondary bridges. Starting from the emission phase of the primary driving pulse signal, the delay action timing of the secondary full-bridge switch relative to the midpoint of the primary diode is configured according to the phase shift time between the primary and secondary bridges to generate the secondary driving pulse signal.

[0048] In a preferred embodiment, the operation of the three-level bridge arm clamped at the midpoint of the primary-side diode is controlled by a primary-side drive pulse signal, and the operation of the secondary-side full bridge is controlled by a secondary-side drive pulse signal, including: The primary-side drive pulse signal is applied to the control terminals of the outer and inner switches of the three-level bridge arm clamped at the midpoint of the primary-side diode, respectively, to synthesize a primary-side high-frequency AC voltage containing positive, zero, and negative level sequences. The secondary-side drive pulse signal is applied to the control terminal of each switch of the secondary-side full bridge, controlling the secondary-side full bridge to perform turn-on and turn-off actions according to the preset asymmetrical conduction sequence, and synthesizing the corresponding secondary-side high-frequency AC voltage. Active power transmission is driven by the voltage difference generated between the primary and secondary windings of the high-frequency AC voltage and the secondary high-frequency AC voltage. In the active power transmission process, the leakage inductance energy of the transformer is used to enable the switching transistors of the three-level bridge arm clamped at the midpoint of the primary diode and the secondary full bridge to achieve zero-voltage turn-on within the range defined by the phase shift working interval, and to suppress the reactive power circulating current between the primary and secondary sides of the high-frequency transformer.

[0049] Specifically, after completing the deep constraint decoupling of the target phase-shifting variable, the execution phase of the control logic enters the stage of pulse generation and physical drive of the underlying hardware. Combined with... Figures 3 to 6 The diagram shows the driving pulses and waveforms of the converter in different phase shift intervals. In order to accurately map the purely mathematical duty cycle to the on and off actions of the physical switching devices, the underlying driving module extracts a preset global switching period carrier as a unified phase reference.

[0050] like Figures 3 to 6 As shown in detail, the horizontal axis uniformly represents a complete high-frequency switching cycle. T s The time span is represented by the vertical axis, which depicts the operating status of each control signal and electrical quantity. The rectangular color blocks at the top of the figure are used to visually identify the drive waveform and conduction intervals of each switch. Different background fill colors are used to strictly distinguish groups of switches that operate alternately. The light blue rectangular color blocks at the top and left represent switches S1, S8, S2, and S7, as well as secondary switches S9 and S1. 12 The light red rectangular blocks, complementary to the active state, represent the switching transistors S4, S5, S3, and S6, as well as the secondary switching transistor S. 10 S 11 In the triggered conduction state; the middle part V h1 and V h2 In the waveform trace, the cyan square wave block corresponds to the duration of the positive and negative high-frequency AC levels output by the primary side bridge arm, and the yellow square wave block corresponds to the duration of the positive and negative high-frequency AC levels output by the secondary side full bridge; the bottom... i L The line graph accurately depicts the continuous physical changes of the equivalent resonant inductor current under multi-step voltage excitation, including forward magnetization, zero-crossing commutation, and reverse freewheeling at different time points.

[0051] Further analysis Figures 3 to 6 The time axis is marked with discrete time nodes on the horizontal axis. t 0 To the time node t 6It completely segments all transient commutation action moments within a single high-frequency switching cycle. Specifically, the time nodes... t 0 Characterized by the starting moment of the current cycle; time node t 1 Time nodes t 2 and time nodes t 3 The time nodes sequentially characterize the multiple physical commutation nodes within the three-level bridge arm clamped at the midpoint of the primary-side diode, caused by the timing difference between the inner and outer switching transistors; t 2m Strictly corresponding to the symmetrical center point of half a high-frequency switching cycle; time node t 4 and time nodes t 5 The commutation nodes of the bridge sequentially represent the actions of the secondary-side full-bridge switches; time nodes t 6 This represents the end time of the current cycle. The aforementioned discrete time nodes strictly divide the continuous inductor current trajectory into multiple linearly evolving physical mode intervals.

[0052] The global switching periodic carrier provides an absolute time zero point for all phase-shifting operations. Based on the unified phase reference provided by the global switching periodic carrier, the underlying drive module first converts the calculated target inner phase-shifting duty cycle into the phase-shifting time of the primary-side bridge arm proportionally. Subsequently, according to the phase-shifting time of the primary-side bridge arm, the timing difference between the inner and outer switches in the three-level bridge arm clamped at the midpoint of the primary-side diode is precisely configured. The timing difference directly determines the physical duration of the zero-level output of the primary-side bridge arm, thereby generating a primary-side drive pulse signal adapted to the target inner phase-shifting duty cycle.

[0053] Synchronously, the underlying driver module converts the target outward phase shift duty cycle into the primary and secondary bridge phase shift time using the same time conversion ratio. To ensure strict phase alignment of the primary and secondary energy interaction, the underlying driver module strictly uses the emission phase of the primary-side drive pulse signal as the starting point for time calculation, and configures the delayed action timing of the secondary-side full-bridge switch relative to the midpoint of the primary-side diode clamping three-level bridge arm according to the primary and secondary bridge phase shift time. By applying the delayed action timing, the underlying driver module generates a secondary-side drive pulse signal carrying inter-bridge phase difference information.

[0054] After generating the drive pulse signal, the underlying drive module applies the primary-side drive pulse signal to the control terminals of the outer and inner switches of the three-level bridge arm clamped at the midpoint of the primary-side diodes. According to the logic high and low levels of the primary-side drive pulse signal, the outer and inner switches perform alternating current switching, synthesizing a primary-side high-frequency AC voltage containing positive, zero, and negative level sequences. Simultaneously, the underlying drive module applies the secondary-side drive pulse signal to the control terminals of each switch in the secondary-side full bridge, controlling the secondary-side full bridge to perform turn-on and turn-off actions according to a preset asymmetric turn-on sequence. This preset asymmetric turn-on sequence effectively balances the thermomechanical stress of the secondary-side semiconductor devices and synthesizes the corresponding secondary-side high-frequency AC voltage.

[0055] Specifically, the preset asymmetric turn-on sequence is an asymmetric cyclic turn-on sequence of the secondary-side full-bridge switches, which is as follows: switches S9 and S11 are turned on, switches S10 and S11 are turned on, switches S10 and S12 are turned on, and switches S10 and S11 are turned on again. Assuming the physical switching period of the switches is Ts, this sequence divides the period into four equal-length intervals, each interval occupying 0.25Ts. This asymmetric central distribution effectively balances the thermomechanical stress of the semiconductor devices and improves the power density of the transformer's magnetic components.

[0056] Combination Figures 7 to 14 The diagram shows the physical operating modes of the hybrid three-level converter in different commutation stages within a single cycle. The high-frequency AC voltage on the primary side interacts with the high-frequency AC voltage on the secondary side, generating a high-frequency alternating voltage difference between the primary and secondary windings of the high-frequency transformer.

[0057] like Figures 7 to 14 The detailed breakdown shows that each sub-diagram reveals the microscopic commutation process within a discrete-time slice of a single high-frequency switching cycle. The red dashed line and its accompanying arrow in the primary side region clearly indicate the actual physical excitation and return path of the inductor current flowing through the primary voltage divider capacitor, the main switch in the corresponding conduction state, the anti-parallel diode or clamping diode, and the primary winding of the high-frequency transformer under the corresponding mode. The blue dashed line and its accompanying arrow in the secondary side region synchronously map the freewheeling energy release path of the induced current in the secondary side of the high-frequency transformer flowing through the full-bridge switch, rectified feedback to the output filter capacitor, or directly to the load. Figures 7 to 14 The periodic switching and reconstruction of the red and blue dashed line trajectories fully demonstrate the objective physical facts of bidirectional energy shuttle transfer inside the converter and the use of leakage inductance energy to complete the charging and discharging of the junction capacitance.

[0058] In order to completely exhaust the physical commutation behavior within a single cycle Figures 7 to 14 The microscopic current connectivity states under each discrete mode were decomposed sequentially according to the time evolution. The specific analysis is as follows: Figure 7In mode 1 shown, the primary side inner switches S3 and S6 are turned on, and the inductor current flows through the clamping diode to form a zero-level freewheeling circuit. The secondary side switch S... 10 With the switching transistor S 11 It is activated and subjected to reverse clamping; Figure 8 In mode 2 shown, the switch on the inner side of the primary side is turned off, and the current is transferred to the anti-parallel diodes of switches S1, S2, S7 and S8 for freewheeling, and the inductor current approaches zero. Figure 9 In the mode 3 shown, the primary-side main switches S1, S2, S7 and S8 are fully turned on, the inductor current completes the zero-crossing reversal and begins to rise in the positive direction. Figure 10 In mode 4 shown, a phasor flip occurs on the secondary side of the full bridge, and the conduction state is controlled by the switch S. 10 With the switching transistor S 11 Switching to switching transistor S9 and switching transistor S 12 This drives the positive injection of active power into the load; Figure 11 In mode 5 shown, primary-side switches S1 and S8 are turned off, and the current is transferred to the anti-parallel diode of switches S4 and S5. The inductor current reaches the positive peak and then begins to fall back. Figure 12 In the mode 6 shown, the primary side is in a negative full-level excitation state, the inductor current continues to rise negatively and extracts leakage inductance energy in the reverse direction; Figure 13 In the mode 7 shown, the primary-side main switch S3, switch S4, switch S5 and switch S6 achieve zero-voltage turn-on, and the inductor current crosses zero and is negatively excited. Figure 14 In mode 8 shown, the secondary-side full bridge flips again, and the switching transistor S... 10 With the switching transistor S 11 Reconnection completes the energy reset for half an electrical cycle and smoothly transitions to the next full high-frequency switching cycle.

[0059] The voltage difference is directly applied across the equivalent leakage inductance of the isolation transformer, driving bidirectional transmission of active power between the primary energy storage side and the secondary load side. The physical process of active power transmission satisfies the following analytical mathematical model: In the formula, This represents the average active power output of the hybrid three-level converter. Indicates the terminal voltage ratio. Indicates the duty cycle of the target's inward phase angle. Indicates the duty cycle of the target outward shift phase angle. This represents the actual input voltage on the primary side. This represents the equivalent leakage inductance of the high-frequency transformer after conversion. This represents the physical switching frequency of the global switching periodic carrier.

[0060] During the physical commutation mode evolution of active power transmission, based on the aforementioned boundary definition of the phase-shifted operating range, the inductor current maintains the correct freewheeling direction at the turn-on nodes of all main switches. Relying on the leakage inductance energy of the transformer, the charge inside the parasitic capacitance of the switching transistors is preferentially extracted and the anti-parallel diodes are turned on, ensuring that the voltage across each switch in the three-level bridge arm clamped at the midpoint of the primary diode and the secondary full-bridge naturally drops to zero volts before the turn-on signal is triggered. A detailed analysis of the zero-voltage turn-on physical commutation evolution process reveals that within the dead time segment when any main switch receives the turn-off drive signal, the equivalent leakage inductance current flowing through the high-frequency isolation transformer cannot undergo transient changes. Due to the physical constraint of the continuity of the inductor current, the equivalent leakage inductance current, maintaining its original flow direction, is forcibly transferred to the parasitic output capacitance circuit of the complementary switch about to be turned on in the same bridge arm. The equivalent leakage inductance current rapidly discharges and extracts the parasitic output capacitance of the complementary switch about to be turned on, simultaneously charging the parasitic output capacitance of the switch that was just turned off. As the dead time progresses, the drain-source voltage across the complementary switch about to be turned on exhibits a linear decreasing trend. When the drain-source voltage across the complementary switch completely drops to zero volts, the anti-parallel diode inside the complementary switch reaches the forward bias condition and conducts first, taking over the task of freewheeling the inductor current, thus clamping the voltage across the complementary switch at a zero voltage level. The underlying drive module must accurately capture the physical time window when the anti-parallel diode is in steady-state conduction and issue a turn-on pulse signal within the specific physical time window to ensure that there is no Joule heat loss due to capacitor discharge when the internal semiconductor channel of the switching device is turned on. Through the above series of rigorous duty cycle decoupling and timing control steps, the hybrid three-level converter completely achieves zero-voltage turn-on of the switching device across the entire operating range, and completely suppresses the reactive power circulating current between the primary and secondary sides of the high-frequency transformer from the underlying physical link, significantly reducing the overall hardware conduction loss and switching thermal stress, and ultimately achieving the ultimate energy conversion efficiency under high-power wide voltage ratio conditions.

[0061] Based on the above method embodiments, the present invention provides corresponding apparatus embodiments.

[0062] like Figure 15 As shown, an embodiment of the present invention provides a hybrid three-level converter control device for suppressing reactive circulating current. The hybrid three-level converter includes a primary-side diode midpoint clamped three-level bridge arm, an isolation transformer, and a secondary-side full bridge. The control device includes: a data acquisition module, a parameter calculation module, a constraint boundary determination module, a duty cycle decoupling calculation module, and a drive control module; The data acquisition module is used to acquire the actual input voltage of the primary side and the actual output voltage of the secondary side of the hybrid three-level converter; The parameter calculation module is used to calculate the terminal voltage ratio based on the actual input voltage of the primary side, the actual output voltage of the secondary side, and the preset isolation transformer ratio; and to calculate the total phase shift requirement based on the actual output voltage of the secondary side and the preset reference output voltage. The constraint boundary determination module is used to determine the upper limit threshold of the outer phase shift duty cycle and the lower limit of the outer phase shift duty cycle relative to the inner phase shift duty cycle based on the preset zero-voltage turn-on critical condition and the terminal voltage ratio. The duty cycle decoupling calculation module is used to calculate the target inner phase shift angle duty cycle and the target outer phase shift angle duty cycle based on the total phase shift requirement, the upper threshold, and the associated lower limit. The target inner phase shift angle duty cycle is a phase shift variable that controls the operation of the internal switching transistor of the three-level bridge arm clamped at the midpoint of the primary diode. The target outer phase shift angle duty cycle is a phase shift variable that controls the cross-bridge delay operation of the switching transistor of the secondary full-bridge relative to the switching transistor of the three-level bridge arm clamped at the midpoint of the primary diode. The drive control module is used to generate primary-side drive pulse signals and secondary-side drive pulse signals according to the target inner phase shift duty cycle and the target outer phase shift duty cycle; the primary-side drive pulse signals are used to control the action of the three-level bridge arm clamped at the midpoint of the primary-side diodes, and the secondary-side drive pulse signals are used to control the action of the secondary-side full bridge.

[0063] like Figure 2 As shown, one embodiment of the present invention provides a hybrid three-level converter for suppressing reactive circulating current, comprising: a primary-side diode midpoint clamped three-level bridge arm, a high-frequency isolation transformer, and a secondary-side full bridge; The hybrid three-level converter has a primary-side DC input terminal. A first voltage-dividing capacitor and a second voltage-dividing capacitor are connected between the positive and negative terminals of the primary-side DC input terminal. The first end of the first voltage-dividing capacitor is connected to the positive terminal, and the second end of the second voltage-dividing capacitor is connected to the negative terminal. The second end of the first voltage-dividing capacitor and the first end of the second voltage-dividing capacitor are connected to form a first common connection point. The primary-side diode midpoint clamping three-level bridge arm includes the first to eighth switching transistors and the corresponding clamping diodes, wherein the first to fourth switching transistors are connected end to end in sequence to form a leading bridge arm, and the fifth to eighth switching transistors are connected end to end in sequence to form a lagging bridge arm. The first common connection point is not only connected to the clamping midpoint of the three-level bridge arm of the primary diode, but also directly extended to the center tap of the primary winding of the high-frequency isolation transformer. The hybrid three-level converter also includes an equivalent resonant inductor, the first end of which is connected to the midpoint of the three-level bridge arm clamped by the midpoint of the primary diode, and the second end of which is connected to the primary winding of the high-frequency isolation transformer. The secondary-side full bridge includes the ninth to twelfth switching transistors; the hybrid three-level converter also includes a DC blocking capacitor, the first end of which is connected to the secondary winding of the high-frequency isolation transformer, and the second end of which is connected to the input midpoint of the secondary-side full bridge. The hybrid three-level converter has a secondary output terminal. A first output filter capacitor and a second output filter capacitor are connected between the positive and negative terminals of the secondary output terminal. The first end of the first output filter capacitor is connected to the positive terminal of the secondary output terminal, and the second end of the second output filter capacitor is connected to the negative terminal of the secondary output terminal for connecting a load.

[0064] Furthermore, the aforementioned asymmetric hybrid three-level hardware topology, by employing a diode-clamped three-level structure on the primary high-voltage side and cleverly utilizing the hardware topology loop of the passive clamping diode, can strictly and stably limit the steady-state voltage stress of all primary-side switching devices to half of the actual primary-side input voltage. This physical voltage reduction mechanism not only allows the system to use high-efficiency semiconductor power devices with low withstand voltage ratings and low on-resistance, fundamentally reducing hardware manufacturing costs and suppressing the accumulation of switching heat, but also effectively eliminates the difficulties in dynamic voltage equalization and the risk of overvoltage breakdown caused by the traditional direct series connection scheme of high-voltage devices.

[0065] Meanwhile, compared to other multi-level topologies such as flying capacitor topologies, the hardware architecture of this embodiment abandons bulky floating passive energy storage components, completely avoiding the challenges of complex pre-charge timing control and the resulting space encroachment problems. This primary-side topology, combined with the compact traditional two-level full-bridge circuit on the secondary low-voltage side, maximizes the power density of the entire isolated bidirectional converter while ensuring the insulation safety of the high-voltage side, making it highly suitable for high-power mobile energy storage operation scenarios with extremely stringent requirements for physical space and weight. More importantly, this asymmetric hardware foundation provides a solid and necessary hardware support boundary at the physical level for the subsequent introduction of extended phase-shifting dual-degree-of-freedom collaborative modulation strategies, including inner and outer phase shift angles, thereby actively reducing reactive power backflow in the high-frequency resonant cavity.

[0066] It should be noted that the embodiments of the device described above correspond to the embodiments of the present invention described above, and can realize the hybrid three-level converter control method for suppressing reactive circulating current as described in any one of the above embodiments of the present invention. Furthermore, the embodiments of the device described above are merely illustrative. The modules described as separate components may or may not be physically separate, and the components shown as modules may or may not be physical units, that is, they may be located in one place or distributed across multiple network units. Some or all of the modules can be selected to achieve the purpose of this embodiment according to actual needs. In addition, in the accompanying drawings of the device embodiments provided by the present invention, the connection relationship between modules indicates that they have a communication connection, which can be specifically implemented as one or more communication buses or signal lines. Those skilled in the art can understand and implement this without creative effort.

[0067] Based on the above-described method embodiments of the present invention, a corresponding embodiment of an electronic device is provided.

[0068] An embodiment of the present invention provides an electronic device, including a processor, a memory, and a computer program stored in the memory and configured to be executed by the processor. When the processor executes the computer program, it implements the hybrid three-level converter control method for suppressing reactive circulating current as described in any one of the present invention, or, when the processor executes the computer program, it implements the functions of each module in the above-described device embodiments.

[0069] For example, the computer program may be divided into one or more modules, which are stored in the memory and executed by the processor to complete the present invention. The one or more modules may be a series of computer program instruction segments capable of performing specific functions, which describe the execution process of the computer program in the terminal device.

[0070] The terminal device may be a desktop computer, laptop, handheld computer, or cloud server, etc. The terminal device may include, but is not limited to, a processor and a memory.

[0071] The processor can be a Central Processing Unit (CPU), or other general-purpose processors, digital signal processors (DSPs), application-specific integrated circuits (ASICs), field-programmable gate arrays (FPGAs), or other programmable logic devices, discrete gate or transistor logic devices, discrete hardware components, etc. A general-purpose processor can be a microprocessor or any conventional processor. The processor is the control center of the terminal device, connecting all parts of the terminal device via various interfaces and lines.

[0072] The memory can be used to store the computer programs and / or modules. The processor implements various functions of the terminal device by running or executing the computer programs and / or modules stored in the memory and by calling data stored in the memory. The memory may mainly include a program storage area and a data storage area. The program storage area may store the operating system, applications required for at least one function, etc.; the data storage area may store data created based on the use of the mobile phone, etc. In addition, the memory may include high-speed random access memory, and may also include non-volatile memory, such as hard disk, memory, plug-in hard disk, smart media card (SMC), secure digital card (SD card), flash card, at least one disk storage device, flash memory device, or other volatile solid-state storage device.

[0073] In the description of this specification, the references to terms such as "one embodiment," "some embodiments," "example," "specific example," or "some examples," etc., indicate that a specific feature, structure, material, or characteristic described in connection with that embodiment or example is included in at least one embodiment or example of this application. Furthermore, the specific features, structures, materials, or characteristics described may be combined in any suitable manner in one or more embodiments or examples. Moreover, without contradiction, those skilled in the art can combine and integrate the different embodiments or examples described in this specification, as well as the features of those different embodiments or examples.

[0074] The above description represents the preferred embodiments of the present invention. It should be noted that those skilled in the art can make various improvements and modifications without departing from the principles of the present invention, and these improvements and modifications are also considered to be within the scope of protection of the present invention.

Claims

1. A control method for a hybrid three-level converter to suppress reactive circulating current, characterized in that, The hybrid three-level converter includes a primary-side diode-midpoint-clamped three-level bridge arm, an isolation transformer, and a secondary-side full bridge; the control method includes: Obtain the actual primary input voltage and actual secondary output voltage of the hybrid three-level converter; Calculate the terminal voltage ratio based on the actual input voltage of the primary side, the actual output voltage of the secondary side, and the preset isolation transformer turns ratio; calculate the total phase shift requirement based on the actual output voltage of the secondary side and the preset reference output voltage. Based on the preset zero-voltage turn-on critical condition and the terminal voltage ratio, determine the upper limit threshold of the outer phase shift duty cycle and the lower limit of the outer phase shift duty cycle relative to the inner phase shift duty cycle; Based on the total phase shift requirement, upper threshold, and associated lower limit, calculate the target inner phase shift duty cycle and the target outer phase shift duty cycle; wherein, the target inner phase shift duty cycle is the phase shift variable that controls the operation of the internal switch of the three-level bridge arm clamped at the midpoint of the primary diode; the target outer phase shift duty cycle is the phase shift variable that controls the cross-bridge delay operation of the switch of the secondary full bridge relative to the switch of the three-level bridge arm clamped at the midpoint of the primary diode; Based on the duty cycle of the target's inner phase shift angle and the duty cycle of the target's outer phase shift angle, primary-side drive pulse signals and secondary-side drive pulse signals are generated; the primary-side drive pulse signals are used to control the action of the three-level bridge arm clamped at the midpoint of the primary-side diodes, and the secondary-side drive pulse signals are used to control the action of the secondary-side full bridge.

2. The hybrid three-level converter control method for suppressing reactive circulating current as described in claim 1, characterized in that, Based on the actual input voltage of the primary side, the actual output voltage of the secondary side, and the preset isolation transformer turns ratio, the terminal voltage ratio is calculated, including: Based on the preset isolation transformer turns ratio, the actual input voltage on the primary side is voltage-reduced to generate an equivalent voltage-reduced voltage. The terminal voltage ratio is determined based on the equivalent converted voltage and the actual output voltage on the secondary side.

3. The hybrid three-level converter control method for suppressing reactive circulating current as described in claim 2, characterized in that, Based on the actual secondary output voltage and the preset reference output voltage, calculate the total phase shift requirement, including: The voltage deviation value is determined based on the preset reference output voltage and the actual output voltage on the secondary side. The voltage deviation value is input to a preset closed-loop voltage regulator for adjustment, generating an adjustment output result; Based on the adjustment output results, determine the total phase shift requirement.

4. The hybrid three-level converter control method for suppressing reactive circulating current as described in claim 3, characterized in that, Based on the preset zero-voltage turn-on critical condition and the terminal voltage ratio, determine the upper limit threshold of the outer phase shift duty cycle and the lower limit of the outer phase shift duty cycle relative to the inner phase shift duty cycle, including: Based on the terminal voltage ratio and the preset zero-voltage turn-on critical condition, a soft-switching boundary constraint model is constructed for the target operating mode. From the soft-switching boundary constraint model, the maximum outward phase shift duty cycle critical value that satisfies the preset power transmission direction and maintains zero voltage turn-on is extracted as the upper limit threshold of the outward phase shift duty cycle; Based on the soft-switching boundary constraint model, the minimum boundary condition is determined to ensure that the duty cycle of the outer phase shift angle dynamically changes with the duty cycle of the inner phase shift angle when the switching device is turned on at zero voltage. The minimum boundary condition is used as the lower limit of the correlation between the duty cycle of the outer phase shift angle and the duty cycle of the inner phase shift angle.

5. The hybrid three-level converter control method for suppressing reactive circulating current as described in claim 4, characterized in that, Based on the total phase shift requirement, upper threshold, and associated lower limit, calculate the target inner phase shift duty cycle and the target outer phase shift duty cycle, including: Based on the upper threshold and the associated lower limit, the boundary of the phase-shifting working range is constructed; The total phase shift demand is compared with the phase shift operating range boundary to determine the target phase shift operating range to which the converter currently belongs; Based on the preset mapping relationship between phase shifting working intervals and allocation rules, the target allocation rule for the corresponding target phase shifting working interval is obtained; Based on the target allocation rules, the total phase shift requirement is decoupled and calculated to generate the target inner phase shift duty cycle and the target outer phase shift duty cycle.

6. The hybrid three-level converter control method for suppressing reactive circulating current as described in claim 5, characterized in that, Based on the target's inner phase shift duty cycle and the target's outer phase shift duty cycle, primary-side drive pulse signals and secondary-side drive pulse signals are generated, including: Extract a preset global switching periodic carrier as a unified phase reference; Based on the phase reference, the duty cycle of the target inner phase shift angle is converted into the inner phase shift time of the primary side bridge arm. The timing difference between the action of the inner and outer switches in the three-level bridge arm is configured according to the inner phase shift time of the primary side bridge arm, and the primary side drive pulse signal is generated. The target phase shift angle duty cycle is converted into the phase shift time between the primary and secondary bridges. Starting from the emission phase of the primary driving pulse signal, the delay action timing of the secondary full-bridge switch relative to the midpoint of the primary diode is configured according to the phase shift time between the primary and secondary bridges to generate the secondary driving pulse signal.

7. The hybrid three-level converter control method for suppressing reactive circulating current as described in claim 6, characterized in that, The operation of the three-level bridge arm is controlled by the primary-side drive pulse signal to clamp the midpoint of the primary-side diode, and the operation of the secondary-side full bridge is controlled by the secondary-side drive pulse signal, including: The primary-side drive pulse signal is applied to the control terminals of the outer and inner switches of the three-level bridge arm clamped at the midpoint of the primary-side diode, respectively, to synthesize a primary-side high-frequency AC voltage containing positive, zero, and negative level sequences. The secondary-side drive pulse signal is applied to the control terminal of each switch of the secondary-side full bridge, controlling the secondary-side full bridge to perform turn-on and turn-off actions according to the preset asymmetrical conduction sequence, and synthesizing the corresponding secondary-side high-frequency AC voltage. Active power transmission is driven by the voltage difference generated between the primary and secondary windings of a high-frequency transformer through the primary-side high-frequency AC voltage and the secondary-side high-frequency AC voltage.

8. A hybrid three-level converter control device for suppressing reactive circulating current, characterized in that, The hybrid three-level converter includes a primary-side diode midpoint clamped three-level bridge arm, an isolation transformer, and a secondary-side full bridge. The control device includes: a data acquisition module, a parameter calculation module, a constraint boundary determination module, a duty cycle decoupling calculation module, and a drive control module; The data acquisition module is used to acquire the actual input voltage of the primary side and the actual output voltage of the secondary side of the hybrid three-level converter; The parameter calculation module is used to calculate the terminal voltage ratio based on the actual input voltage of the primary side, the actual output voltage of the secondary side, and the preset isolation transformer ratio; and to calculate the total phase shift requirement based on the actual output voltage of the secondary side and the preset reference output voltage. The constraint boundary determination module is used to determine the upper limit threshold of the outer phase shift duty cycle and the lower limit of the outer phase shift duty cycle relative to the inner phase shift duty cycle based on the preset zero-voltage turn-on critical condition and the terminal voltage ratio. The duty cycle decoupling calculation module is used to calculate the target inner phase shift angle duty cycle and the target outer phase shift angle duty cycle based on the total phase shift requirement, the upper threshold, and the associated lower limit. The target inner phase shift angle duty cycle is a phase shift variable that controls the operation of the internal switching transistor of the three-level bridge arm clamped at the midpoint of the primary diode. The target outer phase shift angle duty cycle is a phase shift variable that controls the cross-bridge delay operation of the switching transistor of the secondary full-bridge relative to the switching transistor of the three-level bridge arm clamped at the midpoint of the primary diode. The drive control module is used to generate primary-side drive pulse signals and secondary-side drive pulse signals according to the target inner phase shift duty cycle and the target outer phase shift duty cycle; the primary-side drive pulse signals are used to control the action of the three-level bridge arm clamped at the midpoint of the primary-side diodes, and the secondary-side drive pulse signals are used to control the action of the secondary-side full bridge.

9. An electronic device, characterized in that, The device includes a processor, a memory, and a computer program stored in the memory and configured to be executed by the processor, wherein the processor, when executing the computer program, implements the hybrid three-level converter control method for suppressing reactive circulating current as described in any one of claims 1 to 7.

10. A hybrid three-level converter for suppressing reactive circulating current, characterized in that, include: Primary-side diode midpoint clamping three-level bridge arm, high-frequency isolation transformer, and secondary-side full bridge; The hybrid three-level converter has a primary-side DC input terminal. A first voltage-dividing capacitor and a second voltage-dividing capacitor are connected between the positive and negative terminals of the primary-side DC input terminal. The first end of the first voltage-dividing capacitor is connected to the positive terminal, and the second end of the second voltage-dividing capacitor is connected to the negative terminal. The second end of the first voltage-dividing capacitor and the first end of the second voltage-dividing capacitor are connected to form a first common connection point. The primary-side diode midpoint clamping three-level bridge arm includes the first to eighth switching transistors and the corresponding clamping diodes, wherein the first to fourth switching transistors are connected end to end in sequence to form a leading bridge arm, and the fifth to eighth switching transistors are connected end to end in sequence to form a lagging bridge arm. The first common connection point is not only connected to the clamping midpoint of the three-level bridge arm of the primary diode, but also directly extended to the center tap of the primary winding of the high-frequency isolation transformer. The hybrid three-level converter also includes an equivalent resonant inductor, the first end of which is connected to the midpoint of the three-level bridge arm clamped by the midpoint of the primary diode, and the second end of which is connected to the primary winding of the high-frequency isolation transformer. The secondary-side full bridge includes the ninth to twelfth switching transistors; the hybrid three-level converter also includes a DC blocking capacitor, the first end of which is connected to the secondary winding of the high-frequency isolation transformer, and the second end of which is connected to the input midpoint of the secondary-side full bridge. The hybrid three-level converter has a secondary output terminal. A first output filter capacitor and a second output filter capacitor are connected between the positive and negative terminals of the secondary output terminal. The first end of the first output filter capacitor is connected to the positive terminal of the secondary output terminal, and the second end of the second output filter capacitor is connected to the negative terminal of the secondary output terminal for connecting a load.

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