A dual output voltage converter and dual input voltage converter

Abstract

The application relates to the technical field of voltage conversion, and discloses a voltage converter with bipolar output and a double-input voltage converter. The voltage converter comprises a voltage boosting unit and an output capacitor branch. The voltage boosting unit comprises an energy storage inductor L1, a coupling inductor L2, a coupling inductor L3, a voltage boosting branch, a switching tube S1, a fifth unidirectional conduction unit and a sixth unidirectional conduction unit. The coupling inductor L2 is coupled with the coupling inductor L3. In use, the energy storage and release of the energy storage inductor L1, the coupling inductor L2 and the coupling inductor L3, and the voltage superposition of capacitors C1, C2 and C3 make the total output voltage gain of the voltage converter reach 2(1+N) / (1-D) 2 wherein N is the turn ratio of the energy storage inductor L1 and the coupling inductor L2, and D is the duty ratio of a PWM control signal input at the control end of the switching tube S1. Therefore, a higher output voltage can be obtained by reasonably setting the turn ratio under the condition of a smaller duty ratio, and high gain is achieved.

Description

Technical Field

[0001] This invention relates to the field of voltage conversion technology, specifically to a bipolar output voltage converter and a dual-input voltage converter. Background Technology

[0002] Photovoltaic power generation, fuel cells, and energy storage batteries are typically low-voltage DC power outputs, while high-voltage DC buses, grid-connected inverter front-ends, DC microgrid interfaces, and bipolar DC power supply systems often require higher and more stable DC voltages. Therefore, a high-gain DC-DC converter is usually needed between the low-voltage renewable energy source and the high-voltage DC load to achieve voltage boosting, energy regulation, and bus voltage stabilization.

[0003] Existing non-isolated high-gain DC-DC converters mainly include traditional Boost converters, coupled inductor boost converters, switched capacitor / voltage doubler unit boost converters, and coupled inductor and capacitor-diode network combined boost converters.

[0004] Traditional Boost converters have a simple structure, but in high-output-voltage applications they typically require a very large duty cycle, which can easily lead to problems such as increased voltage and current stress on switching devices and diodes, aggravated diode reverse recovery, decreased efficiency, and reduced control margin.

[0005] Coupled inductor boost converters can improve voltage gain by utilizing turns ratio, but voltage spikes caused by leakage inductance, voltage deviations on the coupled secondary side, and device stress still need to be addressed. Switched capacitors or voltage multiplier units can achieve capacitor voltage superposition, but in high-power applications, they are prone to problems such as large charging and discharging pulsating current, output voltage imbalance, and high device losses.

[0006] Furthermore, for applications requiring positive and negative DC buses, some existing solutions typically employ downstream voltage divider capacitors or additional balancing control to achieve bipolar output. When the charging paths of the upper and lower output capacitors are asymmetrical or the load changes, voltage imbalance between the positive and negative buses can easily occur, thereby affecting the operational stability of the downstream inverter or bipolar DC load.

[0007] Based on the above, existing voltage converters cannot simultaneously achieve high gain, bipolar output, balanced output bus voltage, and relatively simple control output with a limited number of components. Summary of the Invention

[0008] In view of the shortcomings of the prior art, the present invention provides a bipolar output voltage converter and a dual-input voltage converter. The technical problem to be solved is that existing voltage converters cannot simultaneously achieve high gain, bipolar output, output bus voltage balance and relatively simple control output with a small number of components.

[0009] To solve the above technical problems, in a first aspect, the present invention provides the following technical solution: a bipolar output voltage converter, comprising a boost unit and an output capacitor branch; The boost unit includes an energy storage inductor L1, a coupling inductor L2, a coupling inductor L3, a boost branch, a switching transistor S1, a fifth unidirectional conductive unit, and a sixth unidirectional conductive unit; the coupling inductor L2 is coupled to the coupling inductor L3; The boost circuit includes a capacitor C1, a first unidirectional conductive unit, a second unidirectional conductive unit, a third unidirectional conductive unit, a capacitor C3, a capacitor C2, and a fourth unidirectional conductive unit connected in series. The input terminal of the first unidirectional conductive unit is electrically connected to the input terminal of the second unidirectional conductive unit, and the output terminal of the second unidirectional conductive unit is electrically connected to the input terminal of the third unidirectional conductive unit. The end of capacitor C2 that is not electrically connected to capacitor C3 is electrically connected to the input terminal of the fourth unidirectional conductive unit. The output terminal of the fourth unidirectional conductive unit is electrically connected to the output terminal of the switching transistor S1. The two ends of the inductor L1 are electrically connected to one end of the capacitor C1 and the input end of the first unidirectional conductive unit, respectively, and the other end of the capacitor C1 is electrically connected to the output end of the first unidirectional conductive unit. The same-name terminal of the coupling inductor L2 is electrically connected to the output terminal of the first unidirectional conductive unit; the opposite-name terminal of the inductor L2 is electrically connected to the output terminal of the second unidirectional conductive unit, the input terminal of the switch S1, and the same-name terminal of the coupling inductor L3, respectively; the coupling inductor L3 and the capacitor C3 are electrically connected to one end of the capacitor C2. The input terminal of the fifth unidirectional conductive unit is electrically connected to the output terminal of the third unidirectional conductive unit, and the output terminal of the fifth unidirectional conductive unit is electrically connected to the first terminal of the output capacitor branch. The input terminal of the sixth unidirectional conductive unit is electrically connected to the second terminal of the output capacitor branch, and the output terminal of the sixth unidirectional conductive unit is electrically connected to the input terminal of the fourth unidirectional conductive unit.

[0010] In one embodiment of the first aspect, the first unidirectional conductive unit, the second unidirectional conductive unit, the third unidirectional conductive unit, the fourth unidirectional conductive unit, the fifth unidirectional conductive unit, and the sixth unidirectional conductive unit are all turned on when the voltage difference between their input and output terminals is greater than the threshold voltage. The first, second, third, fourth, fifth, and sixth unidirectional conductive units are all turned off when the voltage difference between their input and output terminals is less than the threshold voltage.

[0011] In one embodiment of the first aspect, the first unidirectional conductive unit, the second unidirectional conductive unit, the third unidirectional conductive unit, the fourth unidirectional conductive unit, the fifth unidirectional conductive unit, and the sixth unidirectional conductive unit each include a diode, with the anode of the diode serving as the input terminal of the unidirectional conductive unit and the cathode of the diode serving as the output terminal of the unidirectional conductive unit.

[0012] In one embodiment of the first aspect, the diode is a fast recovery diode, a Schottky diode, or a silicon carbide diode.

[0013] In one embodiment of the first aspect, the switching transistor is an NMOS transistor, the drain of the NMOS transistor is the input terminal of the switching transistor S1, the source of the NMOS transistor is the output terminal of the switching transistor S1, and the gate of the NMOS transistor is the control terminal of the switching transistor S1, used to input PWM control signals.

[0014] In one embodiment of the first aspect, the output capacitor branch includes an output capacitor CO1 and an output capacitor CO2; one end of the output capacitor CO1 is the first endpoint of the output capacitor branch, the other end of the output capacitor CO1 is electrically connected to one end of the output capacitor CO2 and is the intermediate node of the output capacitor branch, and the other end of the output capacitor CO2 is the second endpoint of the output capacitor branch.

[0015] In one embodiment of the first aspect, the intermediate node of the output capacitor branch is also electrically connected to the output terminal of the fourth unidirectional conductive unit for grounding.

[0016] In one embodiment of the first aspect, capacitors C1, C2, and C3 are thin-film capacitors.

[0017] In a second aspect, the present invention also provides a dual-input voltage converter, comprising two boost units in the above-described bipolar output voltage converters and an output capacitor branch in the above-described bipolar output voltage converter. The input terminal of the fifth unidirectional conductive unit of the two boost units is electrically connected to the first terminal of the output capacitor branch, and the input terminal of the sixth unidirectional conductive unit of the two boost units is electrically connected to the second terminal of the output capacitor branch. In one embodiment of the second aspect, one end of capacitor C1 of one boost unit is used to be electrically connected to a photovoltaic power generation unit, and one end of capacitor C1 of another boost unit is used to be electrically connected to a fuel cell unit.

[0018] The beneficial effects of this invention compared to the prior art are: First, through the energy storage and release of the energy storage inductors L1, L2, and L3, and the voltage superposition of capacitors C1, C2, and C3, the total output voltage gain of the voltage converter reaches 2. (1+N) / (1-D) 2 Where N is the turns ratio of coupled inductors L1 and L2, and D is the duty cycle of the PWM control signal input to the control terminal of switch S1. Therefore, a higher output voltage can be obtained by reasonably setting the turns ratio with a smaller duty cycle, thus achieving high gain. Secondly, the present invention is simple to control. The circuit only sets one main switch S1. The driving and control method is simpler than that of a multi-switch interleaved structure. There is no problem of uneven power caused by the timing deviation of the driving of multiple main switches. In addition, compared with multi-phase, multi-stage cascaded or multi-switch high-gain solutions, the present invention can achieve high-gain output with a main switch, three inductor / coupling windings and a small number of diodes and capacitors, which is beneficial to improve power density and has fewer overall components; Finally, by transferring energy to the two endpoints of the output capacitor branch respectively, the problem of voltage imbalance between the positive and negative buses can be reduced. Attached Figure Description

[0019] Figure 1 This is a circuit diagram of the bipolar output voltage converter in Example 1; Figure 2 for Figure 1 A schematic diagram of the current flow when the circuit is operating in mode one; Figure 3 for Figure 1 A schematic diagram of the current flow direction when the circuit is operating in mode two; Figure 4 for Figure 1 A schematic diagram of the current flow direction when the circuit is operating in mode three; Figure 5 for Figure 1 A schematic diagram of the current flow when the circuit is operating in mode four; Figure 6 This is a circuit diagram of the dual-input voltage converter in Example 2. Detailed Implementation

[0020] The illustrative embodiments of this application include, but are not limited to, a bipolar output voltage converter and a dual-input voltage converter.

[0021] Exemplary embodiments will now be described in detail, examples of which are illustrated in the accompanying drawings. When the following description relates to the drawings, unless otherwise indicated, the same numbers in different drawings denote the same or similar elements. The embodiments described in the following exemplary embodiments do not represent all embodiments consistent with this application. Rather, they are merely examples of apparatuses and methods consistent with some aspects of this application as detailed in the appended claims.

[0022] The terminology used in this application is for the purpose of describing particular embodiments only and is not intended to be limiting of the application. The singular forms “a,” “the,” and “the” used in this application and the appended claims are also intended to include the plural forms unless the context clearly indicates otherwise. It should also be understood that the term “and / or” as used herein refers to and includes any or all possible combinations of one or more of the associated listed items. Words such as “comprising” or “including” mean that the element or object preceding “comprising” or “including” covers the element or object listed following “comprising” or “including” and its equivalents, and does not exclude other elements or objects. Words such as “connected” or “linked” are not limited to physical or mechanical connections and can include electrical connections, whether direct or indirect.

[0023] Example 1 Existing non-isolated high-gain DC-DC converters mainly include traditional Boost converters, coupled-inductor boost converters, switched-capacitor / voltage doubler boost converters, and combined coupled-inductor and capacitor-diode network boost converters. Traditional Boost converters have a simple structure, but in high-output-voltage applications, they typically require a very large duty cycle, which can easily lead to increased voltage and current stress on switching devices and diodes, exacerbated diode reverse recovery, decreased efficiency, and reduced control margin. Coupled-inductor boost converters can improve voltage gain using turns ratio, but voltage spikes caused by leakage inductance, coupled secondary-side voltage deviation, and device stress still require significant attention. Switched-capacitor or voltage doubler units can achieve capacitor voltage superposition, but in high-power applications, they are prone to problems such as large charging and discharging ripple current, output voltage imbalance, and high device losses.

[0024] Furthermore, for applications requiring positive and negative DC buses, some existing solutions typically employ downstream voltage divider capacitors or additional balancing control to achieve bipolar output. When the charging paths of the upper and lower output capacitors are asymmetrical or the load changes, voltage imbalance between the positive and negative buses can easily occur, thereby affecting the operational stability of the downstream inverter or bipolar DC load.

[0025] As can be seen from the problems of the existing voltage converters mentioned above, the existing voltage converters cannot simultaneously achieve high gain, bipolar output, output bus voltage balance, and relatively simple control output with a small number of components.

[0026] To address the problems existing in current voltage converters, such as Figure 1 As shown, this embodiment provides a bipolar output voltage converter, including a boost unit 1 and an output capacitor branch 2; The boost unit 1 includes an energy storage inductor L1, a coupling inductor L2, a coupling inductor L3, a boost branch 10, a switching transistor S1, a fifth unidirectional conductive unit 11, and a sixth unidirectional conductive unit 12; the coupling inductor L2 is coupled to the coupling inductor L3; The boost circuit 10 includes a capacitor C1, a first unidirectional conductive unit 100, a second unidirectional conductive unit 101, a third unidirectional conductive unit 102, a capacitor C3, a capacitor C2, and a fourth unidirectional conductive unit 103 connected in series. The input terminal of the first unidirectional conductive unit 100 is electrically connected to the input terminal of the second unidirectional conductive unit 101, the output terminal of the second unidirectional conductive unit 101 is electrically connected to the input terminal of the third unidirectional conductive unit 102, and the end of capacitor C2 that is not electrically connected to capacitor C3 is electrically connected to the input terminal of the fourth unidirectional conductive unit 103. The output terminal of the fourth unidirectional conductive unit 103 is electrically connected to the output terminal of the switching transistor S1. The two ends of the inductor L1 are electrically connected to one end of the capacitor C1 and the input end of the first unidirectional conductive unit 100, respectively, and the other end of the capacitor C1 is electrically connected to the output end of the first unidirectional conductive unit 100. In actual use, one end of the capacitor C1 is used to input the voltage to be converted, which can be provided by the photovoltaic power generation unit, the fuel cell unit or the energy storage battery unit. The same-name terminal of the coupling inductor L2 is electrically connected to the output terminal of the first unidirectional conductive unit 100; the opposite-name terminal of the inductor L2 is electrically connected to the output terminal of the second unidirectional conductive unit 101, the input terminal of the switch S1, and the same-name terminal of the coupling inductor L3, respectively; the coupling inductor L3 and the end of the capacitor C3 that is electrically connected to the capacitor C2 are electrically connected. The input terminal of the fifth unidirectional conductive unit 11 is electrically connected to the output terminal of the third unidirectional conductive unit 102, and the output terminal of the fifth unidirectional conductive unit 11 is electrically connected to the first terminal of the output capacitor branch 2. The input terminal of the sixth unidirectional conductive unit 12 is electrically connected to the second terminal of the output capacitor branch 2, and the output terminal of the sixth unidirectional conductive unit 12 is electrically connected to the input terminal of the fourth unidirectional conductive unit 103.

[0027] In practical use, the voltage converter in this embodiment first utilizes the energy storage and release of the energy storage inductor L1, coupling inductor L2, and coupling inductor L3, as well as the voltage superposition of capacitors C1, C2, and C3, to achieve a total output voltage gain of 2. (1+N) / (1-D) 2 Where N is the turns ratio of coupled inductors L1 and L2, and D is the duty cycle of the PWM control signal input to the control terminal of switch S1. Therefore, a higher output voltage can be obtained by reasonably setting the turns ratio with a smaller duty cycle, thus achieving high gain. Secondly, the voltage converter control in this embodiment is simple. The circuit only sets one main switch S1. The driving and control method is simpler than that of the multi-switch interleaved structure. There is no problem of uneven power caused by the timing deviation of the multi-main switch driving. In addition, compared with multi-phase, multi-stage cascaded or multi-switch high-gain solutions, the voltage converter in this embodiment can achieve high-gain output with a main switch, three inductor / coupling windings and a small number of diodes and capacitors, which is beneficial to improve power density and has fewer overall components. Finally, by transferring energy to the two endpoints of the output capacitor branch 2 respectively, the problem of voltage imbalance between the positive and negative buses can be reduced.

[0028] Specifically, in this embodiment, the first unidirectional conductive unit 100 is turned on when the voltage difference between its input terminal and output terminal is greater than the threshold voltage, and turned off when the voltage difference between its input terminal and output terminal is less than the threshold voltage.

[0029] Specifically, in this embodiment, the second unidirectional conductive unit 101 is turned on when the voltage difference between its input terminal and output terminal is greater than the threshold voltage, and turned off when the voltage difference between its input terminal and output terminal is less than the threshold voltage.

[0030] Specifically, in this embodiment, the third unidirectional conductive unit 102 is turned on when the voltage difference between its input terminal and output terminal is greater than the threshold voltage, and turned off when the voltage difference between its input terminal and output terminal is less than the threshold voltage.

[0031] Specifically, in this embodiment, the fourth unidirectional conductive unit 103 is turned on when the voltage difference between its input terminal and output terminal is greater than the threshold voltage, and turned off when the voltage difference between its input terminal and output terminal is less than the threshold voltage.

[0032] Specifically, in this embodiment, the fifth unidirectional conductive unit 11 is turned on when the voltage difference between its input terminal and output terminal is greater than the threshold voltage, and turned off when the voltage difference between its input terminal and output terminal is less than the threshold voltage.

[0033] Specifically, in this embodiment, the sixth unidirectional conductive unit 12 is turned on when the voltage difference between its input terminal and output terminal is greater than the threshold voltage, and turned off when the voltage difference between its input terminal and output terminal is less than the threshold voltage.

[0034] Furthermore, in this embodiment, the first unidirectional conductive unit 100, the second unidirectional conductive unit 101, the third unidirectional conductive unit 102, the fourth unidirectional conductive unit 103, the fifth unidirectional conductive unit 11, and the sixth unidirectional conductive unit 12 each include a diode, with the anode of the diode being the input terminal of the unidirectional conductive unit and the cathode of the diode being the output terminal of the unidirectional conductive unit.

[0035] exist Figure 1In the first unidirectional conductive unit 100, the diode is D1; ​​in the second unidirectional conductive unit 101, the diode is D2; in the third unidirectional conductive unit 102, the diode is D3; in the fourth unidirectional conductive unit 103, the diode is D4; in the fifth unidirectional conductive unit 11, the diode is D5; and in the sixth unidirectional conductive unit 12, the diode is D6.

[0036] Specifically, in this embodiment, the switching transistor is an NMOS transistor, the drain of the NMOS transistor is the input terminal of the switching transistor S1, the source of the NMOS transistor is the output terminal of the switching transistor S1, and the gate of the NMOS transistor is the control terminal of the switching transistor S1, used to input PWM control signals.

[0037] In practical use, the gate of the NMOS transistor is turned on when a high-level signal is input and turned off when a low-level signal is input; the boost gain of the voltage converter in this embodiment can be adjusted by adjusting the magnitude of the PWM control signal.

[0038] For example, the control process of switch S1 is as follows: Taking photovoltaic applications as an example, a maximum power point tracking (MPPT) algorithm can be added to the front end to adjust the duty cycle in real time according to the photovoltaic output voltage and current, so that the output voltage meets the requirements of the downstream DC bus.

[0039] There are two main ways to adjust the duty cycle: Method 1: The MPPT algorithm first calculates a target reference voltage (i.e., the maximum power point voltage under the current environment). Then, it compares this reference voltage with the actual collected photovoltaic panel voltage, and the difference is sent to the PI (Proportional-Integral) controller. The PI controller calculates the required duty cycle based on this difference, and then generates a PWM (Pulse Width Modulation) signal to control the switching transistor.

[0040] Method 2: The MPPT algorithm directly calculates and outputs the duty cycle value, without needing the intermediate step of a PI controller. This method is simple to implement and low in cost, and is currently the method used by the vast majority of MPPT algorithms.

[0041] In practical use, when the voltage to be converted by the voltage converter in this embodiment is provided by a photovoltaic module or a low-voltage DC power supply, the switching transistor S1 is an MSOFET, i.e., a field-effect transistor, and diodes D1-D6 can be fast recovery diodes, Schottky diodes, or silicon carbide diodes. Capacitors C1, C2, and C3 are thin film capacitors or low ESR capacitors.

[0042] In practical applications, when the voltage converter in this embodiment is used in high-frequency, high-efficiency, or high-temperature environments, the switching transistor S1 can be a power switching device such as a GaN HEMT, SiC MOSFET, or IGBT; diodes D5 and D6 can be SiC diodes to reduce the reverse recovery effect.

[0043] In practical applications, coupled inductors L2 and L3 can be wound using the same magnetic core to form a coupled inductor, or they can be integrated into a magnetic component structure. The voltage gain can be adjusted by changing the turns ratio N of coupled inductors L2 and L3. Leakage inductance, winding arrangement, and core air gap can be optimized according to requirements to reduce voltage spikes and improve current stress. Furthermore, if a further power rating is required, the energy storage inductor L1 can be integrated with coupled inductors L2 and L3 into the same magnetic component, or multiple magnetic cores can be connected in parallel.

[0044] Specifically, in this embodiment, in Figure 1 In the output capacitor branch 2, there are output capacitors CO1 and CO2. One end of output capacitor CO1 is the first terminal of output capacitor branch 2, and the other end of output capacitor CO1 is electrically connected to one end of output capacitor CO2, which is the middle node of output capacitor branch 2. The other end of output capacitor CO2 is the second terminal of output capacitor branch 2.

[0045] Furthermore, in this embodiment, the intermediate node of the output capacitor branch 2 is also electrically connected to the output terminal of the fourth unidirectional conductive unit 103 for grounding; this can help reduce the common-mode interference path, improve EMI or EMC characteristics, and make the voltage converter in this embodiment suitable for the subsequent bipolar DC bus or the front stage of the inverter.

[0046] right Figure 1 The working process of the circuit shown is analyzed as follows: Figure 1 The circuit shown includes four modes in actual operation, namely mode one, mode two, mode three and mode four.

[0047] In Mode 1, during the initial turn-on phase, switch S1 freewheels. In this mode, switch S1, diodes D2, D4, and D5 are conducting, while diodes D1, D3, and D6 are cut off. Since the current in coupling inductor L3 in the previous turn-off main mode flows to the right through capacitor C3 and diode D5, the inductor current cannot change abruptly. Therefore, even though switch S1 is already conducting, the right-side branches of diodes D4 and D5 remain conducting, and the current in coupling inductor L3 freewheels in its original direction and gradually decreases. Simultaneously, the input source forms an energy storage circuit through energy storage inductor L1, diode D2, and switch S1, and energy storage inductor L1 begins to charge. Capacitor C1 and coupling inductor L2 also form an energy storage circuit through switch S1. As the rightward current in coupling inductor L3 decays, the current in diodes D4 and D5 drops to zero and they naturally turn off, ending Mode 1 and entering Mode 2. The current flow direction in Mode 1 is as follows: Figure 2 As shown.

[0048] Mode 2: Switch S1, diodes D2, D3, and D6 are turned on, while diodes D1, D4, and D5 are turned off. The input source stores energy in inductor L1 via inductor L1, diode D2, and switch S1. The input source and capacitor C1 work together to store energy in coupling inductor L2. Since the switching node potential decreases after switch S1 is turned on, and the voltage of capacitor C3 cannot change abruptly, diode D3 conducts forward, forming a local loop consisting of diode D3, capacitor C3, coupling inductor L3, and switch S1, causing the current in coupling inductor L3 to build up to the left. Simultaneously, coupling inductor L3 and capacitor C2 charge the output capacitor Co2 to the lower side through diode D6. This mode is the main stage where the negative output capacitor gains energy. The current flow in Mode 2 is as follows: Figure 3 As shown.

[0049] Mode 3: Switch S1 is off. When S1 transitions from on to off, the state of the left-side diodes can switch rapidly: diode D2 is off, and diode D1 is on. On the right side, because the current of the energy storage inductor L3 cannot change abruptly, L3 maintains the leftward current from the previous mode. At this time, diode D6 is off, diode D4 is on, and diode D3 can still conduct briefly. The rightward current of energy storage inductor L2 and the leftward current of energy storage inductor L3 commutate through the branch of diode D3, capacitor C3, and capacitor C2, charging or reconstructing the voltage of capacitor C2. As the commutation process proceeds, the leftward current of energy storage inductor L3 gradually decreases and then increases in the opposite direction. When the current in the diode D3 branch drops to zero, diode D3 naturally turns off, diode D5 is forward biased and conducts, and the circuit enters Mode 4. The current flow in Mode 3 is as follows: Figure 4 As shown.

[0050] Mode 4: Switch S1 remains off. In this mode, diodes D1, D4, and D5 are on, while switches S1, D2, D3, and D6 are off. Energy is released from storage inductors L1, L2, and L3. The voltages across capacitors C1, C2, and C3 are superimposed with the induced voltage of the coupling inductors. Current flows to the right through coupling inductor L3, transferring energy to the output capacitor Co1 and the load via capacitor C3 and diode D5. Simultaneously, diode D4 provides clamping and a return path for the lower end of capacitor C2, allowing C2 to participate in the subsequent boost process. The current flow in Mode 4 is as follows: Figure 5 As shown.

[0051] When the switching transistor S1 turns on again in the next cycle, the current in the right-side energy storage inductor L3 cannot change abruptly, and diodes D4 and D5 will continue to flow freely, allowing the circuit to re-enter mode one. Additionally... Figures 2-5 The arrows in the diagram indicate the direction of current flow.

[0052] The voltage gain is analyzed based on the above four modalities, as follows: To simplify the analysis, we assume that all power devices are ideal devices, the inductor current operates in continuous conduction mode, and the voltage of each capacitor remains approximately constant within one switching cycle.

[0053] Since Mode 1 and Mode 3 are freewheeling and commutation modes during the switching process, and their duration is relatively short, the voltage gain derivation mainly considers the main switching mode and the main switching mode.

[0054] Applying the volt-second balance principle to the energy storage inductor L1, during the conduction period of the switching transistor S1, the voltage across the energy storage inductor L1 is: ; V in The voltage to be converted is the external input. During the turn-off period of switch S1, the energy storage inductor L1 forms an energy transfer circuit with diode D1 and capacitor C1, and its voltage amplitude during the turn-off phase is: ; From the volt-second balance, we can obtain: ; in It is the voltage of the energy storage inductor L1 when the switching transistor S1 is turned on. 1 is the voltage of the energy storage inductor L1 when the switching transistor S1 is turned off; D is the time during which the switching transistor S1 is turned on within a switching cycle; and 1-D is the time during which the switching transistor S1 is turned off within a switching cycle.

[0055] From the above, we can deduce that the voltage across capacitor C1 is: .

[0056] Based on mode two, we can obtain: ; Based on mode four, we can obtain: ; According to the above expression, the two voltages across output capacitor CO1 and output capacitor CO2 are equal, as follows: .

[0057] From the above expression, the gain expression of the voltage converter in this embodiment can be obtained as follows: .

[0058] In the above formula, This is the voltage across the coupling inductor L2 when the switching transistor S1 is turned on. V is the voltage across the coupling inductor L2 when the switch S2 is turned off; Co1 V is the voltage across the output capacitor CO1. Co2 V is the voltage across the output capacitor CO2. C2 V is the voltage across capacitor C2. C3 This is the voltage across capacitor C3.

[0059] From the analysis of the above four modes, we can conclude that: The voltage converter in this embodiment achieves high gain. Through the energy storage and release of the energy storage inductors L1, L2, and L3, and the voltage superposition of capacitors C1, C2, and C3, the total output voltage gain of the voltage converter reaches 2. (1+N) / (1-D) 2 It can achieve high output voltage and high gain by reasonably setting the turns ratio with a small duty cycle.

[0060] The voltage converter in this embodiment is simple to control. The circuit only has one main switch S1. The driving and control method is simpler than that of a multi-switch interleaved structure. There is no power unevenness problem caused by the timing deviation of the driving of multiple main switches. The voltage converter in this embodiment has a small number of components. Compared with multi-phase, multi-stage cascaded or multi-switch high-gain schemes, the voltage converter in this embodiment can achieve high-gain output with a main switch, three inductor / coupling windings and a small number of diodes and capacitors, which is beneficial to improve power density and has fewer components overall.

[0061] In this embodiment, the output capacitors CO1 and CO2 of the voltage converter are connected in series to form positive and negative buses. Mode 2 mainly transfers energy to the output capacitor CO2, and Mode 4 mainly transfers energy to the output capacitor CO1. Ideally, the voltages on the output capacitors CO1 and CO2 are the same, which helps to reduce the voltage imbalance problem of the positive and negative buses.

[0062] In this embodiment, the intermediate node of the output capacitor branch 2 of the voltage converter is directly grounded to the negative terminal of the input power supply, which helps to reduce the common-mode interference path, improve EMI or EMC characteristics, and is suitable for subsequent bipolar DC buses or inverter front-ends.

[0063] In this embodiment, the voltage converter exhibits natural current decay in diodes D4 and D5 in mode two, and completes commutation in the diode D3 branch in mode three. The relevant diodes can naturally turn off when the current drops to near zero, thereby reducing reverse recovery losses and switching shocks.

[0064] Example 2 like Figure 6 As shown, this embodiment provides a dual-input voltage converter, including a boost unit 1 in two bipolar output voltage converters as described in Embodiment 1, and an output capacitor branch 2 in a bipolar output voltage converter as described in Embodiment 1. The input terminal of the fifth unidirectional conductive unit 11 of the two boost units 1 is electrically connected to the first terminal of the output capacitor branch 2, and the input terminal of the sixth unidirectional conductive unit 12 of the two boost units 1 is electrically connected to the second terminal of the output capacitor branch 2.

[0065] for Figure 6 The circuit shown can convert the voltage of two power inputs and only requires one output capacitor branch 2, which simplifies the structure of the two-input voltage converter.

[0066] Specifically, in this embodiment, one end of capacitor C1 of one boost unit 1 is used to be electrically connected to the photovoltaic power generation unit, and one end of capacitor C1 of the other boost unit is used to be electrically connected to the fuel cell unit.

[0067] Based on the above description, those skilled in the art can make various changes and modifications without departing from the technical concept of this invention. The technical scope of this invention is not limited to the contents of the specification, but must be determined according to the scope of the claims.

Claims

1. A bipolar output voltage converter, characterized in that, Includes a boost unit and an output capacitor branch; The boost unit includes an energy storage inductor L1, a coupling inductor L2, a coupling inductor L3, a boost branch, a switching transistor S1, a fifth unidirectional conductive unit, and a sixth unidirectional conductive unit; the coupling inductor L2 is coupled to the coupling inductor L3; The boost circuit includes a capacitor C1, a first unidirectional conductive unit, a second unidirectional conductive unit, a third unidirectional conductive unit, a capacitor C3, a capacitor C2, and a fourth unidirectional conductive unit connected in series. The input terminal of the first unidirectional conductive unit is electrically connected to the input terminal of the second unidirectional conductive unit, and the output terminal of the second unidirectional conductive unit is electrically connected to the input terminal of the third unidirectional conductive unit. The end of capacitor C2 that is not electrically connected to capacitor C3 is electrically connected to the input terminal of the fourth unidirectional conductive unit. The output terminal of the fourth unidirectional conductive unit is electrically connected to the output terminal of the switching transistor S1. The two ends of the inductor L1 are electrically connected to one end of the capacitor C1 and the input end of the first unidirectional conductive unit, respectively, and the other end of the capacitor C1 is electrically connected to the output end of the first unidirectional conductive unit. The same-name terminal of the coupling inductor L2 is electrically connected to the output terminal of the first unidirectional conductive unit; the opposite-name terminal of the inductor L2 is electrically connected to the output terminal of the second unidirectional conductive unit, the input terminal of the switch S1, and the same-name terminal of the coupling inductor L3, respectively; the coupling inductor L3 and the capacitor C3 are electrically connected to one end of the capacitor C2. The input terminal of the fifth unidirectional conductive unit is electrically connected to the output terminal of the third unidirectional conductive unit, and the output terminal of the fifth unidirectional conductive unit is electrically connected to the first terminal of the output capacitor branch. The input terminal of the sixth unidirectional conductive unit is electrically connected to the second terminal of the output capacitor branch, and the output terminal of the sixth unidirectional conductive unit is electrically connected to the input terminal of the fourth unidirectional conductive unit.

2. The bipolar output voltage converter according to claim 1, characterized in that, The first unidirectional conductive unit, the second unidirectional conductive unit, the third unidirectional conductive unit, the fourth unidirectional conductive unit, the fifth unidirectional conductive unit, and the sixth unidirectional conductive unit all conduct when the voltage difference between their input and output terminals is greater than the threshold voltage. The first, second, third, fourth, fifth, and sixth unidirectional conductive units are all turned off when the voltage difference between their input and output terminals is less than the threshold voltage.

3. A bipolar output voltage converter according to claim 2, characterized in that, The first unidirectional conductive unit, the second unidirectional conductive unit, the third unidirectional conductive unit, the fourth unidirectional conductive unit, the fifth unidirectional conductive unit, and the sixth unidirectional conductive unit each include a diode, with the anode of the diode serving as the input terminal of the unidirectional conductive unit and the cathode of the diode serving as the output terminal of the unidirectional conductive unit.

4. A bipolar output voltage converter according to claim 3, characterized in that, The diode is a fast recovery diode, a Schottky diode, or a silicon carbide diode.

5. A bipolar output voltage converter according to claim 1, characterized in that, The switching transistor is an NMOS transistor. The drain of the NMOS transistor is the input terminal of the switching transistor S1, the source of the NMOS transistor is the output terminal of the switching transistor S1, and the gate of the NMOS transistor is the control terminal of the switching transistor S1, used to input PWM control signals.

6. A bipolar output voltage converter according to claim 1, characterized in that, The output capacitor branch includes an output capacitor CO1 and an output capacitor CO2; one end of the output capacitor CO1 is the first endpoint of the output capacitor branch, the other end of the output capacitor CO1 is electrically connected to one end of the output capacitor CO2, which is the intermediate node of the output capacitor branch, and the other end of the output capacitor CO2 is the second endpoint of the output capacitor branch.

7. A bipolar output voltage converter according to claim 6, characterized in that, The intermediate node of the output capacitor branch is also electrically connected to the output terminal of the fourth unidirectional conductive unit for grounding.

8. A bipolar output voltage converter according to claim 1, characterized in that, The capacitors C1, C2, and C3 are thin-film capacitors.

9. A dual-input voltage converter, characterized in that, It includes a boost unit in a bipolar output voltage converter according to any one of claims 1-8 and an output capacitor branch in a bipolar output voltage converter according to any one of claims 1-8; The input terminal of the fifth unidirectional conductive unit of the two boost units is electrically connected to the first terminal of the output capacitor branch, and the input terminal of the sixth unidirectional conductive unit of the two boost units is electrically connected to the second terminal of the output capacitor branch.

10. A dual-input voltage converter according to claim 9, characterized in that, One end of capacitor C1 in one boost unit is used to connect to the photovoltaic power generation unit, and one end of capacitor C1 in another boost unit is used to connect to the fuel cell unit.

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