A differential-fed wideband circularly polarized substrate integrated waveguide back cavity antenna array

Abstract

The application discloses a differential feed wideband circularly polarized substrate integrated waveguide back cavity antenna array, a group of arc microstrip line stubs are introduced, two modes in the cavity can be excited, the resonant frequencies of the two different modes are fused, and thus the working bandwidth of the circularly polarized antenna is widened. Through the arc microstrip line excitation double-mode resonance, in combination with an improved ring-shaped coupler feed network, at least 32.6% impedance bandwidth, at least 13.8% axial ratio bandwidth, at least 41.75% gain bandwidth and at least 10.56dBi peak gain are realized. The array profile height is only 0.19-0.21lambda0, the cross-polarization level is lower than-25dB, and the array is suitable for 5G communication and satellite navigation systems.

Description

[0001] This application claims priority to Chinese invention application No. 2025105509618, filed on April 28, 2025, with the State Intellectual Property Office of China, entitled "A Differential-Fed Broadband Circularly Polarized Substrate Integrated Waveguide Back-Cavity Antenna Array", the entire contents of which are incorporated herein by reference. Technical Field

[0002] This invention relates to the field of microwave antenna technology, specifically to a broadband circularly polarized substrate integrated waveguide (SIW) cavity antenna array based on differential feeding and its design method, applicable to, but not limited to, 5G communication, satellite navigation and millimeter-wave radar system applications. Background Technology

[0003] Circularly polarized antennas offer advantages such as reduced polarization mismatch loss and resistance to multipath fading, making them highly useful in many systems. With increasing operating frequencies and miniaturization of integrated systems, circularly polarized antennas, characterized by wide bandwidth, multi-bandwidth, high gain, low profile, and low cross-polarization, have even broader application prospects.

[0004] Cavity-backed antenna structures can improve the performance of circularly polarized antennas. However, the large thickness of the metal cavity and its high manufacturing cost pose challenges to its integration into compact circuits. To address these limitations, the integration of substrate-integrated waveguide (SIW) circularly polarized cavity-backed antennas offers a more feasible approach.

[0005] Substrate-integrated waveguide (DIWB) structures are renowned for their ability to seamlessly integrate with planar circuits, their compact design, and high Q values. When integrated WBB structures are integrated with circularly polarized antennas instead of metallic cavities, they can exhibit a variety of highly attractive properties. Some prior art utilizes DWB cavity structures to design a broadband circularly polarized antenna based on multimode integration, achieving an impedance bandwidth of 38% and an axial ratio (AR) bandwidth of 24.9%. Another proposed antenna and array achieves dual-frequency circularly polarized radiation at 28 and 38 GHz, with axial ratio bandwidths of 2.8% and 2.6%, respectively. Another DWB cavity antenna achieves high-order mode resonant frequencies by etching a pair of angled strip parasitic patches on top, achieving a peak gain of 26.1 dBic. Furthermore, prior art has developed a circularly polarized substrate-integrated waveguide cavity antenna and array with an extremely low profile height of only 0.035λ0 (λ0 being the free-space wavelength corresponding to the center frequency).

[0006] On the other hand, differential circuits offer numerous advantages in microwave devices, including mitigating higher harmonics, reducing cross-coupling interference, and improving noise immunity. When incorporating differential circuits into RF front-end designs, a balun is typically required to achieve 180° phase shift. However, this component introduces additional losses and increases circuit complexity. Differential antennas eliminate the need for a balun and can be directly integrated with the front-end circuitry. When combined with differential circuitry, broadband circularly polarized substrate-integrated waveguide cavity antennas can further reduce the cross-polarization ratio and improve antenna resolution.

[0007] Existing technology WO2020009834A1 introduces an ultra-widebandwidth, frequency-independent circularly polarized array antenna. This antenna achieves tight coupling between elements by using circuit elements (such as capacitors or resistors) to couple adjacent elements. This tight coupling design helps reduce grating lobes and improve antenna performance. However, this feeding technique also has some drawbacks. For example, the feeding network is complex, requiring precise design and manufacturing processes to ensure antenna performance. Furthermore, this feeding technique may require a large number of circuit elements, increasing the cost and size of the antenna.

[0008] Existing technologies have proposed various differential circularly polarized antenna structures with substrate-integrated waveguide cavity-backed antennas. One design achieves an effective circular polarization bandwidth of 52% and a profile height as low as 0.096λ0. Another design proposes a quad-polarization agile substrate-integrated waveguide cavity-backed patch antenna and array, where the axial ratio bandwidth of circularly polarized radiation exceeds 9.8%. Furthermore, a high-gain differentially fed substrate-integrated waveguide cavity-backed antenna for the millimeter-wave band has been developed and arrayed, achieving a peak gain of 20.5 dBic. To minimize cross-polarization levels, some structures employ differential feed networks, achieving circularly polarized radiation while maintaining cross-polarization levels below -20 dB.

[0009] Existing literature on broadband circularly polarized substrate integrated waveguide cavity-backed antennas with differential feeding methods highlights many attractive characteristics, such as broadband, multi-polarization, high gain, low profile, and low cross-polarization. However, research results show that few of the elements of these broadband circularly polarized substrate integrated waveguide cavity-backed antennas are suitable for array applications, and while array structures exist, their feeding networks are extremely complex. Therefore, research on substrate integrated waveguide cavity-backed antenna arrays with broadband circularly polarized radiation and simplified differential feeding mechanisms is crucial for their application in compact integrated circuits.

[0010] Circularly polarized antennas are widely used in wireless communication due to their advantages of resistance to multipath interference and low polarization mismatch loss. While traditional metallic cavity antennas offer stable performance, they suffer from high profile and low integration density. Substrate integrated waveguide (SIW) technology simulates a metallic cavity through metallized vias, significantly reducing the profile height. However, existing SIW circularly polarized antennas still face the following challenges:

[0011] 1) Bandwidth limitation: For example, the impedance bandwidth of some existing 4×4 arrays is only 3.17%, and the axial bandwidth is less than 1%;

[0012] 2) Complex power supply: Some existing technologies use differential power supply but require an external balun, resulting in additional losses;

[0013] 3) High cross-polarization: In some existing technologies, the cross-polarization level is only below -20dB, which is difficult to meet the requirements of high resolution.

[0014] To address these issues, the art requires improved differentially fed broadband circularly polarized substrate integrated waveguide cavity antenna arrays and their design techniques to overcome the shortcomings of existing technologies and achieve more beneficial technical effects, including simplifying the feed network, reducing antenna cost and size, and simultaneously improving antenna bandwidth and performance. By employing novel feed technologies, construction methods, and design approaches, miniaturization, integration, and high performance of antennas can be achieved, meeting the stringent requirements of modern communication systems.

[0015] The information included in this background section of the specification of this invention, including any references cited in this invention and any descriptions or discussions thereof, is included for technical reference purposes only and is not intended to limit the scope of the invention. Summary of the Invention

[0016] In view of the foregoing and other concepts, the present invention is proposed to solve the above and other technical problems through techniques and inventive concepts such as dual-mode resonant excitation and simplified differential feed network.

[0017] One of the main concepts of this invention is to increase bandwidth, reduce the design complexity of the feed network, and lower the cross-polarization ratio by designing a novel broadband substrate integrated waveguide cavity antenna array and its feeding technology. One advantage of the feed network in this invention is that, by employing a simple and improved microstrip ring coupler, 180-degree power distribution and phase shift can be generated within the required operating frequency band of the antenna. Existing designs generally use broadband baluns, which suffer from significant losses at high frequencies due to the need to weld balancing resistors to the balun structure. This design eliminates the need for balancing resistors, making it more suitable for feeding in 5G / millimeter-wave environments.

[0018] To overcome the challenges in the research of differentially fed substrate integrated waveguide cavity antenna arrays and simplify the differential feeding method, according to one aspect of the present invention, a broadband substrate integrated waveguide cavity antenna array is proposed. This array employs a 2×2 element layout and utilizes an improved differential ring coupler as the feeding network. Adding a pair of arc-shaped microstrip lines facilitates broadband circularly polarized radiation. Compared with existing technologies, the proposed array features wide bandwidth, high gain, simple feeding method, compact design, and low profile, making it particularly suitable for applications in compact integrated circuits.

[0019] Furthermore, one of the core aspects of this invention lies in proposing a broadband, low-profile, low-cross-polarization differential-fed SIW cavity antenna array, the technical features of which are as follows:

[0020] 1) Dual-mode resonant design: A pair of arc-shaped microstrip lines are embedded in the second bottom substrate, and the substrate-integrated waveguide cavity is synchronously excited by differential feeding. and Orthogonal mode; by adjusting the length (l1 = 6.47 mm) and spacing (d1 = 5 mm) of the arc-shaped microstrip line, the resonant frequencies of the two modes partially overlap at 19.04 GHz and 22 GHz, thus achieving broadband circularly polarized radiation.

[0021] 2) Simplified differential feed network: An improved ring coupler is used to replace the traditional balun and is directly integrated into substrate III to reduce signal loss; the feed probe is connected through a copper pad (0.5 mm in diameter) to ensure impedance matching and mechanical stability.

[0022] 3) Low profile optimization: using the formula:

[0023]

[0024] To optimize the substrate height h1+h2, the profile height is reduced to 0.21λ0. Where, v c ε is the speed of light in free space, f0 is the resonant frequency of the substrate-integrated waveguide back cavity, h1 is the height of the first top substrate, h2 is the height of the second bottom substrate, C1 is the diameter of the substrate-integrated waveguide back cavity, and ε is the velocity of light in free space. r denoted as , where is the relative permittivity of the first top substrate and the second bottom substrate.

[0025] 4) Parametric performance control: The ratio of metallized via diameter to spacing is set to 1:1 to suppress high-order mode interference; the matching via diameter is increased from 1.1mm to 1.6mm to broaden the impedance bandwidth to 32.6%.

[0026] More specifically, to address the aforementioned and other problems, according to one aspect of the present invention, the present invention aims to provide a differentially fed broadband circularly polarized substrate integrated waveguide back cavity antenna array, characterized in that it comprises: an array structure composed of 2×2 antenna elements, wherein each antenna element includes a substrate integrated waveguide back cavity, the substrate integrated waveguide back cavity being constructed by stacking a first top substrate and a second bottom substrate, including an array of metallized vias penetrating the first top substrate and the second bottom substrate for forming the electromagnetic boundary of the substrate integrated waveguide back cavity; a pair of arc-shaped microstrip lines located in the second bottom substrate, which synchronously excite the substrate integrated waveguide back cavity through differential feeding. and Orthogonal mode; differential feed network, including four improved ring couplers integrated on a third substrate, for providing differential signals with a phase difference in the range of 175°-185° to the antenna element.

[0027] According to one or more embodiments of the present invention, the length, width and spacing of the pair of arcuate microstrip lines are adjusted such that the operating bandwidths of the first and second resonant frequencies partially overlap.

[0028] According to one or more embodiments of the present invention, the resonant frequency f of the first resonant frequency point c1 =19.04GHz, the resonant frequency of the second resonant frequency point is f c2 =22GHz.

[0029] According to one or more embodiments of the present invention, the diameter and spacing of the metallized via array are adjusted such that the resonant frequency of the substrate integrated waveguide back cavity matches the target frequency band.

[0030] According to one or more embodiments of the present invention, the improved ring coupler is connected to a feed probe via a copper pad on the third substrate, and the feed probe connects the arcuate microstrip line to the differential feed network.

[0031] According to one or more embodiments of the present invention, the sum of the height h1 of the first top substrate and the height h2 of the second bottom substrate, h1+h2, is optimized by the following formula:

[0032]

[0033] Among them, v c f0 is the speed of light in free space, h1 is the height of the first top-layer substrate, h2 is the height of the second bottom-layer substrate, C1 is the diameter of the substrate-integrated waveguide back cavity, and ε is the resonant frequency of the back cavity. r is the relative permittivity of the first top substrate and the second bottom substrate.

[0034] According to one or more embodiments of the present invention, the ratio of the height h1 of the first top substrate to the height h2 of the second bottom substrate is 1:1.

[0035] According to one or more embodiments of the present invention, the improved ring coupler is designed to replace the conventional balun.

[0036] According to one or more embodiments of the present invention, the cross-polarization level of the antenna element is less than -25dB.

[0037] According to one or more embodiments of the present invention, the profile height of the antenna array is 0.19-0.21λ0, where λ0 is the free space wavelength corresponding to the center frequency of the antenna array.

[0038] According to one or more embodiments of the present invention, the ratio of the through-hole spacing to the diameter of the metallized through-hole array is 1:1.

[0039] According to one or more embodiments of the present invention, the antenna array has an impedance bandwidth of at least 32.6% (17.28-24.00 GHz), a 3-dB axial ratio bandwidth of at least 13.8% (17.05-19.57 GHz), and a 3-dB gain bandwidth of at least 41.75% (15.39-23.51 GHz).

[0040] According to one or more embodiments of the present invention, the improved ring coupler is directly integrated on a third substrate.

[0041] According to one or more embodiments of the present invention, each antenna element has a radiation hole on the top layer of the substrate integrated waveguide back cavity and two matching holes on the bottom layer; and each antenna element has a central hole at its geometric center.

[0042] According to one or more embodiments of the present invention, the phase difference is 180°.

[0043] According to one or more embodiments of the present invention, the diameter of the substrate integrated waveguide back cavity is 10 mm, and the thickness of the first top substrate and the second bottom substrate is 1.524 mm.

[0044] According to one or more embodiments of the present invention, the spacing between the antenna elements is 11.1 mm, corresponding to 0.7λ0 of the center frequency of 19.04 GHz, where λ0 is the free space wavelength.

[0045] According to one or more embodiments of the present invention, the thickness of the third substrate of the improved annular coupler is 0.254 mm, and the surface of the substrate is etched with copper pads with a diameter of 0.5 mm to enhance connection reliability.

[0046] According to one or more embodiments of the present invention, the antenna array excites the cavity via differential feeding. and Two modes are used to achieve broadband circularly polarized radiation.

[0047] According to one or more embodiments of the present invention, the first top substrate and the second bottom substrate are both Rogers RO4350 substrates with a relative permittivity of 3.66 and a loss tangent of 0.004.

[0048] To address the aforementioned and other problems, according to another aspect of the present invention, a design method for an antenna array is also provided, comprising the following steps: optimizing the length, width, and spacing of the arc-shaped microstrip line based on ANSYS HFSS simulation to achieve dual-mode resonant frequency overlap; adjusting the diameter and spacing of the metallized vias to match the resonant frequency of the substrate-integrated waveguide back cavity with the target frequency band; designing an improved ring coupler, connecting the feed probe via copper pads to enhance connection reliability, and fixing the multilayer substrate with plastic screws; measuring the reflection coefficient using a vector network analyzer to verify the impedance bandwidth and axial ratio bandwidth; and testing the radiation pattern and cross-polarization level in a microwave anechoic chamber to ensure peak gain and sidelobe suppression performance.

[0049] According to one or more embodiments of the present invention, the antenna array adopts a 2×2 element layout with an element spacing of 11.1 mm, which is approximately equal to 0.7λ0 of the center frequency of 19.04 GHz.

[0050] According to one or more embodiments of the present invention, the differential feed network consists of four improved ring couplers etched on the bottom surface of substrate III, which has a thickness of 0.254 mm.

[0051] According to one or more embodiments of the present invention, the arcuate microstrip line has a length l1 of 6.47 mm, a width w1 of 0.3 mm, and a spacing d1 of 5 mm.

[0052] According to one or more embodiments of the present invention, the differential-fed broadband circularly polarized substrate integrated waveguide cavity antenna array is suitable for feeding 5G networks / millimeter waves, and is suitable for 5G communication networks and satellite navigation systems, etc.

[0053] According to one or more embodiments of the present invention, the antenna array of the present invention is suitable for communication systems in microwave and millimeter-wave bands, and features wide bandwidth, high gain, low profile and low cross-polarization.

[0054] According to one or more embodiments of the present invention, the antenna array of the present invention reduces the complexity of the feeding network, reduces the cross-polarization level of the antenna, and improves the performance of the antenna by simplifying the differential feeding mechanism.

[0055] Further embodiments of the present invention can also achieve other beneficial technical effects not listed one by one. These other beneficial technical effects may be partially described below and can be expected and understood by those skilled in the art after reading the present invention. Attached Figure Description

[0056] The above-described features and advantages, as well as other features and advantages, and the ways in which they are implemented, of the embodiments of the present invention will become more apparent and better understood by referring to the following description in conjunction with the accompanying drawings, in which:

[0057] Figure 1(a)-1(b) An antenna element according to an embodiment of the present invention is schematically shown, wherein FIG1(a) schematically shows the three-dimensional structure of the antenna element, and FIG1(b) schematically shows a top view of the antenna element, wherein the radiation aperture (14 mm), metallized via (1.53 mm spacing) and arcuate microstrip line (6.47 mm × 0.3 mm) of the antenna element are labeled.

[0058] Figure 2 The diagram schematically illustrates the electric field distribution of a substrate-integrated waveguide cavity obtained from simulation according to an embodiment of the present invention at two resonant points, 19.04 GHz and 22 GHz, demonstrating the electric field distribution at the two resonant points (f... c1 =19.04GHz and f c2 =22GHz) The horizontal components of the electric field in the substrate integrated waveguide cavity during different time intervals are (a)t=0, (b)t=T / 4, (c)t=T / 2, (d)t=3T / 4.

[0059] Figure 3 The illustration schematically shows the simulation results obtained at two resonant points (f) according to an embodiment of the present invention. c1 =19.04GHz and f c2 =22GHz), electric field distribution within the cavity (a) Pattern in f c1 =19.04GHz and (b) Pattern in f c2 =22GHz.

[0060] Figure 4The schematically illustrates the (a) differential S-parameters |Sdd11| and (b) axial ratio bandwidth AR and left-hand circular polarization gain LHCP Gain of the proposed antenna element under different arcuate microstrip line lengths l1 according to an embodiment of the present invention (selecting a plane defined by phi = 0° and theta = 0°).

[0061] Figure 5 The schematically illustrates the (a) differential S-parameters |Sdd11| and (b) axial ratio bandwidth AR and left-hand circular polarization gain LHCP Gain of the proposed antenna element under different arcuate microstrip line widths w1 according to an embodiment of the present invention (selecting a plane defined by phi = 0° and theta = 0°).

[0062] Figure 6 The illustration shows the (a) differential S-parameters |Sdd11| and (b) axial ratio bandwidth AR and left-hand circular polarization gain LHCP Gain of the proposed antenna element under different substrate heights h1 and h2 according to an embodiment of the present invention (selecting a plane defined by phi = 0° and theta = 0°).

[0063] Figure 7 The diagram schematically shows the radiation patterns of the E-plane and H-plane at 19.04 GHz according to an embodiment of the present invention, with the main lobe direction and cross-polarization level marked. The antenna element dimensions are l1 = 6.47 mm, w1 = 0.3 mm, and h1:h2 = 1:1.

[0064] Figures 8(a)-(c) schematically show an antenna array with a feed network according to an embodiment of the present invention, wherein Figure 8(a) is a three-dimensional view of the antenna array, Figure 8(b) is a bottom view of the antenna array, and Figure 8(c) is a side view of the antenna array.

[0065] Figure 9 The schematic illustration shows the (a) differential S-parameters |Sdd11| and (b) axial ratio bandwidth AR and left-hand circular polarization gain LHCP Gain obtained from the simulation of an antenna array with and without a feed network according to an embodiment of the present invention (selecting a plane defined by phi = 0° and theta = 0°).

[0066] Figure 10(a) schematically shows the components and assembly of an antenna array according to an embodiment of the present invention, and Figure 10(b) schematically shows the measuring device of the antenna array.

[0067] Figure 11 The illustration schematically shows the simulation and measurement reflection coefficient of port 1 of the proposed antenna array according to an embodiment of the present invention.

[0068] Figure 12The diagram schematically illustrates the simulated and measured axial ratio bandwidth AR and left-hand circular polarization gain LHCP Gain of the proposed antenna array according to an embodiment of the present invention (selected as a plane defined by phi = 0° and theta = 0°).

[0069] Figures 13(a)-(b) schematically show the normalized radiation pattern of the proposed antenna array at a frequency of 19.04 GHz in the simulation and measurement results of the E-plane (Figure 13(a)) and H-plane (Figure 13(b)) according to an embodiment of the present invention. Detailed Implementation

[0070] In the following description of specific embodiments, taken in conjunction with the accompanying drawings, details of one or more embodiments of the invention will be set forth. Other features, objects, and advantages of the invention will become apparent from these descriptions, drawings, and claims.

[0071] It should be understood that the illustrated and described embodiments are not limited in application to the details of the configurations set forth in the following description or illustrated in the accompanying drawings. The illustrated examples may be other embodiments and can be implemented or performed in various ways. The examples are provided by way of explanation rather than limitation of the disclosed embodiments. Indeed, it will be apparent to those skilled in the art that various modifications and variations can be made to the embodiments of the invention without departing from the scope or spirit of the disclosure. For example, features illustrated or described as part of one embodiment may be used with another first embodiment to still produce another embodiment. Therefore, this disclosure covers such modifications and variations that fall within the scope of the appended claims and their equivalents.

[0072] Similarly, it is understood that the phrases and terms used in this invention are for descriptive purposes and should not be considered restrictive. The use of "comprising," "including," or "having" and variations thereof in this invention is intended to include, in an open-ended manner, the items listed thereafter, their equivalents, and any additional items.

[0073] The present invention will now be described in more detail with reference to several embodiments thereof.

[0074] 1. Antenna Unit

[0075] 1.1 Antenna Unit Design and Manufacturing

[0076] Material selection: Two-layer Rogers RO4350 substrate (ε) r =3.66, loss tangent tanδ=0.004), single layer thickness 1.524mm.

[0077] Structure fabrication: an arc-shaped microstrip line with a length l1 = 6.47 mm, a width w1 = 0.3 mm, and a spacing d1 = 5 mm is etched on the bottom substrate; metallized vias (diameter 1.53 mm, spacing 1.53 mm) are drilled to form the SIW cavity boundary; a radiation aperture (14 mm) is opened on the top layer, and a matching aperture (1.6 mm) is opened on the bottom layer.

[0078] Figure 1 illustrates the proposed substrate-integrated waveguide-back cavity antenna element design. This design utilizes two Rogers RO4350 substrates with a relative permittivity of ε0. r =3.66, loss tangent is 0.004. The top layer is substrate I with thickness h1, and the bottom layer is substrate II with thickness h2. To generate circularly polarized radiation within the operating frequency band, a pair of arc-shaped microstrip lines with length l1 and width w1 are inserted into substrate II. The spacing between the arc-shaped microstrip lines is d1. In addition, two diameters d s The feed probe connects the curved microstrip line to the feed network to enable differential feeding from below. The geometric center of the antenna element has a diameter of d. h Holes are used to ensure good impedance matching. Metallized vias with a diameter of d and a spacing of p penetrate all layers of the antenna element to form a substrate-integrated waveguide cavity with a diameter of C1. To allow the cavity to radiate fully, a hole with a diameter of d is etched on the top layer. R Holes with a diameter of d are formed at the bottom layer. M Matching holes are used to adjust the impedance of the feed port. For ease of subsequent analysis, a set of differential ports are labeled as port + and port -. Detailed parameters of the antenna element can be found in Table I.

[0079] Table I lists the dimensions of the antenna elements (unit: millimeters).

[0080] parameter <![CDATA[D1]]> <![CDATA[C1]]> <![CDATA[h1]]> <![CDATA[h2]]> <![CDATA[l1]]> <![CDATA[w1]]> p Specific value 14 10 1.524 1.524 6.47 0.3 1.53 parameter d <![CDATA[d1]]> <![CDATA[d R ]]> <![CDATA[d h ]]> <![CDATA[d s ]]> <![CDATA[d M ]]> Specific value 1 5 4.2 1 0.5 1.1

[0081] 1.2 Antenna Unit Working Principle

[0082] To explain the working principle of the proposed antenna element Figure 2 This shows the two resonant points (f) c1 =19.04GHz and f c2 The horizontal components of the electric field within the substrate-integrated waveguide cavity at different time intervals corresponding to 22 GHz. The analysis assumes a 0° phase for the port + input voltage and a 180° phase for the port - input voltage. Data were obtained through simulation calculations using ANSYS HFSS software.

[0083] At a frequency of 19.04 GHz, when t = 0, the electric field distribution on the horizontal plane has a phase of 0°, indicating that the electric field inside the cavity... The mode is excited, as shown by the black arrow in the figure. At t = T / 4, the electric field distribution on the horizontal plane shifts to a phase of 90°, indicating that... The mode is excited. At t = T / 2, the horizontal electric field distribution of the substrate integrated waveguide cavity is similar to that at t = 0, but in the opposite direction and with a phase difference of 180°. Similarly, at t = 3T / 4, the horizontal electric field distribution of the substrate integrated waveguide cavity is consistent with that at t = T / 4, but in the opposite direction, with a phase difference of 270°. When these two sets of orthogonal modes have equal amplitudes, they can produce left-handed circularly polarized radiation, which is the working principle of this antenna element.

[0084] At a different operating frequency (22 GHz), another mode can be triggered. At t = 0, the electric field distribution on the horizontal plane has a phase of 270°, indicating... The mode was excited. At t = T / 4, the electric field phase on the horizontal surface became 0°, indicating that... The modes are excited. At t = T / 2 and t = 3T / 4, the transverse electric field distributions correspond to the electric field distributions at t = 0 and t = T / 4, respectively, but in opposite directions. Therefore, these two orthogonal modes can produce left-hand circularly polarized radiation at this operating frequency. By optimizing the parameters to make the operating bandwidths of the two resonant frequencies partially overlap, the antenna element can achieve broadband circularly polarized radiation.

[0085] Figure 3 It shows the frequency f respectively c1 =19.04GHz and f c2 The electric field distribution of the substrate-integrated waveguide cavity at 22 GHz. Since the cavity cross-section is circular. and The patterns are indistinguishable. Therefore, the pattern distribution within the cavity is consistent with EHM. 111 A consistent pattern is more reasonable

[21] ,

[22] . At 22 GHz and Patterns will be generated Figure 3 The electric field is shown in (b). Similarly, because the antenna cavity is circular, it is difficult to distinguish between the two modes mentioned above. Therefore, they are referred to as EHM. 211 Pattern or EHM 121 The mode is more reasonable. This design allows for the simultaneous excitation of two modes in a waveguide cavity integrated on the substrate. When the operating bandwidths of the two resonant frequencies partially overlap through parameter optimization, the antenna element can achieve broadband circularly polarized radiation.

[0086] 1.3 Research and Conception of Antenna Element Parameters

[0087] To clarify the impact of specific structural parameters on antenna element performance, the inventors of this invention focused their research on these parameters. The dimensions of the arcuate microstrip line, particularly its length l1 and width w1, as well as the heights h1 and h2 of substrates I and II, are several very important parameters.

[0088] Figure 4 The differential S-parameters S are shown in the simulation results for different arc-shaped microstrip line lengths l1 on the plane defined by phi = 0° and theta = 0°. dd11 axial ratio bandwidth (AR) and left-hand circular polarization gain (LHCP Gain). For example... Figure 4 As shown in (a), the impedance bandwidth increases with the increase of l1 value. When l1 equals 6.47 mm, the axial ratio bandwidth of the antenna element is the largest, as shown in (a). Figure 4 As shown in (b). Therefore, when considering impedance bandwidth, axial ratio bandwidth, and left-hand circular polarization gain, the antenna element exhibits superior performance when l1 = 6.47 mm, with a -10 dB impedance bandwidth (15.78-22.33 GHz), a 3 dB axial ratio bandwidth (17.30-21.28 GHz), and a 3 dB left-hand circular polarization gain bandwidth (15.07-22.57 GHz).

[0089] Figure 5 The differential S-parameters S are described by simulation on a plane defined by phi = 0° and theta = 0° under different arc-shaped microstrip line widths w1. dd11 The impedance bandwidth (AR) and axial ratio bandwidth (LHCP) are observed. Unlike the changes observed along the length l1, the impedance bandwidth and axial ratio bandwidth exhibit a trend of first increasing and then decreasing as the linewidth w1 changes. Regarding the left-hand circular polarization gain bandwidth, it decreases with increasing width w1. Similarly, when considering both impedance bandwidth and axial ratio bandwidth, the antenna element exhibits better performance at w1 = 0.3 mm.

[0090] Figure 6 The differential S-parameters Si obtained by simulation on the plane defined by phi = 0° and theta = 0° under different substrate heights h1 and h2 are described. dd11 The axial ratio bandwidth AR and the left-hand circular polarization gain LHCP Gain. According to the theory of substrate integrated waveguide resonant cavities, the height of the cavity can be determined by the following formula (1):

[0091]

[0092] Among them, v cε is the speed of light in free space, f0 is the resonant frequency of the substrate integrated waveguide back cavity, h1 is the height of the first top layer substrate, h2 is the height of the second bottom layer substrate, and the diameter of the substrate integrated waveguide back cavity is C1. r Let h1 and h2 be the relative permittivity of the first top substrate and the second bottom substrate. Although h1 and h2 must be determined individually, the overall height of the substrate, i.e., the value of h1 + h2, can still be calculated from this formula. Figure 6 As shown in (a), the impedance bandwidth of the antenna element is maximized when the ratio of h1 to h2 is 1:1. Similarly, as... Figure 6 As shown in (b), the antenna element achieves the widest axial ratio bandwidth. In contrast, the axial ratio bandwidth deteriorates in the other two cases, becoming even worse when the ratio of h1 to h2 is 1:3, resulting in a loss of circular polarization radiation characteristics. Regarding the left-hand circular polarization gain bandwidth, variations in substrate height indicate that increasing h1 and decreasing h2 lead to a gradual decrease in the left-hand circular polarization gain bandwidth. Compared to variations in l1 and w1, variations in substrate height have a greater impact on the left-hand circular polarization gain bandwidth.

[0093] To verify the radiation characteristics of the antenna element under study, Figure 7 The radiation patterns in the E-plane and H-plane are shown at a specified center frequency of 19.04 GHz, including left-handed main polarization and right-handed cross-polarization. Consistent with the results of the parameter studies, the selected antenna element dimensions are l1 = 6.47 mm, w1 = 0.3 mm, and h1:h2 = 1:1. Under these parameters, the antenna elements exhibit symmetrical radiation patterns in the E-plane and H-plane, with the right-handed cross-polarization level below -25 dB. This is attributed to the use of a pair of differential feeds, which suppress electromagnetic interference, thereby reducing the cross-polarization level and improving the antenna resolution.

[0094] 2. Antenna Array Design and Assembly

[0095] 2.1 Antenna Array Design and Parameters

[0096] Antenna element arrangement: The four antenna elements are arranged in a 2×2 array with a spacing of 11.1mm (0.7λ0).

[0097] As shown in Figure 8, a 2×2 antenna array was constructed using the antenna elements mentioned above, with the aim of improving the peak gain of the antenna. According to placement principles, the spacing between elements in the antenna array should be set to 0.5λ0 to λ0 of the center frequency. Here, the inventors of this invention chose a spacing L of 11.1 mm between two elements, which is approximately equal to 0.7λ0 at a center frequency of 19.04 GHz. This ensures both high gain for the antenna array and reduces sidelobes.

[0098] 2.2 Differential Feeder Network

[0099] Differential feed network integration: An improved ring coupler was fabricated on substrate III with a thickness of 0.254 mm and connected to the feed probe via copper pads.

[0100] like Figure 8(b) and 8(c) As shown, in order to achieve differential feeding, four improved ring couplers constituting the differential feeding network were added to the antenna array

[24] . The ring couplers were etched on the bottom surface of substrate III, which has a thickness h3 of 0.254 mm. The upper surface of substrate III was plated with copper as a common ground for the antenna array and the ring couplers. In order to strengthen the connection between the antenna array and the differential feeding network, a copper pad with a diameter of 0.7 mm was also etched on substrate III to facilitate good contact between the feed probe and the antenna.

[0101] Mechanical fixing: Use 2mm diameter plastic screws to fix each layer, ensuring that there are no air gaps between the layers.

[0102] Considering the impact of the introduced copper pad on impedance bandwidth and axial ratio bandwidth, as well as the increased processing difficulty, the diameter of the copper pad was determined to be 0.5 mm. Original matching hole diameter d M The diameter d of the feed probe increased from 1.1 mm to 1.6 mm. s The thickness remains constant at 0.5mm. All layers are secured together with 2mm diameter plastic screws. After inputting a signal at port 1, the equal power of the two outputs maintains a 180° phase difference, and the unbalanced phase difference is maintained within ±5° across the entire operating frequency band of the array.

[0103] Figure 9 The difference S-parameters S are displayed. dd11 The axial ratio bandwidth (AR) and left-hand circular polarization gain (LHCP Gain) are respectively applicable to configurations with and without a fed network. Impedance bandwidth starts from 34.4%.

[0104] The gain bandwidth decreased from 15.78-22.33 GHz to 32.6% (17.28-24.00 GHz), a decrease of 1.8%. Similarly, the 3-dB axial ratio bandwidth also decreased from 20.6% (17.30-21.28 GHz) to 16.1% (18.73-22.00 GHz), a decrease of 4.2%. Compared with and without a power supply network, a decrease in left-hand circular polarization gain bandwidth was observed, from 47.8% (14.14-23 GHz) to 42.9% (14.84-22.94 GHz), a reduction of 4.9%. In addition, the peak gain also changed significantly, decreasing from 14.30 dBi at 21.5 GHz to 10.43 dBi at 22.125 GHz under the same conditions.

[0105] The integration of the feed network can significantly affect the resonant characteristics of the antenna array. However, in the antenna array proposed in this invention, these effects can be controlled within a reasonable range, thereby achieving effective feeding of each differential antenna element.

[0106] 3. Testing and Performance Verification

[0107] 3.1 Antenna Array Prototype Manufacturing and Measurement

[0108] The inventors of this invention fabricated a prototype of the designed antenna array and its feed network to evaluate its performance. Figure 10(a) shows a photograph of the antenna array components and assembly. Furthermore, the axial ratio bandwidth AR, left-hand circular polarization gain LHCP Gain, and radiation pattern of the antenna array were measured in a microwave anechoic chamber, with the measurement setup shown in Figure 10(b). Each antenna element of the antenna array is equipped with a ring coupler, and the antenna array is powered via a 1-to-4 power divider during the measurement process.

[0109] A 4-port Ceyear 3672D vector network analyzer was used to measure the reflection coefficient. Since the differential ports of each antenna element are integrated into the antenna array, only the reflection coefficient of each port could be measured. When measuring the reflection coefficient of port 1, the other three ports were connected to 50-ohm loads.

[0110] Figure 11 The simulation and measured reflection coefficients of the designed antenna array under port 1 excitation are shown. The curves show that although the overall trends of the simulation and measurements are consistent, their locations vary, such as the resonant point and the associated resonant frequency decreasing.

[0111] Figure 12 Simulated and measured curves for axial ratio bandwidth (AR) and left-hand circular polarization gain (LHCP Gain) are shown. Overall, the experimental results generally follow the trends of the simulated data. However, some differences exist. The measured axial ratio bandwidth is 13.8% (17.05 to 19.57 GHz), while the simulated axial ratio bandwidth is 16.1% (18.73 to 22.00 GHz), a decrease of 2.3%. The measured left-hand circular polarization gain bandwidth is 41.75% (15.39 to 23.51 GHz), with a peak gain of 10.56 dBi at 23.02 GHz, slightly different from the simulated value. The simulated left-hand circular polarization gain bandwidth is 42.9% (14.84 to 22.94 GHz), with a peak gain of 10.43 dBi at 22.125 GHz.

[0112] Figure 13 shows the simulated and measured radiation patterns of the designed antenna array in the E-plane and H-plane at a frequency of 19.04 GHz. The measured results for left-hand circular polarization are in high agreement with the simulation results. In contrast, the experimental results for right-hand circular polarization are higher than the simulated data. However, they still indicate that the designed array can maintain its optimal radiation direction within the plane defined by phi = 0° and theta = 0°, with the maximum radiation direction along the positive z-axis.

[0113] The test results are summarized as follows:

[0114] Bandwidth test: The impedance bandwidth was measured to be 32.6% (17.28-24.00GHz) and the axial ratio bandwidth was 13.8% using a vector network analyzer.

[0115] Gain measurement: Peak gain 10.56 dBi (23.02 GHz), 3-dB gain bandwidth 41.75%;

[0116] Radiation characteristics: Cross-polarization level below -25dB, main lobe along the z-axis, sidelobe level -15dB.

[0117] 3.2 Error Analysis

[0118] right Figure 11 , Figure 12 The errors between the experimental data and simulation results in Figure 13 were analyzed. Figure 11 As shown, the initial resonant frequency shifted, which may be due to errors in the manufacturing process. As shown in Equation (1), the resonant frequency is affected by the cavity diameter C1; even a slight increase in this diameter will lead to a decrease in the resonant frequency. Furthermore, although the plastic screws were tightened to their maximum extent during measurement, air gaps were created between the layers due to the small antenna size. This may lead to an increase in the relative permittivity between these layers, thus affecting the resonant position. In addition, the feed network of the antenna array is sensitive to curvature because substrate III is very thin. Therefore, the antenna array must be tested very carefully during testing. Considering these factors, there are some differences between the experiments and simulations.

[0119] 4. Technical Effects

[0120] This invention achieves comprehensive performance with wide bandwidth, high gain, low profile, and low cross-polarization through dual-mode resonance and differential feeding technology. Compared with existing technologies, this invention offers a significant bandwidth improvement and eliminates the need for an external balun, allowing direct integration into a compact RF front-end. Table II summarizes the performance and key specifications of the antenna array of this invention compared to other similar designs.

[0121] Table II. Comparison of the antenna array of the present invention with similar antenna structures in other prior art. Note: λ0 refers to the wavelength of free space corresponding to the center frequency, and NG indicates that it is not given in the text.

[0122]

[0123] Of the various feeding methods, only the antenna array and antenna sample [5] of this invention can utilize differential feeding, which helps reduce common-mode interference and cross-polarization. The antenna array developed in this invention exhibits a wider impedance bandwidth, a wider axial ratio bandwidth, a larger gain bandwidth, and a lower profile height, all achieved through a simple microstrip differential feeding technique. Although the design in antenna sample [4] has a higher peak gain and a lower profile height, its bandwidth-related metrics are not as good as those of this invention. Antenna sample [3] shows higher gain and a wider axial ratio bandwidth, but it has more antenna elements, a larger size, and its impedance and gain bandwidths are not as wide as those of this invention. In contrast, antenna samples [1] and [2] have lower profiles, but their bandwidth-related metrics are not comparable to those of this invention.

[0124] In summary, this invention employs a microstrip differential feeding method, which improves the performance of the antenna array, resulting in a simpler structure, wider bandwidth, and smaller size.

[0125] According to the present invention, a novel 2×2 circularly polarized substrate-integrated waveguide cavity antenna array employing differential feeding technology is presented. This design utilizes a pair of arc-shaped microstrip lines, resulting in a very simple structure that can be manufactured using low-cost printed circuit board processes. Simulations and experiments demonstrate excellent bandwidth, gain, and profile performance. Furthermore, modifying the structure of the arc-shaped microstrip lines facilitates the realization of dual-frequency circularly polarized radiation and polarization reconfigurability. This versatility allows this type of antenna to adapt to numerous application scenarios, which is a significant advantage.

[0126] The foregoing description of several embodiments of the invention has been provided for illustrative purposes. This foregoing description is not intended to be exhaustive, nor is it intended to limit the invention to the specific forms and / or configurations disclosed. The invention is not limited to the illustrative embodiments described above. Those skilled in the art will understand that other modifications and alterations can be made to the embodiments disclosed herein without departing from the scope and spirit of the invention, and such modifications and alterations are all within the scope of the invention.

Claims

1. A differentially fed broadband circularly polarized substrate integrated waveguide cavity antenna array, characterized in that, include: An array structure consisting of 2×2 antenna elements, wherein each antenna element includes a substrate integrated waveguide back cavity, the substrate integrated waveguide back cavity being formed by stacking a first top substrate and a second bottom substrate, including an array of metallized vias penetrating the first top substrate and the second bottom substrate, for forming the electromagnetic boundary of the substrate integrated waveguide back cavity; A pair of arc-shaped microstrip lines located in the second bottom substrate are synchronously excited by differential feeding of the back cavity of the substrate integrated waveguide. and Orthogonal mode; and The differential feed network includes four improved ring couplers integrated on a third substrate for providing differential signals with a phase difference in the range of 175°–185° to the antenna element.

2. The antenna array according to claim 1, characterized in that, The length, width, and spacing of the pair of arc-shaped microstrip lines are adjusted such that the operating bandwidths of the first and second resonant frequencies partially overlap.

3. The antenna array according to claim 2, characterized in that, The resonant frequency f of the first resonant frequency point c1 =19.04GHz, the resonant frequency of the second resonant frequency point is f c2 =22GHz.

4. The antenna array according to any one of claims 1-3, characterized in that, The diameter and spacing of the metallized via array are adjusted to match the resonant frequency of the substrate-integrated waveguide back cavity with the target frequency band.

5. The antenna array according to any one of claims 1-3, characterized in that, The sum of the height h1 of the first top substrate and the height h2 of the second bottom substrate, h1+h2, is optimized using the following formula: Among them, v c f0 is the speed of light in free space, h1 is the height of the first top-layer substrate, h2 is the height of the second bottom-layer substrate, C1 is the diameter of the substrate-integrated waveguide back cavity, and ε is the resonant frequency of the back cavity. r is the relative permittivity of the first top substrate and the second bottom substrate.

6. The antenna array according to any one of claims 1-3, characterized in that, The cross-polarization level of the antenna element is below -25dB.

7. The antenna array according to any one of claims 1-3, characterized in that, The profile height of the antenna array is 0.19-0.21λ0, where λ0 is the free space wavelength corresponding to the center frequency of the antenna array.

8. The antenna array according to any one of claims 1-3, characterized in that, The antenna array has an impedance bandwidth of at least 32.6% (17.28-24.00 GHz), a 3-dB axial ratio bandwidth of at least 13.8% (17.05-19.57 GHz), and a 3-dB gain bandwidth of at least 41.75% (15.39-23.51 GHz).

9. The antenna array according to any one of claims 1-3, characterized in that, Each of the antenna elements has a radiation hole on the top layer of the substrate integrated waveguide back cavity and two matching holes on the bottom layer; and each of the antenna elements has a center hole at its geometric center.

10. The antenna array according to any one of claims 1-3, characterized in that, The phase difference is 180°.

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