Current sampling adaptive correction method and electronic device

Abstract

The application discloses a current sampling adaptive correction method and electronic equipment, and relates to the technical field of power electronic control. The method is applied to a totem pole bridgeless power factor correction converter. In the current control cycle, the grid voltage instantaneous value, the DC bus voltage, the inductor current instantaneous value and the PWM duty cycle of the previous control cycle are obtained. The voltage difference value is determined according to the absolute values of the DC bus voltage and the grid voltage, and the current correction coefficient is calculated in combination with the DC bus voltage and the PWM duty cycle of the previous control cycle. The inductor current instantaneous value is corrected by using the current correction coefficient to obtain an equivalent current feedback value. Then, the current control error is determined according to the AC current reference value and the equivalent current feedback value, and the PWM duty cycle of the current control cycle is generated by the current loop adjustment. Thus, without the need for conduction mode detection and controller switching, the matching of the current feedback quantity and the operating state can be improved, and the current tracking precision and the system stability can be improved.

Description

Technical Field

[0001] This application relates to power electronic control technology, and in particular to a current sampling adaptive correction method and electronic device. Background Technology

[0002] Totem-pole bridgeless power factor correction converters are increasingly used in AC-DC conversion applications due to their fewer conducting devices, lower conduction losses, and higher conversion efficiency. In actual operation, totem-pole bridgeless power factor correction converters typically require closed-loop control based on sampled grid voltage, DC bus voltage, and inductor current signals to ensure that the input current varies according to the desired reference current and maintains a stable DC bus voltage.

[0003] However, under operating conditions such as load changes and instantaneous changes in grid voltage, the inductor current state of the converter will change, and there may be a deviation between the sampled instantaneous inductor current value and the effective feedback quantity used for current loop control. If the sampled current is directly used as current feedback for control, it can easily lead to an increase in current control error, making it difficult for the pulse width modulation duty cycle output in the current control cycle to accurately match the actual operating state, thereby affecting the current tracking effect and the stability of converter operation.

[0004] Existing control methods typically adjust the current loop directly based on the deviation between the sampled current and the reference current, or rely on relatively fixed compensation parameters to process the sampled current. When the converter's operating state changes with variations in grid voltage, DC bus voltage, and duty cycle, fixed compensation methods struggle to adapt to different operating conditions, potentially leading to a mismatch between the current feedback and actual control requirements. Therefore, it is necessary to provide a control method that can adaptively correct the current sample value based on the converter's current operating state to improve the effectiveness of the current feedback and the accuracy of the current loop control. Summary of the Invention

[0005] This application aims to address at least one of the technical problems existing in the prior art. To this end, this application proposes a current sampling adaptive correction method and electronic device, which can improve the matching between the current feedback quantity and the operating state, and improve the current tracking accuracy and system stability.

[0006] In a first aspect, embodiments of this application provide a current sampling adaptive correction method and an electronic device.

[0007] According to the first aspect of this application, a current sampling adaptive correction method and electronic device are applied to a totem-pole bridgeless power factor correction converter. The current sampling adaptive correction method includes: acquiring the instantaneous values ​​of the grid voltage, DC bus voltage, and inductor current in the current control cycle, as well as the pulse width modulation duty cycle of the previous control cycle; determining a voltage difference based on the instantaneous grid voltage and the DC bus voltage; calculating a current correction coefficient based on the voltage difference, the DC bus voltage, and the pulse width modulation duty cycle of the previous control cycle; adaptively proportionally correcting the absolute value of the instantaneous inductor current based on the current correction coefficient to obtain an equivalent current feedback value; acquiring an AC current reference value and determining a current control error based on the absolute value of the AC current reference value and the equivalent current feedback value; calculating a current loop adjustment based on the current control error; and obtaining the pulse width modulation duty cycle of the current control cycle based on the current loop adjustment to control the high-frequency switching devices of the totem-pole power factor correction converter.

[0008] The current sampling adaptive correction method according to the first aspect of this application has at least the following beneficial effects: By acquiring the instantaneous value of the grid voltage, the DC bus voltage, the instantaneous value of the inductor current, and the pulse width modulation duty cycle of the previous control cycle in the current control cycle, and determining the voltage difference based on the instantaneous value of the grid voltage and the DC bus voltage, and further calculating the current correction coefficient by combining the voltage difference, the DC bus voltage, and the pulse width modulation duty cycle of the previous control cycle, the current correction coefficient can be adaptively adjusted according to the changes in the operating state represented by the grid voltage, the bus voltage, and the duty cycle. By using the current correction coefficient to adaptively proportionally correct the absolute value of the instantaneous value of the inductor current, an equivalent current feedback value more suitable for current loop control can be obtained, thereby reducing the feedback deviation caused by changes in the converter's operating state when directly using the instantaneous sampled value of the inductor current as the feedback quantity. Furthermore, this application determines the current control error based on the absolute value of the AC current reference value and the equivalent current feedback value, calculates the current loop adjustment based on the current control error, and then obtains the pulse width modulation duty cycle of the current control cycle from the current loop adjustment to control the high-frequency switching devices of the totem-pole power factor correction converter. This enables the duty cycle adjustment of the current control cycle to match the corrected current feedback, improves the current loop's ability to track the reference current, and enhances the current control accuracy and operational stability of the totem-pole bridgeless power factor correction converter under different operating conditions.

[0009] Secondly, embodiments of this application provide an electronic device.

[0010] An electronic device according to a second aspect embodiment of this application includes a processor and a memory, the memory storing a computer program that, when executed by the processor, implements the current sampling adaptive correction method described in any embodiment of the first aspect. The technical problems solved and the beneficial effects of the electronic device in these embodiments all originate from the current sampling adaptive correction method of the first aspect embodiment, and will not be repeated here. Attached Figure Description

[0011] The present application will be further described below with reference to the accompanying drawings and embodiments, wherein: Figure 1 This is a flowchart illustrating the current sampling adaptive correction method according to an embodiment of this application; Figure 2 This is a flowchart illustrating the current correction coefficient calculation method according to an embodiment of this application. Figure 3 This is a schematic flowchart of the pulse width modulation duty cycle generation method according to an embodiment of this application; Figure 4 This is a flowchart illustrating the voltage feedforward calculation method according to an embodiment of this application. Detailed Implementation

[0012] The embodiments of this application are described in detail below. Examples of these embodiments are shown in the accompanying drawings, wherein the same or similar reference numerals denote the same or similar elements or elements having the same or similar functions throughout. The embodiments described below with reference to the accompanying drawings are exemplary and are only used to explain this application, and should not be construed as limiting this application.

[0013] In the description of this application, it should be understood that the orientation descriptions, such as up, down, front, back, left, right, etc., indicate the orientation or positional relationship based on the orientation or positional relationship shown in the accompanying drawings. They are only for the convenience of describing this application and simplifying the description, and do not indicate or imply that the device or element referred to must have a specific orientation, or be constructed and operated in a specific orientation. Therefore, they should not be construed as limitations on this application.

[0014] In the description of this application, "several" means one or more, "multiple" means two or more, "greater than," "less than," and "exceeding" are understood to exclude the stated number, while "above," "below," and "within" are understood to include the stated number. The use of "first" and "second" in the description is merely for distinguishing technical features and should not be construed as indicating or implying relative importance, or implicitly indicating the number of indicated technical features, or implicitly indicating the order of the indicated technical features.

[0015] In the description of this application, unless otherwise expressly defined, terms such as "setup," "installation," and "connection" should be interpreted broadly, and those skilled in the art can reasonably determine the specific meaning of the above terms in this application in conjunction with the specific content of the technical solution.

[0016] In the description of this application, the terms "one embodiment," "some embodiments," "illustrative embodiment," "example," "specific example," or "some examples," etc., refer to specific features, structures, materials, or characteristics described in connection with that embodiment or example, which are included in at least one embodiment or example of this application. In this specification, the illustrative expressions of the above terms do not necessarily refer to the same embodiment or example. Furthermore, the specific features, structures, materials, or characteristics described may be combined in any suitable manner in one or more embodiments or examples.

[0017] It should be noted that in all specific embodiments of this application, when processing data related to user identity or characteristics, such as user information, user behavior data, user historical data, and user location information, user permission or consent is obtained first. Furthermore, the collection, use, and processing of this data comply with relevant laws, regulations, and standards. In addition, when embodiments of this application require access to sensitive personal information of users, separate permission or consent from the user is obtained through pop-ups or redirection to confirmation pages. Only after obtaining the user's separate permission or consent is the necessary user-related data required for the proper functioning of these embodiments acquired.

[0018] In a first aspect, embodiments of this application provide a current sampling adaptive correction method.

[0019] The current sampling adaptive correction method is applied to a totem-pole bridgeless power factor correction converter. The totem-pole bridgeless power factor correction converter may include an AC input terminal, a boost inductor, high-frequency switching devices, a power frequency commutation device, a DC bus capacitor, and a digital controller. The high-frequency switching devices are used to switch on and off at high frequencies under the action of the pulse width modulation signal output by the digital controller to regulate the current in the boost inductor; the DC bus capacitor is used to maintain the DC bus voltage; the digital controller is used to collect operating parameters such as voltage and current, and to execute current loop control based on the collected results. During the operation of this totem-pole bridgeless power factor correction converter, the current in the boost inductor may vary between continuous conduction and intermittent conduction states depending on factors such as load size, instantaneous grid voltage position, and DC bus voltage. For example, when the load is heavy or the grid voltage is near its peak, the inductor current is more likely to remain continuous; while under light load or near the grid voltage zero crossing, the inductor current may drop to zero within one switching cycle, thus entering an intermittent conduction state. Since the inductor current waveform has a zero current range in the intermittent conduction state, if the instantaneous value of the inductor current is directly used as the current feedback input to the current loop, the correspondence between the current feedback and the pulse width modulation duty cycle will deviate from the linear control relationship in the continuous conduction state, resulting in a reduction in the current loop regulation accuracy.

[0020] The current sampling adaptive correction method in this embodiment can be executed by a digital controller in a totem-pole bridgeless power factor correction converter. The digital controller can be a microcontroller, digital signal processor, field-programmable gate array, system-on-a-chip, or other control unit with sampling, computation, and pulse-width modulation output capabilities. The method can be executed cyclically according to a control cycle, which can be the same as the pulse-width modulation cycle or an integer multiple of the pulse-width modulation cycle.

[0021] like Figure 1 As shown, the current sampling adaptive correction method in the embodiment includes, but is not limited to, steps S100 to S700: S100: Obtain the instantaneous values ​​of the grid voltage, DC bus voltage, and inductor current for the current control cycle, as well as the pulse width modulation duty cycle of the previous control cycle. S200. Determine the voltage difference based on the instantaneous value of the grid voltage and the DC bus voltage; S300. The current correction coefficient is calculated based on the voltage difference, the DC bus voltage, and the pulse width modulation duty cycle of the previous control cycle. S400. Based on the current correction coefficient, the absolute value of the instantaneous value of the inductor current is adaptively proportionally corrected to obtain the equivalent current feedback value. S500: Obtain the AC current reference value, and determine the current control error based on the absolute value of the AC current reference value and the equivalent current feedback value. S600. The current loop adjustment amount is calculated based on the current control error. S700: Based on the current loop adjustment, the pulse width modulation duty cycle of the current control cycle is obtained to control the high-frequency switching devices of the totem pole power factor correction converter.

[0022] In step S100, the instantaneous values ​​of the grid voltage, DC bus voltage and inductor current of the current control cycle are obtained, as well as the pulse width modulation duty cycle of the previous control cycle.

[0023] In the current control cycle, the digital controller acquires the instantaneous values ​​of the grid voltage, DC bus voltage, and inductor current. The instantaneous grid voltage is the instantaneous value of the AC-side grid voltage of the totem-pole bridgeless power factor correction converter at the sampling moment corresponding to the current control cycle. It can be positive, negative, or close to zero, and is used to characterize the real-time amplitude and polarity of the AC grid voltage. The DC bus voltage is the voltage across the DC-side bus of the totem-pole bridgeless power factor correction converter, used to characterize the energy state of the DC output or DC input side. The instantaneous inductor current is the instantaneous value of the current flowing through the boost inductor at the sampling moment corresponding to the current cycle, used to reflect the current state of the power stage. In practical implementation, the instantaneous grid voltage, DC bus voltage, and inductor current can be acquired through voltage sampling circuits, current sampling circuits, and analog-to-digital conversion modules. The voltage sampling circuit can include a voltage divider circuit, an isolation sampling circuit, an operational amplifier circuit, or a filter circuit; the current sampling circuit can take the form of a sampling resistor, a Hall current sensor, a current transformer, or a fluxgate current sensor, etc. The digital controller can trigger analog-to-digital conversion at a preset sampling point in the pulse width modulation period to reduce the impact of switching ripple and sampling delay on the control results.

[0024] The pulse width modulation (PWM) duty cycle of the previous control cycle is the duty cycle output by the digital controller to the high-frequency switching devices in the previous control cycle. Since the change in inductor current in the power circuit is closely related to the switching state applied in the previous control cycle, incorporating the PWM duty cycle of the previous control cycle into the calculation of the current correction coefficient allows the current correction process to reflect the current load state and switching adjustment state, thereby improving the adaptiveness of the correction. Here, "previous control cycle" can refer to the control cycle immediately preceding the current control cycle, or, in digital control implementation, the most recent effective PWM duty cycle after delay, filtering, or update-and-hold processing.

[0025] In step S200, after obtaining the instantaneous value of the grid voltage and the DC bus voltage, the digital controller determines the voltage difference based on the two. The voltage difference is used to characterize the voltage margin of the DC bus voltage relative to the instantaneous amplitude of the AC grid voltage. For a totem-pole bridgeless power factor correction converter, the rise and fall of the boost inductor current within a switching cycle is related to the difference between the AC voltage and the DC bus voltage; therefore, the voltage difference can reflect the slope of the inductor current drop and the degree of current discontinuity at the current instantaneous position of the grid voltage.

[0026] In one alternative implementation, the amplitude information of the grid voltage can be determined first based on its instantaneous value, and then the voltage difference can be determined using the difference between the DC bus voltage and this amplitude information. Since the totem-pole bridgeless power factor correction converter can perform boost current shaping control in both the positive and negative half-cycles of the grid, and current control typically focuses on the instantaneous amplitude of the grid voltage rather than its polarity, the absolute value of the instantaneous grid voltage can be used in the voltage difference calculation. The voltage difference can be expressed as a quantity corresponding to the following relationship: Δ V = Vbus - | Vgrid |, where Δ V Indicates the voltage difference. Vbus Indicates the DC bus voltage. Vgrid Represents the instantaneous value of the grid voltage, | Vgrid | represents the absolute value of the instantaneous grid voltage. In practical digital control, to avoid calculation instability when the instantaneous amplitudes of the DC bus voltage and the grid voltage are close, the voltage difference can be further processed by limiting, clamping, or protection. For example, when the amplitudes of the instantaneous DC bus voltage and the grid voltage are very close, the voltage difference can be limited to a preset lower limit. This preset lower limit can be determined based on the voltage sampling accuracy, controller numerical format, system rated voltage, maximum allowable computational gain, and safety margin. This avoids problems such as division by zero, overflow, or sudden changes in correction coefficients when performing division calculations based on the voltage difference.

[0027] In step S300, the current correction coefficient is calculated based on the voltage difference, the DC bus voltage, and the pulse width modulation duty cycle of the previous control cycle. The current correction coefficient is used to proportionally correct the instantaneous value of the inductor current, so that the current feedback quantity can adaptively change with the instantaneous value of the grid voltage, the DC bus voltage, and the duty cycle state.

[0028] In a totem-pole bridgeless power factor correction converter, the change in inductor current within a switching cycle is related to the on-time and off-time of the high-frequency switching devices. The pulse width modulation duty cycle of the previous control cycle reflects the relative on-time of the high-frequency switching devices within that cycle, thus reflecting the strength of the inductor's energy storage process. The DC bus voltage and voltage difference reflect the voltage conditions when the inductor current releases energy. Therefore, combining the DC bus voltage, the pulse width modulation duty cycle of the previous control cycle, and the voltage difference to calculate the current correction coefficient allows the correction coefficient to simultaneously reflect changes in bus voltage, instantaneous changes in grid voltage position, and changes in load or control output.

[0029] In one implementation, the current correction factor can be determined according to the following relationship:

[0030] in, KIndicates the current correction factor. Vbus Indicates the DC bus voltage. Duty Δ represents the pulse width modulation duty cycle of the previous control cycle. V This represents the voltage difference determined based on the instantaneous value of the grid voltage and the DC bus voltage. In this relationship... Vbus × Duty This can be understood as a quantity related to the inductor's energy storage and the strength of its control action during the switching cycle, Δ V This can be understood as a quantity related to the freewheeling current or energy release conditions of the inductor. By calculating the ratio of these two factors, the current correction coefficient can be made to change synchronously with the AC side voltage position, DC bus status, and actual duty cycle status.

[0031] To accommodate the real-time operation of the digital controller and the safe operation of the power converter, the current correction coefficient can be limited according to the system operating conditions. For example, when the voltage difference is small or the pulse width modulation duty cycle of the previous control cycle is large, the current correction coefficient calculated according to the above relationship may have a large value. In this case, the current correction coefficient can be limited to a preset range to avoid undesirable abrupt changes in the current feedback value. The preset range can be determined based on the current sampling ratio, controller gain, PI controller parameters, allowable current feedback dynamic range, and system stability requirements.

[0032] In another alternative implementation, to reduce the impact of noise on the current correction coefficient, low-pass filtering, moving average, or rate limiting can be applied to the instantaneous grid voltage, DC bus voltage, pulse width modulation duty cycle of the previous control cycle, or the calculated current correction coefficient. Rate limiting refers to limiting the rate of change of the current correction coefficient between adjacent control cycles to avoid rapid jumps in the current correction coefficient caused by sampling noise or transient interference.

[0033] In step S400, after obtaining the current correction coefficient, the digital controller performs adaptive proportional correction on the absolute value of the instantaneous inductor current based on the current correction coefficient to obtain the equivalent current feedback value. The absolute value of the instantaneous inductor current is used to eliminate the influence of the different current directions during the positive and negative half-cycles of the power grid on the current amplitude control, enabling the current loop to make unified adjustments based on the current amplitude. The equivalent current feedback value is a feedback quantity used to compare with the absolute value of the AC current reference value. It does not necessarily represent the actual current at a single physical sampling point, but rather a feedback value that has been corrected to be more suitable for current loop control calculations.

[0034] In one implementation, the equivalent current feedback value can be obtained according to the following relationship: ,in, I fb This represents the equivalent current feedback value. K Indicates the current correction factor.I raw Represents the instantaneous value of the inductor current, | I raw | represents the absolute value of the instantaneous inductor current. This proportional correction process can automatically adjust the current feedback value based on the changes in the instantaneous grid voltage, DC bus voltage, and pulse width modulation duty cycle of the previous control cycle, thereby improving the matching relationship between the sampled inductor current value and the current control target under intermittent conduction conditions.

[0035] Because the power transmission paths of the totem-pole bridgeless power factor correction converter differ in the positive and negative half-cycles, but the amplitude control law of the boost inductor current is similar, using the absolute value of the instantaneous inductor current for correction is beneficial for employing a consistent current loop control structure throughout the entire AC cycle. Especially in the current zero-crossing region, the inductor current may exhibit discontinuity, increased ripple, or enhanced dispersion of sampled values. After proportional correction using the current correction coefficient, the equivalent current feedback value obtained by the current loop can better reflect the current regulation requirements within the current control cycle. In practical implementation, the instantaneous inductor current value can be a single-point sampled value or a processed value obtained after multiple samplings within a control cycle. For example, a sampled value located at a position with low switching ripple can be used, or the average, median, or digitally filtered sampled value from multiple sampling points can be used as the instantaneous inductor current value. Regardless of the sampling method used, as long as it serves as the basic quantity for inductor current feedback in the current control cycle and undergoes adaptive proportional correction based on the current correction coefficient, it can be used to obtain the equivalent current feedback value.

[0036] In step S500, the digital controller acquires an AC current reference value. This AC current reference value characterizes the desired current waveform on the AC side of the totem-pole bridgeless power factor correction converter. To achieve power factor correction, the AC current reference value is typically correlated with the phase information of the grid voltage, enabling the AC-side current to track the grid voltage according to a predetermined phase relationship. The AC current reference value can be generated jointly with the current amplitude output from the voltage outer loop, power control loop, DC bus voltage regulation loop, or other upper-level control loops, along with a grid synchronization signal. The grid synchronization signal can be obtained from a phase-locked loop, zero-crossing detection circuit, or other synchronization algorithms, and is used to characterize the phase of the grid voltage.

[0037] After acquiring the AC current reference value, the digital controller determines the current control error based on the absolute value of the AC current reference value and the equivalent current feedback value. The absolute value of the AC current reference value represents the desired current amplitude, and the equivalent current feedback value represents the corrected actual current amplitude feedback. The difference between the two values ​​yields the tracking deviation of the current amplitude for the current control cycle. In one implementation, the current control error can be expressed as: ,in, Ierr Indicates current control error. I ref Indicates the reference value of alternating current, | I ref | represents the absolute value of the AC current reference value. I fb This represents the equivalent current feedback value. This current control error is used to drive subsequent current loop regulation, causing the pulse width modulation duty cycle to change in the direction of reducing the error.

[0038] By comparing the absolute value of the AC current reference with the equivalent current feedback value, the current loop can employ a unified amplitude control method during both the positive and negative half-cycles of the power grid. The selection of switching devices corresponding to the grid voltage polarity, the power frequency commutation logic, or the current direction processing can be handled by the power stage modulation logic, while the current loop itself is primarily adjusted based on the current amplitude error. This reduces the risk of current distortion caused by inconsistencies in the control logic during the positive and negative half-cycles.

[0039] In step S600, after obtaining the current control error, the digital controller calculates the current loop adjustment amount based on the current control error. The current loop adjustment amount is used to characterize the adjustment result of the current loop output based on the current error, and it can serve as the basis for subsequently generating the pulse width modulation duty cycle of the current control cycle.

[0040] In one implementation, the current control error can be input into a proportional-integral (PI) controller to obtain the current loop adjustment. The proportional element in the PPI controller is used to quickly generate an adjustment based on the current current control error, while the integral element is used to eliminate steady-state errors. The current loop adjustment can be expressed as: ,in, U i This indicates the current loop adjustment amount. PI (•) indicates proportional integral operation. I err This indicates the current control error. The proportional coefficient, integral coefficient, integral limit value, and output limit value of the proportional-integral operation can be tuned according to the boost inductor parameters, switching frequency, current sampling ratio, DC bus voltage range, grid voltage range, and desired dynamic response speed.

[0041] In practical implementation, to facilitate controller parameter tuning and adaptation to sampling circuits with different ranges, the current control error can be normalized before being input to the proportional-integral (PI) controller. Normalization converts the error in the actual current dimension into a dimensionless error suitable for digital controller operation, reducing the impact of current sampling range variations on controller parameters. Furthermore, to avoid integral saturation, the PI controller can be configured with measures such as integral separation, integral limiting, anti-saturation feedback, or output limiting. When the system is in startup, shutdown, fault protection, or before the voltage loop is established, integral calculations can be paused or the current loop adjustment can be preset to improve the stability of the system during startup and protection processes.

[0042] It should be noted that the current loop regulation is not limited to being obtained from a proportional-integral (PI) controller. In other embodiments, a proportional controller, a proportional-integral-derivative (PID) controller, a resonant controller, a repetitive controller, a model predictive controller, or other control algorithms capable of generating regulation results based on current control errors can also be used. However, in the digital control of totem-pole bridgeless power factor correction converters, the PI controller has the advantages of low computational complexity, simple implementation, and good engineering adaptability, and is therefore a preferred implementation method.

[0043] In step S700, after obtaining the current loop adjustment amount, the digital controller obtains the pulse width modulation duty cycle of the current control cycle based on the current loop adjustment amount. The pulse width modulation duty cycle of the current control cycle is used to determine the on-time ratio of the high-frequency switching device in the current control cycle, thereby adjusting the current rise time and energy transfer process of the boost inductor.

[0044] In one implementation, the pulse width modulation (PWM) duty cycle for the current control cycle can be directly generated based on the current loop regulation. For example, the current loop regulation can be proportionally converted, biased, or limited to obtain the PWM duty cycle for the current control cycle. To ensure reliable driving of high-frequency switching devices and avoid abnormal operating states, the PWM duty cycle for the current control cycle is typically limited to a preset duty cycle range. The lower limit of this preset duty cycle range can be determined based on the minimum on-time of the switching devices, the response time of the drive circuit, and the resolution of the PWM module; the upper limit of this preset duty cycle range can be determined based on the boost inductor saturation risk, the safe operating area of ​​the switching devices, dead time requirements, and the DC bus safety margin.

[0045] In another implementation, to improve the current loop response speed and reduce the current loop regulation burden, a compensation amount related to the grid voltage or DC bus voltage can be introduced when generating the pulse width modulation duty cycle of the current control cycle based on the current loop regulation. For example, the current loop regulation can be synthesized with a feedforward amount obtained based on the voltage relationship, and then the synthesis result can be amplitude-limited to obtain the pulse width modulation duty cycle of the current control cycle. This feedforward amount can provide a duty cycle reference adapted to the instantaneous amplitude of the grid voltage, allowing the current loop regulation to be mainly used to correct model errors, sampling errors, load disturbances, and parameter deviations. Using this method, the pulse width modulation duty cycle of the current control cycle can be expressed as: , Duty now This indicates the pulse width modulation duty cycle of the current control cycle. U i This indicates the current loop adjustment amount. U ff The input represents the feedforward quantity, and sat(•) represents the limiting process. The specific calculation method for the feedforward quantity can be determined based on the target value of the DC bus voltage, the instantaneous value of the grid voltage, the grid voltage delay value, or other voltage state quantities; the limiting process is used to restrict the synthesized control quantity within the allowable duty cycle range of the high-frequency switching devices and power stage.

[0046] After the pulse width modulation (PWM) duty cycle for the current control cycle is generated, the digital controller can load it into the PWM module, which then generates control signals to drive the high-frequency switching devices. For a totem-pole bridgeless power factor correction converter, the high-frequency switching devices can include high-speed switching transistors located in the totem-pole arms, such as gallium nitride (GaN), silicon carbide (SiC), or fast silicon-based power devices. The digital controller can also generate complementary drive signals or control the power frequency commutation devices based on the grid voltage polarity, dead-time requirements, and synchronous rectification strategy, thereby enabling the totem-pole bridgeless power factor correction converter to achieve current shaping and energy transfer throughout the entire AC cycle.

[0047] Through steps S100 to S700, the current sampling adaptive correction method of this embodiment can adaptively calculate the current correction coefficient based on the instantaneous grid voltage, DC bus voltage, and pulse width modulation duty cycle of the previous control cycle, without explicitly determining whether the totem-pole bridgeless power factor correction converter is in a continuous or intermittent conduction state. This current correction coefficient is then used to proportionally correct the instantaneous inductor current. Consequently, the equivalent current feedback value obtained by the current loop can better adapt to the current control requirements under different conduction states, reducing current control errors caused by intermittent conduction, improving AC-side current tracking capability, and helping to reduce current waveform distortion. Furthermore, since this method mainly relies on the instantaneous grid voltage, DC bus voltage, instantaneous inductor current, and pulse width modulation duty cycle already present in the totem-pole bridgeless power factor correction converter, it does not require additional conduction mode detection circuits or current zero-crossing detection circuits, thus reducing hardware complexity. This method also eliminates the need to switch between different current controller structures between continuous and intermittent conduction states, thereby reducing the risk of current surges and discontinuous control parameters caused by mode switching.

[0048] In some embodiments, in step S200, determining the voltage difference based on the instantaneous value of the grid voltage and the DC bus voltage may further include, but is not limited to, steps S210 to S230: S210. Take the absolute value of the instantaneous grid voltage to obtain the absolute value of the grid voltage; S220. Calculate the difference between the absolute values ​​of the DC bus voltage and the grid voltage to obtain the initial voltage difference. S230. Compare the initial voltage difference with the preset minimum voltage difference, and determine the larger of the two values ​​as the voltage difference.

[0049] In step S210, the digital controller takes the absolute value of the instantaneous grid voltage to obtain the absolute value of the grid voltage. The instantaneous grid voltage is a sampled value of the AC grid voltage within the current control cycle, which varies with the AC grid cycle, typically being positive during the positive half-cycle and negative during the negative half-cycle. Since the totem-pole bridgeless power factor correction converter needs to perform boost current control based on the instantaneous amplitude of the grid voltage during both the positive and negative half-cycles, and the current correction coefficient is mainly related to the magnitude of the grid voltage and does not depend on the polarity of the grid voltage, the absolute value of the instantaneous grid voltage is taken before calculating the voltage difference to obtain the absolute value of the grid voltage representing the current amplitude of the grid voltage. The absolute value of the grid voltage can be expressed as: |Vgrid|.

[0050] In step S220, the digital controller calculates the difference between the absolute values ​​of the DC bus voltage and the grid voltage to obtain the initial voltage difference. The DC bus voltage is the voltage across the DC-side bus capacitor of the totem-pole bridgeless power factor correction converter, which is typically higher than the instantaneous amplitude of the AC-side grid voltage to meet the operating conditions of the boost power factor correction. The initial voltage difference characterizes the voltage margin of the DC-side bus voltage relative to the current amplitude of the AC-side grid voltage. This voltage margin is related to the voltage stress of the boost inductor during the high-frequency switching device turn-off or freewheeling phase and the current drop slope. In one implementation, the initial voltage difference can be obtained according to the following relationship: Vbus Vgrid ， Where, Δ V 0 represents the initial voltage difference. Vbus Indicates DC bus voltage, | Vgrid | represents the absolute value of the grid voltage. The larger this initial voltage difference, the greater the margin of the DC bus voltage relative to the current grid voltage amplitude; the smaller this initial voltage difference, the closer the instantaneous amplitude of the grid voltage is to the DC bus voltage, which makes it easier to have a smaller denominator and a larger correction factor when calculating the current correction factor later.

[0051] In actual operation, the DC bus voltage may have ripple, the grid voltage sampling may have noise or delay, and under certain abnormal or transient conditions, the absolute value of the grid voltage may be close to or even slightly higher than the DC bus voltage. Therefore, if the initial voltage difference is directly used in subsequent calculations, it may lead to unstable division operations, excessively large calculation results, or abnormal signs. To improve the stability of digital control, step S230 is further executed.

[0052] In step S230, the digital controller compares the initial voltage difference with a preset minimum voltage difference and determines the larger of the two as the voltage difference. The preset minimum voltage difference is a pre-set positive threshold value used to limit the minimum voltage difference value used in subsequent current correction coefficient calculations. Through this process, even if the initial voltage difference is close to zero, equal to zero, or negative due to sampling errors, subsequent calculations can still use an effective voltage difference value not less than the preset minimum voltage difference, thereby avoiding division by zero errors and numerical overflow. In one implementation, the voltage difference can be determined according to the following relationship: Δ V =max( Vbus - | Vgrid |,Δ V min ), where ΔV represents the voltage difference ultimately used to calculate the current correction factor. Vbus Indicates DC bus voltage, | Vgrid | represents the absolute value of the grid voltage, Δ V `min` represents the preset minimum voltage difference, and `max(•)` represents taking the larger value. The preset minimum voltage difference can be set according to the calculation accuracy of the digital controller, voltage sampling noise, DC bus voltage level, maximum allowable correction gain, and system protection margin. For example, in one specific implementation, the preset minimum voltage difference can be set to a voltage value close to zero but greater than zero, to avoid the denominator being zero or too small without significantly affecting the correction accuracy under normal operating conditions. The preset minimum voltage difference is not limited to a fixed constant. In other implementations, the preset minimum voltage difference can also be adaptively adjusted according to the operating state of the totem-pole bridgeless power factor correction converter. For example, when the DC bus voltage is high, the voltage sampling noise is low, or the controller uses floating-point arithmetic, a smaller preset minimum voltage difference can be selected; when the DC bus voltage is low, the voltage sampling noise is high, or the controller uses fixed-point arithmetic, the preset minimum voltage difference can be appropriately increased to improve calculation stability. The preset minimum voltage difference can also be calibrated according to the grid voltage level, switching frequency, current loop bandwidth, or safety protection strategy.

[0053] Through steps S210 to S230, the voltage difference obtained in this embodiment reflects both the actual voltage margin between the absolute values ​​of the DC bus voltage and the grid voltage, and avoids the problem of an abnormally large increase in the subsequent current correction coefficient due to an excessively small voltage difference. Since the voltage difference is a key input in the calculation of the current correction coefficient, its stability directly affects the smoothness of the equivalent current feedback value and the stability of the current loop. Therefore, the above processing helps improve the control reliability of the totem-pole bridgeless power factor correction converter under conditions such as near the peak grid voltage, DC bus voltage fluctuations, and light-load intermittent conduction.

[0054] like Figure 2 As shown, in some embodiments, in step S300, the current correction coefficient is calculated based on the voltage difference, the DC bus voltage, and the pulse width modulation duty cycle of the previous control cycle. This may further include, but is not limited to, steps S310 to S330: S310. Calculate the product of the DC bus voltage and the pulse width modulation duty cycle of the previous control cycle to obtain the intermediate current correction quantity. S320. Calculate the ratio of the intermediate current correction value to the voltage difference to obtain the initial current correction coefficient; S330. Compare the initial current correction coefficient with the preset maximum correction coefficient, and determine the smaller of the two values ​​as the current correction coefficient.

[0055] In step S310, the digital controller calculates the product of the DC bus voltage and the pulse width modulation duty cycle of the previous control cycle to obtain the intermediate current correction quantity. This intermediate current correction quantity characterizes the intermediate calculation result related to both the DC bus voltage level and the control output state of the previous control cycle when performing current correction in the current control cycle. Specifically, the DC bus voltage reflects the voltage state on the DC side of the totem-pole bridgeless power factor correction converter; the pulse width modulation duty cycle of the previous control cycle reflects the relative on-time of the high-frequency switching devices in the previous control cycle, and can also reflect the control strength applied by the current loop to the energy storage process of the boost inductor in the previous control cycle.

[0056] Since the change in inductor current in the current control cycle is related to the pulse width modulation (PWM) duty cycle applied in the previous control cycle, using the PWM duty cycle from the previous control cycle in the current correction coefficient calculation allows the current correction process to adaptively adjust to changes in the control output. In other words, a large PWM duty cycle in the previous control cycle indicates a relatively long conduction time for the high-frequency switching devices, resulting in a stronger energy storage process for the boost inductor during that cycle; conversely, a small PWM duty cycle indicates a relatively short conduction time for the high-frequency switching devices, resulting in a weaker energy storage process for the boost inductor. Introducing this information into the intermediate current correction values ​​helps to make the subsequent current correction coefficients more consistent with the actual power stage state.

[0057] In step S320, the digital controller calculates the ratio of the intermediate current correction value to the voltage difference to obtain the initial current correction coefficient. The voltage difference is a quantity determined based on the instantaneous value of the grid voltage and the DC bus voltage, used to characterize the voltage margin of the DC bus voltage relative to the instantaneous amplitude of the grid voltage. By calculating the ratio of the intermediate current correction value to the voltage difference, the initial current correction coefficient, which is simultaneously affected by the DC bus voltage, the pulse width modulation duty cycle of the previous control cycle, and the instantaneous position of the grid voltage, can be obtained.

[0058] In one implementation, the initial current correction coefficient can be expressed as:

[0059] in, K 0 indicates the initial current correction factor. Vbus Indicates the DC bus voltage. Duty Δ represents the pulse width modulation duty cycle of the previous control cycle. VThis represents the voltage difference. This formula corresponds to the relationship between the boost inductor voltage and the voltage of the totem-pole bridgeless power factor correction converter: the smaller the voltage difference, the closer the instantaneous amplitude of the grid voltage is to the DC bus voltage, and the more significant the changes in the freewheeling condition and current waveform characteristics of the inductor current, resulting in a corresponding increase in the initial current correction coefficient; conversely, the larger the voltage difference, the greater the margin of the DC bus voltage relative to the instantaneous amplitude of the grid voltage, resulting in a corresponding decrease in the initial current correction coefficient. Therefore, the initial current correction coefficient can adaptively change with the voltage position within the AC grid cycle. In practical digital control, to avoid unnecessary high-frequency changes in the initial current correction coefficient between adjacent control cycles due to sampling noise or control output jitter, filtering can be performed before or after calculating the initial current correction coefficient. For example, first-order low-pass filtering, moving average, or speed limiting can be applied to the DC bus voltage, voltage difference, pulse width modulation duty cycle of the previous control cycle, or the initial current correction coefficient.

[0060] In step S330, the digital controller compares the initial current correction coefficient with the preset maximum correction coefficient and determines the smaller of the two as the current correction coefficient. The preset maximum correction coefficient is used to limit the maximum value of the current correction coefficient, preventing the initial current correction coefficient from becoming too large due to small voltage differences, transient changes in DC bus voltage, sudden changes in the pulse width modulation duty cycle of the previous control cycle, or sampling anomalies, which could cause sudden changes in the equivalent current feedback value and affect the stability of the current loop. In one implementation, the current correction coefficient can be expressed as: K =min( K 0, K max ),in, K This represents the current correction coefficient ultimately used for adaptive proportional correction of the absolute value of the instantaneous inductor current. K 0 indicates the initial current correction factor. K max This represents the preset maximum correction factor, and min(•) indicates taking the smaller value. The preset maximum correction factor can be determined based on the operating mode of the totem-pole bridgeless power factor correction converter, the allowable current feedback gain, the current sampling range, the PI controller parameters, and the system stability requirements. In one specific implementation, the preset maximum correction factor can be set to 1.0, ensuring that the current correction factor does not exceed unity gain, thereby avoiding excessive amplification of the absolute value of the instantaneous inductor current under specific operating conditions.

[0061] Through steps S310 to S330, this embodiment combines the DC bus voltage, the pulse width modulation duty cycle of the previous control cycle, and the voltage difference into a current correction coefficient, enabling the current correction coefficient to adaptively update with changes in the instantaneous grid voltage, DC bus voltage, and control output. Since this calculation relies only on control quantities typically acquired or generated in the totem-pole bridgeless power factor correction converter, there is no need to add an additional inductor current zero-crossing detection circuit, nor is it necessary to explicitly determine whether the current operation is in a continuous or intermittent conduction state. This provides a real-time correction basis for the proportional correction of the subsequent instantaneous inductor current value.

[0062] In some embodiments, in step S400, the absolute value of the instantaneous inductor current is adaptively proportionally corrected according to the current correction coefficient to obtain an equivalent current feedback value, which may further include, but is not limited to, steps S410 to S420: In step S410, the digital controller takes the absolute value of the instantaneous inductor current to obtain the absolute value of the inductor current. The instantaneous inductor current is the sampled value of the current flowing through the boost inductor of the totem-pole bridgeless power factor correction converter at the sampling moment corresponding to the current control cycle. Since the AC current has different directions in the positive and negative half-cycles of the power grid, the instantaneous inductor current may be positive or negative. In the current loop control of this embodiment, the AC current reference value can also be further used in error calculation using its absolute value. Therefore, taking the absolute value of the instantaneous inductor current can unify the current feedback of the positive and negative half-cycles to the current amplitude level for processing. The absolute value of the inductor current can be expressed as: | I raw |

[0063] In a totem-pole bridgeless power factor correction converter, the positive and negative half-cycles of the power grid are typically formed by different switching branches or commutation devices participating in the current path. However, the target current amplitude control of the boost inductor remains consistent across both half-cycles. Therefore, converting the instantaneous value of the inductor current to its absolute value helps to ensure that the current loop uses a uniform error calculation and adjustment method throughout the entire AC cycle, reducing current distortion caused by differences in control logic between the positive and negative half-cycles.

[0064] In step S420, the digital controller obtains the equivalent current feedback value based on the product of the current correction coefficient and the absolute value of the inductor current. The current correction coefficient is a proportionality coefficient calculated based on the voltage difference, the DC bus voltage, and the pulse width modulation duty cycle of the previous control cycle. It can adaptively update with changes in the instantaneous position of the grid voltage, the state of the DC bus voltage, and the state of the control output. Multiplying this current correction coefficient by the absolute value of the inductor current can proportionally correct the original current amplitude feedback, making the obtained equivalent current feedback value more suitable as the feedback input of the current loop.

[0065] In one implementation, the equivalent current feedback value can be obtained according to the following relationship: I fb = K× | I raw |, among which, I fb This represents the equivalent current feedback value. K Indicates the current correction factor. I raw Represents the instantaneous value of the inductor current, | I raw | represents the absolute value of the inductor current. The current correction coefficient in this formula serves as a proportional correction factor, used to adjust the scale of the current feedback value based on the current voltage conditions of the power stage and the control state of the previous control cycle; the absolute value of the inductor current provides information on the current amplitude at the current sampling moment. The equivalent current feedback value obtained by multiplying the two can be directly compared with the absolute value of the AC current reference value to generate the current control error.

[0066] Because totem-pole bridgeless power factor correction converters may vary between continuous and intermittent conduction states, the correspondence between the instantaneous inductor current and the average current over a control cycle is not entirely consistent under different operating conditions. Especially in the intermittent conduction state, there may be a zero-current interval within a control cycle. If the absolute value of the inductor current is directly used as the feedback quantity, the feedback obtained by the current loop may not fully reflect the desired current control relationship. Adaptive proportional correction of the absolute value of the inductor current using a current correction coefficient allows the equivalent current feedback value to automatically adjust according to changes in operating conditions, thereby improving the accuracy of current control error calculation.

[0067] Through steps S410 to S420, this embodiment converts the instantaneous value of the inductor current into its absolute value, and further uses a current correction coefficient for proportional correction to obtain an equivalent current feedback value. This equivalent current feedback value includes both the current sampling information of the current control cycle and the operating status information reflected by the instantaneous grid voltage, DC bus voltage, and pulse width modulation duty cycle of the previous control cycle. Therefore, using the equivalent current feedback value in the current loop control helps reduce the error caused by the nonlinearity of current sampling under intermittent conduction conditions, and improves the current tracking performance of the totem-pole bridgeless power factor correction converter over a wide load range and the entire AC cycle.

[0068] In some embodiments, obtaining the AC current reference value in step S500 may further include, but is not limited to, steps S510 to S520: S510: Obtain current amplitude and grid synchronization sinusoidal quantity; S520: Generate AC current reference value based on current amplitude and grid synchronization sinusoidal quantity.

[0069] In steps S510 and S520, the current amplitude is used to characterize the expected current magnitude generated on the AC side by the totem-pole bridgeless power factor correction converter, which can be generated by the upper-level control loop according to the system operating target. For example, when the totem-pole bridgeless power factor correction converter is used for rectifier power factor correction, the current amplitude can be calculated by the DC bus voltage regulation loop based on the deviation between the DC bus voltage and the target value. When the DC bus voltage is lower than the target value, the DC bus voltage regulation loop can increase the current amplitude to increase the power absorbed from the AC side; when the DC bus voltage is higher than the target value, the DC bus voltage regulation loop can decrease the current amplitude to reduce the input power. Thus, the current amplitude is used to determine the envelope size of the AC current reference value.

[0070] A grid synchronization sinusoidal quantity is used to characterize a sinusoidal reference signal that is phase-synchronized with the grid voltage. It can be obtained from the instantaneous grid voltage value by a phase-locked loop (PLL), zero-crossing detection algorithm, quadrature signal generator, or other grid synchronization algorithms. The grid synchronization sinusoidal quantity is typically a normalized quantity, with an amplitude ranging from -1 to 1, used to represent the phase and waveform variation trend of the grid voltage. By generating an AC current reference value using the grid synchronization sinusoidal quantity, the AC current reference value can maintain a predetermined phase relationship with the grid voltage, thereby achieving power factor correction or bidirectional power control. In one implementation, the grid synchronization sinusoidal quantity can be calculated by a single-phase PLL based on the instantaneous grid voltage value. The PLL can track the phase and frequency of the grid voltage, and can output a sinusoidal signal synchronized with the grid voltage even if there is a certain deviation in the grid frequency. To reduce the impact of grid voltage harmonics or sampling noise on the synchronization result, the PLL input can be filtered, or a notch filter, low-pass filter, or quadrature decoupling structure can be set inside the PLL. The purpose of the above processing is to ensure that the synchronous sinusoidal quantity of the power grid has stable phase information, thereby providing a reliable reference for the AC current reference value.

[0071] The digital controller generates an AC current reference value based on the current amplitude and the grid synchronization sinusoidal quantity. The AC current reference value is the target current that the current loop aims to track; its waveform is determined by the grid synchronization sinusoidal quantity, and its magnitude is determined by the current amplitude. The generated AC current reference value can adjust its amplitude according to grid phase changes throughout the entire AC grid cycle, and based on load demand or DC bus energy status. In one implementation, the AC current reference value can be generated according to the following relationship: I ref = I amp × S grid ,in, I refIndicates the reference value of alternating current. I amp Indicates the current amplitude. S grid This represents the grid synchronization sinusoidal quantity. The current amplitude is used to determine the peak value or magnitude of the AC current reference value, while the grid synchronization sinusoidal quantity is used to determine the phase position and polarity of the AC current reference value in the current control cycle. When the grid synchronization sinusoidal quantity is positive, the AC current reference value corresponds to the current direction of the positive half-cycle of the grid; when the grid synchronization sinusoidal quantity is negative, the AC current reference value corresponds to the current direction of the negative half-cycle of the grid; when the grid synchronization sinusoidal quantity is close to zero, the AC current reference value is also close to zero, indicating that the AC current reference value is in the zero-crossing region.

[0072] Understandably, the absolute value of the AC current reference can be taken and compared with the equivalent current feedback value to determine the current control error. Since the AC current reference value is generated jointly by the current amplitude and the grid synchronization sinusoidal quantity, the current control error can not only reflect the deviation of the current amplitude but also enable the current loop to adjust around the sinusoidal target of grid voltage synchronization. Therefore, using this AC current reference value is beneficial for making the AC side current of the totem-pole bridgeless power factor correction converter approach a sinusoidal waveform and maintain the desired phase relationship with the grid voltage.

[0073] Through steps S510 to S520, this embodiment uses the current amplitude to determine the magnitude of the AC current reference value and uses the grid synchronization sinusoidal quantity to determine the phase and waveform of the AC current reference value, thereby generating an AC current reference value suitable for current loop tracking in a totem-pole bridgeless power factor correction converter. This AC current reference value, used in conjunction with the aforementioned equivalent current feedback value, enables the current loop to be adjusted according to a uniform amplitude error during the positive half-cycle, negative half-cycle, and zero-crossing region of the grid, which is beneficial for improving the quality of the AC side current waveform and the power factor.

[0074] In some embodiments, in step S600, the current loop adjustment amount is calculated based on the current control error, which may further include, but is not limited to, steps S610 to S630: S610, Obtain the full-scale current value; S620. Normalize the current control error based on the full-scale current value to obtain the normalized current control error. S630: Input the normalized current control error into the PI controller to obtain the current loop adjustment.

[0075] In step S610, the digital controller acquires the full-scale current value. The full-scale current value characterizes the maximum reference current scale corresponding to current sampling or current control calculations. It can be related to the range of the current sampling circuit, the input range of the analog-to-digital converter module, the software calibration ratio, or the maximum allowable inductor current of the totem-pole bridgeless power factor correction converter. By introducing the full-scale current value, the current control error with actual current units can be converted into a normalized quantity suitable for digital controller calculations, thereby improving the versatility of the control algorithm across different power levels, sampling ratios, or hardware platforms. In practical implementation, the full-scale current value can be a fixed parameter or a calibration parameter read according to the operating mode or hardware configuration. For example, when the current sampling circuit uses a fixed-ratio sampling resistor and amplifier, the full-scale current value can be determined based on the maximum input voltage of the analog-to-digital converter module, the sampling resistor value, and the amplification factor; when using a Hall current sensor or current transformer, the full-scale current value can be determined based on the sensor's rated measurement range and signal conditioning ratio. The full-scale current value is usually selected to be greater than or equal to the system's allowable peak current value to ensure that the normalized current control error is within a reasonable calculation range.

[0076] In step S620, the digital controller normalizes the current control error based on the full-scale current value to obtain the normalized current control error. The current control error is the difference between the absolute value of the AC current reference value and the equivalent current feedback value, representing the deviation in current amplitude tracking during the current control cycle. Since different systems may have different current ranges, if the current control error is directly input to the controller, the PI controller parameters may need to be readjusted according to the hardware range. Through normalization, the current control error can be converted into a proportional quantity relative to the full-scale current value, ensuring that the input processed by the subsequent PI controller is on a uniform scale.

[0077] In one implementation, the normalized current control error can be obtained according to the following relationship:

[0078] in, e i This represents the normalized current control error. I err Indicates current control error. I FS This indicates the full-scale value of the current.

[0079] In practical digital control, the normalized current control error can be further limited or filtered. For example, when abnormal sampling or sudden load changes cause excessive current control error, the normalized current control error can be limited to a preset range to avoid sudden changes in the PI controller output. When high-frequency ripple exists in the current sampling, the normalized current control error can be appropriately filtered to reduce the impact of high-frequency noise on the current loop regulation. The amplitude and bandwidth of the above processing can be set according to the dynamic response requirements of the current loop to avoid excessive filtering that slows down the current loop response.

[0080] In step S630, the digital controller inputs the normalized current control error into the PI controller to obtain the current loop adjustment. The PI controller can be a proportional-integral controller, used to calculate the current loop adjustment based on the normalized current control error. The proportional component provides immediate adjustment based on the current normalized current control error to improve the current loop response speed; the integral component provides adjustment based on the accumulation of the normalized current control error to reduce or eliminate steady-state error. The current loop adjustment is used to subsequently generate the pulse width modulation duty cycle for the current control cycle.

[0081] In one implementation, the PI controller can be updated once per control cycle using a discrete control approach. Its output can be expressed as: ,in, U i This indicates the current loop regulation; PI(•) indicates that the PI controller performs proportional-integral operations on the input. e i This represents the normalized current control error. The proportional and integral coefficients of the PI controller can be tuned based on the boost inductor parameters, switching frequency, control cycle, current sampling delay, DC bus voltage range, and desired current loop bandwidth. Generally, a larger proportional coefficient results in a faster response of the current loop to instantaneous errors, but an excessively large coefficient may lead to oscillations; a larger integral coefficient results in faster elimination of steady-state errors, but an excessively large coefficient may lead to integral saturation or low-frequency oscillations.

[0082] To improve the stability of the totem-pole bridgeless power factor correction converter during startup, load surges, and limiting conditions, the PI controller can be configured with output limiting, integral limiting, or anti-integral saturation strategies. Output limiting keeps the current loop regulation within the allowable range for duty cycle synthesis; integral limiting prevents the integral term from accumulating excessively when errors persist over a long period; the anti-integral saturation strategy can pause integration, recalculate the integral term, or reduce the integral accumulation rate when the subsequent pulse width modulation duty cycle reaches the upper or lower limit, thereby preventing large overshoot when the system recovers from saturation.

[0083] In some implementations, the PI controller can also be selectively enabled based on the system status. For example, when the totem-pole bridgeless power factor correction converter has not yet completed its soft start, grid synchronization has not been established, the DC bus voltage is abnormal, or the protection status is active, the PI controller output can be kept at a preset value, the integral term can be cleared, or the integral term can be frozen. Once the system enters normal operating condition, the normalized current control error is then input to the PI controller for closed-loop regulation. This approach avoids the PI controller's integral term from becoming uncontrollable due to error input under abnormal operating conditions.

[0084] Through steps S610 to S630, this embodiment normalizes the current control error using the full-scale current value and inputs the normalized current control error into the PI controller to obtain the current loop adjustment. This process ensures that the error signal faced by the PI controller has a uniform scale, which helps to reduce the impact of different current sampling ranges and different power levels on the control parameter tuning, and enables the aforementioned equivalent current feedback value to participate in the current closed-loop regulation of the totem-pole bridgeless power factor correction converter in a stable and controllable manner.

[0085] like Figure 3 As shown, in some embodiments, in step S700, the pulse width modulation duty cycle of the current control cycle is obtained based on the current loop adjustment amount, which may further include, but is not limited to, steps S710 to S750: S710. Obtain the DC bus voltage reference value and the delayed grid voltage instantaneous value, wherein the delayed grid voltage instantaneous value is the grid voltage instantaneous value delayed by one control cycle relative to the current control cycle; S720. Obtain the voltage proportionality value based on the ratio of the absolute value of the instantaneous value of the delayed grid voltage to the reference value of the DC bus voltage. S730. Calculate the voltage feedforward quantity based on the voltage proportional quantity; S740. Based on the current loop regulation and voltage feedforward, the synthesized control quantity is obtained; S750: Limit the amplitude of the synthesized control quantity to obtain the pulse width modulation duty cycle of the current control cycle.

[0086] In step S710, the digital controller acquires the DC bus voltage reference value and the delayed instantaneous grid voltage value. The DC bus voltage reference value is the target value for maintaining the DC bus voltage, which can be determined by the system's rated output voltage, downstream load demand, energy storage unit voltage range, or higher-level control commands. For totem-pole bridgeless power factor correction converters, the DC bus voltage reference value is typically used to characterize the target voltage level for boost control and is also the benchmark used to normalize the grid voltage amplitude when generating voltage feedforward quantities.

[0087] The delayed grid voltage instantaneous value is the grid voltage instantaneous value delayed by one control cycle relative to the current control cycle. Since digital controllers typically experience delays in sampling, analog-to-digital conversion, computation, and pulse width modulation updates during actual operation, the grid voltage information used for calculation in the current control cycle may correspond to the previous sampling moment or the previous control cycle. Therefore, using the delayed grid voltage instantaneous value in the voltage feedforward calculation ensures that the feedforward calculation matches the actual sampling link and duty cycle update timing of the digital controller, avoiding feedforward phase deviations caused by timing inconsistencies. In one implementation, the delayed grid voltage instantaneous value can be obtained by storing the grid voltage instantaneous value of the previous control cycle in a register, cache array, or delay unit.

[0088] In step S720, the digital controller obtains a voltage proportionality value based on the ratio of the absolute value of the delayed grid voltage instantaneous value to the DC bus voltage reference value. The absolute value of the delayed grid voltage instantaneous value represents the amplitude of the AC-side grid voltage at the delayed sampling time, eliminating the influence of the polarity difference between the positive and negative half-cycles of the grid on the feedforward calculation. The voltage proportionality value characterizes the proportional relationship between the AC-side grid voltage amplitude and the DC bus voltage reference value. In one implementation, the voltage proportionality value can be expressed as:

[0089] in, R v Indicates a voltage proportionality. Vgrid delay Indicates the instantaneous value of the delayed grid voltage, | Vgrid delay | represents the absolute value of the delayed instantaneous value of the grid voltage. Vdc ref This represents the reference value of the DC bus voltage. As a normalized benchmark, the DC bus voltage reference value should be a positive effective voltage value. In actual software implementation, the validity of the DC bus voltage reference value can be checked to avoid abnormal division calculations. The larger the voltage ratio, the closer the instantaneous amplitude of the grid voltage is to the DC bus voltage reference value; the smaller the voltage ratio, the lower the instantaneous amplitude of the grid voltage.

[0090] In step S730, the digital controller calculates the voltage feedforward based on the voltage proportional quantity. The voltage feedforward is used to provide pre-compensation to the duty cycle generation process based on the instantaneous amplitude of the AC grid voltage. For boost-type power factor correction control, when the instantaneous amplitude of the grid voltage is low, a larger pulse width modulation (PWM) duty cycle is typically required to maintain energy transfer to the DC bus; when the instantaneous amplitude of the grid voltage is high, a smaller PWM duty cycle is typically required. Therefore, the voltage feedforward can be set to decrease as the voltage proportional quantity increases and increase as the voltage proportional quantity decreases.

[0091] like Figure 4 As shown, in some embodiments, the voltage feedforward is calculated based on the voltage proportionality, which may include, but is not limited to, steps S731 to S733: S731. Obtain the voltage feedforward base quantity based on the difference between the preset normalized reference value and the voltage proportional quantity. S732. Obtain the initial voltage feedforward quantity by multiplying the voltage feedforward base quantity and the preset feedforward coefficient; S733. Compare the initial voltage feedforward with the preset maximum voltage feedforward, and determine the smaller of the two values ​​as the voltage feedforward.

[0092] In step S731, the digital controller obtains the voltage feedforward base quantity based on the difference between the preset normalized reference value and the voltage proportional quantity. The preset normalized reference value is used to represent the upper limit or reference level in the normalization calculation. In one specific implementation, the preset normalized reference value can be set to 1.0, which means that the DC bus voltage reference value is used as the reference. When the absolute value of the delayed grid voltage instantaneous value is close to the DC bus voltage reference value, the voltage proportional quantity is close to the preset normalized reference value, and the voltage feedforward base quantity decreases accordingly; when the absolute value of the delayed grid voltage instantaneous value is low, the voltage proportional quantity is small, and the voltage feedforward base quantity increases accordingly. In one implementation, the voltage feedforward base quantity can be expressed as:

[0093] in, B ff This represents the basic quantity of voltage feedforward. R v This represents the voltage proportionality, and 1.0 represents the preset normalized reference value. This expression reflects the basic relationship between the duty cycle and the instantaneous voltage amplitude on the AC side in boost control, that is, the lower the instantaneous voltage amplitude on the AC side, the larger the required feedforward duty cycle reference.

[0094] In step S732, the digital controller obtains the initial voltage feedforward quantity based on the product of the voltage feedforward base quantity and the preset feedforward coefficient. The preset feedforward coefficient is used to adjust the influence of the voltage feedforward base quantity on the duty cycle generation process. If the preset feedforward coefficient is large, the voltage feedforward quantity responds more strongly to instantaneous changes in the grid voltage, reducing the adjustment amount required by the PI controller; if the preset feedforward coefficient is small, the voltage feedforward effect is weak, and the system relies more on the current loop closed-loop regulation. The preset feedforward coefficient can be tuned according to the rated voltage, switching frequency, current loop bandwidth, PI controller parameters, and current waveform distortion requirements of the totem-pole bridgeless power factor correction converter. In one implementation, the initial voltage feedforward quantity can be expressed as: ,in, U ff0This represents the initial voltage feedforward. k ff This represents the preset feedforward coefficient. R v It represents a proportional voltage.

[0095] In step S733, the digital controller compares the initial voltage feedforward with the preset maximum voltage feedforward and determines the smaller of the two values ​​as the voltage feedforward. The preset maximum voltage feedforward is used to limit the maximum value of the voltage feedforward, preventing the voltage feedforward from being too large near the zero-crossing of the grid voltage or when the voltage ratio is small, thus avoiding the synthesized control quantity from approaching or exceeding the upper limit of the allowable duty cycle. This limiting process can improve the safety and stability of the duty cycle generation process. In one implementation, the voltage feedforward can be expressed as: ,in, U ff This represents the voltage feedforward quantity. U ff0 This represents the initial voltage feedforward. U ffmax This represents the preset maximum voltage feedforward amount, and min(•) indicates taking the smaller value. In one specific implementation, the preset maximum voltage feedforward amount can be the same as or close to the preset feedforward coefficient, for example, set to 0.80, so that the voltage feedforward amount does not exceed the preset duty cycle feedforward upper limit. The preset maximum voltage feedforward amount can also be adjusted according to different power levels, different grid voltage levels, or different startup stages.

[0096] It should be noted that when the voltage proportional quantity exceeds the preset normalized reference value due to grid disturbances, sampling deviations, or a low DC bus voltage reference value, the initial voltage feedforward obtained according to the above relationship may be negative. In practical engineering implementation, this negative value can be retained to reduce the synthesized control quantity, depending on the control strategy. Alternatively, a lower limit can be set to ensure that the voltage feedforward is not less than zero. This lower limit does not change the basic idea of ​​this embodiment: calculating the voltage feedforward based on the voltage proportional quantity and limiting its maximum value.

[0097] In step S740, the digital controller obtains the synthesized control quantity based on the current loop regulation and the voltage feedforward. The current loop regulation is the closed-loop regulation result output by the PI controller based on the current control error, mainly used to compensate for the deviation between the current reference and the current feedback; the voltage feedforward is the open-loop pre-compensation result obtained based on the delayed instantaneous grid voltage value and the DC bus voltage reference value, mainly used to provide a duty cycle reference that varies with the instantaneous amplitude of the grid voltage. Combining the two allows the control quantity to simultaneously include feedforward prediction components and closed-loop correction components. In one implementation, the synthesized control quantity can be expressed as: ,in, U sumIndicates the synthetic control quantity. U i This indicates the current loop adjustment amount. U ff This represents the voltage feedforward quantity. Since the voltage feedforward quantity already provides the main duty cycle variation trend, the current loop regulation quantity can be mainly used to correct deviations caused by factors such as inductor parameter errors, sampling errors, load disturbances, voltage drops of switching devices, and control delays. This can improve the dynamic response speed of the current loop while maintaining closed-loop stability.

[0098] In step S750, the digital controller performs amplitude limiting on the synthesized control quantity to obtain the pulse width modulation duty cycle of the current control cycle. The amplitude limiting process restricts the synthesized control quantity to the range allowed by the high-frequency switching devices and pulse width modulation module of the totem-pole bridgeless power factor correction converter. If the synthesized control quantity is too small, the high-frequency switching devices may fail to form effective inductor energy storage due to insufficient conduction time, or the pulse width modulation pulse may be smaller than the minimum effective pulse width of the drive circuit. If the synthesized control quantity is too large, it may lead to excessive boost inductor current, DC bus overvoltage, increased stress on the switching devices, or insufficient control margin. Therefore, amplitude limiting can improve the safety of power stage operation.

[0099] In one implementation, the pulse width modulation duty cycle of the current control cycle can be expressed as: ,in, Duty now This indicates the pulse width modulation duty cycle of the current control cycle. U sum This represents the synthesized control quantity, and `sat(•)` indicates the limiting process. The upper and lower limits of the limiting process can be determined based on the minimum and maximum on-time of the switching devices, the dead time, the propagation delay of the drive circuit, the saturation current of the boost inductor, the safe range of the DC bus voltage, and the resolution of the pulse width modulation module. In one specific implementation, the pulse width modulation duty cycle of the current control cycle can be limited to a range close to 0 to close to 1, for example, limited to a safe range greater than 0 and less than 1, to avoid extreme duty cycles causing power stage anomalies.

[0100] After the pulse width modulation (PWM) duty cycle for the current control cycle is generated, the digital controller can write it into the comparison register or duty cycle register of the PWM module, enabling the PWM module to output the corresponding drive signal in the current control cycle or at the next update time. The PWM duty cycle of the current control cycle can also be saved and used in the next control cycle as the PWM duty cycle of the previous control cycle in the calculation of the current correction coefficient, thus forming a closed-loop correlation between current sampling correction and duty cycle control.

[0101] Through steps S710 to S750 and S731 to S733, this embodiment introduces a voltage feedforward based on the delayed instantaneous value of the grid voltage and the reference value of the DC bus voltage, in addition to the current loop regulation. This voltage feedforward can vary with the AC grid voltage over one power frequency cycle, providing a larger duty cycle reference when the instantaneous amplitude of the grid voltage is low and a smaller duty cycle reference when the instantaneous amplitude of the grid voltage is high. Therefore, the PI controller does not need to independently undertake the entire duty cycle regulation task, which helps reduce current control error, improves the dynamic response near the current zero crossing, and enhances the current waveform quality of the totem-pole bridgeless power factor correction converter.

[0102] Understandably, in some embodiments, the totem-pole bridgeless power factor correction converter can operate in either PFC rectification mode or inverter inversion mode. PFC rectification mode refers to the operating mode where power primarily flows from the AC side to the DC side. In this mode, the totem-pole bridgeless power factor correction converter absorbs energy from the AC grid and maintains the DC bus voltage within the target range. Inverter inversion mode refers to the operating mode where power primarily flows from the DC side to the AC side. In this mode, the totem-pole bridgeless power factor correction converter can feed energy from the DC bus side back to the AC side or supply power to AC loads. Because the totem-pole bridgeless structure has a bidirectional conduction and bidirectional energy transfer topology, the same power stage can achieve both operating modes under different control commands.

[0103] Regardless of whether the totem-pole bridgeless power factor correction converter is in PFC rectification mode or inverter inverter mode, the aforementioned current sampling adaptive correction process can be executed. This means that the instantaneous values ​​of the grid voltage, DC bus voltage, and inductor current for the current control cycle, as well as the pulse width modulation duty cycle of the previous control cycle, can be obtained. Furthermore, the voltage difference is determined, the current correction coefficient is calculated, the equivalent current feedback value is obtained, the current control error is determined, the current loop regulation is calculated, and the pulse width modulation duty cycle for the current control cycle is generated. In other words, the current feedback correction mechanism based on the current correction coefficient can be shared between PFC rectification mode and inverter inverter mode, without needing to set different current feedback paths or controller parameters for continuous conduction and intermittent conduction states.

[0104] Specifically, when the totem-pole bridgeless power factor correction converter is in PFC rectification mode, the digital controller acquires a first AC current reference value that is in phase with the instantaneous grid voltage value, which serves as the AC current reference value. The first AC current reference value being in phase with the instantaneous grid voltage value means that when the instantaneous grid voltage value is in the positive half-cycle, the first AC current reference value corresponds to a positive current reference; when the instantaneous grid voltage value is in the negative half-cycle, the first AC current reference value corresponds to a negative current reference; and the zero-crossing time and waveform phase of the first AC current reference value are basically consistent with the instantaneous grid voltage value. By adopting a first AC current reference value that is in phase with the instantaneous grid voltage value, the AC side current can be controlled to be basically in phase with the grid voltage, thereby enabling the totem-pole bridgeless power factor correction converter to absorb active power from the AC side with a higher power factor. In one implementation, the first AC current reference value can be obtained by multiplying the current amplitude by a grid synchronization sinusoidal quantity. When the current amplitude is positive in PFC rectification mode and the grid synchronization sinusoidal quantity is in phase with the instantaneous grid voltage value, the obtained first AC current reference value is also in phase with the instantaneous grid voltage value. The current amplitude can be generated by the DC bus voltage regulation loop and used to adjust the AC side power absorption based on the deviation between the DC bus voltage and the target value; the grid synchronization sinusoidal quantity can be generated by the phase-locked loop based on the instantaneous grid voltage value and used to provide a phase reference synchronized with the grid voltage. Therefore, the AC current reference value in PFC rectification mode can take into account both DC bus voltage regulation and AC side power factor correction.

[0105] When the totem-pole bridgeless power factor correction converter is in inverter mode, the digital controller acquires a second AC current reference value that is out of phase with the instantaneous grid voltage value, which serves as the AC current reference value. The second AC current reference value being out of phase with the instantaneous grid voltage value means that when the instantaneous grid voltage value is in the positive half-cycle, the second AC current reference value corresponds to a negative current reference; when the instantaneous grid voltage value is in the negative half-cycle, the second AC current reference value corresponds to a positive current reference; and the second AC current reference value has a phase difference of approximately 180 degrees relative to the instantaneous grid voltage value. By using a second AC current reference value that is out of phase with the instantaneous grid voltage value, the power direction can be shifted from the DC side to the AC side, thereby achieving inverter output or energy feedback. In one implementation, the second AC current reference value can also be obtained by multiplying the current amplitude and the grid synchronization sinusoidal quantity. When the current amplitude is negative in inverter mode, while the grid synchronization sinusoidal quantity is still in phase with the instantaneous grid voltage value, the second AC current reference value obtained by multiplying the two is out of phase with the instantaneous grid voltage value. In other words, in both PFC rectification mode and inverter inverter mode, the same grid synchronization sinusoidal quantity generation mechanism can be used. By simply changing the sign of the current amplitude or altering the phase command, in-phase or out-of-phase AC current reference values ​​can be obtained. In inverter inverter mode, the current amplitude can be determined by DC-side energy management commands, power output commands, DC bus voltage deviation, or energy storage system control commands.

[0106] It should be noted that although the phase relationship of the AC current reference value differs between PFC rectification mode and inverter inverter mode, the method for determining the current control error based on the absolute value of the AC current reference value and the equivalent current feedback value can remain consistent. For example, the digital controller can take the absolute value of either the first or second AC current reference value to obtain the desired current amplitude, and compare it with the equivalent current feedback value obtained by correction using a current correction coefficient. In this way, the current loop can perform closed-loop control based on the current amplitude in both power flow directions, while the current direction and power direction are jointly determined by the phase of the AC current reference value, the bridge arm modulation logic, and the power frequency commutation logic.

[0107] Since the current correction coefficient is calculated based on the instantaneous grid voltage, DC bus voltage, and the pulse width modulation duty cycle of the previous control cycle, it primarily reflects the current sampling correction requirements of the totem-pole bridgeless power factor correction converter under current voltage conditions and switching states. This correction requirement is closely related to the continuity of the boost inductor current, the instantaneous amplitude of the grid voltage, and the bus voltage state, but does not depend on the power flow direction itself. Therefore, the same current correction coefficient calculation method can be used to correct the absolute value of the instantaneous inductor current in both PFC rectification mode and inverter inverter mode, thereby obtaining an equivalent current feedback value under a unified scale.

[0108] In practical control, the digital controller can determine whether it is currently operating in PFC rectification mode or inverter inverter mode based on mode commands, power commands, DC bus voltage status, or external communication commands. Once the operating mode is determined, the digital controller selects either a first or second AC current reference value as the AC current reference value and continues executing the aforementioned current loop process. During mode switching, the current amplitude can be ramped up or speed-limited to ensure a smooth transition between the first and second AC current reference values, avoiding inductor current surges caused by sudden changes in the AC current reference value. Simultaneously, the calculation of the current correction coefficient, the generation of the equivalent current feedback value, and the PI controller parameters can remain unchanged during switching to reduce transient disturbances caused by control structure switching.

[0109] Through the current sampling adaptive correction method implemented in this embodiment, the totem-pole bridgeless power factor correction converter can achieve rectified power factor correction using a first AC current reference value that is in phase with the instantaneous grid voltage value in PFC rectification mode, and achieve inverter power output using a second AC current reference value that is out of phase with the instantaneous grid voltage value in inverter inverter mode. Since the current sampling adaptive correction, current control error calculation, and current loop regulation process can be shared in both modes, it helps to simplify the software architecture of the bidirectional AC-DC converter, reduce the problem of discontinuous control parameters during mode switching, and improve the consistency of current control under mixed operation conditions of continuous and intermittent conduction states.

[0110] Secondly, embodiments of this application provide an electronic device. The electronic device includes: at least one processor; and a memory communicatively connected to the at least one processor; wherein the memory stores a computer program executable by the at least one processor, and when the computer program is executed, it implements the current sampling adaptive correction method of any embodiment of the first aspect.

[0111] The processor and memory can be connected via a bus or other means.

[0112] Memory, as a non-transitory computer-readable storage medium, can be used to store non-transitory software programs and non-transitory computer-executable programs, such as the current sampling adaptive correction method of any embodiment of the first aspect of this application. The processor implements the above-described method by running the non-transitory software program and instructions stored in the memory.

[0113] The memory may include a program storage area and a data storage area, wherein the program storage area may store the operating system and application programs required for at least one function; the data storage area may store the methods described above. Furthermore, the memory may include high-speed random access memory and may also include non-transitory memory, such as at least one storage device, flash memory, or other non-transitory solid-state memory. In some embodiments, the memory may optionally include memory remotely located relative to the processor, which can be connected to the electronic device via a network. Examples of such networks include, but are not limited to, the Internet, intranets, local area networks, mobile communication networks, and combinations thereof.

[0114] The non-transient software program and instructions required to implement the current sampling adaptive correction method of any of the first aspects are stored in memory. When executed by one or more processors, the current sampling adaptive correction method of any of the first aspects is executed.

[0115] The device embodiments described above are merely illustrative. The units described as separate components may or may not be physically separate; that is, they may be located in one place or distributed across multiple network units. Some or all of the modules can be selected to achieve the purpose of this embodiment according to actual needs.

[0116] The embodiments of this application have been described in detail above with reference to the accompanying drawings. However, this application is not limited to the above embodiments. Within the scope of knowledge possessed by those skilled in the art, various changes can be made without departing from the spirit of this application. Furthermore, unless otherwise specified, the embodiments and features described in the embodiments of this application can be combined with each other.

Claims

1. A current sampling adaptive correction method, characterized in that, Applied to totem-pole bridgeless power factor correction converters; the method includes: Obtain the instantaneous values ​​of the grid voltage, DC bus voltage, and inductor current for the current control cycle, as well as the pulse width modulation duty cycle for the previous control cycle; The voltage difference is determined based on the instantaneous value of the grid voltage and the DC bus voltage; The current correction coefficient is calculated based on the voltage difference, the DC bus voltage, and the pulse width modulation duty cycle of the previous control cycle. Based on the current correction coefficient, the absolute value of the instantaneous value of the inductor current is adaptively proportionally corrected to obtain the equivalent current feedback value. Obtain an AC current reference value, and determine the current control error based on the absolute value of the AC current reference value and the equivalent current feedback value; The current loop adjustment amount is calculated based on the current control error. Based on the current loop adjustment, the pulse width modulation duty cycle of the current control cycle is obtained to control the high-frequency switching devices of the totem pole bridgeless power factor correction converter.

2. The current sampling adaptive correction method according to claim 1, characterized in that, The step of determining the voltage difference based on the instantaneous value of the grid voltage and the DC bus voltage includes: The absolute value of the grid voltage is obtained by taking the absolute value of the instantaneous grid voltage. Calculate the difference between the absolute values ​​of the DC bus voltage and the grid voltage to obtain the initial voltage difference; The initial voltage difference is compared with a preset minimum voltage difference, and the larger of the two values ​​is determined as the voltage difference.

3. The current sampling adaptive correction method according to claim 1, characterized in that, The current correction coefficient is calculated based on the voltage difference, the DC bus voltage, and the pulse width modulation duty cycle of the previous control cycle, including: The intermediate current correction quantity is obtained by calculating the product of the DC bus voltage and the pulse width modulation duty cycle of the previous control cycle. Calculate the ratio of the intermediate current correction value to the voltage difference to obtain the initial current correction coefficient; The initial current correction coefficient is compared with the preset maximum correction coefficient, and the smaller of the two values ​​is determined as the current correction coefficient.

4. The current sampling adaptive correction method according to claim 1, characterized in that, The step of adaptively proportionally correcting the absolute value of the instantaneous inductor current based on the current correction coefficient to obtain the equivalent current feedback value includes: The absolute value of the inductor current is obtained by taking the absolute value of the instantaneous value of the inductor current. The equivalent current feedback value is obtained by multiplying the current correction coefficient by the absolute value of the inductor current.

5. The current sampling adaptive correction method according to claim 1, characterized in that, The acquisition of the AC current reference value includes: Obtain the current amplitude and the grid synchronization sinusoidal quantity; An AC current reference value is generated based on the current amplitude and the grid synchronization sinusoidal quantity.

6. The current sampling adaptive correction method according to claim 1, characterized in that, The step of calculating the current loop adjustment amount based on the current control error includes: Obtain the full-scale current value; The current control error is normalized based on the full-scale value of the current to obtain the normalized current control error. The normalized current control error is input into the PI controller to obtain the current loop adjustment amount.

7. The current sampling adaptive correction method according to claim 1, characterized in that, The step of obtaining the pulse width modulation duty cycle of the current control cycle based on the current loop adjustment includes: Obtain the reference value of the DC bus voltage and the instantaneous value of the delayed grid voltage, wherein the instantaneous value of the delayed grid voltage is the instantaneous grid voltage value delayed by one control cycle relative to the current control cycle; The voltage ratio is obtained by comparing the absolute value of the instantaneous value of the delayed grid voltage with the reference value of the DC bus voltage. The voltage feedforward is calculated based on the voltage proportionality. The synthesized control quantity is obtained based on the current loop adjustment amount and the voltage feedforward amount; The synthesized control quantity is subjected to amplitude limiting processing to obtain the pulse width modulation duty cycle of the current control cycle.

8. The current sampling adaptive correction method according to claim 7, characterized in that, The calculation of the voltage feedforward based on the voltage proportionality includes: The voltage feedforward base quantity is obtained based on the difference between the preset normalized reference value and the voltage proportional quantity. The initial voltage feedforward quantity is obtained by multiplying the voltage feedforward base quantity by the preset feedforward coefficient; The initial voltage feedforward is compared with the preset maximum voltage feedforward, and the smaller of the two values ​​is determined as the voltage feedforward.

9. The current sampling adaptive correction method according to any one of claims 1 to 8, characterized in that, The totem pole bridgeless power factor correction converter operates in either PFC rectification mode or inverter inverter mode. The acquisition of the AC current reference value includes: When the totem-pole bridgeless power factor correction converter is in PFC rectification mode, a first AC current reference value that is in phase with the instantaneous value of the grid voltage is obtained and used as the AC current reference value. When the totem-pole bridgeless power factor correction converter is in inverter mode, a second AC current reference value that is inversely phase to the instantaneous value of the grid voltage is obtained and used as the AC current reference value.

10. An electronic device, characterized in that, It includes a processor and a memory, the memory storing a computer program that, when executed by the processor, implements the current sampling adaptive correction method as described in any one of claims 1 to 9.

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