Mine equipment built-in permanent magnet motor position sensorless control method and device

Abstract

The application discloses a kind of mine equipment with built-in permanent magnet motor position sensorless control method and device.The method obtains the estimated value of stator flux linkage based on the third-order generalized integrator hybrid flux linkage observer, and obtains the rotor position estimation value in combination with effective flux linkage model and phase-locked loop;At the same time, according to the relationship between stator current and rotor position error, an adaptive minimum current tracking compensation link is constructed, the rotor position compensation angle is generated online, and the rotor electric angle estimation value output by the phase-locked loop is corrected.The method can suppress the direct current bias and harmonic disturbance caused by measurement noise, inverter nonlinearity and parameter mismatch, and compensate the position estimation error caused by factors such as digital control delay, loop filter and motor parameter mismatch, thereby improving the estimation accuracy, dynamic performance and operation reliability of the coal mine underground transport equipment permanent magnet motor position sensorless control system.

Description

Technical Field

[0001] This invention belongs to the field of permanent magnet motor control, and particularly relates to a sensorless control method and device for a built-in permanent magnet motor used in mining equipment. Background Technology

[0002] Coal mine underground transportation equipment typically operates in dusty, humid, highly vibrating, and shock-prone environments with complex electromagnetic interference and limited maintenance space. This places high demands on the reliability, anti-interference capabilities, and low-maintenance characteristics of the drive motor control system. Using mechanical position sensors would not only increase system costs and installation and maintenance difficulty, but would also make the system more susceptible to vibration, contamination, and electromagnetic interference under harsh underground conditions, thus reducing system reliability.

[0003] Permanent magnet motors (PMMs) are widely used in industrial drives, rail transportation, aerospace, and new energy vehicles due to their advantages such as high power density, high efficiency, and simple structure. High-performance control of PMMs relies on accurate rotor position and speed information. Traditional control systems typically obtain rotor position information through mechanical position sensors such as photoelectric encoders, rotary transformers, or Hall effect sensors. However, mechanical position sensors increase system cost and size, and their reliability decreases in harsh environments such as high temperature, high humidity, strong vibration, and strong electromagnetic interference. Therefore, sensorless control technology for PMMs is of significant research importance.

[0004] In sensorless control methods for permanent magnet motors, the flux linkage observer method has attracted widespread attention due to its advantages such as simple structure, low computational cost, and ease of implementation. A hybrid flux linkage observer typically consists of a voltage model and a current model. The voltage model is used to reconstruct the flux linkage based on the stator voltage and stator current, while the current model is used to provide feedback correction to the voltage model output. Traditional voltage models often employ structures such as pure integrators, equivalent low-pass filters, or second-order generalized integrators. However, pure integrators are susceptible to integral saturation due to DC bias, equivalent low-pass filters cause amplitude attenuation and phase shift in the estimated flux linkage, and second-order generalized integrators have insufficient suppression capabilities against DC bias and low-frequency disturbances. Therefore, traditional hybrid flux linkage observers are prone to flux linkage observation errors and rotor position estimation errors when factors such as measurement noise, inverter nonlinearity, and parameter mismatch exist. In practical applications, DC bias and harmonics are unavoidable in the observer input, resulting in insufficient disturbance rejection capabilities of traditional hybrid flux linkage observers.

[0005] Furthermore, in sensorless control systems, non-ideal factors such as digital control delay, loop filters, and motor parameter mismatch can also lead to deviations in rotor position estimation. Position estimation errors result in inaccurate current decomposition, leading to current fluctuations, torque pulsation, increased motor losses, and decreased efficiency. While traditional minimum current tracking compensation methods do not rely on precise motor parameters, they are typically applicable to surface-mounted permanent magnet motors, and the fixed step size approach struggles to balance rapid compensation with steady-state oscillation-free operation.

[0006] In summary, improving the ability of the hybrid flux observer to suppress DC bias and harmonic disturbances, and providing online compensation for rotor position estimation errors in sensorless control, are urgent problems to be solved in achieving high-precision sensorless control of permanent magnet motors. Summary of the Invention

[0007] The purpose of this invention is to provide a sensorless control method and device for a built-in permanent magnet motor used in mining equipment, which solves the problems of insufficient reliability of mechanical position sensors, limited anti-interference capability of traditional flux linkage observers, and difficulty in online compensation of rotor position estimation errors in the permanent magnet motor drive system of underground coal mine transportation equipment.

[0008] This invention provides the following technical solution:

[0009] In a first aspect, the present invention provides a sensorless control method for a built-in permanent magnet motor used in mining equipment, the method comprising: Obtain the stator voltage and current in the αβ coordinate system of the built-in permanent magnet motor; The stator voltage and current are input into the hybrid flux observer, which uses a resonant integrator to suppress DC bias and harmonic components. The back electromotive force is corrected based on the deviation between the actual current and the estimated current before being input into the resonant integrator, and the estimated stator flux is output. Based on the estimated stator flux linkage and stator current, the effective flux linkage model, which includes direct-axis and quadrature-axis inductance parameters, is used to eliminate the salient pole effect caused by inductance asymmetry, thereby obtaining the estimated effective flux linkage. The effective flux linkage estimate is input into the phase-locked loop to obtain the preliminary rotor electrical angle and rotor angular velocity; Using the initial rotor electrical angle as the initial search center, without injecting an external high-frequency excitation signal, and with the goal of minimizing the current amplitude, the rotor position compensation angle is generated through iterative optimization using a dual-state machine. The dual-state machine includes a tracking state and a disturbed state. In the tracking state, the tracking step size decreases as the current amplitude decreases. In the disturbed state, the step size is forcibly decayed until the minimum tracking step size is reached. The rotor position compensation angle is superimposed on the initial rotor electrical angle to obtain the compensated electrical angle; Vector control is performed on the built-in permanent magnet motor based on the compensated electrical angle and the rotor angular velocity.

[0010] In one implementation, the corrected back electromotive force is: , in, This is the back electromotive force after feedback correction. Stator voltage, For stator resistance, This is the actual stator current. The estimated stator current is obtained from the current model calculation. This is the feedback correction coefficient.

[0011] In one embodiment, the resonant integrator is a third-order generalized integrator with a continuous-domain transfer function as follows: , , In the formula, For in-phase signal transfer function, For orthogonal signal transfer functions, For the feedback-corrected back electromotive force e s Laplace transform, This is the in-phase output signal of the generalized integrator. This is the quadrature output signal of the generalized integrator. Forward gain coefficient For feedback gain coefficient, The center angular frequency, For the Laplace operator; The third-order generalized integrator is discretized using a bilinear transform, and the bilinear transform relationship is as follows: , Define intermediate variables: , The discrete realization coefficients are obtained as follows: , In the formula, The sampling period is For discrete domain variables, The discrete equivalent operator obtained by bilinear transformation. , , and The coefficients are intermediate variables used to simplify discrete expressions. , and The feedback coefficients are those of the discrete implementation of the generalized integrator. These are the input coefficients for the discrete implementation of the generalized integrator.

[0012] In one implementation, the effective flux linkage estimate is: , In the formula, , These are the effective flux linkage estimates in the αβ coordinate system, respectively. , These are the estimated values ​​of the stator flux linkage in the αβ coordinate system, respectively. It is the q-axis inductance. , These are the stator currents in the αβ coordinate system, respectively; The compensated electrical angle is: , in, For the compensated rotor electrical angle, This is the estimated rotor electrical angle output by the phase-locked loop. The rotor position compensation angle generated for iterative optimization.

[0013] In one implementation, the process of iteratively optimizing to generate the rotor position compensation angle includes: The stator current amplitude is sampled or filtered to obtain the stator current amplitude for the current compensation cycle. Calculate the change in stator current amplitude based on the stator current amplitude of the current compensation cycle and the stator current amplitude of the previous compensation cycle. The update direction of the compensation angle is determined based on the change in stator current amplitude and the change direction of the compensation angle in adjacent compensation cycles. The tracking step size is determined by the absolute value of the ratio of the change in stator current amplitude to the change in compensation angle. The rotor position compensation angle is updated based on the update direction of the compensation angle and the tracking step size.

[0014] In one implementation, the rule for determining the compensation angle update direction is: When the stator current amplitude decreases after the compensation angle is updated along the current direction, the current compensation direction is maintained. When the stator current amplitude increases after the compensation angle is updated along the current direction, the compensation direction is changed.

[0015] In one implementation, the state switching logic of the dual-state machine is as follows: The state direction variable is determined based on the change in stator current amplitude and the change in compensation angle, and the continuous direction is calculated. t The absolute value of the sum of the state direction variables within each compensation cycle, where,t It is an integer greater than 3; When the absolute value is greater than or equal to t At that time, the dual-state machine is in the tracking state; When the absolute value is less than t At that time, the dual-state machine is in a disturbed state.

[0016] In one implementation, during the tracking state, the tracking step size satisfies: , Or meet the requirements of the limited-amplitude format: , In the formula, For the first The tracking step size for each compensation cycle. For adaptive step-size gain coefficient, This represents the change in stator current amplitude during the previous compensation cycle. This represents the change in compensation angle in the previous compensation cycle. The ratio of these two values ​​is used to characterize the local slope of the stator current amplitude as a function of the compensation angle. For the maximum tracking step size, This is the minimum tracking step size.

[0017] Secondly, the present invention provides a sensorless control device for a built-in permanent magnet motor in mining equipment to implement the above-mentioned method, the device comprising: The acquisition module is used to acquire the stator voltage and current in the αβ coordinate system of the built-in permanent magnet motor; The hybrid flux linkage observation module is used to input stator voltage and current into the hybrid flux linkage observer. The hybrid flux linkage observer uses a resonant integrator to suppress DC bias and harmonic components, and corrects the back electromotive force according to the deviation between the actual current and the estimated current before inputting it into the resonant integrator to output the estimated stator flux linkage value. The rotor position estimation module is used to eliminate the salient pole effect caused by inductance asymmetry by using an effective flux model that includes direct-axis and quadrature-axis inductance parameters, based on the stator flux estimate and stator current, to obtain the effective flux estimate, and to obtain the preliminary rotor electrical angle and angular velocity through a phase-locked loop; The minimum current tracking module is used to generate the rotor position compensation angle by iterative optimization through a dual-state machine, with the initial rotor electrical angle as the initial search center and without injecting an external high-frequency excitation signal, aiming to minimize the current amplitude. The dual-state machine includes a tracking state and a disturbed state. In the tracking state, the tracking step size decreases as the current amplitude decreases. In the disturbed state, the step size is forcibly decayed until the minimum tracking step size is reached. The compensation module is used to superimpose the rotor position compensation angle onto the initial rotor electrical angle to obtain the compensated electrical angle. The motor control module is used to perform vector control on the built-in permanent magnet motor based on the compensated electrical angle and the rotor angular velocity.

[0018] Thirdly, the present invention provides a sensorless control system for a built-in permanent magnet motor for mining equipment, including the aforementioned sensorless control device for a built-in permanent magnet motor for mining equipment.

[0019] Beneficial effects of this invention:

[0020] (1) The present invention uses a third-order generalized integrator to replace the integrator in the traditional hybrid flux observer, which can effectively suppress the DC bias and harmonic disturbance in the input back EMF, and avoid problems such as easy integration saturation of pure integrator, easy amplitude attenuation and phase shift of equivalent low-pass filter, and insufficient DC bias suppression capability of second-order generalized integrator.

[0021] (2) The present invention obtains the stator flux estimate by using a hybrid flux observer based on a third-order generalized integrator and further constructs an effective flux for rotor position and speed estimation, which can improve the flux observation accuracy and position estimation accuracy in the sensorless control of the built-in permanent magnet motor for underground coal mine transportation equipment.

[0022] (3) The present invention uses an adaptive minimum current tracking method to compensate the rotor electrical angle estimate output by the phase-locked loop online, which can reduce the position estimation deviation caused by digital control delay, loop filter, motor parameter mismatch and other non-ideal factors.

[0023] (4) The present invention adaptively adjusts the tracking step size according to the change of stator current, increases the compensation speed when the position error is large, and reduces steady-state oscillation when approaching the optimal compensation angle, thereby taking into account both the speed and stability of position compensation.

[0024] (5) This invention combines a hybrid flux observer based on a third-order generalized integrator with an adaptive minimum current tracking compensation circuit, which can simultaneously improve the system’s anti-disturbance performance and position error compensation capability, thereby improving the estimation accuracy, dynamic performance and operational reliability of the sensorless control system for permanent magnet motors in underground coal mine transportation equipment. Attached Figure Description

[0025] The accompanying drawings, as part of this invention, are provided to further illustrate the invention. The illustrative embodiments and descriptions of the invention are used to explain the invention, but do not constitute an undue limitation thereof. Clearly, the drawings described below are merely some embodiments, and those skilled in the art can obtain other drawings based on these drawings without any creative effort.

[0026] Figure 1 This is a flowchart of a sensorless control method for a built-in permanent magnet motor in mining equipment provided in one embodiment of the present invention.

[0027] Figure 2 This is a block diagram of a hybrid flux observer provided in one embodiment of the present invention;

[0028] Figure 3 This is a block diagram of a third-order generalized integrator structure provided in one embodiment of the present invention;

[0029] Figure 4 This is a diagram of the discretized structure of a third-order generalized integrator provided in one embodiment of the present invention;

[0030] Figure 5 This is a block diagram of a phase-locked loop structure provided in one embodiment of the present invention;

[0031] Figure 6 This is a schematic diagram illustrating the influence of rotor position error on maximum torque-current ratio control in one embodiment of the present invention;

[0032] Figure 7 This is a graph showing the relationship between stator current amplitude and rotor position error in one embodiment of the present invention.

[0033] Figure 8 A flowchart illustrating the generation of rotor position compensation angle using the adaptive minimum current tracking method provided in one embodiment of the present invention;

[0034] Figure 9 This is a block diagram of a sensorless control system for a built-in permanent magnet motor used in mining equipment, provided in one embodiment of the present invention.

[0035] Figure 10 This is an experimental comparison diagram of different position compensation methods provided in one embodiment of the present invention, wherein, Figure 10 (a) shows the experimental results of the fixed step size MCT method with a tracking step size of 5°. Figure 10 (b) shows the experimental results of the fixed-step MCT method with a tracking step size of 0.25°. Figure 10 (c) shows the experimental results of the state-switching MCT method. Figure 10 In the middle (d), the experimental results of the adaptive MCT method are shown.

[0036] It should be noted that these accompanying drawings and textual descriptions are not intended to limit the scope of the invention in any way, but rather to illustrate the concept of the invention to those skilled in the art by referring to specific embodiments. Detailed Implementation

[0037] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments will be clearly and completely described below with reference to the accompanying drawings. The following embodiments are used to illustrate the present invention, but are not intended to limit the scope of the present invention.

[0038] To address the issues in sensorless control of permanent magnet motors in underground coal mine transportation equipment, where traditional hybrid flux observers are susceptible to DC bias and harmonic disturbances leading to decreased flux observation accuracy, and where non-ideal factors such as digital control delay, loop filter, and motor parameter mismatch make it difficult to compensate for rotor position estimation errors online, this invention proposes a sensorless control method for embedded permanent magnet motors in mining equipment.

[0039] Reference Figure 1 As shown in the figure, the present invention discloses a sensorless control method for a built-in permanent magnet motor used in mining equipment, which includes the following steps:

[0040] Step S100: Obtain the stator voltage and current in the αβ coordinate system of the built-in permanent magnet motor.

[0041] Step S200: Input the stator voltage and current into the hybrid flux observer. The hybrid flux observer uses a resonant integrator to suppress DC bias and harmonic components. After correcting the back electromotive force based on the deviation between the actual current and the estimated current, it is input into the resonant integrator to output the estimated value of the stator flux.

[0042] The resonant integrator is a third-order generalized integrator (TOGI), as shown in the reference. Figure 2 As shown, the hybrid flux observer based on the third-order generalized integrator includes: a TOGI voltage model, a current model feedback correction stage, an effective flux calculation stage, and a phase-locked loop stage.

[0043] The TOGI voltage model takes the stator voltage and current signals in the αβ coordinate system as inputs, performs flux linkage integration based on the corrected back electromotive force, and outputs the estimated stator flux linkage in the αβ coordinate system. The effective flux linkage calculation stage calculates the effective flux linkage estimate based on the stator flux linkage estimate and the stator current signal, and inputs the effective flux linkage estimate into the phase-locked loop (PLL) stage. The PLL stage constructs a position error signal based on the effective flux linkage estimate, obtains the rotor electrical angular velocity estimate through proportional-integral (PI) regulation, and then obtains the rotor electrical angle estimate through an integration stage. The rotor electrical angle estimate is fed back to the current model feedback correction stage, where the current model calculates the estimated stator current based on the stator flux linkage estimate and the rotor electrical angle estimate. Subsequently, the estimated stator current is compared with the actual stator current to obtain the current error signal, which is then corrected by a feedback correction coefficient. The weighted values ​​are used to generate a back EMF compensation. This compensation is then superimposed with the back EMF calculated by the voltage model to obtain the corrected back EMF, which is used as the input to the TOGI voltage model, thus forming a closed-loop feedback correction structure.

[0044] Furthermore, the corrected back electromotive force is:

[0045]

[0046] in, This is the back electromotive force after feedback correction. Stator voltage, For stator resistance, This is the actual stator current. The estimated stator current is obtained from the current model calculation. This is the feedback correction coefficient.

[0047] Figure 3 This is a block diagram of a third-order generalized integrator, used to replace the pure integrator in traditional voltage models. The third-order generalized integrator includes an error calculation stage, a proportional gain stage, a feedback gain stage, an integration stage, and a frequency adjustment stage. The input is a back electromotive force signal. The error signal is obtained by comparing it with the feedback signal. Error signal Two integration channels are input separately, one of which passes through the feedback gain coefficient sequentially. , center angular frequency And the points system One path is used to generate the feedback compensation signal; the other path passes through the forward gain coefficient. Addition and subtraction operations, center angular frequency And points process Obtain the in-phase output signal of the generalized integrator. At the same time, the output signal The frequency adjustment stage generates a quadrature output signal. And participate in error signal conditioning. Among them, This is the Laplace transform of the stator flux estimation signal output by the third-order generalized integrator.

[0048] As can be seen from the above structure, the third-order generalized integrator can generate mutually orthogonal output signals based on the input back electromotive force signal, and achieve the suppression of DC bias and harmonic disturbances in the input signal.

[0049] Furthermore, the continuous-domain transfer function of the third-order generalized integrator is:

[0050] ,

[0051] ,

[0052] In the formula, For in-phase signal transfer function, For orthogonal signal transfer functions, For the feedback-corrected back electromotive force e s Laplace transform, This is the in-phase output signal of the generalized integrator. This is the quadrature output signal of the generalized integrator. Forward gain coefficient For feedback gain coefficient, The center angular frequency, For the Laplace operator.

[0053] To facilitate the implementation of the TOGI structure in a digital controller, refer to Figure 4 As shown, the transfer function of the third-order generalized integrator is discretized using a bilinear transform. The discretized third-order generalized integrator includes an input signal processing stage, a delay stage, a feedback coefficient calculation stage, and an output combination stage. The feedback-corrected back electromotive force signal is used as an example. After inputting coefficients After weighting, the mixture enters the discrete filtering calculation stage, which involves a three-stage delay process. Historical output data is stored. The historical data is processed by feedback coefficients. , and The weighted signal is then added together with the input signal to obtain the output signal of the generalized integrator for the current sampling period. The output combination stage combines the current input signal and the historical feedback signal to obtain the in-phase output signal of the third-order generalized integrator. .coefficient , and Used to implement discrete feedback regulation, coefficient To achieve input signal gain adjustment, the above discrete structure realizes the equivalent implementation of a third-order generalized integrator in the continuous domain in a digital controller.

[0054] Furthermore, the third-order generalized integrator is discretized using a bilinear transform, and the bilinear transform relationship is as follows:

[0055]

[0056] Substituting this into the TOGI continuous-domain transfer function, we get:

[0057]

[0058]

[0059] To facilitate the implementation of the digital controller, intermediate variables are defined as follows:

[0060]

[0061] The discrete realization coefficients are obtained as follows:

[0062]

[0063] Therefore, the discrete-domain transfer function of TOGI can be expressed as:

[0064]

[0065]

[0066] In the formula, The sampling period is For discrete domain variables, The discrete equivalent operator obtained by bilinear transformation. , , and The coefficients are intermediate variables used to simplify discrete expressions. , and The feedback coefficients are those of the discrete implementation of the generalized integrator. These are the input coefficients for the discrete implementation of the generalized integrator.

[0067] Discrete TOGI calculations were performed on the feedback-corrected back electromotive force along the α-axis and β-axis respectively, and the following values ​​were taken: Figure 4 China Economic After the phase, multiply by the center angular frequency The previous output is used as the stator flux linkage estimate for the corresponding axis. When the center angular frequency is fixed and non-zero, the flux linkage estimate satisfies the quadrature output condition. .

[0068] Based on the above discretized expression, the digital controller can implement the TOGI voltage model through differential operations, thereby obtaining the stator flux linkage estimate. When the center angular frequency coincides with the input fundamental angular frequency, the continuous domain TOGI flux linkage channel exhibits the same amplitude and phase response at the fundamental frequency as an ideal integrator. The approximate accuracy of the discretized implementation is related to the sampling period. TOGI can also suppress DC bias and harmonic disturbances in the input back EMF, thereby improving the stator flux linkage estimation accuracy.

[0069] Step S300: Based on the stator flux estimate and stator current, use the effective flux model, which includes direct-axis and quadrature-axis inductance parameters, to eliminate the salient pole effect caused by inductance asymmetry and obtain the effective flux estimate.

[0070] For a built-in permanent magnet synchronous motor, the rotor position information is not only contained in the permanent magnet flux linkage, but also related to the stator flux linkage and the inductor salient polarity. To facilitate the extraction of rotor position using a flux linkage observer, this embodiment employs an effective flux linkage model.

[0071] Furthermore, the effective flux linkage estimate is:

[0072] ,

[0073] In the formula, , These are the effective flux linkage estimates in the αβ coordinate system, respectively. , These are the estimated values ​​of the stator flux linkage in the αβ coordinate system, respectively. It is the q-axis inductance. , These are the stator currents in the αβ coordinate system, respectively.

[0074] The compensated rotor electrical angle is:

[0075] ,

[0076] in, For the compensated rotor electrical angle, This is the estimated rotor electrical angle output by the phase-locked loop. The rotor position compensation angle generated for iterative optimization.

[0077] Step S400: Input the effective flux linkage estimate into the phase-locked loop to obtain the preliminary rotor electrical angle and rotor angular velocity.

[0078] Reference Figure 5 As shown, the phase-locked loop includes a sine wave operation stage, a cosine wave operation stage, a multiplication operation stage, an error calculation stage, a PI controller, and an integral stage. The two components of the effective flux linkage in the αβ coordinate system are respectively... and Rotor electrical angle estimation value The results are fed back to the sine and cosine calculation stages respectively, yielding... and .in, and Multiply, and The two product signals are multiplied and combined through an error calculation stage to obtain the phase-locked loop input error signal. Error signal The input is a PI controller, and the output of the PI controller is an estimated value of the rotor's electrical angular velocity. The rotor electric angular velocity estimate is integrated by a 1 / s element to obtain the rotor electric angle estimate. The estimated rotor electrical angle is fed back to the sine and cosine calculation stages, thus forming a closed-loop rotor position tracking structure.

[0079] The error signal of the phase-locked loop is expressed as:

[0080]

[0081] in, This is the input error signal for the phase-locked loop. This is the estimated value of the rotor electrical angle.

[0082] When the phase-locked loop enters a steady state, the error signal gradually approaches zero, the estimated angle gradually converges to the actual rotor electrical angle, the PI controller outputs the estimated electrical angular velocity value, and after integration, the estimated rotor electrical angle value is obtained.

[0083] Reference Figure 6 As shown, under maximum torque-to-current ratio (MTPA) control, when there is an error in the rotor position estimation, the synchronous rotating coordinate system established based on the estimated rotor position will deflect relative to the actual synchronous rotating coordinate system. Here, the d-axis and q-axis are the direct axis and quadrature axis of the actual synchronous rotating coordinate system, respectively. shaft and The axes are the direct axis and the quadrature axis of the estimated synchronous rotating coordinate system established based on the rotor position estimate. This represents the rotor position estimation error, which is also the angular deviation between the actual synchronous rotating coordinate system and the estimated synchronous rotating coordinate system. This represents the stator current vector corresponding to the ideal MTPA operating point when there is no rotor position estimation error. This represents the stator current vector corresponding to the offset MTPA operating point when there is a rotor position estimation error.

[0084] Since the coordinate transformation in MTPA control uses the rotor electrical angle estimate, when there is a rotor position estimation error, the estimation... The shaft cannot be accurately aligned with the actual rotor flux direction, causing the current vector to deviate relative to the ideal MTPA operating point. This angular deviation of the current vector is related to the rotor position estimation error. Correspondingly, under the same torque conditions, the stator current amplitude required when there is a rotor position estimation error is greater than the stator current amplitude required for the ideal MTPA operating point when there is no position error.

[0085] Reference Figure 7 As shown, under the motor parameters and operating conditions used in this embodiment, the relationship between the stator current amplitude and the rotor position estimation error is fitted. When the rotor position estimation error... When varying within the range of -30° to 30°, the resulting relationship is approximately:

[0086]

[0087] In the formula, This is the stator current amplitude, in amperes (A). The rotor position estimation error is expressed in degrees (°) in the above fitting relationship. Figure 7 It can be seen that the stator current amplitude has an approximately even function relationship with respect to the rotor position estimation error, and in The value is minimized in the vicinity of the rotor position estimation error. As the absolute value of the rotor position estimation error increases, the stator current amplitude increases.

[0088] Furthermore, to generalize the above relationship, the relationship between stator current and rotor position error is expressed as:

[0089]

[0090] in, This refers to the stator current when there is a position error. This refers to the stator current when there is no position error. For rotor position estimation error, These are the coefficients used to characterize the relationship between motor parameters and the influence of the maximum torque-current ratio on the operating point. The functional relationship is used to characterize the variation of stator current with rotor position error. The adaptive minimum current tracking method does not rely on... Instead of using the online identification value as the compensation input, the rotor position compensation angle is updated online based on the stator current change trend and the amount of stator current change. According to the above relationship, the stator current increases with the increase of the absolute value of the position error. Therefore, this embodiment uses the stator current change trend to determine the direction of the compensation angle adjustment and adaptively adjusts the tracking step size based on the amount of stator current change.

[0091] Step S500: Using the initial rotor electrical angle as the initial search center, without injecting an external high-frequency excitation signal, the rotor position compensation angle is generated through iterative optimization using a dual-state machine with the goal of minimizing the current amplitude.

[0092] The dual-state machine includes a tracking state and a disturbed state. In the tracking state, the tracking step size decreases as the current amplitude decreases. In the disturbed state, the step size is forcibly decayed until the minimum tracking step size is reached.

[0093] Specifically, the update direction of the compensation angle and the adaptive tracking step size are determined based on the change in stator current amplitude, and the rotor position compensation angle is updated accordingly. The TOGI observer follows the sampling period. Execution. The adaptive minimum current tracking algorithm is updated according to the compensation cycle. T comp Execute, and T comp = MT s ,in M The value is a positive integer. Each compensation cycle performs current amplitude acquisition, state judgment, tracking step size calculation, and compensation angle update, maintaining the compensation angle constant between adjacent update times. The compensation update cycle should allow sufficient time for the current response and filter output to stabilize after the previous compensation angle adjustment. (The compensation algorithm...) This indicates the compensation cycle number.

[0094] Furthermore, the process of iteratively optimizing and generating the rotor position compensation angle includes: Step S510: Sample or filter the stator current amplitude to obtain the stator current amplitude for the current compensation cycle.

[0095] Step S520: Calculate the change in stator current amplitude based on the stator current amplitude of the current compensation cycle and the stator current amplitude of the previous compensation cycle.

[0096] Step S530: Determine the update direction of the compensation angle based on the change in stator current amplitude and the change direction of the compensation angle in adjacent compensation cycles.

[0097] Furthermore, the rule for determining the direction of the compensation angle update is as follows:

[0098] When the stator current amplitude decreases after the compensation angle is updated along the current direction, the current compensation direction is maintained.

[0099] When the stator current amplitude increases after the compensation angle is updated along the current direction, the compensation direction is changed.

[0100] Step S540: Determine the tracking step size based on the absolute value of the ratio of the change in stator current amplitude to the change in compensation angle.

[0101] Furthermore, in tracking mode, the tracking step size satisfies:

[0102] Or meet the requirements of the limited-amplitude format:

[0103] In the formula, For the first The tracking step size for each compensation cycle. For adaptive step-size gain coefficient, This represents the change in stator current amplitude during the previous compensation cycle. This represents the change in compensation angle in the previous compensation cycle. The ratio of these two values ​​is used to characterize the local slope of the stator current amplitude as a function of the compensation angle. For the maximum tracking step size, This is the minimum tracking step size.

[0104] Step S550: Update the rotor position compensation angle according to the update direction of the compensation angle and the tracking step size.

[0105] Furthermore, the state transition logic of the dual-state machine is as follows: The state direction variable is determined based on the change in stator current amplitude and the change in compensation angle, and the continuous direction is calculated. t The absolute value of the sum of state direction variables within each compensation period, where, t It is an integer greater than 3; When the absolute value is greater than or equal to t At this time, the dual-state machine is in tracking state; When the absolute value is less than t At this time, the dual-state machine is in a disturbed state.

[0106] The stator current and rotor position error satisfy a preset functional relationship, which includes a relationship coefficient characterizing the influence of motor parameters on the maximum torque current ratio operating point. The adaptive minimum current tracking method does not use the online identification value of the relationship coefficient as the compensation input, but updates the rotor position compensation angle online based on the stator current change trend and the amount of stator current change.

[0107] Step S600: Superimpose the rotor position compensation angle onto the initial rotor electrical angle to obtain the compensated electrical angle.

[0108] Reference Figure 8 As shown, in step S500, the rotor position compensation angle is generated by the adaptive minimum current tracking method, which specifically includes initialization, state interval judgment, tracking step size update and compensation angle update process.

[0109] First, obtain the number Stator current in each compensation cycle Calculate its amplitude and initialize the adaptive step size gain coefficient. Decreasing step size and minimum tracking step size .

[0110] Subsequently, based on the stator current amplitude and rotor position compensation angle of adjacent compensation cycles, the stator current change and compensation angle change are calculated respectively:

[0111]

[0112]

[0113] In the formula, For the first The change in stator current amplitude over each compensation cycle For the first The change in compensation angle over one compensation cycle For the first Rotor position compensation angle for each compensation cycle.

[0114] according to and The sign relation determines the state direction variable. :when season Otherwise, let .

[0115] Further calculation of continuity The cumulative value of the state direction variable within each compensation cycle:

[0116]

[0117] In the formula, The preset number of state judgment cycles, and ; For continuous The absolute cumulative value of the state direction variable within each compensation cycle. When At that time, set the status flag bit Setting it to 0 indicates that the compensation algorithm is in a tracking state; when At that time, set the status flag bit Setting it to 1 indicates that the compensation algorithm is in a disturbed state.

[0118] In this embodiment, when At that time, the first step is adaptively determined based on the local slope of the stator current amplitude as a function of the compensation angle. The tracking step size for each compensation cycle is expressed as:

[0119]

[0120] In the formula, For the first The tracking step size for each compensation cycle. This is the adaptive step-size gain coefficient.

[0121] when At that time, the tracking step size of the previous compensation cycle is decreased according to the preset decreasing step size:

[0122]

[0123] when At that time, the tracking step size is limited to and order This is to keep the compensation algorithm in a disturbed state and avoid erroneous state switching due to sampling errors or external disturbances.

[0124] Finally, the rotor position compensation angle is updated based on the direction of change of stator current amplitude and the direction of change of compensation angle. At that time, according to Update the rotor position compensation angle; otherwise, follow the instructions. The process involves updating the compensation angle iteratively along the direction of decreasing stator current amplitude, gradually approaching the position that minimizes the stator current amplitude, thereby achieving adaptive compensation for rotor position estimation error.

[0125] In tracking mode, the system determines the adjustment direction of the compensation angle based on the changes in stator current amplitude and compensation angle within adjacent compensation cycles. When the stator current amplitude decreases after the compensation angle is updated along the current direction, the current compensation direction is maintained; when the stator current amplitude increases after the compensation angle is updated along the current direction, the compensation direction is changed. Simultaneously, the tracking step size is adaptively adjusted based on the degree of local change in stator current amplitude with the compensation angle. A larger tracking step size is used when the position error is large and the stator current change is significant, thereby improving the position error compensation speed. As the compensation angle gradually approaches the position that minimizes the stator current amplitude, the stator current change gradually decreases, and the tracking step size decreases accordingly, thus reducing oscillations of the compensation angle near the optimal position.

[0126] The state interval is based on the state direction variable within a continuous compensation period. Judge based on the changes. When continuous The absolute cumulative value of the state direction variable within each compensation cycle Greater than or equal to At that time, set the status flag bit When the absolute cumulative value is set to 0, the compensation algorithm is in tracking mode; when the absolute cumulative value is set to 0, the compensation algorithm is in tracking mode. Less than At that time, set the status flag bit When set to 1, the compensation algorithm enters the disturbed state. Once in the disturbed state, the tracking step size decreases according to a preset step size. The compensation period is gradually reduced until the minimum tracking step size is reached. When the tracking step size reaches the minimum tracking step size, the state direction variable is changed. The value is set to 0 to avoid incorrect state switching caused by sampling errors or external disturbances, thus keeping the compensation algorithm in a disturbed state. Through the above state judgment and step size adjustment mechanism, while ensuring rapid compensation of position errors, the continuous oscillation of stator current and compensation angle near the optimal operating point is reduced.

[0127] Although the adaptive minimum current tracking compensation method is based on the functional relationship between stator current and position error under maximum torque-current ratio control, in actual compensation, it does not require online identification or calculation of the coefficient ξ in this functional relationship, nor does it require separate models for digital control delay, loop filter, dead-zone effect, or parameter mismatch. Instead, it searches for the rotor position compensation angle online based on the stator current minimization principle, through the stator current variation trend and the amount of stator current change. Therefore, this method has strong robustness to parameter variations and non-ideal factors.

[0128] Step S700: Perform vector control on the built-in permanent magnet motor based on the compensated electrical angle and rotor angular velocity.

[0129] This invention replaces the integrator in the traditional hybrid flux observer with a third-order generalized integrator, thereby improving the flux observer's ability to suppress DC bias and harmonic disturbances in the permanent magnet motor drive system of underground coal mine transportation equipment. At the same time, through an adaptive minimum current tracking compensation method, the compensation angle is adjusted online according to the stator current change to achieve adaptive compensation for rotor position estimation error, thereby improving the estimation accuracy and operational reliability of the sensorless control system for permanent magnet motors in underground coal mine transportation equipment.

[0130] In this embodiment, the permanent magnet motor can be a permanent magnet motor for underground coal mine transportation equipment, preferably a built-in permanent magnet synchronous motor.

[0131] The following is an embodiment of a sensorless control device for a built-in permanent magnet motor used in mining equipment according to the present invention, which can be used to execute an embodiment of a sensorless control method for a built-in permanent magnet motor used in mining equipment according to the present invention. For details not disclosed in the embodiment of the sensorless control device for a built-in permanent magnet motor used in mining equipment according to the present invention, please refer to the embodiment of the sensorless control method for a built-in permanent magnet motor used in mining equipment according to the present invention.

[0132] Reference Figure 9 As shown, in one embodiment, a sensorless control device for a built-in permanent magnet motor in mining equipment is disclosed. The device includes:

[0133] The acquisition module is used to acquire the stator voltage and current in the αβ coordinate system of the built-in permanent magnet motor;

[0134] The hybrid flux monitoring module is used to input stator voltage and current into the hybrid flux monitoring device. The hybrid flux monitoring device uses a resonant integrator to suppress DC bias and harmonic components, and corrects the back electromotive force according to the deviation between the actual current and the estimated current before inputting it into the resonant integrator to output the estimated stator flux.

[0135] The rotor position estimation module is used to eliminate the salient pole effect caused by inductance asymmetry by using an effective flux model that includes direct-axis and quadrature-axis inductance parameters, based on the stator flux estimate and stator current, to obtain the effective flux estimate, and to obtain the preliminary rotor electrical angle and angular velocity through a phase-locked loop;

[0136] The minimum current tracking module is used to generate the rotor position compensation angle by iteratively optimizing the current amplitude through a dual-state machine, with the initial rotor electrical angle as the initial search center and without injecting an external high-frequency excitation signal.

[0137] The dual-state machine includes a tracking state and a disturbed state. In the tracking state, the tracking step size decreases as the current amplitude decreases. In the disturbed state, the step size is forcibly decayed until the minimum tracking step size is reached.

[0138] The compensation module is used to superimpose the rotor position compensation angle onto the initial rotor electrical angle to obtain the compensated electrical angle.

[0139] The motor control module is used to perform vector control on the built-in permanent magnet motor based on the compensated electrical angle and rotor angular velocity.

[0140] Specifically, refer to Figure 9 As shown, the acquisition module acquires the stator voltage signal in the αβ coordinate system of the permanent magnet motor. , and stator current signal , The stator voltage and stator current signals are input to the generalized integrator hybrid flux linkage observation module. The generalized integrator hybrid flux linkage observation module outputs the estimated stator flux linkage value. , The stator flux linkage estimate and stator current signal are input to the rotor position estimation module, which then constructs the effective flux linkage estimate. , And obtain the rotor electrical angle estimate through a phase-locked loop. and estimated electric angular velocity .

[0141] The adaptive minimum current tracking module generates the rotor position compensation angle based on the change in stator current amplitude. Rotor position compensation angle Input the compensation module and compare it with the rotor electrical angle estimate. By superimposing the values, the compensated rotor electrical angle is obtained. Compensated rotor electrical angle Estimated electrical angular velocity output of a phase-locked loop (PLL) used for Park transform and inverse Park transform. Used for closed-loop speed feedback.

[0142] The rotational speed ring is based on a given electrical angular velocity. With the estimated electric angular velocity The deviation between the two values ​​generates the stator current setpoint, and the MTPA module further generates the d-axis and q-axis current setpoints. , The current loop is based on the current setpoint and feedback current. , The error between them generates d-axis and q-axis voltage commands. , The voltage command in the αβ coordinate system is obtained through inverse Park transformation. , The inverter switching control signal is generated through the SVPWM module to drive the permanent magnet motor.

[0143] It should be noted that the functional modules in the above device embodiments can be implemented through software, hardware, or a combination of both. In practical applications, these functional modules can be integrated into the same control chip, or they can be distributed across different control units, depending on the hardware resources and control cycle requirements of the digital controller.

[0144] This invention discloses a sensorless control system for a permanent magnet motor in underground coal mine transportation equipment. The system includes a sensorless control device for a built-in permanent magnet motor used in mining equipment.

[0145] The sensorless control device for a built-in permanent magnet motor used in mining equipment is described in the above embodiments and will not be repeated here.

[0146] In one specific embodiment, the sensorless control system for a built-in permanent magnet motor used in mining equipment includes a permanent magnet motor for underground coal mine transportation equipment, an inverter, a current sampling unit, a voltage acquisition unit, a digital controller, and the aforementioned control device. The current sampling unit is used to collect the three-phase stator current of the permanent magnet motor. The voltage acquisition unit is used to obtain the stator voltage signal of the permanent magnet motor in the αβ coordinate system based on the inverter output voltage command, the DC bus voltage, or the voltage sampling signal. The digital controller is used to operate the aforementioned sensorless control device for the built-in permanent magnet motor used in mining equipment and to perform MTPA current setting, current loop control, coordinate transformation, and SVPWM modulation.

[0147] The system works as follows: First, the three-phase stator current of the permanent magnet motor is acquired through the current sampling unit. , , The stator current in the αβ coordinate system is obtained by Clark transformation. , The voltage acquisition unit obtains the stator voltage signal in the αβ coordinate system based on the inverter output voltage command, DC bus voltage, or voltage sampling signal. , The acquisition module inputs the stator voltage signal and stator current signal into the generalized integrator hybrid flux observation module to obtain the stator flux estimate; the stator flux estimate and stator current signal are further input into the rotor position estimation module, which constructs the effective flux estimate, and obtains the rotor electrical angle estimate and electrical angular velocity estimate through the phase-locked loop.

[0148] After obtaining the estimated rotor electrical angle, the adaptive minimum current tracking module generates a rotor position compensation angle based on the stator current variation trend and the amount of stator current change. This compensation angle is then superimposed on the estimated rotor electrical angle to obtain the compensated rotor electrical angle. The compensated rotor electrical angle is used for Park transform and inverse Park transform, making the current vector decomposition more accurate.

[0149] The speed loop outputs the stator current setpoint based on the deviation between the given speed and the estimated speed. The MTPA current setpoint module generates the d-axis and q-axis current setpoints based on the stator current setpoint. The current loop generates the dq-axis voltage command based on the current setpoint and the current feedback value. The dq-axis voltage command is transformed by inverse Park to obtain the αβ-axis voltage command, which is then modulated by SVPWM to drive the inverter, thereby controlling the operation of the permanent magnet motor used in underground coal mine transportation equipment.

[0150] Verification was conducted through simulation or experimentation. (Refer to...) Figure 10 The control system of built-in permanent magnet motor without position sensor for mining equipment was compared using different position compensation methods, including fixed step minimum current tracking (MCT) method, state switching MCT method and adaptive MCT method.

[0151] Specifically, Figure 10 (a) shows the experimental results of the fixed step size MCT method with a tracking step size of 5°. Figure 10 (b) shows the experimental results of the fixed-step MCT method with a tracking step size of 0.25°. Figure 10 (c) shows the experimental results of the state-switching MCT method. Figure 10 Figure (d) shows the experimental results of the adaptive MCT method. The initial rotor position error for each method was set to -20°, the stator current waveform scale was 0.5 A / div, and the position error waveform scale was 14.4° / div. Figure 10 Middle (a) to Figure 10 The time scales corresponding to (d) are 500 ms / div, 5 s / div, 1 s / div, and 200 ms / div, respectively. In this embodiment, the compensation update period... T compSet to 0.4 s.

[0152] In the fixed-step MCT method, a larger tracking step size results in faster compensation speed but more pronounced steady-state oscillations; a smaller tracking step size reduces steady-state oscillations but slows down compensation speed. The state-switching MCT method can improve the shortcomings of the fixed-step method to some extent, but it still struggles to simultaneously achieve both dynamic compensation speed and steady-state performance. With the adaptive minimum current tracking compensation method of this invention, the compensation step size can be adjusted online according to the stator current change. When the position error is large, a larger compensation step size is beneficial for quickly approaching the optimal compensation angle; when approaching the optimal compensation angle, the compensation step size automatically decreases, which helps reduce steady-state oscillations. Therefore, this invention can balance compensation speed and steady-state stability. Simultaneously, the TOGI hybrid flux observer can effectively suppress DC bias and harmonic disturbances in the input back EMF, avoiding the shortcomings of traditional pure integrators, equivalent low-pass filters, and second-order generalized integrators in terms of disturbance suppression, amplitude attenuation, and phase shift. Thus, this invention can simultaneously improve flux observation accuracy, position estimation accuracy, and position error compensation capability.

[0153] In summary, this invention can improve the operational reliability and control accuracy of the sensorless control system for built-in permanent magnet motors used in mining equipment when there are non-ideal factors such as measurement noise, inverter nonlinearity, parameter mismatch, digital control delay, and loop filter.

[0154] The above description is merely a preferred embodiment of the present invention and is not intended to limit the invention. Various modifications and variations can be made to the present invention by those skilled in the art. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A sensorless control method for a built-in permanent magnet motor used in mining equipment, characterized in that, The method includes: Obtain the stator voltage and current in the αβ coordinate system of the built-in permanent magnet motor; The stator voltage and current are input into the hybrid flux observer, which uses a resonant integrator to suppress DC bias and harmonic components. The back electromotive force is corrected based on the deviation between the actual current and the estimated current before being input into the resonant integrator, and the estimated stator flux is output. Based on the estimated stator flux linkage and stator current, the effective flux linkage model, which includes direct-axis and quadrature-axis inductance parameters, is used to eliminate the salient pole effect caused by inductance asymmetry, thereby obtaining the estimated effective flux linkage. The effective flux linkage estimate is input into the phase-locked loop to obtain the preliminary rotor electrical angle and rotor angular velocity; Using the initial rotor electrical angle as the initial search center, without injecting an external high-frequency excitation signal, and with the goal of minimizing the current amplitude, the rotor position compensation angle is generated through iterative optimization using a dual-state machine. The dual-state machine includes a tracking state and a disturbed state. In the tracking state, the tracking step size decreases as the current amplitude decreases. In the disturbed state, the step size is forcibly decayed until the minimum tracking step size is reached. The rotor position compensation angle is superimposed on the initial rotor electrical angle to obtain the compensated electrical angle; Vector control is performed on the built-in permanent magnet motor based on the compensated electrical angle and the rotor angular velocity.

2. The sensorless control method for a built-in permanent magnet motor in mining equipment according to claim 1, characterized in that, The corrected back electromotive force is: ， in, This is the feedback-corrected back electromotive force. Stator voltage, For stator resistance, This is the actual stator current. The estimated stator current is obtained from the current model calculation. This is the feedback correction coefficient.

3. The sensorless control method for a built-in permanent magnet motor in mining equipment according to claim 1, characterized in that, The resonant integrator is a third-order generalized integrator, and its continuous-domain transfer function is: ， ， In the formula, For in-phase signal transfer function, For orthogonal signal transfer functions, For the feedback-corrected back electromotive force e s Laplace transform, This is the in-phase output signal of the generalized integrator. The quadrature output signal of the generalized integrator. Forward gain coefficient For feedback gain coefficient, The center angular frequency, For the Laplace operator; The third-order generalized integrator is discretized using a bilinear transform, and the bilinear transform relationship is as follows: ， Define intermediate variables: ， The discrete realization coefficients are obtained as follows: ， In the formula, The sampling period is For discrete domain variables, The discrete equivalent operator obtained by bilinear transformation. , , and The coefficients are intermediate variables used to simplify discrete expressions. , and The feedback coefficients are those of the discrete implementation of the generalized integrator. These are the input coefficients for the discrete implementation of the generalized integrator.

4. The sensorless control method for a built-in permanent magnet motor in mining equipment according to claim 1, characterized in that, The effective flux linkage estimate is: ， In the formula, , These are the effective flux linkage estimates in the αβ coordinate system, respectively. , These are the estimated values ​​of the stator flux linkage in the αβ coordinate system, respectively. It is the q-axis inductance. , These are the stator currents in the αβ coordinate system, respectively; The compensated electrical angle is: ， in, For the compensated rotor electrical angle, This is the estimated rotor electrical angle output by the phase-locked loop. The rotor position compensation angle generated for iterative optimization.

5. The sensorless control method for a built-in permanent magnet motor in mining equipment according to claim 1, characterized in that, The iterative optimization process for generating the rotor position compensation angle includes: The stator current amplitude is sampled or filtered to obtain the stator current amplitude for the current compensation cycle. Calculate the change in stator current amplitude based on the stator current amplitude of the current compensation cycle and the stator current amplitude of the previous compensation cycle. The update direction of the compensation angle is determined based on the change in stator current amplitude and the change direction of the compensation angle in adjacent compensation cycles. The tracking step size is determined by the absolute value of the ratio of the change in stator current amplitude to the change in compensation angle. The rotor position compensation angle is updated based on the update direction of the compensation angle and the tracking step size.

6. The sensorless control method for a built-in permanent magnet motor in mining equipment according to claim 5, characterized in that, The rule for determining the direction of compensation angle update is as follows: When the stator current amplitude decreases after the compensation angle is updated along the current direction, the current compensation direction is maintained. When the stator current amplitude increases after the compensation angle is updated along the current direction, the compensation direction is changed.

7. The sensorless control method for a built-in permanent magnet motor in mining equipment according to claim 5, characterized in that, The state transition logic of the dual-state machine is as follows: The state direction variable is determined based on the change in stator current amplitude and the change in compensation angle, and the continuous direction is calculated. t The absolute value of the sum of the state direction variables within each compensation cycle, where, t It is an integer greater than 3; When the absolute value is greater than or equal to t At that time, the dual-state machine is in the tracking state; When the absolute value is less than t At that time, the dual-state machine is in a disturbed state.

8. The sensorless control method for a built-in permanent magnet motor in mining equipment according to claim 7, characterized in that, In the tracking state, the tracking step size satisfies: ， Or meet the requirements of the limited-amplitude format: ， In the formula, For the first The tracking step size for each compensation cycle. For adaptive step-size gain coefficient, This represents the change in stator current amplitude during the previous compensation cycle. The ratio of the two values ​​represents the change in the compensation angle in the previous compensation cycle and is used to characterize the local slope of the stator current amplitude as a function of the compensation angle. For the maximum tracking step size, This is the minimum tracking step size.

9. A sensorless control device for a built-in permanent magnet motor used in mining equipment, characterized in that, The apparatus for implementing the method of any one of claims 1 to 8 comprises: The acquisition module is used to acquire the stator voltage and current in the αβ coordinate system of the built-in permanent magnet motor; The hybrid flux linkage observation module is used to input stator voltage and current into the hybrid flux linkage observer. The hybrid flux linkage observer uses a resonant integrator to suppress DC bias and harmonic components, and corrects the back electromotive force according to the deviation between the actual current and the estimated current before inputting it into the resonant integrator to output the estimated stator flux linkage value. The rotor position estimation module is used to eliminate the salient pole effect caused by inductance asymmetry by using an effective flux model that includes direct-axis and quadrature-axis inductance parameters, based on the stator flux estimate and stator current, to obtain the effective flux estimate, and to obtain the preliminary rotor electrical angle and angular velocity through a phase-locked loop; The minimum current tracking module is used to generate the rotor position compensation angle by iterative optimization through a dual-state machine, with the initial rotor electrical angle as the initial search center and without injecting an external high-frequency excitation signal, aiming to minimize the current amplitude. The dual-state machine includes a tracking state and a disturbed state. In the tracking state, the tracking step size decreases as the current amplitude decreases. In the disturbed state, the step size is forcibly decayed until the minimum tracking step size is reached. The compensation module is used to superimpose the rotor position compensation angle onto the initial rotor electrical angle to obtain the compensated electrical angle. The motor control module is used to perform vector control on the built-in permanent magnet motor based on the compensated electrical angle and the rotor angular velocity.

10. A sensorless control system for a built-in permanent magnet motor used in mining equipment, characterized in that, The device includes the sensorless control device for a built-in permanent magnet motor for mining equipment as described in claim 9.

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