A DC overcurrent protection circuit suitable for pulse width modulation control

Abstract

This utility model discloses a DC overcurrent protection circuit suitable for pulse width modulation control. It is applied to a main power circuit comprising a power supply, a load, and a main power switch connected in series. The main power circuit has a control node for controlling the on / off state of the main power switch. The overcurrent protection circuit includes: a current sampling device located in the main power circuit, used to convert the current flowing through the main power circuit into a voltage signal proportional to it; an operational amplifier, whose input terminal is connected to the output terminal of the current sampling device, used to amplify the voltage signal; and a thyristor latch module, whose gate is connected to the output terminal of the operational amplifier, and whose anode and cathode form a path connected to the control node, to trigger a self-locking conduction state when the voltage signal exceeds a preset threshold, thereby controlling the main power switch to turn off. This overcurrent protection circuit has the advantages of fast response, low loss, and reliable latching protection.

Description

Technical Field

[0001] This utility model relates to the field of overcurrent protection technology, specifically to a DC overcurrent protection circuit suitable for pulse width modulation control. Background Technology

[0002] In fields such as DC power supplies, motor drives, and LED lighting, where pulse width modulation (PWM) technology is widely used, overcurrent protection circuits are essential to ensure the safety and reliability of the main power switching transistors and the entire system. However, existing overcurrent protection circuits generally suffer from the following drawbacks in practical applications, especially when dealing with complex operating conditions under PWM control: First, existing protection circuits generally exhibit slow response speeds and are prone to missed detections when dealing with instantaneous overcurrents (such as current spikes generated during load short circuits), especially under low duty cycle PWM control. Second, traditional current sampling methods often introduce significant power losses into the main power circuit to achieve current detection, which reduces the overall system efficiency and may cause additional thermal management problems. Finally, some existing protection mechanisms employ automatic recovery modes after a fault occurs, which can cause repeated electrical shocks to power devices under permanent short-circuit faults, reducing the overall reliability of the system. Utility Model Content

[0003] The purpose of this invention is to overcome the aforementioned defects or problems in the prior art and to provide a DC overcurrent protection circuit suitable for pulse width modulation control, which has the advantages of fast response, low loss and reliable latching protection.

[0004] To achieve the above objectives, the present invention adopts the following technical solution:

[0005] Technical Solution 1: A DC overcurrent protection circuit suitable for pulse width modulation control, applied to a main power circuit including a power supply, a load, and a main power switch connected in series. The main power circuit has a control node for controlling the on and off of the main power switch. The overcurrent protection circuit includes: a current sampling device located in the main power circuit, used to convert the current flowing through the main power circuit into a voltage signal proportional to it; an operational amplifier, whose input terminal is connected to the output terminal of the current sampling device, used to amplify the voltage signal; and a silicon controlled rectifier (SCR) latch module, whose gate is connected to the output terminal of the operational amplifier, and whose anode and cathode form a path connected to the control node, so as to trigger a self-locking conduction state when the voltage signal exceeds a preset threshold, thereby controlling the main power switch to turn off.

[0006] Technical Solution 2 based on Technical Solution 1: The current sampling device is a sampling resistor connected in series with the main power circuit to generate the voltage signal at its two ends when the current flows through the main power circuit.

[0007] Technical Solution 3 based on Technical Solution 1: The current sampling device is the on-resistance of the main power switch, the voltage signal is the drain-source voltage of the main power switch in the on-state, and the input terminal of the operational amplifier is connected to the drain of the main power switch to obtain this voltage.

[0008] Technical solution four, based on technical solution one, also includes a low-pass filter unit composed of resistors and capacitors, which is connected in series between the current sampling device and the input terminal of the operational amplifier to filter out high-frequency noise in the voltage signal.

[0009] Technical solution five based on technical solution one: also includes a shutdown execution switch, the control node is the control terminal of the shutdown execution switch, the conduction path of the shutdown execution switch is connected between the gate and source of the main power switch, so as to turn off the main power switch when the shutdown execution switch is turned on.

[0010] Technical solution six based on technical solution one: also includes a drive circuit that provides a drive signal to the main power switch, the control node is the power input terminal of the drive circuit, and the conduction path of the thyristor latch module is used to cut off the power supply to the drive circuit to turn off the main power switch.

[0011] Technical solution seven based on technical solution one: The operational amplifier is configured as a non-inverting amplifier, and the voltage signal is applied to the non-inverting input terminal of the operational amplifier.

[0012] Technical solution eight, based on technical solution seven, further includes a first feedback resistor and a second feedback resistor. The first feedback resistor is connected between the output terminal and the inverting input terminal of the operational amplifier, and the second feedback resistor is connected between the inverting input terminal and ground. The resistance ratio of the first and second feedback resistors is used to set the amplification factor of the operational amplifier.

[0013] Technical solution nine based on technical solution five further includes a pull-up resistor connected between the control terminal of the shutdown execution switch and the power supply, so as to provide a high level to the control terminal of the shutdown execution switch when the thyristor latch module has not entered the self-locking conduction state, so as to keep it off.

[0014] Technical solution ten, based on technical solution one, also includes a current-limiting resistor connected in series between the output of the operational amplifier and the gate of the thyristor latch module to limit the trigger current flowing into the gate.

[0015] As can be seen from the above description of this utility model, compared with the prior art, this utility model has the following beneficial effects:

[0016] Technical solution one provides a DC overcurrent protection circuit suitable for pulse width modulation control. This protection circuit combines a current sampling device, an operational amplifier, and a thyristor latch module in a specific structural combination, which effectively solves the technical problems of slow response speed, high power loss, and repeated electrical shocks in the existing technology under pulse width modulation control.

[0017] First, in the main power circuit of PWM control, especially under low duty cycle conditions, the conduction time of the main power switch may only last for a few microseconds. When a transient overcurrent fault such as a short circuit occurs in the load, the resulting overcurrent signal is also in the form of a brief pulse, which disappears once the PWM cycle ends. This solution utilizes the high-speed response characteristics of an operational amplifier. As an analog device, the operational amplifier can respond to changes in the input signal on a timescale of nanoseconds to microseconds, thus ensuring timely detection of the overcurrent signal during the brief PWM conduction time. Simultaneously, this solution uses a thyristor as a latching module; the gate only requires a brief trigger pulse to enter a stable and continuous self-locking conduction state. The high-speed response of the operational amplifier and the fast latching characteristics of the thyristor work together: the operational amplifier detects the fleeting overcurrent signal and immediately outputs a signal sufficient to trigger the thyristor; the thyristor then converts this brief trigger signal into a stable and continuous turn-off control state. This purely hardware mechanism, which transforms an instantaneous electrical signal into a steady-state physical property, overcomes the shortcomings of existing technologies that miss transient fault signals due to processing delays, and achieves a fast and reliable response to transient overcurrents.

[0018] Secondly, traditional current sampling methods typically require large-value sampling resistors to obtain a sufficiently large and easily detectable voltage signal, which results in significant power loss in the main circuit. This invention introduces an operational amplifier to amplify the sampled signal. This structure significantly reduces the amplitude requirements of the front-end sampling signal. Therefore, the current sampling device can use extremely low-value sampling resistors, or even directly utilize the on-resistance of the main power switch for lossless sampling. Even if the voltage signal generated by these methods is extremely weak, such as at the millivolt level, the subsequent operational amplifier can effectively amplify it to a level sufficient to drive the thyristor. The amplification function of the operational amplifier and the low-resistance characteristic of the current sampling device allow for the simultaneous achievement of high detection sensitivity and low power loss, overcoming the technical deficiency of prior art where these two aspects are difficult to balance.

[0019] Furthermore, some existing protection circuits employ an automatic recovery mode after detecting overcurrent. In this mode, if faults such as short circuits persist, the protection circuit will cause the main power switch to repeatedly attempt to turn on and off. Each turn-on attempt generates a large current surge in the main power circuit, causing continuous electrical stress accumulation on the power devices, thereby reducing device lifespan and system reliability. This solution uses a thyristor as a latching module. Once the thyristor is triggered into a self-locking conduction state, it will remain on as long as the current flowing through its anode and cathode is uninterrupted. In DC circuit applications, this means that once the protection is triggered, the circuit will be permanently shut down unless the entire system power supply is cut off and restarted. This non-self-recovering hardware latching mechanism eliminates the possibility of repeated restart attempts when the fault has not been resolved, thus avoiding repeated electrical shocks to the power devices and ensuring system safety after a fault occurs.

[0020] In technical solution two, by connecting a sampling resistor in series in the main power circuit, the current flowing through the main power circuit can be directly and linearly converted into a voltage signal generated across its terminals. The high gain characteristic of the operational amplifier allows for the use of a sampling resistor with a very small resistance value, thereby ensuring reliable current signal conversion while significantly reducing the power loss generated by the resistor itself.

[0021] In technical solution three, the drain-source voltage is directly detected as the sampling signal, effectively utilizing the switching transistor itself as a current sampling element. Since no additional discrete resistors are introduced, the potential power loss from the current sampling stage is eliminated. The voltage signal generated in this way is typically very weak, but when combined with a high-gain operational amplifier, the amplifier can effectively amplify this weak signal, making it usable. Therefore, this synergistic relationship enables completely lossless current sampling, significantly improving the overall system efficiency and reducing the number of components and cost.

[0022] In technical solution four, a low-pass filter unit composed of resistors and capacitors is added, significantly improving the reliability and anti-interference capability of the protection circuit. In PWM-controlled circuits, the high-speed switching action of the main power transistor inevitably introduces high-frequency noise into the circuit. This noise may be superimposed on the current sampling signal, forming erroneous voltage spikes. If a noisy signal is directly fed into a high-sensitivity operational amplifier, the noise itself may be amplified, leading to erroneous triggering of the thyristor and causing the protection circuit to malfunction even when no actual overcurrent has occurred. The low-pass filter unit in this solution, based on the physical characteristic that capacitors present low impedance to high-frequency signals, can effectively attenuate and filter out high-frequency noise components in the sampling signal, while retaining the DC or low-frequency signal that reflects the true overcurrent state, avoiding false triggering caused by noise and ensuring the accuracy of overcurrent protection action.

[0023] In technical solution five, the thyristor latch module acts as a control signal source, while the shutdown execution switch serves as the final execution element. When an overcurrent occurs, the conducting thyristor latch module outputs a control signal to activate the shutdown execution switch. The conduction path of this shutdown execution switch is connected between the gate and source of the main power switch, forming an extremely low impedance path once turned on. This path can quickly discharge the charge on the gate of the main power switch, forcing its gate-source voltage clamped to zero. This two-stage execution structure allows the shutdown execution switch to be specifically optimized to provide extremely fast switching speed and extremely strong current discharge capability, thereby ensuring a faster and more thorough shutdown of the main power switch.

[0024] In technical solution six, the conduction path of the thyristor latch module is connected to the power input terminal of the drive circuit. When an overcurrent occurs, the thyristor is triggered and conducts, directly short-circuiting the power supply of the drive circuit to ground. This causes the drive circuit to immediately stop operating due to the loss of power, and its output capability disappears, thus preventing it from continuing to provide the turn-on voltage to the main power switch, causing the switch to turn off. This method of cutting off the power supply to the front-end control unit is an indirect but equally effective shutdown method. In some complex drive systems, this method can avoid potential electrical conflicts between the protection circuit and the output terminal of the drive circuit, providing a more fundamental and safer protection execution mechanism.

[0025] In technical solution seven, the operational amplifier is configured as a co-inverting amplifier. This structure has a high input impedance, which ensures that the protection circuit will not generate a load effect on the main power circuit during sampling, thereby ensuring the accuracy of the sampled signal.

[0026] In technical solution eight, a negative feedback network is formed by the first and second feedback resistors. The voltage gain of the in-phase amplifier is precisely determined by the resistance ratio of these two feedback resistors, thereby enabling the overcurrent protection trigger threshold to be precisely and stably designed and adjusted by selecting the resistor values, providing predictable and high-precision performance for the entire protection circuit.

[0027] In technical solution nine, a pull-up resistor is added to establish a defined and safe default operating state for the shutdown execution switch of the protection circuit. Without this pull-up resistor, when the SCR latch module is not triggered, its output is in a high-impedance state, causing the control terminal of the shutdown execution switch to be electrically floating. A floating control terminal is highly susceptible to circuit noise or static electricity, which may cause its voltage to drift unexpectedly, thus incorrectly turning on the shutdown execution switch and triggering a malfunction of the entire protection circuit. In this solution, one end of the pull-up resistor is connected to the power supply, and the other end is connected to the control terminal of the shutdown execution switch. According to its circuit connection, this pull-up resistor provides a stable and continuous high level to the control terminal of the shutdown execution switch when the circuit is operating normally. For the P-channel MOSFET acting as the shutdown execution switch, this high level ensures that it is in a reliable off state.

[0028] In technical solution ten, a current-limiting resistor is added in series between the operational amplifier output and the thyristor gate, enhancing the long-term operational reliability of the circuit. The output stage of an operational amplifier typically has a certain current-driving capability, while the gate of the thyristor only requires a very small current to be reliably triggered. Without current limiting measures, when the operational amplifier outputs a high voltage, a large instantaneous current may flow into the thyristor gate. Although this current is brief, it may exceed the upper limit of the operational amplifier's output current or the current limit that the thyristor gate can withstand. Repeated occurrences over a long period could damage both devices. The current-limiting resistor limits the trigger current to a safe range for both the operational amplifier and the thyristor, ensuring reliable triggering while protecting critical components and improving the durability and reliability of the entire protection circuit. Attached Figure Description

[0029] To more clearly illustrate the technical solutions of the embodiments of this utility model, the drawings used in the following description of the embodiments will be briefly introduced. Obviously, the drawings described below are some embodiments of this utility model. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0030] Figure 1 This is a circuit diagram of the overcurrent protection circuit involved in Embodiment 1 of this utility model;

[0031] Figure 2 This is a circuit diagram of the overcurrent protection circuit involved in Embodiment 2 of this utility model.

[0032] Explanation of key figure labels:

[0033] Bootstrap conduction module 10; operational amplifier module 20; operational amplifier 21; low-pass filter unit 22; thyristor latch module 30; power supply module 40; drive module 50; load 60. Detailed Implementation

[0034] The technical solutions of the present utility model will be clearly and completely described below with reference to the accompanying drawings of the embodiments. Obviously, the described embodiments are preferred embodiments of the present utility model and should not be considered as excluding other embodiments. All other embodiments obtained by those skilled in the art based on the embodiments of the present utility model without creative effort are within the scope of protection of the present utility model.

[0035] Unless otherwise expressly defined, the use of terms such as "first," "second," or "third" in the claims, description, and drawings of this utility model is for distinguishing different objects and not for describing a specific order.

[0036] Unless otherwise expressly defined, in the claims, description, and accompanying drawings of this utility model, the use of directional terms such as "center," "lateral," "longitudinal," "horizontal," "vertical," "top," "bottom," "inner," "outer," "upper," "lower," "front," "rear," "left," "right," "clockwise," and "counterclockwise" to indicate orientation or positional relationships is based on the orientation and positional relationships shown in the accompanying drawings and is only for the convenience of describing this utility model and simplifying the description. It does not indicate or imply that the device or element referred to must have a specific orientation or be constructed and operated in a specific orientation, and therefore should not be construed as limiting the specific protection scope of this utility model.

[0037] Unless otherwise expressly defined, the terms "fixed connection" or "fixed connection" used in the claims, description and drawings of this utility model shall be interpreted broadly to refer to any connection in which there is no displacement or relative rotation relationship between the two parties, including non-removable fixed connection, detachable fixed connection, integral connection and fixed connection through other devices or components.

[0038] In the claims, description and accompanying drawings of this utility model, the terms "comprising", "having", and variations thereof are used to mean "including but not limited to".

[0039] Example 1

[0040] Reference Figure 1 This embodiment relates to a DC overcurrent protection circuit suitable for pulse width modulation control. It is applied to a main power circuit comprising a power supply V24+, a load 60, and a main power switch transistor connected in series. The main power circuit has a control node for controlling the on / off state of the main power switch transistor. Specifically, the main power switch transistor Q33 can be a controllable semiconductor switching device such as an N-channel MOSFET, a P-channel MOSFET, or an IGBT. The main power circuit constitutes the main path of the system's operating current. For example, in a DC motor drive application, this circuit is composed of a DC power supply V24+, a load 60, and the switch transistor Q33 connected in series.

[0041] In this embodiment, the overcurrent protection circuit may include a bootstrap conduction module 10, an operational amplifier module 20, and a common latch module. The bootstrap conduction module 10 mainly includes a current sampling device located in the main power circuit, used to convert the current flowing through the main power circuit into a voltage signal proportional to it. The operational amplifier module 20 mainly includes an operational amplifier 21, the input of which is connected to the output of the current sampling device, used to amplify the voltage signal. The gate of the thyristor latch module 30 is connected to the output of the operational amplifier 21, and the path formed by its anode and cathode is connected to the control node, so as to trigger a self-locking conduction state when the voltage signal exceeds a preset threshold, thereby controlling the main power switch to turn off.

[0042] In general, the current sampling device is used to monitor the current in the main power circuit in real time. Operational amplifier 21, acting as a high-gain comparator, rapidly amplifies the sampled signal once it indicates an overcurrent. The amplified signal triggers the thyristor latch module 30. Once triggered, the thyristor enters a self-locking conduction state, providing a stable and continuous turn-off signal to the control node of the main power switch, cutting off the main circuit current to protect the load 60 and power devices. This latching characteristic ensures that the circuit will not automatically recover until the fault is cleared, requiring manual intervention such as a power-off restart to unlock, thus avoiding repeated electrical shocks to the devices under continuous fault conditions.

[0043] The current sampling device is a sampling resistor connected in series in the main power circuit to generate a voltage signal across its terminals when current flows through the main power circuit. Specifically, the sampling resistor RA is preferably a high-precision, low-temperature-drift milliohm-level current sensing resistor, with a resistance value selectable between 1 milliohm and 100 milliohm. The resistor RA can be connected in series between the source of the main power switch Q33 and system ground (i.e., low-side sampling), or between the power supply V24+ and the load 6060 (i.e., high-side sampling). In this embodiment, low-side sampling is chosen because the voltage signal generated by low-side sampling is referenced to ground, making the signal processing circuit design relatively simple. Choosing an extremely low resistance value is to minimize its own power loss and reduce its impact on the overall efficiency of the main power circuit. Simultaneously, the weak voltage signal it generates is sufficient for effective detection by the subsequent high-gain operational amplifier 21U6.

[0044] The bootstrap turn-on module 10 also includes a turn-off execution switch. The control node is the control terminal of the turn-off execution switch. The conduction path of the turn-off execution switch is connected between the gate and source of the main power switch to turn off the main power switch when the turn-off execution switch is turned on. Furthermore, a pull-up resistor is included, connected between the control terminal of the turn-off execution switch and the power supply, to provide a high level to the control terminal of the turn-off execution switch when the thyristor latch module 30 is not in the self-locking turn-on state, thus keeping it off.

[0045] Specifically, the shutdown execution switch Q6 can be a P-channel enhancement-mode MOSFET or a transistor circuit with similar logic (high-level input turns off, low-level input turns on). Its conduction path is connected between the gate and source of the main power switch Q33; for example, its source is connected to the gate of the main power switch Q33, and its drain is connected to the source of the main power switch Q33 or system ground. One end of the pull-up resistor R61 is connected to the control terminal (i.e., the gate) of the shutdown execution switch Q6, and the other end is connected to a suitable positive power supply, such as V24+. In this embodiment, the control node specifically refers to the control terminal of the shutdown execution switch Q6, i.e., its gate.

[0046] When the system is operating normally, i.e., the SCR latch module 30 is not triggered, the anode of the SCR is in a high-resistance state. At this time, the pull-up resistor R61 stably pulls the gate level of the shutdown execution switch Q6 high to the power supply level. Since the shutdown execution switch Q6 is a P-channel MOSFET, the high gate voltage (gate-source voltage close to zero or positive) reliably keeps it in the off state, its conduction path is broken, and therefore it will not affect the normal PWM drive signal of the main power switch Q33. Once an overcurrent is detected, the SCR latch module 30 is triggered and enters a self-locking conduction state, forming a stable low-resistance path to ground between its anode and cathode. This will quickly pull the gate level of the shutdown execution switch Q6 low to near ground level through its anode connection. The low gate level (gate-source voltage negative) causes the P-channel MOSFET Q6 to conduct quickly, thereby forming a low-resistance path between its source and drain. This path, connected between the gate and source of the main power switch Q33, immediately discharges the charge from the gate of Q33, forcing its gate-source voltage to zero, thus achieving fast and reliable turn-off protection. This "two-stage execution" structure (SCR triggers Q6, Q6 turns off Q33) ensures the thoroughness and instantaneity of the turn-off action.

[0047] The thyristor latch module 30SCR is the key component for realizing the "latch" function. The gate of the thyristor latch module 30SCR serves as the control input, receiving the trigger signal from the output of the operational amplifier 21U6. Once a brief, sufficiently large current pulse is applied to its gate, the anode and cathode of the SCR quickly switch from a high-impedance off state to a low-impedance on state. Most importantly, once on, even if the gate trigger signal disappears, the SCR will remain on as long as the current flowing between its anode and cathode is not lower than its holding current (this condition is usually constantly met in a DC power supply environment), i.e., it is self-locking.

[0048] In this embodiment, the operational amplifier 21 is configured as a non-inverting amplifier, and the voltage signal is applied to the non-inverting input terminal of the operational amplifier 21. Specifically, in the overcurrent protection circuit, the operational amplifier 21U6, for example, model LM321, serves as the core signal amplification and comparison unit. The non-inverting input terminal IN+ of the operational amplifier 21U6 is the signal sensing entry point of the entire protection circuit. It is connected to the output terminal of the current sampling device, specifically, after the low-pass filter unit 22 (composed of resistor R63 and capacitor C64). The weak voltage signal from the current sampling resistor RA, which is proportional to the main circuit current, first passes through this RC low-pass filter to filter out high-frequency switching noise, and then this relatively pure DC or low-frequency voltage signal is applied to the IN+ pin. By using the non-inverting input method, the extremely high input impedance characteristics of the operational amplifier 21 can be utilized to ensure that the protection circuit does not generate a load effect on the main power circuit when sampling the signal, thereby ensuring the accuracy of the sampled signal. The inverting input terminal IN- of operational amplifier 21U6 is the connection point of the negative feedback network, used to set the amplification factor of the circuit. It does not directly receive external input signals, but is connected at the intersection of the first feedback resistor R65 and the second feedback resistor R64. One end of the first feedback resistor R65 is connected to the output terminal OUT of operational amplifier 21U6, and the other end is connected to the inverting input terminal IN-; one end of the second feedback resistor R64 is connected to the inverting input terminal IN-, and the other end is connected to circuit ground GND. This connection method constitutes a classic non-inverting amplifier negative feedback structure. The output terminal OUT of operational amplifier 21U6 is the output port of the decision result. After the input signal (i.e., the sampled voltage) is amplified, the output terminal OUT will generate a significantly enhanced voltage signal. This output terminal is connected to the gate (control electrode) of the silicon controlled rectifier (SCR) latch module 30SCR through a current-limiting resistor R66. Its function is as follows: Under normal operating conditions, the output is low, insufficient to trigger the SCR; once an overcurrent occurs, the sampling voltage rises, and the output of U6 quickly jumps to a high level, providing a trigger current sufficient to turn on the SCR gate through the current-limiting resistor R66, thereby initiating subsequent protection actions. To ensure the normal operation of the internal circuitry of operational amplifier 21U6, its power supply pins need to be correctly connected. The positive power supply pin VDD is connected to a stable low-voltage DC power supply, such as V3.3+ in this embodiment. The negative power supply or ground pin VSS is connected to the circuit's common ground GND. This provides the operational amplifier 21 with the required bias voltage and reference level for operation.

[0049] In addition, the operational amplifier circuit also includes a low-pass filter unit 22 composed of resistors and capacitors, which is connected in series between the current sampling device and the input terminal of the operational amplifier 21 to filter out high-frequency noise in the voltage signal. This low-pass filter unit 22 and the operational amplifier 21 constitute the main components of the operational amplifier circuit. Specifically, the low-pass filter unit 22 is typically an RC filter, consisting of a resistor R63 connected in series in the signal path and a capacitor C64 connected between the signal path and ground, and is located close to the input terminal of the operational amplifier 21U6. Its main function is to filter out high-frequency noise and voltage spikes generated by the high-speed switching of the main power switch Q33 under PWM control. The cutoff frequency of the filter needs to be set by appropriately selecting the values ​​of the resistors and capacitors. This cutoff frequency should be low enough to effectively suppress switching noise, while also being high enough to ensure that the actual overcurrent signal can pass through without delay. For example, a 1kΩ resistor R63 and a 470pF capacitor C64 can be used to construct this filter.

[0050] The operational amplifier circuit also includes a first feedback resistor and a second feedback resistor. The first feedback resistor is connected between the output terminal and the inverting input terminal of the operational amplifier 21, and the second feedback resistor is connected between the inverting input terminal and ground. The ratio of the resistance values ​​of the first and second feedback resistors is used to set the amplification factor of the operational amplifier 21. One end of the first feedback resistor R65 is connected to the output terminal OUT of the operational amplifier 21U6, and the other end is connected to its inverting input terminal IN-. One end of the second feedback resistor R64 is connected to the inverting input terminal IN-, and the other end is connected to the circuit ground GND. This negative feedback network allows the closed-loop voltage gain of the entire operational amplifier circuit to be precisely set. Its gain value is mainly determined by the ratio of the resistance values ​​of these two external resistors, calculated using the formula: Gain G = 1 + (R65 / R64). For example, when R65 is 39kΩ and R64 is 1kΩ, the voltage gain of the circuit is precisely set to 40 times. The purpose of this design is to stably and reliably amplify the weak overcurrent voltage signal from the current sampling resistor RA, which is typically only tens to hundreds of millivolts, to a voltage level (typically several volts) sufficient to trigger the gate of the subsequent SCR latch module 30. By selecting high-precision resistors, the trigger threshold of the overcurrent protection can be ensured to be accurate and stable, unaffected by changes in the open-loop gain of the operational amplifier 21 itself.

[0051] The operational amplifier circuit also includes a current-limiting resistor, which is connected in series between the output of the operational amplifier 21 and the gate of the silicon controlled rectifier (SCR) latch module 30 to limit the trigger current flowing into the gate. This current-limiting resistor is R66. When the operational amplifier 21U6 detects an overcurrent signal and outputs a high level, its output directly drives the gate of the SCR. Since the gate-cathode voltage drop of the SCR during triggering is approximately the forward voltage drop of a diode, without limitation, the output of the operational amplifier 21U6 may output a current far exceeding its rated capacity, potentially damaging the operational amplifier 21 itself or causing performance degradation. Simultaneously, excessive gate current may also impact the gate of the SCR. The current-limiting resistor R66 effectively solves this problem. Based on Ohm's law, it limits the trigger current to a range that ensures reliable SCR triggering while also protecting the preceding and following stages. For example, when U6 outputs a high level of 3.3V and the SCR gate trigger voltage is approximately 0.7V, the trigger current will be limited to I = (3.3V - 0.7V) / 1kΩ = 2.6mA through a 1kΩ resistor R66. This current value is far lower than the maximum output current of a typical operational amplifier 21, while being far higher than the minimum gate trigger current required by most thyristors. This ensures reliable triggering while greatly enhancing the long-term operational reliability of the circuit.

[0052] This embodiment relates to a DC overcurrent protection circuit suitable for pulse width modulation control. The protection circuit combines a current sampling device, an operational amplifier 21, and a thyristor latch module 30 in a specific structural combination, which effectively solves the technical problems of slow response speed, large power loss, and repeated electrical shocks in the prior art under pulse width modulation control.

[0053] First, in the main power circuit of PWM control, especially under low duty cycle conditions, the conduction time of the main power switch may only last for a few microseconds. When a short circuit or other transient overcurrent fault occurs in the load 60, the resulting overcurrent signal is also a brief pulse, which disappears once the PWM cycle ends. This solution utilizes the high-speed response characteristics of operational amplifier 21. As an analog device, operational amplifier 21 can respond to changes in the input signal on a timescale of nanoseconds to microseconds, thus ensuring timely detection of the overcurrent signal during the brief PWM conduction time. Simultaneously, this solution uses a thyristor as a latching module; the gate only requires a brief trigger pulse to enter a stable and continuous self-locking conduction state. The high-speed response of operational amplifier 21 and the fast latching characteristics of the thyristor work together: operational amplifier 21 detects the fleeting overcurrent signal and immediately outputs a signal sufficient to trigger the thyristor; the thyristor then converts this brief trigger signal into a stable and continuous turn-off control state. This purely hardware mechanism, which transforms an instantaneous electrical signal into a steady-state physical property, overcomes the shortcomings of existing technologies that miss transient fault signals due to processing delays, and achieves a fast and reliable response to transient overcurrents.

[0054] Secondly, traditional current sampling methods typically require a large-value sampling resistor to obtain a sufficiently large and easily detectable voltage signal, which results in significant power loss in the main circuit. This invention introduces an operational amplifier 21 to amplify the sampled signal. This structure significantly reduces the amplitude requirement of the front-end sampling signal. Therefore, the current sampling device can use a sampling resistor with extremely low resistance, or even directly utilize the on-resistance of the main power switch for lossless sampling. Even if the voltage signal generated by these methods is extremely weak, such as at the millivolt level, the subsequent operational amplifier 21 can effectively amplify it to a level sufficient to drive the thyristor. The amplification function of the operational amplifier 21 and the low-resistance characteristic of the current sampling device allow for the simultaneous achievement of high detection sensitivity and low power loss, overcoming the technical deficiency of prior art where these two aspects are difficult to balance.

[0055] Furthermore, some existing protection circuits employ an automatic recovery mode after detecting overcurrent. In this mode, if faults such as short circuits persist, the protection circuit will cause the main power switch to repeatedly attempt to turn on and off. Each turn-on attempt generates a large current surge in the main power circuit, causing continuous electrical stress accumulation on the power devices, thereby reducing device lifespan and system reliability. This solution uses a thyristor as a latching module. Once the thyristor is triggered into a self-locking conduction state, it will remain on as long as the current flowing through its anode and cathode is uninterrupted. In DC circuit applications, this means that once the protection is triggered, the circuit will be permanently shut down unless the entire system power supply is cut off and restarted. This non-self-recovering hardware latching mechanism eliminates the possibility of repeated restart attempts when the fault has not been resolved, thus avoiding repeated electrical shocks to the power devices and ensuring system safety after a fault occurs.

[0056] Example 2

[0057] The difference between Example 2 and Example 1 lies in the current sampling method and the specific execution mechanism for subsequently turning off the main power switch Q33.

[0058] Reference Figure 2 In Embodiment 2, the current sampling device is the on-resistance of the main power switch, the voltage signal is the drain-source voltage of the main power switch in the on-state, and the input terminal of the operational amplifier 21 is connected to the drain of the main power switch to obtain this voltage.

[0059] Specifically, in this embodiment, a separate sampling resistor RA is no longer used to detect the current. Instead, the current sampling device is configured to use the on-resistance (Rds(on)) of the main power switch Q33 itself for lossless current sampling. Specifically, when the main power switch Q33 (e.g., an N-channel MOSFET) is turned on under the drive of a PWM signal, the main circuit current I flows through its channel, generating a small voltage drop Vds between its drain and source, which is approximately equal to I*Rds(on). Since the source of the main power switch Q33 is directly grounded, the voltage across its drain is this voltage drop Vds. The input terminal of the operational amplifier circuit, i.e., the input terminal of the low-pass filter unit 22, is directly connected to the drain of the main power switch Q33. In this way, this Vds voltage signal, which is proportional to the current, is sent as the original sampling signal to the subsequent circuit for processing. This sampling method avoids power loss and heat generation issues caused by the sampling process because it does not require an additional series sampling resistor, thus improving the energy efficiency of the entire system. Furthermore, it reduces external components (sampling resistors), thereby lowering material costs and PCB layout space.

[0060] Furthermore, unlike the method in Embodiment 1 where the gate of the main power switch Q33 is directly clamped by turning off the execution switch Q6, this embodiment employs another indirect but more fundamental turn-off method: cutting off the power supply to the drive circuit. Specifically, as shown... Figure 2 As shown, the system includes a power supply module 40 that provides power to the drive module 50. The drive module 50 is responsible for generating the PWM signal required to drive the main power switch Q33. The output terminal (anode of the SCR) of the thyristor latch module 30 is connected to a power supply control terminal of the drive module 50. In this embodiment, the control node specifically refers to a power control input terminal of the drive module 50.

[0061] During normal operation, the SCR is in the off state, with its anode in a high-resistance state, which does not affect the normal power supply from the power supply module 40 to the drive module 50. The drive module 50 operates normally, outputting a PWM signal to drive Q33. When an overcurrent occurs, the SCR is triggered and latched on, its anode clamped to a low level. This low-level "power control" signal acts on the drive module 50, cutting off its internal power path or putting it into a power-down protection state. The drive module 50 immediately stops working due to the loss of power, and its PWM signal output capability also disappears. Since the drive module 50 can no longer provide a turn-on voltage to the gate of the main power switch Q33, the gate charge of Q33 will be discharged through the gate pull-down resistor or internal mechanism, thus quickly turning it off. This turn-off method achieves protection by removing the drive source, which is a safer and more thorough protection mechanism, effectively avoiding potential electrical conflicts that may occur when the protection circuit and the drive circuit compete for control of the Q33 gate.

[0062] In this embodiment, the low-pass filter unit 22 (composed of R63 and C64), the operational amplifier circuit (U6 and its feedback resistors R64 and R65), and the silicon controlled rectifier latch module 30 (SCR) operate on the same principle as described in Embodiment 1. They work together to achieve fast response and reliable latching of the sampled signal based on the on-state voltage drop of Q33, and finally complete overcurrent protection by controlling the power supply of the drive module 50.

[0063] The foregoing description of the specifications and embodiments is intended to explain the scope of protection of this utility model, but does not constitute a limitation on the scope of protection of this utility model. Modifications, equivalent substitutions, or other improvements to the embodiments of this utility model or a portion thereof that can be obtained by those skilled in the art through logical analysis, reasoning, or limited experimentation, based on the teachings of this utility model or the foregoing embodiments, should all be included within the scope of protection of this utility model.

Claims

1. A DC overcurrent protection circuit suitable for pulse width modulation control, applied to a main power circuit comprising a power supply, a load (60), and a main power switch transistor connected in series, wherein the main power circuit has a control node for controlling the on and off of the main power switch transistor, characterized in that, The overcurrent protection circuit includes: A current sampling device is provided in the main power circuit and is used to convert the current flowing through the main power circuit into a voltage signal that is proportional to it. An operational amplifier (21), whose input terminal is connected to the output terminal of the current sampling device, is used to amplify the voltage signal; and The thyristor latch module (30) has its gate connected to the output terminal of the operational amplifier (21), and its anode and cathode are connected to the control node to trigger the self-locking conduction state when the voltage signal exceeds a preset threshold, thereby controlling the main power switch to turn off.

2. A DC overcurrent protection circuit for use in a pulse width modulation control as defined in claim 1, wherein The current sampling device is a sampling resistor connected in series with the main power circuit to generate the voltage signal across its terminals when current flows through the main power circuit.

3. A DC overcurrent protection circuit for use in pulse width modulation control as defined in claim 1, wherein The current sampling device is the on-resistance of the main power switch, the voltage signal is the drain-source voltage of the main power switch in the on state, and the input terminal of the operational amplifier (21) is connected to the drain of the main power switch to obtain the voltage.

4. A DC overcurrent protection circuit for use in pulse width modulation control as defined in claim 1, wherein It also includes a low-pass filter unit (22) consisting of resistors and capacitors, which is connected in series between the current sampling device and the input terminal of the operational amplifier (21) to filter out high-frequency noise in the voltage signal.

5. A DC overcurrent protection circuit for use in pulse width modulation control as defined in claim 1, wherein It also includes a shutdown execution switch, wherein the control node is the control terminal of the shutdown execution switch, and the conduction path of the shutdown execution switch is connected between the gate and source of the main power switch, so as to turn off the main power switch when the shutdown execution switch is turned on.

6. A DC overcurrent protection circuit for use in pulse width modulation control as defined in claim 1, wherein It also includes a drive circuit that provides a drive signal to the main power switch, the control node is the power input terminal of the drive circuit, and the conduction path of the thyristor latch module (30) is used to cut off the power supply to the drive circuit to turn off the main power switch.

7. A DC overcurrent protection circuit suitable for pulse width modulation control as described in claim 1, characterized in that, The operational amplifier (21) is configured as a non-inverting amplifier, and the voltage signal is applied to the non-inverting input of the operational amplifier (21).

8. A DC overcurrent protection circuit suitable for use in a pulse width modulation control as defined in claim 7, characterized in that It also includes a first feedback resistor and a second feedback resistor. The first feedback resistor is connected between the output terminal and the inverting input terminal of the operational amplifier (21), and the second feedback resistor is connected between the inverting input terminal and ground. The resistance ratio of the first and second feedback resistors is used to set the amplification factor of the operational amplifier (21).

9. A DC overcurrent protection circuit for use in pulse width modulation control as defined in claim 5, wherein It also includes a pull-up resistor connected between the control terminal of the shutdown execution switch and the power supply, so as to provide a high level to the control terminal of the shutdown execution switch when the thyristor latch module (30) is not in the self-locking conduction state, so as to keep it off.

10. A DC overcurrent protection circuit for use in pulse width modulation control as defined in claim 1, wherein It also includes a current-limiting resistor, which is connected in series between the output of the operational amplifier (21) and the gate of the thyristor latch module (30) to limit the trigger current flowing into the gate.

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