High linearity current source circuit for laser driver

Abstract

The utility model discloses a kind of high linearity current source circuit for laser driver, belong to electronic circuit and laser driving technical field.The circuit aims at solving the problem of low output precision and poor stability caused by inherent nonlinearity and temperature drift of semiconductor device of existing laser driving current source.Its technical scheme main points include: error control module, for comparing input control voltage with feedback voltage and output error control signal;Power output module, according to error control signal driving load;Current detection module, in series in output loop to obtain detection voltage proportional to output current;Differential feedback module, differential amplification is carried out to detection voltage and generates feedback voltage, constitutes closed-loop control;And overcurrent protection module, when detection voltage exceeds preset threshold, limit driving signal.The circuit is mainly used to provide high-precision, high linearity and have fast overcurrent protection driving current for laser, applicable to precision optical communication, measurement and other fields.

Description

Technical Field

[0001] This utility model relates to the field of electronic circuits and laser driving technology, specifically to a high linearity current source circuit for laser driving. Background Technology

[0002] In fields such as laser driving and precision measurement, high-precision, high-linearity current sources are critical components, directly impacting system stability and measurement accuracy. Currently, mainstream current source solutions have inherent limitations in their basic architecture, making them unsuitable for high-end applications. Specifically: First, inherent problems with the basic common-ground VI conversion circuit. Principle flaws: The most common solution uses an operational amplifier to drive a MOSFET or BJT, with the load connected in series with a current sensing resistor (current sensing resistor) and grounded. The output current formula is: Nonlinearity and temperature drift: This circuit appears ideal, but in reality, it is severely limited by the input offset voltage of the op-amp. And its temperature drift. The actual current is... , (Typically in the µV~mV range) and its temperature drift (typically µV / °C) directly introduce errors. For driving low-power lasers (e.g., requiring 10mA, a 1Ω resistor will produce a 10mV voltage drop), a 100µV offset voltage will introduce 1% error, and the device's temperature drift characteristics will cause the output current to "drift" with temperature. Limited output impedance: The output impedance of this circuit is approximately... ≈ ,in This is the open-loop gain of the op-amp. Although the value may be large, the output impedance decreases at high frequencies. The sharp decrease in current leads to a poorer ability to suppress high-frequency load changes. Secondly, there are limitations of simple current mirrors. If a basic current mirror is constructed using two transistors, its accuracy is limited by the transistor's current gain β and the Erlich voltage V. A The image error is inversely proportional to β, and the output impedance... Limited performance leads to poor load regulation. Third, compromises in commercial products. Many integrated laser driver chips, for versatility, integrate DACs and modulation circuits internally, but their core analog output stage remains based on the aforementioned simple architecture. Their noise performance, linearity, and maximum output current often cannot simultaneously meet the demands of high-end applications. Users are confined to the chip's fixed performance framework, lacking flexibility.

[0003] For example, Chinese patent CN101859980A (laser driver) uses a differential current generation circuit to improve the rise / fall time characteristics of a vertical cavity surface-emitting laser, but its linearity is still limited by the thermal noise of the feedback resistor and the bandwidth of the operational amplifier at high currents, resulting in an output current distortion rate as high as 3%-5%, which cannot meet the requirements of precision optical communication applications. Another example is US patent US6587490B2 (Low-noise current source driver for laser diodes), which reduces noise through a shunt regulator, but neglects dynamic linear optimization. When the modulation depth is >80%, the current nonlinearity error exceeds 2%, leading to uneven laser output power.

[0004] Ultimately, the common flaw in existing solutions lies in their inability to completely eliminate the base-emitter voltage drop (Vo) in semiconductor devices (such as bipolar transistors). be Offset voltage V of the operational amplifier os The introduced nonlinearity, temperature drift, and instability are inherent error sources that severely limit the performance of current sources in precision driving and measurement applications. Therefore, there is an urgent need for a high-linearity current source solution that can fundamentally isolate these errors and achieve output current determined solely by the resistance ratio. Utility Model Content

[0005] This invention provides a high linearity current source circuit for laser driving, which can effectively suppress the influence of inherent nonlinearity and temperature drift of semiconductor devices on output accuracy. Through a closed-loop feedback structure based on pure resistance ratio, it achieves high precision and high stability current output, and integrates fast and reliable overcurrent protection function to ensure the safety and performance stability of sensitive loads such as lasers under a wide range of operating conditions.

[0006] To achieve these objectives and other advantages of this invention, a high linearity current source circuit for laser driving is provided, comprising: The error control module has a non-inverting input terminal for receiving an input control voltage and an inverting input terminal for receiving a feedback voltage. It is used to compare the input control voltage with the feedback voltage and output an error control signal. A power output module, the input of which is connected to the output of the error control module, is used to drive the load according to the error control signal; A current detection module, which is connected in series in the current output circuit of the power output module, is used to obtain a detection voltage that is proportional to the output current. A differential feedback module, whose input terminal is connected across the two ends of the current detection module, is used to differentially amplify the detected voltage and generate the feedback voltage. The output terminal of the differential feedback module is connected to the inverting input terminal of the error control module. An overcurrent protection module has its input terminal connected to one end of the current detection module connected to the power output module, and its output terminal connected to the drive terminal of the power output module. The overcurrent protection module includes a comparison unit and a current limiting execution unit, which is used to limit the drive signal of the power output module through the current limiting execution unit when the detected voltage exceeds a preset threshold.

[0007] Preferably, the error control module includes a first operational amplifier U1A, a first compensation capacitor C2, and a first bias resistor R12; Wherein, the non-inverting input terminal of the first operational amplifier U1A constitutes the non-inverting input terminal of the error control module, used to receive the input control voltage Vi; the inverting input terminal constitutes the inverting input terminal of the error control module, used to receive the feedback voltage; and the output terminal constitutes the output terminal of the error control module, outputting the error control signal. The first compensation capacitor C2 is connected between the output terminal and the inverting input terminal of the first operational amplifier U1A. The first bias resistor R12 is connected to the non-inverting input terminal of the first operational amplifier U1A, used to provide a DC path for the input bias current of the first operational amplifier U1A.

[0008] Preferably, the power output module includes a second operational amplifier U1B, a second compensation capacitor C1, and an N-channel MOS field-effect transistor Q1; In this configuration, the non-inverting input of the second operational amplifier U1B is connected to the output of the first operational amplifier U1A; the second compensation capacitor C1 is connected between the output of the second operational amplifier U1B and its inverting input; the output of the second operational amplifier U1B is connected to the gate of the N-channel MOS field-effect transistor Q1; the drain of the N-channel MOS field-effect transistor Q1 is connected to one end of the load Rload, the other end of the load Rload is used to connect to the power supply, and the source of the N-channel MOS field-effect transistor Q1 is connected to the current detection module.

[0009] Preferably, the current detection module is a high-precision current detection resistor R1; the differential feedback module includes a third operational amplifier U1C and an external matching resistor network. The current sensing resistor R1 has its first terminal connected to the source of the N-channel MOS field-effect transistor Q1, and its second terminal connected to the first negative power supply VEE. The matching resistor network includes a second resistor R2, a third resistor R3, a fourth resistor R4, and a fifth resistor R5. The non-inverting input of the third operational amplifier U1C is connected to the first terminal of the current sensing resistor R1 through the second resistor R2, and grounded through the third resistor R3. The inverting input of the third operational amplifier U1C is connected to the second terminal of the current sensing resistor R1 through the fourth resistor R4, and connected to the output of the third operational amplifier U1C through the fifth resistor R5. The resistance values ​​of the second resistor R2 and the fourth resistor R4 are equal, and the resistance values ​​of the third resistor R3 and the fifth resistor R5 are equal.

[0010] Preferably, the comparator unit in the overcurrent protection module is a fourth operational amplifier U1D, and the current limiting execution unit is a diode D1; The fourth operational amplifier U1D forms a voltage comparator, with its inverting input connected to the first terminal of the current detection module and its non-inverting input connected to a negative preset threshold voltage. The anode of the diode D1 is connected to the second negative power supply VEE and the gate of the N-channel MOS field-effect transistor Q1, and the cathode of the diode D1 is connected to the output terminal of the fourth operational amplifier U1D. The voltage values ​​of the first negative power supply VEE and the second negative power supply VEE are lower than the preset threshold voltage. When no current flows through the current detection module, its first terminal voltage is equal to the voltage of the first negative power supply VEE, so that at the initial power-on or when the current detection module has no current, the output of the fourth operational amplifier U1D is high, the diode D1 is reverse biased and cut off, and the overcurrent protection module does not interfere with the operation of the main control loop. When the output current increases, causing the voltage at the first terminal of the current detection module to be higher than the preset threshold voltage, the output of the fourth operational amplifier U1D flips to a low level, causing the diode D1 to conduct in the forward direction and limiting the output current.

[0011] Preferably, the resistance values ​​of the second resistor R2 and the fourth resistor R4 are equal, and the resistance values ​​of the third resistor R3 and the fifth resistor R5 are equal, and the following relationship is satisfied: the gain of the differential feedback module is... The feedback voltage I out This is the driving current flowing through the laser load Rload.

[0012] Preferably, it also includes a gate drive current limiting resistor R10, which is connected in series between the output terminal of the second operational amplifier U1B and the gate of the N-channel MOS field-effect transistor Q1, to limit the drive current of the power output module and improve loop stability.

[0013] Preferably, it also includes an interstage coupling and biasing network, which includes a sixth resistor R6 and a seventh resistor R7; the sixth resistor R6 is connected between the output terminal of the first operational amplifier U1A and the non-inverting input terminal of the second operational amplifier U1B; the seventh resistor R7 is connected between the non-inverting input terminal of the second operational amplifier U1B and the second negative power supply VEE.

[0014] Preferably, the error control module, power output module, differential feedback module, and overcurrent protection module are integrated into the same quad operational amplifier chip.

[0015] This utility model has at least the following beneficial effects: First, by constructing a complete closed-loop system encompassing error control, power output, differential feedback, and overcurrent protection, high-linearity current output is achieved. The error control module compares the input voltage with the feedback voltage in real time, ensuring the output current accurately follows the command. The differential feedback module accurately detects the output current and converts it into a feedback voltage, effectively suppressing common-mode noise and improving detection accuracy. The independent overcurrent protection module can quickly limit the drive signal when abnormally large currents are detected, effectively protecting the load and power devices, and improving the system's robustness and safety. The entire architecture is clearly structured and works collaboratively, maintaining excellent static and dynamic performance under a wide range of load and temperature variations.

[0016] Secondly, by employing an operational amplifier combined with a compensation capacitor and a bias resistor, the error control module achieves stable and accurate error amplification. The first bias resistor R12 provides the necessary DC path for the operational amplifier input stage, completely eliminating the input potential drift problem caused by the lack of a bias current path and reducing DC offset error. The first compensation capacitor C2 introduces dominant pole compensation, effectively widening the loop's phase margin, suppressing the risk of high-frequency oscillation, and ensuring the stability of the error amplification stage and even the entire current source loop, laying a solid foundation for high-precision control.

[0017] Third, this power output module utilizes the second operational amplifier U1B to drive the N-channel MOS field-effect transistor Q1, forming a high-gain, fast-response voltage buffer and power amplification stage. The second compensation capacitor C1 further optimizes the frequency characteristics of this local loop, improving the stability and response speed of the driver stage. The N-channel MOS field-effect transistor, as an output device, has the advantages of simple driving, low on-resistance, and fast switching speed. This structure ensures that the error control signal can be converted into a large current driving the load with low distortion and high efficiency. Simultaneously, due to the deep negative feedback of the operational amplifier, the nonlinear characteristics of the power stage are greatly suppressed, thus guaranteeing excellent linearity of the overall circuit.

[0018] Fourth, a high-precision current sensing resistor R1 is used in conjunction with a differential amplifier circuit composed of a matching resistor network to achieve precise, low-noise detection of the output current. This differential structure effectively suppresses ground noise and other common-mode interference across the sensing resistor, converting the minute current signal into a clean differential voltage signal. Strictly requiring equal resistance values ​​for R2 and R4, and R3 and R5, ensures the differential amplifier has an extremely high common-mode rejection ratio and precise differential-mode gain. This detection method fundamentally improves the accuracy and signal-to-noise ratio of the feedback signal, and is one of the core guarantees for achieving high linearity current output.

[0019] Fifth, the overcurrent protection mechanism provided by this utility model is ingenious and reliable. By connecting the inverting input of the comparator to the current detection point and the non-inverting input to a preset threshold, direct and rapid comparison of the current signal is achieved. Using diode D1 as the current limiting execution unit, it is reverse-biased and cut off during normal operation due to the high level output of the comparator, having no impact on the main drive signal path and ensuring the integrity of the main loop. Once an overcurrent occurs, the comparator output flips to a low level, D1 conducts forward, clamping the gate voltage of the N-channel MOS field-effect transistor Q1 to a low level, thereby quickly limiting the current. This scheme has a rapid response, accurate threshold, and is decoupled from the main loop, achieving intelligent protection that is "undisturbed when nothing is happening and decisive when something happens."

[0020] Sixth, the gain relationship and resistance matching requirements of the differential feedback module are clearly defined, thus establishing a precise and stable mathematical relationship between the feedback voltage and the output current: This indicates the final output current. Controlled solely by the input control voltage V i The ratio R3 / R2 is determined by the sensing resistor R1 and the resistance ratio R3 / R2. By selecting a high-precision, low-temperature-drift matching resistor, this ratio can be ensured to remain constant under temperature changes and long-term operation. This theoretically eliminates the nonlinearity and temperature drift errors introduced by active devices (such as operational amplifier offset and transistor parameters), achieving ultra-high linearity current conversion that is close to the ideal ratio.

[0021] Other advantages, objectives and features of this invention will be partly apparent from the following description, and partly understood by those skilled in the art through study and practice of this invention. Attached Figure Description

[0022] Figure 1 This is a schematic diagram of the high linearity current source circuit for laser driving according to this utility model. Detailed Implementation

[0023] The present invention will now be described in further detail so that those skilled in the art can implement it based on the description.

[0024] It should be understood that terms such as “having,” “comprising,” and “including” as used herein do not exclude the presence or addition of one or more other elements or combinations thereof.

[0025] like Figure 1 As shown, this utility model embodiment provides a high linearity current source circuit for laser driving, comprising: an error control module, whose non-inverting input terminal is used to receive an input control voltage and its inverting input terminal is used to receive a feedback voltage, for comparing the input control voltage with the feedback voltage and outputting an error control signal; a power output module, whose input terminal is connected to the output terminal of the error control module, for driving a load according to the error control signal; a current detection module, which is connected in series in the current output circuit of the power output module, for acquiring a detection voltage proportional to the output current; a differential feedback module, whose input terminal is connected across the two ends of the current detection module, for differentially amplifying the detection voltage and generating the feedback voltage, the output terminal of the differential feedback module being connected to the inverting input terminal of the error control module; and an overcurrent protection module, whose input terminal is connected to one end of the current detection module connected to the power output module, and whose output terminal is connected to the driving terminal of the power output module, the overcurrent protection module including a comparison unit and a current limiting execution unit, for limiting the driving signal of the power output module through the current limiting execution unit when the detection voltage exceeds a preset threshold.

[0026] In the above embodiment, the high linearity current source circuit for laser driving comprises five cooperating modules forming a complete closed-loop control system, achieving precise, stable, and safe current driving of the laser load. First, the error control module is the core of the entire circuit's instruction processing and comparison. This error control module has a non-inverting input terminal for receiving an external input control voltage, typically provided by a precision reference source or CNC circuit, used to set the desired output current magnitude. The inverting input terminal receives a feedback voltage reflecting the actual output current condition. Internally, the error control module uses a comparison circuit composed of a high-precision operational amplifier to continuously compare these two voltages in real time. The principle is to utilize the high open-loop gain of the operational amplifier to amplify the minute voltage difference into an error control signal capable of driving subsequent stages. The error control signal directly reflects the deviation between the set value and the actual value and is transmitted as a correction command to the subsequent power stage.

[0027] Secondly, the power output module, acting as the circuit's actuator, is responsible for converting the error control signal into an actual current capable of driving the laser load. Its input is directly connected to the output of the error control module, receiving the aforementioned error control signal. The power output module typically employs a voltage-to-current conversion stage composed of field-effect transistors (such as MOSFETs) driven by operational amplifiers. The operational amplifier ensures that the gate voltage of the driving transistor strictly follows the error control signal, while the power transistor operates in the linear or saturation region, undertaking the task of current amplification. Its output is directly connected to the laser load, forming a current output loop. Through this design, the error signal is rapidly and with low distortion converted into a driving current, achieving precise control of the load.

[0028] A current sensing module is connected in series in the current output circuit of the power output module to monitor the actual current flowing through the load in real time. The core of the current sensing module is a high-precision, low-temperature-drift sampling resistor. When the output current flows through this sampling resistor, Ohm's law generates a small voltage drop proportional to the current magnitude; this voltage drop is the sensing voltage. The current sensing module is placed on the "low" side of the output circuit, close to the negative terminal of the power supply or the reference potential. This layout helps simplify detection and reduce common-mode interference. The acquired sensing voltage is the direct basis for subsequent feedback and protection decisions.

[0029] The main function of the differential feedback module is to amplify and condition the minute detected voltage obtained from the current sensing module, and convert it into a feedback voltage suitable for processing by the error control module. The input of the differential feedback module is connected across the current sensing resistor, extracting the detected voltage in a differential form. Internally, it typically consists of an instrumentation amplifier or a differential amplifier circuit composed of a precision-matched resistor network and an operational amplifier. This differential structure effectively suppresses common-mode noise (such as ground interference) that may exist across the current sensing resistor, thus extracting a pure current signal with high fidelity. The amplified signal is the feedback voltage, which is directly fed back to the inverting input of the error control module and compared with the input control voltage, thereby closing the entire control loop. By precisely setting the gain of the differential amplifier, a stable and linear proportional relationship can be established between the feedback voltage and the actual output current.

[0030] The overcurrent protection module is an independent and fast-response safety monitoring mechanism used to prevent excessive current from damaging lasers or power devices due to load short circuits, abnormal transients, or other faults. Its input is connected to the end of the current sensing module connected to the power output module, thus directly monitoring the voltage at a critical node reflecting the current magnitude. Internally, the overcurrent protection module contains a comparator unit and a current-limiting execution unit. The comparator unit continuously compares the voltage at the monitored point (representing the current magnitude) with a preset threshold voltage. This threshold voltage can be set using a precision voltage divider network or a reference source. When the circuit is operating normally, the monitored voltage is below the threshold, and the comparator unit output keeps the current-limiting execution unit in the off state, completely not interfering with the main loop. Once an overcurrent occurs, the monitored voltage exceeds the threshold, the comparator unit quickly flips, driving the current-limiting execution unit to operate. The execution unit typically intervenes directly at the drive end of the power output module (e.g., pulling down the gate voltage of the power MOSFET) using parallel clamping or series current limiting, thereby forcibly limiting or cutting off the output current, achieving millisecond-level or even faster protection response.

[0031] During circuit operation, the five modules mentioned above work together. After power-on, the input control voltage sets the target current value. The error control module compares this set value with the initial zero feedback value, generating an initial drive signal. The power output module then drives the load, generating output current. The current flows through the sensing resistor, creating a voltage drop, which is amplified by the differential feedback module and sent back to the error control module to form negative feedback. The entire closed-loop system adjusts rapidly until the feedback voltage equals the input control voltage, at which point the output current stabilizes precisely at the set value.

[0032] Compared to traditional laser driving schemes that employ simple common-ground switching or basic current mirrors, this circuit, by introducing differential feedback detection with a high common-mode rejection ratio and a complete closed-loop control architecture, significantly suppresses errors introduced by inherent nonlinear factors in semiconductor devices such as operational amplifier offset voltage and transistor parameter temperature drift. This results in a fundamental improvement in the accuracy and long-term stability of the output current. Simultaneously, an independent overcurrent protection mechanism provides fast and reliable fault protection capabilities without excessively impacting the main loop performance, enhancing the system's robustness against abnormal conditions such as sudden load changes or short circuits. Overall, this scheme represents a substantial improvement over existing common technologies in terms of output current linearity, adaptability to load and temperature variations, and system safety.

[0033] In one specific embodiment, the error control module includes a first operational amplifier U1A, a first compensation capacitor C2, and a first bias resistor R12; Wherein, the non-inverting input terminal of the first operational amplifier U1A constitutes the non-inverting input terminal of the error control module, used to receive the input control voltage Vi; the inverting input terminal constitutes the inverting input terminal of the error control module, used to receive the feedback voltage; and the output terminal constitutes the output terminal of the error control module, outputting the error control signal. The first compensation capacitor C2 is connected between the output terminal and the inverting input terminal of the first operational amplifier U1A. The first bias resistor R12 is connected to the non-inverting input terminal of the first operational amplifier U1A, used to provide a DC path for the input bias current of the first operational amplifier U1A.

[0034] In the above embodiment, the first operational amplifier U1A is essentially a high-gain, low-offset voltage comparison and error amplification device. In this circuit, the non-inverting input of U1A is configured to receive an external input control voltage Vi, typically provided by a precision digital-to-analog converter or an adjustable reference source, used to set the target current value flowing through the laser load. The inverting input of U1A receives a feedback voltage from a subsequent differential feedback module, which accurately reflects the actual output current. U1A, with its extremely high open-loop gain, amplifies the minute difference between these two input voltages and outputs a corresponding error control signal. The error control signal represents the correction amount required by the system to eliminate the deviation between the setpoint and the actual value, and is the command source for the entire closed-loop control system. To ensure that U1A can operate normally in the linear region and avoid input stage saturation, its non-inverting input terminal provides a DC path to power ground through the first bias resistor R12. The value of this resistor needs to be selected by comprehensively considering the input bias current of the op-amp and the allowable input voltage drop. Typically, a value of tens to hundreds of kilohms can be selected. For example, 100kΩ is a common and effective choice. It can provide a suitable bias voltage for µA-level input bias current without introducing excessive noise or offset.

[0035] To ensure the dynamic stability of the error amplification stage and suppress possible high-frequency oscillations, a first compensation capacitor C2 is connected between the output of the first operational amplifier U1A and its inverting input. The first compensation capacitor C2 introduces dominant pole compensation into the circuit, reducing the amplifier's gain in the high-frequency range, thereby expanding the phase margin of the entire control loop and preventing self-oscillation caused by accumulated loop phase shift. The value of the first compensation capacitor C2 needs to be selected based on a trade-off between the operational amplifier's gain-bandwidth product, the overall feedback depth of the circuit, and the desired transient response speed; preferably, a value of 1000pF is selected.

[0036] The three technical features mentioned above—the first operational amplifier U1A, the first compensation capacitor C2, and the first bias resistor R12—constitute a complete error control unit through specific electrical connections. Its operation is as follows: After power-on, the input control voltage Vi is applied to the non-inverting input of U1A, while the initial feedback voltage is zero. U1A then detects that the voltage at the positive input terminal is higher than that at the negative input terminal, thus outputting a positively rising error voltage. This voltage is amplified by the subsequent power stage and drives the load, generating an output current. Simultaneously, the bias resistor R12 ensures the stability of the operating point of the U1A input stage transistor, preventing floating and uncontrolled input potential due to insufficient bias current discharge. The first compensation capacitor C2 smooths the output changes of U1A throughout the adjustment process, suppressing any possible rapid jumps or ringing phenomena, making the generation of the error signal both fast and stable. These three components work together to achieve accurate and stable detection and amplification of the deviation between the target current setpoint and the actual current feedback value, laying a crucial first link for the precise control of the entire current source.

[0037] like Figure 1 The schematic diagram shown is based on Multisim software version 14.0. In this high linearity current source circuit, the symmetrical dual power supply set of VDD +5V and VEE -5V is the core foundation for the system's operation. Among them, VDD +5V supplies power to the positive power terminals of all operational amplifiers and the laser load, constituting the energy source for the current output.

[0038] In one specific embodiment, the power output module includes a second operational amplifier U1B, a second compensation capacitor C1, and an N-channel MOS field-effect transistor Q1; In this configuration, the non-inverting input of the second operational amplifier U1B is connected to the output of the first operational amplifier U1A; the second compensation capacitor C1 is connected between the output of the second operational amplifier U1B and its inverting input; the output of the second operational amplifier U1B is connected to the gate of the N-channel MOS field-effect transistor Q1; the drain of the N-channel MOS field-effect transistor Q1 is connected to one end of the load Rload, the other end of the load Rload is grounded, and the source of the N-channel MOS field-effect transistor Q1 is connected to the current detection module.

[0039] Specifically, it also includes a gate drive current limiting resistor R10, which is connected in series between the output terminal of the second operational amplifier U1B and the gate of the N-channel MOS field-effect transistor Q1, to limit the drive current of the power output module and improve loop stability.

[0040] It also includes an interstage coupling and biasing network, which includes a sixth resistor R6 and a seventh resistor R7. The sixth resistor R6 is connected between the output terminal of the first operational amplifier U1A and the non-inverting input terminal of the second operational amplifier U1B. The seventh resistor R7 is connected between the non-inverting input terminal of the second operational amplifier U1B and the second negative power supply VEE. The inverting input terminal of the second operational amplifier U1B is grounded through an eighth resistor R8.

[0041] In the above embodiment, the core drive link of the power output module consists of a second operational amplifier U1B, a second compensation capacitor C1, and an N-channel MOS field-effect transistor Q1. U1B acts as a voltage buffer and driver with high input impedance and low output impedance. Its non-inverting input receives the error control signal output from the preceding error control module. The error control signal is a precisely amplified voltage command representing the amount of drive required to correct the output current deviation. The output of U1B directly drives the gate of the N-channel MOS field-effect transistor Q1, rapidly establishing or changing the gate voltage through its powerful output capability. Q1, as the final current regulation element, operates in the linear or saturation region, and its source current, i.e., the load current, is precisely controlled by the gate-source voltage. The second compensation capacitor C1 is connected between the output of U1B and the inverting input, forming a local voltage follower negative feedback loop. Its main function is to introduce phase lead or lag to offset the phase lag introduced by the gate capacitance of Q1 and the wiring distribution parameters, thereby effectively suppressing possible high-frequency oscillations in the drive stage and ensuring its response to the control signal is both fast and stable. For example, the capacitance value of C1 is selected as 100 picofarads. During operation, when the error signal increases, the output of U1B rises, the gate voltage of Q1 rises accordingly, and the conduction degree deepens, thereby driving a larger current to flow through the load and the sense resistor; and vice versa. This core link realizes low-distortion and high-efficiency conversion of voltage command to power current.

[0042] To further optimize drive performance and enhance reliability, a gate drive current-limiting resistor R10 is connected in series between the output of U1B and the gate of Q1. This current-limiting resistor R10 serves a dual function: firstly, it limits the instantaneous peak current of U1B charging and discharging the gate capacitor of Q1, preventing overload or stability degradation of the op-amp output due to driving excessive capacitive loads, thus protecting both the op-amp and the MOSFET; secondly, it forms a low-pass filter together with the gate input capacitor of Q1, filtering out high-frequency noise from the preceding stage or power supply, and moderately slowing down the edge change rate of the drive voltage. This helps eliminate ringing phenomena that may be caused by overly steep switching processes, thereby significantly improving the stability of the entire power loop. The value of R10 needs to be selected to balance drive speed and current-limiting filtering effect. Too high a value will slow down the switching speed, while too low a value will have limited current-limiting filtering effect. A moderate resistance value is typically selected; for example, a value of 4.99 kΩ is chosen.

[0043] To ensure effective and reliable signal transmission from the preamplifier stage to the power drive stage and to set the correct DC operating point, the circuit employs an interstage coupling and biasing network consisting of resistors R6 (sixth) and R7 (seventh). R6 is connected between the preamplifier output and the non-inverting input of U1B. As a coupling resistor, it isolates the DC operating points of the preamplifier and U1B to some extent and limits potential abnormal currents. R7 is connected between the non-inverting input of U1B and the second negative power supply VEE (e.g., -5V). Its core function is to form a voltage divider network with R6, providing a suitable DC bias voltage slightly below ground for the non-inverting input of U1B. This ensures that even when the input error command voltage is zero, U1B and Q1 operate in the linear amplification region, avoiding crossover distortion and guaranteeing a smooth start-up from zero current. The matching values ​​of R6 and R7 require careful design. For example, R6 and R7 can both be 20 kΩ. The eighth resistor is also 20 kΩ.

[0044] By constructing a complete power output stage that includes precision driving, local compensation, gate protection, and DC bias, the minute error signal from the front-end is reliably and linearly amplified into the large current required to drive the laser load. This scheme effectively suppresses the nonlinearity and oscillation risks that may be introduced by the driving stage itself, significantly improving the stability and reliability of power conversion. At the same time, a reasonable bias design ensures good linearity and fast response capability of the circuit over a wide dynamic range, providing a solid guarantee for achieving the final output performance of the entire high-linearity current source.

[0045] In one specific embodiment, the current detection module is a high-precision current detection resistor R1; the differential feedback module includes a third operational amplifier U1C and an external matching resistor network. The current sensing resistor R1 has its first terminal connected to the source of the N-channel MOS field-effect transistor Q1, and its second terminal connected to the first negative power supply VEE. The matching resistor network includes a second resistor R2, a third resistor R3, a fourth resistor R4, and a fifth resistor R5. The non-inverting input of the third operational amplifier U1C is connected to the first terminal of the current sensing resistor R1 through the second resistor R2, and grounded through the third resistor R3. The inverting input of the third operational amplifier U1C is connected to the second terminal of the current sensing resistor R1 through the fourth resistor R4, and connected to the output of the third operational amplifier U1C through the fifth resistor R5. The output of the third operational amplifier U1C is connected to the inverting input of the first operational amplifier U1A through an eleventh resistor R11. The resistance values ​​of the second resistor R2 and the fourth resistor R4 are equal, and the resistance values ​​of the third resistor R3 and the fifth resistor R5 are equal.

[0046] Specifically, the resistance values ​​of the second resistor R2 and the fourth resistor R4 are equal, and the resistance values ​​of the third resistor R3 and the fifth resistor R5 are equal, satisfying the following relationship: the gain of the differential feedback module is... The feedback voltage I out This is the driving current flowing through the laser load Rload.

[0047] In the above implementation, the current detection module and the differential feedback module, which constitute the core of signal detection and feedback, work together to achieve high-precision, low-noise sensing and conversion of the output current.

[0048] The core of the current sensing module is a high-precision current sensing resistor R1 directly connected in series in the power output circuit. One end of this current sensing resistor is connected to the source of the N-channel MOS field-effect transistor Q1, and the other end is connected to the system's first negative power supply VEE (e.g., -5V). Its working principle is based on Ohm's law: when the driving current flows through the laser load and Q1, it all flows through R1, generating a small voltage drop across it that is strictly proportional to the current magnitude. This voltage drop is the original detection voltage signal. The resistance value of R1 needs to be balanced between detection sensitivity and power loss: too large a resistance value will generate unnecessary heat loss and affect output efficiency; too small a resistance value will result in a too small detection voltage, easily drowned out by circuit noise. For example, the resistance value of R1 is chosen to be 2 ohms.

[0049] To accurately extract and amplify the minute differential voltage across R1 while effectively suppressing common-mode interference, the circuit employs a differential amplifier consisting of a third operational amplifier U1C and external resistors as the differential feedback module. U1C is configured in differential amplification mode. Its non-inverting input is connected to the high-potential end (Q1 source terminal) of the sensing resistor R1 via the second resistor R2, and to signal ground via the third resistor R3. Its inverting input is connected to the low-potential end (VEE terminal) of R1 via the fourth resistor R4, and to its own output via the fifth resistor R5, forming negative feedback. This connection method allows U1C to amplify only the voltage difference across R1, while significantly suppressing potential fluctuations common to both ends (such as fluctuations caused by VEE power supply ripple or ground noise). During operation, the voltage difference signal representing the output current is applied to the differential input of U1C. After high-gain amplification, it outputs a ground-referenced, significantly increased, and "clean" feedback voltage V. f .

[0050] The key to ensuring the superior performance of this differential amplifier lies in the precise symmetry of its external matching resistor network. Specifically, the second resistor R2 and the fourth resistor R4 must maintain strictly equal resistance values, and the third resistor R3 and the fifth resistor R5 must also maintain strictly equal resistance values. This strict matching requirement is the cornerstone of the differential amplifier's high common-mode rejection ratio and precise differential-mode gain. When these four resistors are selected from the same batch of precision thin-film resistors with extremely low tolerances (e.g., 0.1%) and similar temperature coefficients (e.g., ≤25ppm / °C), for example, setting R2 and R4 to 25kΩ and R3 and R5 to 50.0kΩ, the amplifier's common-mode rejection capability will reach a very high level, and its differential-mode gain will be precisely set to R3 / R2. The coordinated operation of the entire module is as follows: Output current I out A pressure drop I is generated when the current flows through R1. out ×R1. This differential voltage is applied to the network consisting of R2-R5 and U1C, and the output voltage V of U1C is... f Accurately magnified to I out ×R1×(R3 / R2). This voltage is then sent back to the inverting input of the preceding error amplifier and compared with the set voltage, thereby closing the entire high-precision control loop.

[0051] According to the calculation results, the current magnitude is entirely determined by the input terminal Vi and the sensing resistor R1, and is independent of the drive voltage and load resistor. VEE and Q1 are irrelevant. This eliminates the impact of Q1 leakage current on accuracy, current stability caused by driving voltage fluctuations, and the influence of thermal noise from the load operation. The output of U1C serves as a conversion voltage for monitoring the load current and is simultaneously input to the inverting input of U1A. This is the constant current (ACC) mode. To achieve constant power (APC) mode, simply connect the inverting input of U1A to the voltage converted from the photocurrent.

[0052] This implementation combines a high-precision sampling resistor with a differential amplifier featuring a tightly matched resistor network to construct a current sensing and feedback channel with extremely strong anti-interference capabilities and high conversion accuracy. This scheme fundamentally suppresses the impact of common-mode interference such as power supply noise and ground fluctuations on the detection signal, linearly and stably converting weak current signals into high-quality feedback voltages. This provides the most critical signal guarantee for the entire closed-loop current source to achieve ultra-high linearity and extremely low temperature drift.

[0053] In one specific embodiment, the comparison unit in the overcurrent protection module is a fourth operational amplifier U1D, and the current limiting execution unit is a diode D1; The fourth operational amplifier U1D constitutes a voltage comparator. Its inverting input is connected to the first terminal of the current detection module through the thirteenth resistor R13, and is also connected to the source of the N-channel MOS field-effect transistor Q1. The source of the N-channel MOS field-effect transistor Q1 is also connected to the inverting input of the second operational amplifier U1B through the ninth resistor R9. The non-inverting input is connected to a negative preset threshold voltage. The anode of the diode D1 is connected to the gate of the N-channel MOS field-effect transistor Q1, and is connected to the second negative power supply VEE through the sixteenth resistor R16. The cathode of the diode D1 is connected to the output of the fourth operational amplifier U1D. The voltage values ​​of the first negative power supply VEE and the second negative power supply VEE are lower than the preset threshold voltage. The specific setting method of the negative preset threshold voltage is as follows: the non-inverting input of the fourth operational amplifier U1D is connected to the first negative power supply VEE through the fourteenth resistor R14, and is also connected to ground through the fifteenth resistor R15.

[0054] When no current flows through the current detection module, its first terminal voltage is equal to the voltage of the first negative power supply VEE, so that at the initial power-on or when the current detection module has no current, the output of the fourth operational amplifier U1D is high, the diode D1 is reverse biased and cut off, and the overcurrent protection module does not interfere with the operation of the main control loop. When the output current increases, causing the voltage at the first terminal of the current detection module to be higher than the preset threshold voltage, the output of the fourth operational amplifier U1D flips to a low level, causing the diode D1 to conduct in the forward direction and limiting the output current.

[0055] In the above implementation, the core of the overcurrent protection module's monitoring and decision-making is the fourth operational amplifier U1D, which is configured as a voltage comparator responsible for continuously determining whether the output current is within a safe range. The inverting input of U1D is connected via a thirteenth resistor R13 to a key node characterizing the actual output current, namely the first end of the current sensing resistor R1. This point is also directly related to the source potential of the power MOSFET Q1. R13 here serves as a current limiter and a slight filter, preventing potential voltage spikes from directly impacting the op-amp's input. For example, its resistance value can be chosen as 10 kΩ. It should be noted that the source of Q1 is also connected to the inverting input of the second operational amplifier U1B via a ninth resistor R9 (for example, with a value of 20 kΩ). This connection constitutes a local feedback loop in the power output stage, but this feedback signal is also simultaneously presented at the overcurrent protection monitoring point, allowing U1D to perceive the most accurate output stage state.

[0056] The non-inverting input of the U1D is set with a fixed negative voltage as the threshold reference for protection operation. This threshold voltage is generated by a precision voltage divider network, specifically formed by connecting the fourteenth resistor R14 and the fifteenth resistor R15 in series between the first negative power supply VEE (e.g., -5V) and ground. The threshold voltage value is taken from the connection point of these two resistors. By adjusting the resistance ratio of R14 and R15, the overcurrent trigger point can be set flexibly and precisely. For example, if VEE is -5V and a threshold of -3V is desired, R14 can be selected as 20 kΩ and R15 as 30 kΩ. This pure resistor voltage divider method provides a stable threshold and is inexpensive.

[0057] The protection mechanism is implemented using diode D1 and its associated circuitry. The anode of D1 is directly connected to the gate of the N-channel MOS field-effect transistor Q1, a critical control point for the main power drive signal; its cathode is connected to the output of comparator U1D. A key design detail is the presence of a sixteenth resistor, R16, between the gate of Q1 and the second negative power supply VEE. During normal operation, U1D outputs a high level (close to the positive power supply voltage) because the voltage at its inverting input (generated by a small current) is lower than the threshold voltage at its non-inverting input. At this time, diode D1 is reverse-biased and cut off because its cathode potential is higher than its anode potential, completely isolating the protection circuit from the main drive path and ensuring that the performance of the main loop is unaffected. The sixteenth resistor R16, working in conjunction with the drive stage circuitry, provides a static negative bias path for the gate of Q1, contributing to improved gate potential stability.

[0058] During power-on or normal operation, the current flowing through the sensing resistor R1 is small, and its first terminal voltage is close to VEE (e.g., -5V), far below the set threshold (e.g., -3V). Comparator U1D continuously outputs a high level, and D1 is cut off. Once the output current increases sharply due to a load short circuit or abnormality, the voltage drop across R1 increases accordingly, causing the voltage at the inverting input of U1D (i.e., the source voltage of Q1) to rise from negative to zero volts. When this voltage exceeds the negative threshold voltage set at the non-inverting input, the output state of U1D flips, jumping from high level to low level (close to its negative supply voltage). This transition causes the cathode potential of diode D1 to momentarily drop below the anode potential, and D1 immediately switches from cutoff to forward conduction. After conduction, D1 acts like a rapidly closing switch, pulling the gate potential of power transistor Q1 towards the low level output by U1D. At the same time, the sixteenth resistor R16 limits the peak current at this instant, protecting diode D1 and the output stage of U1D. As a result of this action, the gate drive voltage of Q1 is forcibly pulled down, and its conduction level is rapidly reduced, thereby effectively clamping the increase of output current and preventing further runaway current.

[0059] Through a simple yet ingenious independent circuit module, fast, reliable, and interference-free overcurrent protection is provided for the entire high-linearity current source. This solution can continuously monitor without the main loop noticing and intervene extremely quickly in the early stages of a fault, actively limiting the drive signal. This significantly improves the robustness and safety of the system in the face of dangerous situations such as abnormal loads or short circuits, effectively preventing damage to sensitive laser loads and core power devices due to overcurrent.

[0060] In one specific embodiment, the error control module, power output module, differential feedback module, and overcurrent protection module are integrated into the same quad operational amplifier chip.

[0061] In the above implementation, the core operational amplifiers of the four major modules—error control, power output, differential feedback, and overcurrent protection—are integrated into a single quad operational amplifier chip. Firstly, by ensuring that all critical amplification units enjoy completely consistent process and temperature environments, parameter drift synchronization and mutual compensation are achieved, thereby significantly suppressing the overall temperature drift and nonlinear errors of the system at the physical level. Secondly, it greatly simplifies the circuit layout, reduces the number of discrete components and wiring complexity, not only significantly saving space and cost but also reducing parasitic interference and improving system stability and reliability. Finally, this highly integrated design, while retaining complete closed-loop control and fast protection functions, enables the entire current source system to possess both excellent electrical performance and superior engineering feasibility.

[0062] In summary, this invention constructs a voltage-controlled current source that is completely unaffected by the semiconductor junction voltage drop and is determined by the pure resistance ratio.

[0063] The number of devices and processing scale described herein are for the purpose of simplifying the description of this utility model. Applications, modifications, and variations of this utility model will be readily apparent to those skilled in the art.

[0064] Although the embodiments of this utility model have been disclosed above, they are not limited to the applications listed in the specification and embodiments. They can be applied to various fields suitable for this utility model. For those skilled in the art, other modifications can be easily made. Therefore, without departing from the general concept defined by the claims and their equivalents, this utility model is not limited to the specific details.

Claims

1. A high linearity current source circuit for laser driving, characterized in that, include: The error control module has a non-inverting input terminal for receiving an input control voltage and an inverting input terminal for receiving a feedback voltage. It is used to compare the input control voltage with the feedback voltage and output an error control signal. A power output module, the input of which is connected to the output of the error control module, is used to drive the load according to the error control signal; A current detection module, which is connected in series in the current output circuit of the power output module, is used to obtain a detection voltage that is proportional to the output current. A differential feedback module, whose input terminal is connected across the two ends of the current detection module, is used to differentially amplify the detected voltage and generate the feedback voltage. The output terminal of the differential feedback module is connected to the inverting input terminal of the error control module. An overcurrent protection module has its input terminal connected to one end of the current detection module connected to the power output module, and its output terminal connected to the drive terminal of the power output module. The overcurrent protection module includes a comparison unit and a current limiting execution unit, which is used to limit the drive signal of the power output module through the current limiting execution unit when the detected voltage exceeds a preset threshold.

2. The high linearity current source circuit for laser driving as described in claim 1, characterized in that, The error control module includes a first operational amplifier U1A, a first compensation capacitor C2, and a first bias resistor R12; Wherein, the non-inverting input terminal of the first operational amplifier U1A constitutes the non-inverting input terminal of the error control module, used to receive the input control voltage Vi; the inverting input terminal constitutes the inverting input terminal of the error control module, used to receive the feedback voltage; and the output terminal constitutes the output terminal of the error control module, outputting the error control signal. The first compensation capacitor C2 is connected between the output terminal and the inverting input terminal of the first operational amplifier U1A. The first bias resistor R12 is connected to the non-inverting input terminal of the first operational amplifier U1A, used to provide a DC path for the input bias current of the first operational amplifier U1A.

3. The high linearity current source circuit for laser driving as described in claim 2, characterized in that, The power output module includes a second operational amplifier U1B, a second compensation capacitor C1, and an N-channel MOS field-effect transistor Q1; In this configuration, the non-inverting input of the second operational amplifier U1B is connected to the output of the first operational amplifier U1A; the second compensation capacitor C1 is connected between the output of the second operational amplifier U1B and its inverting input; the output of the second operational amplifier U1B is connected to the gate of the N-channel MOS field-effect transistor Q1; the drain of the N-channel MOS field-effect transistor Q1 is connected to one end of the load Rload, the other end of the load Rload is used to connect to the power supply, and the source of the N-channel MOS field-effect transistor Q1 is connected to the current detection module.

4. The high linearity current source circuit for laser driving as described in claim 3, characterized in that, The current detection module is a high-precision current detection resistor R1; the differential feedback module includes a third operational amplifier U1C and an external matching resistor network. The current sensing resistor R1 has its first terminal connected to the source of the N-channel MOS field-effect transistor Q1, and its second terminal connected to the first negative power supply VEE. The matching resistor network includes a second resistor R2, a third resistor R3, a fourth resistor R4, and a fifth resistor R5. The non-inverting input of the third operational amplifier U1C is connected to the first terminal of the current sensing resistor R1 through the second resistor R2, and grounded through the third resistor R3. The inverting input of the third operational amplifier U1C is connected to the second terminal of the current sensing resistor R1 through the fourth resistor R4, and connected to the output of the third operational amplifier U1C through the fifth resistor R5. The resistance values ​​of the second resistor R2 and the fourth resistor R4 are equal, and the resistance values ​​of the third resistor R3 and the fifth resistor R5 are equal.

5. The high linearity current source circuit for laser driving as described in claim 4, characterized in that, The comparison unit in the overcurrent protection module is the fourth operational amplifier U1D, and the current limiting execution unit is the diode D1; The fourth operational amplifier U1D forms a voltage comparator, with its inverting input connected to the first terminal of the current detection module and its non-inverting input connected to a negative preset threshold voltage. The anode of the diode D1 is connected to the second negative power supply VEE and the gate of the N-channel MOS field-effect transistor Q1, and the cathode of the diode D1 is connected to the output terminal of the fourth operational amplifier U1D. The voltage values ​​of the first negative power supply VEE and the second negative power supply VEE are lower than the preset threshold voltage. When no current flows through the current detection module, its first terminal voltage is equal to the voltage of the first negative power supply VEE, so that at the initial power-on or when the current detection module has no current, the output of the fourth operational amplifier U1D is high, the diode D1 is reverse biased and cut off, and the overcurrent protection module does not interfere with the operation of the main control loop. When the output current increases, causing the voltage at the first terminal of the current detection module to be higher than the preset threshold voltage, the output of the fourth operational amplifier U1D flips to a low level, causing the diode D1 to conduct in the forward direction and limiting the output current.

6. The high linearity current source circuit for laser driving as described in claim 4, characterized in that, The second resistor R2 and the fourth resistor R4 have the same resistance value, and the third resistor R3 and the fifth resistor R5 have the same resistance value, and satisfy the following relationship: the gain of the differential feedback module is... The feedback voltage I out This is the driving current flowing through the laser load Rload.

7. The high linearity current source circuit for laser driving as described in claim 3, characterized in that, It also includes a gate drive current limiting resistor R10, which is connected in series between the output terminal of the second operational amplifier U1B and the gate of the N-channel MOS field-effect transistor Q1, and is used to limit the drive current of the power output module and improve loop stability.

8. The high linearity current source circuit for laser driving as described in claim 5, characterized in that, It also includes an interstage coupling and biasing network, which includes a sixth resistor R6 and a seventh resistor R7; the sixth resistor R6 is connected between the output terminal of the first operational amplifier U1A and the non-inverting input terminal of the second operational amplifier U1B; the seventh resistor R7 is connected between the non-inverting input terminal of the second operational amplifier U1B and the second negative power supply VEE.

9. The high linearity current source circuit for laser driving as described in claim 1, characterized in that, The error control module, power output module, differential feedback module, and overcurrent protection module are integrated into the same quad operational amplifier chip.

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