Inverter device

Abstract

Inverter device comprising: a positive terminal (80) connected to a positive terminal of a power source; a negative terminal (81) connected to a negative terminal of the power source; a three-phase inverter circuit connected to a motor (300) to drive the motor (300); a smoothing capacitor (160); a positive terminal connection point (90) connecting the positive terminal (80), the inverter circuit (130), and the smoothing capacitor (160); a negative terminal connection point (91) connecting the negative terminal (81), the inverter circuit (130), and the smoothing capacitor (160); a PWM generator (220) generating three-phase PWM waves used to drive the inverter circuit (130);a pulse shift unit (230) which, based on the PWM waves, generates a first command in response to which the PWM generator (220) performs a pulse shift, or generates a second command in response to which the PWM generator (220) does not perform a pulse shift; a current detector (120) which detects a DC bus current flowing between the positive-pole connection point (90) and the inverter circuit (130) or between the negative-pole connection point (91) and the inverter circuit (130) when the inverter circuit (130) is driven;an arithmetic operation unit that calculates an average DC bus current (Ical) during a PWM period by performing an arithmetic operation based on instantaneous values ​​of the DC bus current (Idc) detected at predetermined times occurring during contiguous output intervals corresponding to predetermined voltage vectors within the PWM period; and an input source current estimation unit that provides an estimated value for an input source current flowing between the positive terminal (80) and the positive terminal connection point (90) or between the negative terminal (81) and the negative terminal connection point (91) during a PWM period by determining the average DC bus current (Ical) flowing over the PWM period as the estimated value;characterized in that the arithmetic operation unit includes a motor current estimation unit that estimates a motor current based on instantaneous values ​​of the DC bus current (Idc) and the predetermined voltage vectors in accordance with which the instantaneous values ​​are detected; the arithmetic operation unit eliminates current errors (Ips) resulting from the pulse shift corresponding to the predetermined voltage vectors from the motor current in order to correct the motor current; and when the pulse shift is applied, the arithmetic operation unit calculates the mean value of the DC bus current (Ical) during the PWM period based on the corrected motor current and pulse widths of the PWM waves corresponding to the individual phases;where the current errors (Ips) attributable to pulse shifting and contained in the motor current are represented by instantaneous values ​​of the DC bus current (Idc) detected by the current detector when pulse shifting is performed by the PWM generator with PWM duty cycles corresponding to the individual phases, which are balanced with each other.

Description

INCLUDED BY REFERENCE

[0001] The disclosure of the following priority application is included here by reference: Japanese patent specification no. 2014-086095, filed on April 18, 2014. BACKGROUND OF THE INVENTION 1. Field of the invention

[0002] The present invention relates to an inverter device. 2. Description of the state of the art

[0003] In an inverter device containing an inverter through which a motor is driven, the current value of a current flowing on the source side (hereinafter referred to as an input source current) relative to the smoothing capacitor is obtained either by detecting the input source current with a current sensor or by estimating an input source current based on the value detected for an electric current flowing on the inverter circuit side relative to the smoothing capacitor, i.e., based on the detection value obtained for the inverter DC bus current (hereinafter referred to simply as a DC bus current) without the use of a current sensor.

[0004] Japanese patent no. 5069882 discloses an invention that enables the detection of the input source current (it is noted that the term "bus DC current in the converter circuit" is used to refer to this current in the patent identified above). However, adding a current sensor to detect the input source current results in increased manufacturing costs and also increases the required installation space.

[0005] To address the problems discussed above, for example, Japanese patent no. 3843391 discloses an invention relating to a technique by which, instead of detecting the input source current via a current sensor, an input source current is estimated by detecting the DC bus current and filtering the detection signal which indicates the detected DC bus current, and the motor is controlled based on the estimated input source current.

[0006] A sufficient sampling period over which the DC bus current is sampled must be ensured to enable the detection of the DC bus current.

[0007] Japanese patent publication No. H11-004594 describes a method for ensuring a sufficient DC bus current sampling period when PWM pulses are detected at an inverter device and a pulse DC bus current flowing into an inverter is detected under conditions in which the signal difference (pulse width of a line voltage) between PWM pulses corresponding to at least two phases from the three phases is less than a predetermined time period by increasing the signal difference between the PWM pulses (the pulse width of the line voltage) to achieve the required line voltage pulse width (hereinafter referred to simply as a pulse shift method).

[0008] However, when the pulse shifting method described above is used in the invention disclosed in Japanese Patent No. 3843391, a problem arises in that the magnitude of the current ripple in the DC bus current that has been subjected to filter processing increases. Summary of the invention

[0009] An inverter device that allows the input source current to be estimated with high accuracy using the pulse shift method, without the need to use a current sensor for input source current detection, has been actively pursued.

[0010] An inverter device according to the present invention comprises: a positive terminal connected to a positive terminal of a power source; a negative terminal connected to a negative terminal of the power source; a three-phase inverter circuit connected to a motor to drive the motor; a smoothing capacitor; a positive terminal connection point connecting the positive terminal, the inverter circuit, and the smoothing capacitor; a negative terminal connection point connecting the negative terminal, the inverter circuit, and the smoothing capacitor; a PWM generator generating three-phase PWM waves used to drive the inverter circuit;a pulse shift unit which, based on the PWM waves, generates a first command in response to which the PWM generator performs a pulse shift, or generates a second command in response to which the PWM generator does not perform a pulse shift; a current detector which detects a DC bus current flowing between the positive-pole connection point and the inverter circuit or between the negative-pole connection point and the inverter circuit when the inverter circuit is driven; an arithmetic operation unit which calculates an average DC bus current during a PWM period by means of an arithmetic operation performed based on instantaneous values ​​of the DC bus current detected at predetermined times occurring during contiguous output periods corresponding to predetermined voltage vectors within the PWM period;and / or an input source current estimation unit that provides an estimated value for an input source current flowing between the positive terminal and the positive terminal connection point or between the negative terminal and the negative terminal connection point during a PWM period by determining the average of the DC bus current flowing through the PWM period as the estimated value. Brief description of the drawings

[0011] Fig. Figure 1 is a block diagram showing the structure used in relation to the inverter device and similar equipment according to the present invention.

[0012] Fig. 2A to Fig. 2C indicates the relationship between the DC bus current and the motor currents as observed in a first embodiment.

[0013] Fig. Figure 3 is a waveform diagram showing the time course with which the current is detected in the first embodiment.

[0014] Fig. Figure 4 is a waveform diagram that shows how the momentum shift operation is performed in the first embodiment.

[0015] Fig. Figure 5 is a characteristic curve that presents the input source current detection results obtained in the first embodiment.

[0016] Fig. Figure 6 is a waveform diagram that shows how the input source current calculation operation is performed in a second embodiment.

[0017] Fig. Figure 7 is a diagram showing the configuration of an electric power steering system that uses the inverter device in accordance with the present invention.

[0018] Fig. Figure 8 is a diagram showing the configuration of an electric braking system that uses the inverter device in accordance with the present invention.

[0019] Fig. Figure 9 is a waveform diagram that shows how the current faults are detected in the second embodiment.

[0020] Fig. 10 is a representation that belongs to a function of the momentum displacement unit, which relates to the momentum displacement. DESCRIPTION OF EXECUTIONS – First embodiment –

[0021] Fig. Figure 1 is a block diagram showing the structure relating to the inverter device according to the present invention and elements present in its environment. A motor system 500It is suitable for use in applications where a motor is driven with high efficiency by shifting PWM wave pulses in an inverter device in accordance with the motor power, thereby improving the accuracy with which the DC bus current is detected at the inverter circuit. The motor system 500 contains a motor 300 and an inverter device 100 .

[0022] The inverter device 100 includes an inverter circuit 130 , a shunt resistor Rsh, through which the DC bus current in the inverter circuit 130 is detected, a current detector 120 , a momentum displacement unit 230 , a source flow calculation unit 240 , a three-phase calculation unit 121 , a dq converter 111 , a power control unit 110 , a PWM generator 220, a rotary position detector 150 , a smoothing capacitor 160 , a positive terminal connection 80 , a negative terminal connection 81 , a positive-pole-side connection point 90 and a connection point on the negative pole 91 The positive pole-side connection point 90 connects the positive terminal 80 , the smoothing capacitor 160 and the inverter circuit 130 The negative pole-side connection point 91 connects the negative terminal 81 , the smoothing capacitor 160 and the inverter circuit 130 .

[0023] A battery 200 is a DC voltage source for the inverter device 100 The positive electrode of the battery 200 is with the inverter device 100 via the positive terminal 80connected. Additionally, the negative electrode of the battery is 200 with the inverter device 100 via the negative terminal 81 connected. A DC voltage VB at the battery. 200 Three-phase alternating current with variable voltage and variable frequency is converted via the inverter circuit 130 converted, and the three-phase alternating current resulting from the conversion is then fed to the motor 300 created.

[0024] The engine 300 It is a synchronous motor that is driven by the applied three-phase alternating current. A rotary position sensor 320 is on the engine 300 installed to control the phases of the three-phase alternating voltage applied to it, in accordance with the phases of the voltage in the motor 300 is induced, to enable. The rotary position detector 150A detected position θ is calculated by an arithmetic operation based on a signal from the rotary position sensor. 320 The input is sent to it, and it also calculates the rotational speed ωr. Although it would be more desirable to use the rotational position sensor. 320 , which is used for these purposes to configure with a resolver built with cores and windings, the present invention can also be used together with a rotary position sensor built with a GMR sensor or a Hall element.

[0025] The inverter device 100 It features a current control function that allows it to control the motor's output. 300 to control, and detects a pulsed DC bus current that is fed into the inverter circuit 130 flows as current (DC bus current Idc) through the shunt resistor Rsh, which is located between the smoothing capacitor. 160and the inverter circuit 130 Once inserted, the flow continues. Although the shunt resistor Rsh in this example is connected to the negative terminal of the battery. 200 If it is mounted, it can instead be mounted on the positive terminal side of the battery. 200 be mounted.

[0026] The current detector 120 detects at least two detection values ​​(Id1, Id2) as DC bus current values ​​during a single PWM period, where the trigger time Trig is determined by the pulse shift unit 230 is generated (see Fig. 1 and Fig. 4).

[0027] The three-phase calculation unit 121 It estimates three-phase motor current values ​​(Iu, Iv, Iw) based on the DC bus current values ​​(Id1, Id2) and voltage vectors determined in accordance with a PWM pulse pattern. The estimation of the three-phase motor current value (Iu, Iv, Iw) is later compared with the Fig. 2A to Fig. 2C and Fig. 3 described.

[0028] The dq converter 111 It calculates estimated current values ​​(Id, Iq) by dq conversion, which is performed based on the three-phase motor current values ​​(Iu, Iv, Iw) and the rotational position θ.

[0029] The power control unit 110 It calculates a voltage setpoint (Vd*, Vq*) to compare the estimated current values ​​(Id, Iq) with the current setpoints (Id*, Iq*) generated based on a target torque. It should be noted that a constraint applies to the voltage setpoint (Vd*, Vq*) to ensure that an input source current Isrc, calculated by the source current calculation unit, is not exceeded. 240 The calculated input current must not exceed a limit value (a setpoint) of the input source current. This limitation is imposed taking into account the battery load condition and the estimated battery lifespan.

[0030] The PWM generator 220It generates PWM waves, which are to be used as a control signal, through pulse width modulation (PWM) based on the voltage setpoint (Vd*, Vq*) and the rotation angle θ. It then sends the PWM waves to the pulse shift unit. 230 .

[0031] The momentum displacement unit 230 , if they are a PWM pulse pattern (PWM waves) generated by the PWM generator 220 Once generated and received, it sends a command to the PWM generator indicating whether or not a pulse shift should be performed. 220 .

[0032] With reference to Fig. 10 becomes a function of the momentum displacement unit 230 The momentum shift is described in more detail in step S1. Fig. 10 receives the momentum shift unit 230 the PWM wave signal generated by the PWM generator 220 is generated. In step S2, the momentum displacement unit decides. 230, whether two zones of predetermined stress vectors (see V1 and V2 on the side of the sloping flanks in the Fig. 2A to Fig. 2C) both extend over a predetermined time period (TPS) or longer. After making an affirmative decision, the operation proceeds to step S4, while after making a negative decision, the operation proceeds to step S3. In step S4, the momentum shift unit sends 230 a command indicating that no pulse shift should be performed, to the PWM generator 220 The operation in the sequence then ends. In step S3, the pulse shift unit sends 230 a command indicating that a pulse shift should be executed, to the PWM generator 220 The operation in the schedule then ends. It is noted that the operation in the schedule, which is in Fig. 10 is presented, which is executed in each PWM period.

[0033] The PWM generator 220 , when it has received the command that was transmitted by the momentum shift unit 230 The output, which indicates that no pulse shift should occur, sends the PWM waves that go to the pulse shift unit. 230 have been sent to the inverter circuit 130 On the other hand, when the command indicating that a pulse shift should occur is received, the PWM waves that go to the pulse shift unit are 230 have been sent, subjected to a pulse shift, and the PWM generator 220 sends the PWM waves resulting from the pulse shift to the inverter circuit 130 .

[0034] The inverter circuit 130 receives the PWM waves from the PWM generator 220and adjusts the output voltage by controlling the ON / OFF of the semiconductor switching elements based on the PWM waveform.

[0035] The source flow calculation unit 240 It includes an arithmetic operation unit (not shown) that calculates a current average for a single PWM period based on the instantaneous values ​​Id1 and Id2 of the DC bus current Idc, which are measured by the current detector. 120 are detected, the PWM pulse pattern generated by the PWM generator 220 is provided, and a momentum displacement amount that indicates the magnitude of the momentum displacement and is determined by the momentum displacement unit 230 is provided, calculated, and an input source current estimation unit (not shown) that provides an estimate by determining the current mean Ical as an input source current (source current) Isrc. The source current calculation unit 240sends information specifying the average current value Ical to the current control unit 110 It is noted that the arithmetic operation performed by the source stream calculation unit 240 will be explained in more detail later with reference to Fig. 4 and similar descriptions.

[0036] It is pointed out that when the rotational speed of the motor 300 in the engine system 500When the motor is controlled, the motor speed ωr is calculated based on the change in rotational position θ over time, and a voltage or current setpoint is generated so that it matches the motor speed ωr, with a speed setpoint output by a higher-order control unit. Additionally, when the motor power torque is controlled, the current setpoint (Id*, Iq*) is generated using a mathematical expression or mapping that specifies the relationship between the motor currents (Id, Iq) and the motor torque.

[0037] Fig. 2A specifies the relationship between the inverter output voltage vectors (PWM pattern), the motor currents (Iu, Iv, Iw) and the DC bus current Idc. Fig. 2B and Fig. 2C shows how electrical currents are applied to the inverter device. 100 or the engine 300with a stress vector V1 and a stress vector V2. Fig. Figure 3 shows the PWM and the pulse DC bus current corresponding to a single period (single PWM period) of the carrier frequency.

[0038] With regard to the Fig. 2A to Fig. 2C and Fig. Section 3 describes the operation performed to estimate the three-phase motor currents by detecting the DC bus current.

[0039] The open / closed states of the switching elements (switches) in the inverter circuit 130 will be in accordance with each of the in Fig. The voltage vectors shown in 2A (V0 to V7) are determined. The U, V and W in the PWM pattern column in Fig. 2A indicates the open / closed states of the U-phase switches U+ and U–, V-phase switches V+, V– and W-phase switches W+ and W– (see Fig. 1) A value of 0, entered in the PWM pattern column, indicates that the positive-pole switch is closed (OFF) and the negative-pole switch is open (ON). A value of 1, entered in the PWM pattern column, indicates that the positive-pole switch is open and the negative-pole switch is closed. For example, if 0 is entered in accordance with the U-phase in the PWM pattern column, the positive-pole switch U+ is closed, and the negative-pole switch U– is open.

[0040] Accordingly, the open / closed states of the switches corresponding to the voltage vector V1, i.e., the PWM pattern (1, 0, 0), are such that U+ is open, U– is closed, V+ is closed, V– is open, W+ is closed, and W– is open, as shown in Fig. 2B is indicated. Therefore, power is applied, as shown. Fig. 2B is specified in accordance with these switch open / closed states. Under these circumstances, a current equal to the current Iu flowing through switch U+ flows through the shunt resistor Rsh.

[0041] Similarly, the open / closed states of the switches corresponding to the voltage vector V2, i.e., the PWM pattern (1, 1, 0), are such that U+ is open, U– is closed, V+ is open, V– is closed, W+ is closed, and W– is open, as shown in Fig. 2C is indicated. Therefore, current is applied, as shown. Fig. 2C is specified in accordance with these switch open / closed states. Under these circumstances, a current equal to the current –Iw – flowing through switch W– flows through the shunt resistor Rsh.

[0042] As described above, in accordance with each PWM pattern, the three-phase motor current and the DC bus current achieve a specific ratio, and thus the three-phase motor current values ​​can be estimated by detecting the DC bus current.

[0043] Fig. Figure 3 shows the PWM and the pulse DC bus current for a single period (a single PWM period) of the carrier frequency. In the PWM generation timer operation, which is in Fig. As shown in 3, the PWM pulses shown in (b) are Fig. 3 are shown, generated with a timing that allows a sawtooth shaft or a triangular shaft, as shown in (a) of Fig. Figure 3 shows the voltage setpoint. In the figure, the U-phase PWM pulse rises at time T1, causing the voltage setpoint Vu1 and the time counter to match the sawtooth waveform, resulting in a voltage Vu as the U-phase inverter output. The U-phase PWM pulse then falls over time, causing the voltage setpoint Vu2 and a time counter to match the sawtooth waveform. PWM pulses are generated in much the same way for the V-phase and W-phase.

[0044] The DC bus current Idc in this situation is in (c) of Fig. 3. Motor currents corresponding to two phases (Iu and –Iw in the figure) can, as shown in Fig. 2B and Fig. The current specified as 2C can be determined by sampling the current twice during each PWM period (for V1 and V2 on the falling edge side of the figure). The motor current for the remaining phase can be determined by an arithmetic operation, since the three-phase motor currents have a relationship expressed as: Iu + Iv + Iw = 0.

[0045] It should be noted that, although in the description provided above “the current is sampled twice during a single period”, the two instances of current sampling do not occur at exactly the same time, as in (c) of Fig. 3 is specified. However, compared to the inverter device, the motor is a system with a sufficiently large time constant, and accordingly, the current conditions in the motor can be adjusted. 300They are viewed as showing hardly any changes, even if the switching conditions at the inverter device change. 100 In other words, the two instances of current sensing can be considered to occur simultaneously. Additionally, the change over time in the DC bus current Idc in each voltage vector is not significant and can be neglected when considering the degree of motor inductance, which is another factor that enables current sensing as described above.

[0046] A minimum pulse width (TPS) is required to ensure that the peak of the pulsed DC bus current (Idc) is reliably detected by this process. Additionally, to improve detection accuracy for narrower PWM pulses, the signal difference (line voltage pulse width) between PWM pulses corresponding to the two phases is calculated beforehand using the pulse shift unit. 230 is executed, determined, and the current is measured by the current detector. 120The current is sampled with an optimal trigger time (Trg). Factors determining the smallest pulse width (TPS) include the degree of main circuit inductance at the inverter, the throughput rate and response of the detection circuit, the sampling time in the A / D converter, and similar parameters. Preferably, the current should be sampled through vectors V0 and V7 to allow for the detection of any offset error in the current detection circuit and ultimately to enable offset correction for the current sensor. It should be noted that the “predetermined time period” in step S2 of Fig. 10 is synonymous with the smallest pulse width TPS.

[0047] Next, with reference to Fig. 4 the arithmetic operation performed by the source stream calculation unit 240 is carried out, as described. In (a) of Fig. 4 is a time counter value represented by a sawtooth waveform indicating a PWM pulse generation carrier period. In (b) of Fig. Figure 4 shows three-phase PWM pulses in a standard inverter over a single PWM period corresponding to an instantaneous voltage setpoint. In (c) of Fig. 4 is the DC bus current Idc', which (b) from Fig. 4 corresponds to, as stated. In (d) of Fig. Figure 4 shows PWM pulses that have undergone pulse shifting. In (e) of Fig. 4 is the DC bus current Idc, which (d) of Fig. 4 corresponds, as shown. It is pointed out that I_filt' in (c) of Fig. 4 and I_filt in (e) of Fig. 4 belong to the prior art, and reference will be made to them later when the present invention is compared with the prior art.

[0048] First, a procedure for calculating the DC bus current Idc' by arithmetic operation in (c) of Fig. 4 and the input source current Isrc together with the standard PWM pulses in (b) of Fig. 4 described. The differences between different phase pulse widths, between a U-phase PWM pulse width Upw', a V-phase PWM pulse width Vpw' and a W-phase PWM pulse width Wpw', are expressed as line voltage to the motor. 300 Applied, which causes motor currents to flow. The DC bus current (waveform) Idc is the current that flows from the smoothing capacitor to the inverter circuit in this situation. 130 flows.

[0049] A pulse width t1, representing the difference between the U-phase PWM pulse and the V-phase PWM pulse observed on the rising and falling edges of the PWM pulse, is calculated as: (U-phase PWM pulse width Upw' - V-phase PWM pulse width VPw') / 2.

[0050] It is noted that since t1 occurs twice, once on the rising edge and again on the falling edge, the difference is divided by 2. Additionally, the DC bus current value assumed in this zone (voltage vector V1) is equal to Id1'. The motor current Iu is assumed to be equal to Id1'.

[0051] Similarly, a pulse width t2, representing the difference between the V-phase PWM pulse and the W-phase PWM pulse observed on the rising and falling edges of the PWM pulse, is calculated as: (V-phase PWM pulse width Vpw' – W-phase PWM pulse width WPw') / 2.

[0052] It is noted that since t2 occurs twice, once on the rising edge and again on the falling edge, the difference is divided by 2. Additionally, the DC bus current value assumed in this zone (voltage vector V2) is equal to Id2'. The motor current –Iw is assumed to be equal to Id2'.

[0053] It should be noted that Id1' and Id2' can be sampled as described above, since the change occurring in the DC bus current Idc across the zones represented by the pulse widths t1 and t2 is not significant relative to the motor inductance and can be neglected.

[0054] Thus, if the PWM pulses are not subjected to pulse shifting, the average current Ical of the DC bus current Idc during a single PWM period can be determined by an arithmetic operation performed as expressed below in (A): Ical = {(Id1' × t1 + Id2' × t2) × 2} / Tpwm (A)

[0055] This average current Ical is then referred to as an estimated input source current Iscr.

[0056] With reference to (d) and (e) of Fig. Section 4 describes the waveform of the DC bus current that results when a pulse shift is applied.

[0057] If the instantaneous values ​​Id1 and Id2 of the DC bus current Idc are detected via an A / D converter such as a microcomputer, the smallest pulse width TPS derived from the pulse widths t1 and t2 must be greater than the sampling time Tad. Accordingly, the detection of the DC bus current Idc is achieved by shifting the phase of the waveform of a PWM pulse in (b) of Fig. 4 relative to the phase of the waveform of another PWM pulse (pulse shift) allows to ensure the smallest pulse width TPS.

[0058] For example, the instantaneous value Id1 of the DC bus current Idc on the falling edge side of the PWM pulses can be sampled by delaying the phase of the U-phase pulse by an amount equal to a pulse shift magnitude Tt1 with respect to the V-phase pulse, and increasing the pulse width (the time difference between the falling edge of the U-phase pulse and the falling edge of the V-phase pulse, i.e., the time over which the output is continuously provided with the voltage vector V1 on the falling edge side) to obtain a pulse width TPS1 equal to or greater than the smallest pulse width TPS without exciting the voltage vector region V1.

[0059] However, on the rising edge side of the PWM pulse, the relationship between the U-phase pulse and the V-phase pulse changes, causing a switching to the voltage vector V4 and resulting in a pulse current (with an instantaneous value –Iu, i.e., the U-phase current value with an added opposite sign) flowing in accordance with the instantaneous value Id2'' of the DC bus current Idc.

[0060] The voltage vector V1, which extends on the side of the falling edge of the PWM pulse, is canceled out by the voltage vector V4 on the side of the rising edge of the PWM pulse to balance the pre-pulse shift voltage average for a single PWM period.

[0061] Similarly, the phase of the W-phase pulse is advanced by an amount equal to a pulse shift Tt2 relative to the V-phase pulse, and the pulse width (the time difference between the falling edge of the W-phase pulse and the falling edge of the V-phase pulse) is thereby increased to achieve a pulse width TPS2 equal to or greater than the smallest pulse width TPS without exciting the vector voltage region V2. On the rising edge side of the PWM pulses, the voltage vector is switched to V5, causing a pulse current (with an instantaneous value Iw for the W-phase current) to flow in accordance with an instantaneous value Id1'' of the DC bus current Idc.

[0062] The operation described above allows the motor to be controlled by the voltage applied to the motor and its phases, set by generating a sufficient sampling time on the falling edge side of the PWM pulse, without changing the average value of the voltage applied to the motor over a single PWM period.

[0063] It is noted that fixed current control is implemented for the motor currents corresponding to the three phases with PWM pulse widths Upw, Vpw, and Wpw, whereby the motor current values ​​are controlled so that they remain essentially the same regardless of whether the PWM pulses are subjected to pulse shifting or not. For example, the following relationship is obtained with respect to the U-phase PWM pulse and the V-phase PWM pulse: Instantaneous value Id1' of the DC bus current Idc when no pulse shift is applied ≈ Instantaneous value Id1 of the DC bus current Idc when a pulse shift is applied.

[0064] Next, with reference to (d) and (e) of Fig. Section 4 describes a method for determining the average current Ical for a single PWM period in connection with the PWM pulses that have undergone pulse shifting. It is noted that the average current Ical over a single PWM period, calculated in connection with the PWM pulses that have undergone pulse shifting, is obviously referred to as an estimated input source current Isrc corresponding to the PWM pulses that have undergone pulse shifting.

[0065] The pulse width TPS1 in (d) of Fig. 4, which represents the difference between the applied U-phase voltage and the applied V-phase voltage on the side of the falling edge of the PWM pulse, is calculated as expressed below: TPS = (P-phase pulse edge time – V-phase pulse edge time)

[0066] It should be noted that, although it is desirable for TPS1 to assume a value greater than the value calculated as above, to an extent that corresponds to the inverter dead time Td, the dead time Td is not taken into account here.

[0067] The current for the P-phase PWM pulse and the V-phase PWM pulse on the falling side is calculated as TPS1 × Id1. The pulse width TPS1 is equal to the sum (pulse width t1 + pulse shift magnitude Tt1), and accordingly: (t1 + Tt1) × Id1 = t1 × Id1' + Tt1 × Id1.

[0068] The PWM pulses Upw, Vpw, and Wpw are achieved by a pulse shift of an amount equal to the pulse shift magnitude Tt1, while the three-phase PWM pulse widths Upw', Vpw', and Wpw' remain unchanged. Accordingly, the pulse width corresponding to the instantaneous value Id2'' is equal to t1 – Tt1 (t1 < Tt1 in (d) of Fig. 4), and Id2'' ≈ –Id1'.

[0069] As a result, the current for the U-phase PWM pulse and the V-phase PWM pulse on the rising side is calculated as: (t1 – Tt1) × Id2'' = t1 × Id1' – Tt1 × Id1.

[0070] Thus, the sum of the current values ​​for the U-phase and V-phase PWM pulses on the rising and falling sides can be calculated as; (t1 – Tt1) × Id2'' + (t1 + Tt1) × Id1 = 2 × t1 × Id1' = 2 × t1 × Id1.

[0071] Since t1 = (TPS1 – –Tt1), the average current Ical1 over a single PWM period can be calculated as follows based on the relationship between the U-phase and the V-phase: Ical1 = 2 × (TPS1 – Tt1) × Id1 / Tpwm (B)

[0072] A similar principle applies to the W-phase and the V-phase. The current for the W-phase and V-phase PWM pulses on the falling side is calculated as TPS2 × Id2, the pulse width TPS2 is equal to the sum (pulse width t2 + pulse shift magnitude Tt2), and Id2' ≈ Id2. Accordingly, the following applies: (t2 + Tt2) × Id2 = t2 × Id2' + Tt2 × ΔId2.

[0073] The PWM pulses Upw, Vpw, and Wpw are achieved by a pulse shift of an amount equal to the pulse shift magnitude Tt2, while the three-phase PWM pulse widths Upw', Vpw', and Wpw' remain unchanged. Accordingly, the pulse width corresponding to the instantaneous value Id2'' is t2 – Tt2 (t2 < Tt2 in (d) of Fig. 4), and Id1'' ≈ –Id2'.

[0074] As a result, the current for the W-phase PWM pulse and the V-phase PWM pulse on the rising side is calculated as; (t2 – Tt2) × Id1'' = t2 × Id2' – Tt2 × Id2.

[0075] Thus, the sum of the current values ​​for the V-phase and W-phase PWM pulses on the rising and falling sides can be calculated as; (t2 – Tt2) × Id1'' + (t2 + Tt2) × Id2 = 2 × t2 × Id2' = 2 × t2 × Id2.

[0076] Since t2 = (TPS2 – Tt2) holds true, the average current Ical2 over a single PWM period can be calculated as below based on the relationship between the W-phase and the V-phase. Ical2 = 2 × (TPS2 – Tt2) × Id2 / Tpwm (C)

[0077] Accordingly, the average current Ical over a single PWM period, which needs to be determined, can be calculated based on expressions (B) and (C) as: Ical1 = Ical1 + Ical2 = 2 × {(TPS1 – Tt1) × Id1 + (TPS2 – Tt2) × Id2} / Tpwm (D)

[0078] The calculated value Ical is then referred to as an estimated input source current Iscr corresponding to the PWM pulses that have undergone pulse shifting. It should be noted that, since TPS1 – Tt1 = t1, TPS2 – Tt2 = t2, Id1 ≈ Id1' and Id2 ≈ Id2', expression (A) and expression (C) are essentially equivalent. Specifically, the average current Ical calculated for a single PWM period in accordance with the PWM pulses without pulse shifting is essentially the same as the average current Ical calculated for a single PWM period in accordance with the PWM pulses that have undergone pulse shifting.

[0079] Assuming that the inverter's dead time is not significant and can be neglected, Isrc can be calculated as: Isrc = ((Upw – Vpw) × Id1 + (Vpw – Wpw) × Id2) / Tpwm

[0080] As described above, the present invention makes it possible to calculate the current values ​​for a single PWM pulse period with a high degree of accuracy and thus achieves an improvement in the accuracy with which the average current Ical for a single PWM period is calculated as an estimated input source current, even if the PWM pulses are subjected to pulse shifting.

[0081] While the engine 300 If the system is driven, the arithmetic operation should be performed by including the relationship between the individual PWM pulse stages to accurately estimate the input source current Isrc, unaffected by the frequency characteristics, which can be attributed to analog filtering.

[0082] It is pointed out that the motor current Iu and the motor current –Iw can be accurately detected, regardless of whether a pulse shift is applied or not, or regardless of the pulse shift time width, by sampling the instantaneous values ​​Id1 and Id2 of the DC bus current at a time set relative to the falling edge of the V-phase pulse, which is not shifted even in the case of a pulse shift.

[0083] The measures described above can improve the source flow calculation unit. 240 Calculate the average current Ical with a high level of accuracy and thus estimate a highly accurate input source current Iscr, regardless of whether pulse shifting is applied or not, i.e., over the entire operating range of the motor. 300 Furthermore, the source flow calculation unit can 240 via the inverter circuit 130in accordance with a battery charge state and the estimated battery lifespan, to ensure that the estimated input source current Isrc does not exceed the limit (input source current setpoint) for the input source current Isrc.

[0084] Fig. Figure 5 presents the results obtained by comparing the mean current Ical (estimated input source current Isrc), calculated in relation to the DC bus current Idc using the arithmetic operation logic according to the present invention, with the input source current Isrc determined by actual measurement. The results, which indicate the current detection accuracy, with the maximum rated current set to 100%, show that the input source current Isrc is accurately estimated over the range from zero current up to and including the maximum current.

[0085] The following advantages are achieved by the first embodiment. (1) The inverter device 100 , which is achieved in the first embodiment, is configured as follows.

[0086] It includes the positive terminal connection. 80 , the negative terminal 81 , the inverter circuit 130 , the smoothing capacitor 160 , the positive-pole-side connection point 90 , which connects to the positive terminal 80 , the positive terminal side of the inverter circuit 130 and the smoothing capacitor 160 electrically connects the negative pole-side connection point 91 , which connects to the negative terminal 81 , the negative terminal side of the inverter circuit 130 and the smoothing capacitor 160 electrically connects the PWM generator 220, which generates PWM waves corresponding to the three phases to be used to control the inverter circuit 130 to control the impulse shift unit 230 , which is described below, the current detector 120 , which carries the DC bus current Idc, which is between the negative terminal connection point 91 and the inverter circuit 130 flows, is detected, and the source flow calculation unit 240 , which is described below.

[0087] Based on the time intervals over which the output conforms to two predetermined voltage vectors (the voltage vectors V1 and V2 on the falling edge side, see Fig. 4) under different voltage vectors, which are generated based on the temporal position relationship (PWM pulse pattern) under the different phases of the PWM waves, the pulse displacement unit is continuously provided. 230a first command, in response to which the PWM generator 220 performs a pulse shift if at least one of the consecutive output time intervals corresponding to the two predetermined voltage vectors is smaller than the respective time interval TPS1 or TPS2, and generates a second instruction indicating that the PWM generator 220 no pulse shift should be performed if the contiguous output time spans corresponding to the two predetermined voltage vectors are both equal to or greater than the respective predetermined time spans TPS1 and TPS2.

[0088] The source flow calculation unit 240It includes the arithmetic operation unit (not shown), which calculates the average current Ical of the DC bus current Idc over a PWM period based on the instantaneous values ​​Id1 and Id2 of the DC bus current Idc, detected at predetermined times occurring during continuous operating intervals corresponding to the voltage vectors V1 and V2 within the PWM period, and the input source current estimation unit (not shown), which estimates an input source current Isrc by determining the average current Ical of the DC bus current Idc over the PWM period as the input source current Isrc between the positive terminal 80 and the positive pole-side connection point 90 or between the negative terminal 81 and the negative pole-side connection point 91 during the PWM period, it is estimated.

[0089] The inverter device 100, which is achieved in the first embodiment configured as described above, achieves the following advantages.

[0090] Even when a pulse shift is applied, the input source current Isrc can be accurately estimated, and the response can be improved, as described later. Additionally, apart from the shunt resistor Rsh, across which the DC bus current Idc is applied to the inverter circuit... 130 The inverter circuit contains the detected signal. 100 No additional shunt resistors are required. This means that, compared to an invention that includes shunt resistors used to detect the motor currents Iu, Iv and Iw and the input source current Isrc, the present invention requires fewer components, achieves a reduction in installation space, and ultimately reduces manufacturing costs.

[0091] The present invention is compared with a comparative example in which the input source current is estimated based on a current value obtained by filtering a detection value for the DC bus current Idc. In (c) and (e) of Fig. Figure 4 shows a filtered current value I_filt, which is to be compared with the present invention. The ripple in the filtered current value I_filt detected in the comparison example increases due to a pulse shift applied to the PWM pulses. This means that the ripple in the filtered current value I_filt is affected by the motor voltage application state, i.e., whether a pulse shift is applied or not. Consequently, the DC bus current Idc cannot be sampled with good detection accuracy over the entire motor operating range, making it difficult to accurately estimate the input source current Isrc. Furthermore, if stability is to be achieved by increasing the filter time constant, a compromise must be made regarding the response of the DC bus current Idc, and thus the required characteristics are not guaranteed.

[0092] In the inverter device 100According to the present invention, the average current Ical for a single PWM period is determined by an arithmetic operation performed based on the PWM pulse widths, the pulse shift magnitude, and the DC bus current Idc, and Ical is then determined as an estimated input source current Isrc. As described below, the average current Ical for a single PWM period calculated when no pulse shift is applied and the average current Ical for a single PWM period calculated when a pulse shift is applied are essentially the same, and thus the input source current Isrc can be estimated with high accuracy when a pulse shift is applied, comparable to the accuracy with which the input source current Isrc is estimated when no pulse shift is applied.As a result, the input source current Isrc can be estimated with better accuracy compared to the reference example. Additionally, since the present invention does not require filter processing, the problem of having to find a compromise in the response due to an increase in the filter time constant does not arise. In other words, the present invention ensures a better response than the reference example. (2) The source flow calculation unit 240calculates the mean value Ical (see expression (D)) for the DC bus current flowing over one PWM period when a pulse shift is applied, based on the instantaneous values ​​detected for the DC bus current, the contiguous output time spans corresponding to the two predetermined voltage vectors during which the instantaneous values ​​are detected, and the pulse shift magnitude, which determines the extent of the pulse shift at the PWM generator. 220 indicates.

[0093] The average current Ical calculated for the DC bus current when a pulse shift is applied by the process described above is essentially the same as the average current Ical (see expression (A)) for the DC bus current calculated when no pulse shift is applied. Thus, the input source current Isrc can be estimated with the pulse shift having little effect on the result. (3) The power control unit 110 gives a command to the PWM generator 220 based on the estimated input source current Isrc. In response to the command, the PWM generator sends out 220 PWM waves to the inverter circuit 130 The inverter circuit 130 controls the actual input source current using the PWM waves.

[0094] As described above in (1) and (2), the input source current Isrc is estimated with high accuracy, which in turn allows the inverter circuit 130 The actual input source current Isrc is precisely controlled. As a result, the motor can also be controlled. 300 can be precisely controlled. (4) In addition, the source flow calculation unit calculates 240 the input source current Isrc by detecting the current values ​​that correspond to specific PWM pulse patterns (see Fig. 4).

[0095] Thus, even in a fault operating condition (phase failure operating condition) where energy cannot be supplied in accordance with one of the three phases due to a break in the motor wiring or an inverter fault, the current detection value corresponding to the failed (lost) phase in the power supply pattern can be correctly calculated as 0. Consequently, the input source current Isrc can be accurately calculated, even in the case of a phase failure fault. – Second embodiment –

[0096] Next, an arithmetic operation performed to calculate the input source current in the second embodiment of the present invention will be described, mainly with reference to Fig. 6 and Fig. 9 described.

[0097] Fig. Figure 6 shows a portion of the U-phase motor current. How Fig. As stated in section 6, a motor current value IU' before correction contains a current error Ips which can be attributed to a pulse shift, while a motor current value IU is a corrected current value obtained by correcting the current error Ips which can be attributed to the pulse shift.

[0098] Fig. Figure 9 specifies a PWM pulse shift applied at a PWM duty cycle of 50% and the corresponding DC bus current. If no pulse shift is applied and the PWM duty cycle corresponding to the three phases is all 50% (not shown), no DC bus current Idc (zero current) flows because the pulse widths for the three phases are equal, and the rising and falling edges corresponding to the three phases coincide. Under these conditions, current detection cannot be performed in conjunction with the single side-current detection logic, which detects a current with a time course that generates an interphase potential among the three phases (e.g., a pulse width TPS1' representing the difference between the U-phase and the V-phase, or a pulse width TPS2' representing the difference between the W-phase and the V-phase).

[0099] Accordingly, the detection of zero current is made possible by a pulse shift. As in (c) of Fig. As specified in Figure 9, although the mean value of the DC bus current Idc within a single PWM period is essentially zero (the mean value of the motor currents over a single PWM period is also zero), current errors Ips1 and Ips2 are detected as instantaneous values ​​Id3'' and Id4'' of the DC bus current Idc. In particular, the effect of the current errors Ips1 and Ips2 must be significant over a range where the motor currents are close to zero, and thus these current errors Ips1 and Ips2 should be pre-detected by calibration, for example, at the time of power-up, in order to detect the motor currents with a high degree of accuracy. In the present embodiment, the input source current Isrc can be estimated by arithmetic operations based on the relationship between the vectors of the applied voltage, the motor currents, and the DC bus current, which are specified in the Fig. 2A to Fig. The execution is carried out as specified in 2C.

[0100] For example, the current value for the voltage vector zone V1 (see (d) and (e) of Fig. 4) Based on the difference in PWM pulse width (Upw – Vpw) between the U-phase and the V-phase, the current value for zone V1 can be determined as: IU·(Upw – Vpw). The current value for another voltage vector zone can be calculated in much the same way. By calculating the sum of these current values ​​and dividing the sum by the individual PWM period Tpwm to determine a time average, a current average Ical is calculated, and thus an estimated input source current Isrc is obtained.

[0101] Since the current errors Ips1 and Ips2, which can be attributed to pulse shifting, are eliminated from the values ​​used to calculate the motor currents (IU, IV, IW) by the correction, it is not necessary to include the current errors Ips1 and Ips2 when calculating the input source current Isrc. According to the present invention, since the current errors Ips1 and Ips2, which can be attributed to pulse shifting, are corrected beforehand, the advantage is gained of reducing the load on the microcomputer arithmetic processing performed to calculate the input source current Isrc.

[0102] It is pointed out that, although an explanation has been given with reference to the present embodiment under the assumption that the PWM duty cycle (pulse width) is 50%, the current errors Ips1 and Ips2 can be obtained by pulse shifting applied with duty cycles corresponding to the individual phases which are set to be equal to each other.

[0103] The following advantage is achieved by the second embodiment.

[0104] The source flow calculation unit 240The system estimates the motor current IU based on an instantaneous value detected for the DC bus current Idc and the voltage vector corresponding to which the instantaneous value is detected. It corrects the motor current IU by eliminating a current error Ips (Ips1), which can be attributed to a pulse shift applied in the voltage vector zone at the time of instantaneous value detection. It then calculates a current value for the VI zone based on the corrected motor current IU and the pulse widths Upw and Vpw, corresponding to the different phases of the PWM waves. A current value is similarly determined in accordance with another voltage vector zone. The sum of the current values ​​is calculated, and the sum is then divided by the PWM period Tpwm to calculate the average Ical value for the DC bus current Idc over the PWM period in the case of a pulse shift.

[0105] A current fault Ips, contained in a motor current and attributable to a pulse shift, is represented by an instantaneous value of the DC bus current Idc, which is measured by the current detector. 120 It is detected when a pulse shift is applied, via the PWM generator. 220 , by adjusting the PWM duty cycles that correspond to the individual phases, making them equal to each other.

[0106] These measures eliminate the current errors Ips in the motor currents, and the input source current Isrc can be estimated with a higher degree of accuracy, which is achieved by eliminating the errors. – Inverter circuit installation example 1 –

[0107] Fig. Figure 7 is a diagram showing the configuration of an electric power steering system ( 600 ) shows that the inverter device 100in accordance with the present invention. With reference to Fig. 7 will have an electric power steering system ( 600 ) described.

[0108] The electric power steering system ( 600 ) includes an electric actuator 610 , a steering wheel 900 , a steering detector 901 and an actuation command generator 903 , and assumes a structure in which the actuating force on the steering wheel 900 , which is operated by the driver, provides torque support via the electric actuator.

[0109] The electric actuator 610 includes a motor system 500 , which is a torque transmission mechanism 902 , an engine 300 the inverter device 100 , which is installed therein, exhibits, as in Fig. 7 is shown.

[0110] A torque command τ*, generated by the actuation command generator 903 is generated as a steering assistance torque command for the steering wheel 900 , results in an instruction to reduce the required steering force to be exerted by the driver by using the output from the electric actuator 610 .

[0111] In response to the torque command τ* entered into it, the inverter device controls 100 the motor currents based on the motor's torque constant 300 and the torque command τ*, to match a torque setpoint.

[0112] The motor power τm, which is generated by an output shaft that is directly connected to the rotor of the motor. 300 The connection is made available and is transmitted as torque via the torque transmission mechanism. 902, which is configured, for example, with a speed reduction mechanism such as a worm gear, a gear and a planetary gear, or a hydraulic mechanism, onto a rack 910 of the steering system, and the steering angles of the wheels 920 and 921 are adjusted while the steering force (actuating force) that the driver applies to the steering wheel 900 The force required to exert this force is reduced (supported) by the electrical force.

[0113] A support parameter, representing the level of support to be provided, is generated by the Actuation Parameter Command Generator. 903 by detecting, via the steering detector 901, which is built into the control shaft to detect the steering states, an actuation quantity represented by a steering angle or steering torque, and then taking into account the state variables such as vehicle speed and road surface conditions, and the support quantity thus determined is specified in the torque command τ*.

[0114] As explained previously, the inverter device calculates 100The input source current Isrc is determined by detecting the current values ​​that correspond to specific PWM pulse patterns. Thus, even in a fault operating condition (phase failure operating condition) where energy cannot be supplied in accordance with one of the three phases due to a break in the motor wiring or an inverter fault, the current detection value corresponding to the failed (lost) phase in the power supply pattern can be correctly calculated as 0. Consequently, the input source current Isrc can be accurately calculated even in the case of a phase failure fault.Thus, even if the electric power steering system is continuously operating in a phase failure fault condition during an emergency situation to maneuver the vehicle up onto a curb while maintaining the current steering angle, excessive consumption of battery power for the purpose of generating high engine power can be prevented.

[0115] Furthermore, since the inverter device 100Since the actual input source current can be controlled based on the estimated input source current Isrc, as previously explained, smooth steering assistance can be provided even when the steering wheel is turned back and forth to repeatedly switch from a low-load operation to a high-load operation, in response to an input source current that results in an instruction to keep the assistance magnitude low when there is a risk of low battery voltage, e.g., when the vehicle speed is low and the steering magnitude is significant in the low battery state. – Inverter circuit installation example 2 –

[0116] Fig. Figure 8 is a system block diagram showing the configuration of a vehicle braking system.

[0117] A support control unit 706 in Fig. 8, which has a function similar to that of the inverter device 100The feature is programmed into its microcomputer so that it can perform braking operations for a vehicle. Additionally, the motor differs. 731 from the engine 300 by being an integrated component of a brake support device 700 is mounted. Through the covering 712 is an integrated structure through the engine 731 and the support control unit 706 configured.

[0118] The vehicle's braking system includes a brake pedal. 701 , a brake support device 700 , an amplifier unit 800 and wheel mechanisms 850a until 850d The brake assist device 700 includes a support mechanism 720 , a primary fluid chamber 721a , a secondary fluid chamber 721b and a reservoir container 714 The actuation force on the brake pedal 701, which is activated by the driver, is integrated into the support mechanism 720 via an input bar 722 entered and then goes to the primary fluid chamber 721 transmitted.

[0119] Additionally, the brake actuation size, which is determined by a stroke sensor, is 702 is detected, which is located at the brake pedal 701 is mounted in the support control unit 706 , which the support mechanism 720 controls, entered. The support control unit 706 controls the engine 731 to achieve a rotational position corresponding to the brake actuation value input. The rotational torque generated at the motor is reduced via a speed reduction unit. 723 to a ball winch drive 725 transferred, which configures a rotation-translation conversion device that converts rotational force into translational force, to a primary piston726 presses to increase the hydraulic pressure in the primary fluid chamber 721a to increase, and also a secondary piston 727 pressurizes to increase the hydraulic pressure in the secondary fluid chamber 721b to increase.

[0120] The hydraulic pressure of the hydraulic fluid that is in the primary fluid chamber 721a and the secondary fluid chamber 721b Once pressurized, it is fed into the amplifier unit. 800 via the main lines 750a and 750b entered, and in response to a command issued by an amplifier control unit 803 The amplifier unit transmits the output. 800 the hydraulic pressure on the wheel mechanisms 850a until 850d , in order to generate a braking force for the vehicle.

[0121] The support control unit 706 controls the extent of the primary piston's displacement 726, to adjust the degree of pressure applied to the primary piston. Instead of the displacement of the primary piston. 726 To detect directly, the displacement of the primary piston will be detected. 726 through arithmetic operation in accordance with the extent to which the ball wind drive 725 is driven, calculated by calculating the rotation angle of the motor. 731 based on a signal provided by the rotary position sensor (not shown) installed in the motor.

[0122] It is pointed out that, even if the engine 731 due to an error, it shuts down and the reset control for the ball winch drive 725 When deactivated, the braking action performed by the driver is never hindered when the ball winch drive is engaged. 725It is reset to its initial position, with the reaction force being transmitted by a return spring. This ensures that the vehicle's behavior is never destabilized, for example, by brake dragging.

[0123] An amplifier mechanism 801 contains two hydraulic pressure adjustment mechanisms 810a and 810b Each of these mechanisms is installed to adjust the hydraulic fluid pressure for two of the four wheels, positioned diagonally to each other. Thus, even if one of the hydraulic pressure adjustment mechanisms fails, the vehicle can be brought to a stable stop, and the braking forces at the wheel mechanisms can be applied. 850a and 850b The two wheels, which are diagonal to each other, can be individually adjusted. This is due to the two hydraulic pressure adjustment mechanisms. 810a and 810bSince they are used in a similar way in operation, the following description focuses on the operation of one of the mechanisms, i.e., the hydraulic pressure adjustment mechanism. 810a The hydraulic pressure adjustment mechanism 810a includes an OFF shut-off valve 811 , via which hydraulic fluid is supplied to the wheel cylinders 851 is controlled, and an ON shut-off valve, via which the fluid supply to pumps is controlled, ON valves 814a and 814b , via which the hydraulic fluid supply to the individual wheel cylinders 851 from the main line 750a or the pumps are controlled, OFF valves 813a and 813b , about the pressure reduction control for wheel cylinders 851 This is achieved by pumps 853 , which increase the main pressure, which is combined with the hydraulic pressure from the main line 750a is generated, and pump motors 852 , which the pumps 853drive. For example, when hydraulic pressure control is implemented for the purpose of anti-lock braking control, signals from the wheel rotation sensors within the wheel mechanisms are used. 850 be provided in the amplifier control unit 830 After the system detects that a wheel has locked during braking, the corresponding ON / OFF valves (electromagnetic valves) and the pump are activated to adjust the hydraulic pressure to a level that releases the wheel lock. It should be noted that this mechanism can also be used in hydraulic pressure control systems for vehicle stability control.

[0124] In the vehicle braking system, which is structured as described above, the motor device is used to provide stable support at all times and is also used to control the displacement of the primary piston. 726 to control. For this reason, it must also ensure the accurate detection of any anomaly in order to be able to operate continuously and stably while maintaining a high degree of accuracy. Additionally, if the energy stored in the battery used as the power source, 200 As the battery charge decreases, the amount of support that can be provided must decrease. Under such circumstances, brake assist operation is achieved by using an auxiliary power source. 400 as an energy source in the system, continuously executed.

[0125] In an electrically assisted braking system, the operation must be particularly careful, even if, for example, the wiring is connected to the battery. 200 This is solved by switching to the auxiliary power source. 400 It must be operated without interruption for emergency reserve operation. The auxiliary power source 400 It is often designed in such a way that only a relatively small current is available in an emergency, and therefore the motor device must be operated in a power-saving mode with a lower motor output in order to regulate the current accordingly during emergency reserve operation. Under such circumstances, the small current that comes from the auxiliary power source can 400This is provided when the input source current Isrc is accurately detected, and as a result, a braking system can be provided that, by controlling the maximum current value, can generate a braking support force to maintain an essentially constant value close to the limit. Specifically, the support control unit can 706 According to the present invention, the maximum amount of brake support force can be provided even in the event of a battery failure.

[0126] It should be noted that the present invention allows the following variations.

[0127] While the DC bus current Idc is measured by sampling the current on the rising PWM edge side and the falling PWM edge side a total of four times within a single PWM period in (b) and (c) of Fig. 4. If a motor current can be detected, it can be estimated as long as the DC bus current Idc can be determined in accordance with two different voltage vectors, as explained earlier.

[0128] If the carrier period Tpwm (PWM period Tpwm) in (b) and (c) of Fig.4. If the pulse width is sufficiently short compared to the inverter output frequency, an instantaneous value of the DC bus current detected on the rising edge side of the PWM pulse and an instantaneous value of the DC bus current detected on the falling edge side of the PWM pulse corresponding to the same voltage vector are essentially the same. For example, an instantaneous value detected on the rising edge corresponding to voltage vector V1 is the same as an instantaneous value detected on the falling edge corresponding to voltage vector V1. Therefore, the motor current can be estimated based on either two instantaneous values ​​(corresponding to V1 and V2) of the DC bus current on the rising edge side of the PWM pulse or based on two instantaneous values ​​(corresponding to V1 and V2) of the DC bus current on the falling edge side of the PWM pulse.

[0129] Although the current that flows between the negative terminal connection point 91 and the inverter circuit 130 When the DC bus current Idc is detected as described above, the current flowing between the positive-pole-side connection point can instead be measured. 90 and the inverter circuit 130 flows when the direct current bus current Idc is detected.

[0130] Although the input source current Isrc, which is between the battery 200 and the positive pole-side connection point 90 instead of the current flowing as described above, the current between the negative-pole-side connection point can be estimated. 91 and the battery 200 flows, be appreciated.

[0131] Although instantaneous values ​​are sampled over the contiguous output time spans (sampling periods) corresponding to two voltage vectors on the falling edge side as described above, instantaneous values ​​can instead be sampled over the contiguous output time spans corresponding to two voltage vectors on the rising edge side.

[0132] Although a pulse shift is applied to lengthen the contiguous output time spans (sampling periods) corresponding to two voltage vectors on the falling edge side, as described above, a pulse shift can instead be applied to lengthen the contiguous output time spans corresponding to two voltage vectors on the rising edge side.

[0133] As long as the features characterizing the present invention are not impaired, the present invention is in no way limited to the specific features of the embodiments described above, and other modes of operation or combinations that are conceivable within the technical scope of protection of the present invention are also within the scope of protection of the invention. QUOTES INCLUDED IN THE DESCRIPTION

[0134] This list of documents cited by the applicant was automatically generated and is included solely for the reader's convenience. The list is not part of the German patent or utility model application. The DPMA accepts no liability for any errors or omissions. Cited patent literature

[0135] JP 2014-086095

[0001] JP 5069882

[0004] JP 3843391 [0005, 0008] JP 11-004594

[0007]

Claims

[1] Inverter device comprising: a positive terminal connection ( 80 ), which is connected to a positive terminal of a power source; a negative terminal connection ( 81 ), which is connected to a negative terminal of the power source; a three-phase inverter circuit connected to a motor ( 300 ) is connected to power the engine ( 300 ) to drive; a smoothing capacitor ( 160 ); a positive-pole connection point that connects the positive-pole terminal ( 80 ), the inverter circuit ( 130 ) and the smoothing capacitor ( 160 ) connects; a negative-pole connection point that connects the negative-pole terminal ( 81 ), the inverter circuit ( 130 ) and the smoothing capacitor ( 160 ) connects; a PWM generator ( 220), which generates three-phase PWM waves that are used to control the inverter circuit ( 130 ) to target; a momentum displacement unit ( 230 ), which, based on the PWM waves, generates an initial command in response to which the PWM generator ( 220 ) performs a pulse shift, or generates a second command in response to which the PWM generator does not perform a pulse shift; a current detector ( 120 ), which provides a DC bus current between the positive terminal connection point and the inverter circuit ( 130 ) or between the negative terminal connection point and the inverter circuit ( 130 ) flows when the inverter circuit ( 130 ) is targeted, detected; an arithmetic operation unit that calculates an average value of the DC bus current during a PWM period by means of an arithmetic operation performed based on instantaneous values ​​of the DC bus current detected at predetermined times occurring during contiguous output intervals corresponding to predetermined voltage vectors within the PWM period; and an input source current estimation unit that provides an estimated value for an input source current passing between the positive terminal ( 80 ) and the positive terminal connection point or between the negative terminal connection ( 81 ) and the negative pole-side connection point during a PWM period, by determining the average DC bus current flowing over the PWM period as the estimated value. [2] Inverter device according to claim 1, wherein, based on the instantaneous values, the contiguous output intervals corresponding to the predetermined voltage vectors during which the instantaneous values ​​are detected, and a pulse shift amount representing a range of pulse shift applied via the PWM generator, the arithmetic operation unit calculates the mean value of the DC bus current during the PWM period when pulse shift is applied. [3] Inverter device according to claim 1 or 2, wherein: The arithmetic operation unit includes a motor current estimation unit that estimates a motor current based on instantaneous values ​​of the DC bus current and the predetermined voltage vectors in accordance with which the instantaneous values ​​are detected; The arithmetic operation unit eliminates current errors resulting from pulse shifts corresponding to predetermined voltage vectors from the motor current in order to correct the motor current; and When pulse shift is applied, the arithmetic operation unit calculates the mean value of the DC bus current during the PWM period based on the corrected motor current and pulse widths of the PWM waves corresponding to the individual phases. [4] Inverter device according to claim 3, wherein the current errors attributable to pulse shifting and contained in the motor current are represented by instantaneous values ​​of the DC bus current detected by the current detector when pulse shifting is performed by the PWM generator with PWM duty cycles corresponding to the individual phases balanced with each other. [5] Inverter device according to any one of claims 1 to 4, wherein the inverter circuit is controlled based on the estimated input source current.

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