UPWARD WALKER DEVICE

Abstract

Boost converter device (10) comprising: - a boost converter (20) comprising an upper switching element (Q1), a lower switching element (Q2) and an inductor (L1) connected to a node of the upper and lower switching elements (Q1, Q2) and capable of amplifying and outputting a voltage input from a battery (B); - a current sensor (21) that detects an inductor current flowing through the inductor (L1);and- a control unit (50) that controls the boost converter (20) by utilizing a control of the inductor current, wherein- the control unit (50):- reduces a carrier frequency used in controlling the boost converter (20) and / or reduces a duty cycle command value used in controlling the boost converter (20),- detects the amplitude of a current ripple via the current sensor (21) when the reduction of the carrier frequency or the reduction of the duty cycle command value occurs, and- detects the current sensor (21) as abnormal when the amplitude of the current ripple is below a predetermined current fluctuation range that is stored in advance, and- reduces a duty cycle by decreasing an ON period of the upper switching element (Q1) and increasing an ON period of the lower switching element (Q2).;

Description

[0001] This application claims priority over Japanese patent application No. 2014-97799, filed on May 9, 2014, the disclosure of which is hereby incorporated by full reference.

[0002] The present invention relates to a boost converter device for amplifying and outputting a voltage provided by a battery.

[0003] A boost converter control device is known, for example, from JP 2006-311635 A. In this control device, a control signal for switching a switching element contained in a boost converter is generated by controlling a gain voltage, which is an output voltage of the boost converter, and by controlling an inductor current flowing through an inductor contained in the boost converter.

[0004] In the boost converter control device described in JP 2006-311635 A, the inductor current is detected by a current sensor and fed to the control device for use in boost converter control. Therefore, if an abnormality occurs in the current sensor that causes a state where the output value fluctuates only within a narrow current bandwidth (a state where it is fixed at a substantially constant value), it is not possible to control the inductor current appropriately. This also affects the gain voltage control, creating a situation where it is no longer possible to approximate the gain voltage to a gain voltage command value. Consequently, there is a need for a technology capable of appropriately detecting an abnormality in the current sensor detecting the inductor current.

[0005] The object of the present invention is to detect an abnormality of a current sensor detecting a choke coil current in a boost converter device that controls a gain voltage by utilizing a control of the choke coil current in a suitable manner.

[0006] According to one aspect of the present invention, a boost converter device is provided comprising: a boost converter having an inductor and a switching element, capable of amplifying and outputting a voltage input from a battery; a current sensor that detects an inductor current flowing through the inductor;and a control unit that controls the boost converter by utilizing a control of the inductor current, wherein the control unit performs at least either a reduction of a carrier frequency used in controlling the boost converter or a reduction of a duty cycle command value used in controlling the boost converter, detects an amplitude of a current ripple via the current sensor when the reduction of the carrier frequency or the reduction of the duty cycle command value occurs, and detects the current sensor as abnormal when the amplitude of the current ripple falls below a predetermined current fluctuation range during an abnormality of the current sensor.

[0007] In the boost converter device of the present invention, it is possible that the reduction of the carrier frequency or the reduction of the duty cycle command value only takes place in one period in order to detect the amplitude of the current ripple via the current sensor.

[0008] Furthermore, in the upconverter device of the present invention, it is possible that the abnormality detection of the current sensor only occurs when a vehicle speed is greater than or equal to a predetermined value.

[0009] According to the boost converter device of the present invention, abnormality detection is achieved by at least either reducing the carrier frequency used in controlling the boost converter or reducing the duty cycle command value used in controlling the boost converter, and detecting the amplitude of the current ripple of the inductor current via the current sensor. This makes the difference between the amplitudes during normal operation and during abnormality of the inductor current ripple significant during current sensor detection, so that the abnormality can be appropriately detected.

[0010] The present invention will become more apparent from the following detailed description with reference to the accompanying drawings, in which identical parts are shown in different views with the same reference numerals. The drawings show:

[0011] Fig. 1 a schematic figure to illustrate an overall configuration of a motor control device comprising a boost converter device according to an embodiment of the present invention;

[0012] Fig. 2. A diagram illustrating the flow of a current when a switching element of an upper arm of a converter is in an ON state;

[0013] Fig. 3. An illustration to demonstrate the flow of a current when a switching element of a lower arm of the converter is in an ON state;

[0014] Fig. 4 a functional block diagram of a section relating to a converter control in a control unit of the boost converter device;

[0015] Fig. 5. An illustration showing, by means of a bold solid line, a waveform of an inductor current as detected by a current sensor during a normal state, and, by means of a thin solid line, this during an abnormal state;

[0016] Fig. 6. A flowchart illustrating a processing procedure for detecting an abnormality of the current sensor, which is executed in the control unit;

[0017] Fig. 7 a figure to illustrate the waveform of the choke current in the case where a carrier frequency is reduced for current sensor abnormality detection;

[0018] Fig. 8 a figure to illustrate the waveform of the choke current in the case where a duty cycle command value is reduced for current sensor abnormality detection; and

[0019] Fig. Figure 9 illustrates the execution of a processing operation to temporarily increase the ripple of the choke current for abnormality detection of the current sensor when the converter output voltage is in a maximum gain state. (Forms of execution)

[0020] An embodiment of the present invention is described below with reference to the accompanying drawings. In this description, a particular shape, material, numerical value, direction, and the like serve as examples to simplify the understanding of the present invention and can be modified appropriately in accordance with a use, purpose, specifications, or the like. Furthermore, in cases where several embodiments, modifications, or the like are included, the use of characterizing sections of these in a suitable combination is assumed from the outset.

[0021] Fig. Figure 1 shows a circuit diagram of a motor control device 100 , which is a boost converter device 10 according to one embodiment of the present invention. The motor control device 100indicates how in Fig. Figure 1 shows a battery B, which is a DC power source, and a converter (boost converter). 20 , an inverter 30 , positive lines 12a and 12b , a negative line 14 , current sensors 13 and 21 , voltage sensors 16 , 18 and 22 , a filter capacitor C1, a smoothing capacitor C2 and a control unit 50 up. Of these, the boost converter device 10 configured in this embodiment to use the converter 20 , the current sensor 21 and a section of the control unit 50 to demonstrate.

[0022] The motor control device 100is installed in an electric vehicle, such as a hybrid vehicle or an electric car (including a fuel cell vehicle). Consequently, a vehicle-mounted motor M is mechanically connected to drive wheels (not shown) and the motor M generates torque to propel the vehicle. Alternatively, the motor M can be integrated into a hybrid vehicle as an electric motor mechanically connected to an internal combustion engine (not shown) to act as an electric generator, producing electrical energy using the energy from the internal combustion engine and performing the starting of the internal combustion engine.

[0023] Battery B is a rechargeable energy storage device, and preferably a secondary battery, such as a nickel-hydrogen battery or a lithium-ion battery, is used. Instead of or in addition to battery B, a large-capacity capacitor can be used as an energy storage device.

[0024] The positive leadership 12a is connected to a positive terminal of battery B, and the negative lead 14 is connected to the negative terminal of battery B. Then, a system main relay SMR1 is connected to the positive line. 12a and a system main relay SMR2 ​​in the negative line 14 Provided. Each of the relays SMR1 and SMR2 ​​is activated in response to a signal from the control unit. 50 ON / OFF controlled.

[0025] The voltage sensor 16is a sensor that detects the voltage between the terminals of battery B. One of the voltage sensor 16 The detected battery voltage VB is sent to the control unit. 50 Given. Below is the voltage sensor. 16 appropriately described as a VB sensor.

[0026] The current sensor 13 is in positive management 12a provided, which is connected to the positive terminal of battery B. The current sensor 13 is a sensor that detects the current flowing into and out of battery B. A current sensor 13 The detected battery current IB is sent to the control unit 50 Given. Below is the current sensor. 13 appropriately described as an IB sensor.

[0027] The converter 20 The device comprises an inductor L1, switching elements Q1 and Q2, and diodes D1 and D2. One end on one side of the inductor L1 is connected to the positive lead. 12aconnected to the positive terminal of battery B, and one end on the other side is connected to a node 11 The switching element Q1 is connected to the switching element Q2. The switching elements Q1 and Q2 are connected between the positive line. 12b and the negative line 14 The diodes are connected in series. Subsequently, diodes D1 and D2 are connected antiparallel to switching elements Q1 and Q2.

[0028] The converter 20 is a voltage conversion device that converts the voltage between the positive line 12b and the negative line 14 based on a signal from the control unit 50 can amplify to a voltage greater than or equal to the battery voltage VB. In a signal of the converter 20The circuit includes a gate signal S1 for controlling the ON duty cycle of switching element Q1 and a gate signal S2 for controlling the ON duty cycle of switching element Q2. During amplification operation of the converter... 20 The gate signals S1 and S2 are correlated in such a way that the switching elements Q1 and Q2 assume opposite states (i.e., when the switching element Q1 is switched ON, the switching element Q2 is switched OFF, and when the switching element Q1 is switched OFF, the switching element Q2 is switched ON).

[0029] The current sensor 21 detects a choke current IL, which is passed through the choke coil L1 of the converter 20 flows, and transmits the measured value to the control unit. 50 The current sensor 21The current sensor detects a current flowing from battery B to choke coil L1 as a positive value, and a current flowing from choke coil L1 to battery B as a negative value. The current sensor is shown below. 21 appropriately described as an IL sensor.

[0030] The filter capacitor C1 is located between the positive line 12a and the negative line 14 switched. The voltage sensor 18 The voltage VL between the two ends of the filter capacitor C1 is detected as an input voltage of the converter. 20 and sends the measured value to the control unit 50 The voltage sensor is shown below. 18 appropriately referred to as a VL sensor.

[0031] The smoothing capacitor C2 is connected between the positive line 12b and the negative line 14 switched. The smoothing capacitor C2 smooths a DC voltage from the converter. 20and sends the smoothed DC voltage to the inverter. 30 The voltage sensor 22 A voltage VH between the two ends of the smoothing capacitor C2 is detected as an output voltage of the converter. 20 and sends the measured value to the control unit. The voltage sensor is shown below. 22 appropriately described as a VH sensor.

[0032] The inverter 30 exhibits a U-phase arm 32 , a V-phase arm 34 and a W-phase arm 36 up. The U-phase arm 32 , the V-phase arm 34 and the W-phase arm 36 are between the positive line 12b and the negative line 14 connected in parallel. The phase arms 32 , 34 and 36 each has two switching elements Q3 and Q4 connected in series, two switching elements connected in series 05and Q6 and two series-connected switching elements Q7 and Q8. Diodes D3 and D8 are connected antiparallel to switching elements Q3 to Q8. Subsequently, an intermediate point is determined for each of the phase arms. 32 , 34 and 36 connected to one of the phase coils of the motor M.

[0033] The inverter 30 converts direct current energy supplied by the positive line 12b and the negative line 14 is provided based on signals S3 to S8 from the control unit 50 in a three-phase current and supplies the three-phase current to the motor M to drive the motor M. In this way, the motor M is operated to generate a torque that is determined by a torque command value TR. Furthermore, the inverter converts 30 , during regenerative braking of a vehicle connected to the engine control unit 100equipped electric vehicle, three-phase alternating current energy generated by motor M based on a signal from the control unit 50 in direct current energy and transfers the direct current energy to the positive line 12b and the negative line 14 on the side of the converter 20 In this case, the converter 20 the direct current energy supplied by the inverter 30 The voltage is reduced to a level at which battery B can be charged, and the converter outputs... 20 The reduced DC energy is then sent to side B of the battery. Consequently, the converter 20 also known as an upward / downward converter.

[0034] Motor M has a rotary angle sensor. 38 provided. The rotary angle sensor 38 detects a rotor position of the motor M and sends a detection value θ to the control unit. 50 The control unit 50The measured value θ can be used during a coordinate transformation or the like to generate a control signal for the inverter. 30 use. Furthermore, the control unit 50 Obtain an engine speed MRN or a vehicle speed based on the detected value θ.

[0035] The control unit 50 It is built with an electronic control unit (ECU) with an integrated central processing unit (CPU) (not shown) and memory (not shown). The control unit 50 is designed to execute predetermined computational processing based on a characteristic map and a program stored in memory, and to manage the state of battery B or control the converter. 20 and the inverter 30to execute. Alternatively, at least one section of the ECU can be designed to perform a predetermined numerical or logical calculation using hardware, such as an electronic circuit.

[0036] The control unit 50 generates a PWM signal to control the converter 20 using a pulse width modulation method, it outputs the generated PWM signal as signals S1 and S2 to the converter. 20 Furthermore, the control unit generates 50 a PWM signal for controlling the motor M based on the torque command value TR of the motor M, which is received from an external ECU (not shown), and the motor speed MRN, which is derived from the sensing value θ of the rotary angle sensor 38 is obtained, and outputs the generated PWM signal as signals S3 to S8 to the inverter. 30 .

[0037] Furthermore, this embodiment describes the state management of battery B and the control of the converter. 20 and the inverter 30 from a single control unit 50 The present invention is not limited to this, however. For example, a configuration is conceivable in which the state management of battery B and the amplification operation of the converter are carried out. 20 and the DC / AC conversion operation of the inverter 30 each is managed or controlled by separate electronic control units (ECUs).

[0038] Fig. Figure 2 shows the flow of a (positive) current IL when the switching element Q1 of the converter is switched on. 20 exhibits an ON state; i.e., during an ON state of an upper arm. In this case, the current IL flows, as shown in Fig. 2 shown, through diode D1. When the voltage between the node 11of the switching element Q1 with the switching element Q2 and the negative line 14 Given that Vm is determined, an induction value of the choke coil L1 is determined as L, and a slope (a rate of change per time) of the current IL is determined as dIL / dt, a voltage equation in this state is obtained according to the following equation (1). VL – L(dIL / dt) – Vm = 0 (1)

[0039] If the voltage between the two ends of the smoothing capacitor Q2 is determined as VH while the current IL flows through the diode D1, the voltage Vm becomes equal to VH, so that when equation (1) is modified by substituting VH for Vm in equation (1), the following equation (2) is obtained, and then when equation (2) is further modified, equation (3) is obtained. VL-L(dIL / dt)-VH = 0 (2) dIL / dt = (VL – VH) / L (3)

[0040] Equation (3) shows that when the current IL is positive during an ON state of the switching element Q1, the slope dIL / dt of the current IL becomes (VL – VH) / L. Normally, since VL is less than VH, the slope dIL / dt of the current IL is negative.

[0041] Fig. Figure 3 shows the flow of the (positive) current IL when the switching element Q2 of the converter is switched on. 20 exhibits an ON state; i.e., during the ON state of a lower arm. In this case, the current IL flows, as in Fig. 3 shown, by the switching element Q2. A voltage equation in this state yields the following equation (4). VL – L(dIL / dt) – Vm = 0 (4)

[0042] Equation (4) itself is equal to equation (1) described above. However, while the current IL flows through the switching element Q2, the voltage Vm is not VH, but 0, so that when equation (4) is modified by substituting 0 for Vm in equation (4), the following equation (5) is obtained, and then when equation (5) is further modified, equation (6) is obtained. VL – L(dIL / dt) – 0 = 0 (5) dIL / dt = VL / L (6)

[0043] Equation (6) shows that when the current IL is positive during the ON state of switching element Q2, the slope dIL / dt of the current IL becomes VL / L. Normally, since VL is greater than 0, the slope dIL / dt of the current IL is positive.

[0044] Consequently, when the current IL is positive, the current IL decreases with the slope (VL – VH) / L during the ON state of switching element Q1, and the slope dIL / dt of the current IL increases with VL / L during the ON state of switching element Q2. Accordingly, during amplification operation by the converter 20 , a ripple (ripple) is generated in which the choke current IL repeatedly increases and decreases due to the switching operations of the switching elements Q1 and Q2 (see Fig. 5).

[0045] The following is the control of the amplification operation in the converter. 20 with reference to the Fig. 4 described. Fig. Figure 4 shows a functional block diagram of a section relating to the control of the converter. 20 the control unit 50 The control unit 50 indicates how in Fig. 4 shows a voltage command generation section 52 , subtraction sections 54 and58 , a voltage control calculation section 56 , a current control calculation section 60 , a control signal generation section 62 , a carrier generation section 64 and a duty cycle command value change section 66 on.

[0046] The voltage command generation section 52 generates a voltage command value VR, which indicates a setpoint of the voltage VH, the output voltage of the converter. 20 is. The voltage command generation section 52 The voltage command value VR is generated, for example, based on the requested power of the motor M, which is calculated from the torque command value TR of the motor M and the motor speed MRN.

[0047] The subtraction section 54 subtracts the value from the VH sensor 22The input voltage VH value is taken from the voltage command value VR, and the calculation result is sent to the voltage control calculation section. 56 .

[0048] The voltage control calculation section 56 The system executes a control loop (such as a proportional-integral controller) to adjust the voltage VH to the voltage command value VR using the voltage sensing value VL and a value obtained by subtracting the voltage sensing value VH from the voltage command value VR. The voltage control calculation section then outputs the result. 56 a calculated tax amount is output as a current command value IR.

[0049] The subtraction section 58 subtracts the current IL measurement value from the current command value IR, which is determined by the voltage control calculation section. 56is output, and passes the calculation result to the current control calculation section. 60 .

[0050] The current control calculation section 60 receives a value obtained by measuring the current IL through the IL sensor. 21 from the current command value IR from the subtraction section 58 The subtraction takes place, and a control mechanism (such as a proportional-integral controller) is executed to adjust the current IL to the current command value IR. The current control calculation section then outputs the following. 60 a calculated tax amount as a duty cycle command value d to the control signal generation section 62 .

[0051] The carrier generation section 64 generates a carrier signal CR from a triangle wave to generate the PWM signals S1 and S2 in the control signal generation section. 62(described below) and outputs the generated carrier signal CR to the control signal generation section. 62 Furthermore, the carrier generation section exhibits 64 a function for temporarily changing the frequency fcr of the carrier signal CR when processing to detect an abnormality of the IL sensor 21 in the control unit 50 This occurs when a predetermined condition is met. This will be discussed in more detail below.

[0052] The control signal generation section 62 compares the duty cycle command value d, which is determined by the current control calculation section 60 is received with the carrier signal CR, which is generated by the carrier generation section 64 The signal received, with respect to a certain amount, generates the gate signals S1 and S2 in accordance with the comparison result. The control signal generation section 62For example, it sets the gate signal S1 to ON (and the gate signal S2 to OFF) if the carrier signal CR is below the duty cycle instruction value d, and sets the gate signal S2 to ON (and the gate signal S1 to OFF) if this is not the case.

[0053] The duty cycle command value change section 66 includes a function for temporarily changing the duty cycle command value d when processing for abnormality detection of the IL sensor. 21 in the control unit 50 This occurs when a predetermined condition is met. This will be discussed in more detail below.

[0054] At the control unit 50 With the configuration described above, the regulation to adjust the voltage VH to the voltage command value VR is carried out by the voltage control calculation section. 56 (Voltage control).

[0055] The current control calculation section then regulates the adjustment of the current IL to the current command value IR. 60 with the control output of the voltage control calculation section 56 as the current command value IR of the current IL (current control). In this way, the voltage VH and the current IL are regulated so that a fluctuation in the voltage VH due to a dead time or the like, which is defined between the ON times (ON states) of the switching elements Q1 and Q2, can be suppressed.

[0056] Furthermore, the subtraction section 54 , the voltage control calculation section 56 , the subtraction section 58 and the current control calculation section 60 a main loop 68 to match the voltage VH to the voltage command value VR and form the subtraction section 58 and the current control calculation section60 a side loop 70 to match the current IL to the current command value IR.

[0057] To cause the boost converter device 10 , which the converter 20 , the IL sensor 21 and the control unit 50 If the sensor is functioning normally, the VH sensor must be working. 22 , the VL sensor 18 and the IL sensor 21 They operate normally. This assumes that an abnormality, such as a fluctuation in the measured value, occurs only within a narrow current range in the IL sensor. 21 This occurs because it is not possible to regulate the choke current IL appropriately, so that the regulation of the output voltage VH after amplification is also affected, resulting in a situation where the output voltage VH cannot be approximated to a gain voltage command value. From this point of view, an abnormality of the IL sensor must be assumed. 21The device that detects the choke current IL must be detected in a suitable manner. Accordingly, in the boost converter device 10 In this embodiment, a sensor abnormality is detected in a suitable manner by means of processing to detect the abnormality of the IL sensor. 21 The process, which is described below, is executed as follows. The following is the processing procedure for detecting an abnormality of the IL sensor. 21 described, which is in the control unit 50 is executed.

[0058] Fig. Figure 5 shows an illustration of the processing for detecting the abnormality of the IL sensor. 21 . In the Fig. 5 is the waveform of the IL sensor 21 The measured choke coil current IL is shown as a thick solid line during a normal state and as a thin solid line during an abnormal state.

[0059] During the normal state of the IL sensor 21 The choke current IL exhibits an essentially sawtooth waveform, with triangular ripples R resulting from the switching operations of the switching elements Q1 and Q2 of the converter. 20are regular. Here, when focusing on a ripple R, a period T2, in which the inductor current IL increases, corresponds to an ON period of the switching element Q2, and the slope dIL / dt of this becomes VL / L, as shown in equation (6) described above. In contrast, for the current ripple R, a period T1, in which the inductor current IL decreases, corresponds to an ON period of the switching element Q1, and the slope dIL / dt of this becomes (VL – VH) / L, as shown in equation (3) described above. Subsequently, the sum of periods T1 and T2 equals a control period Tcyc, and the duty cycle command value d can be described by the following equation (7) as the ratio of the ON period T1 of the switching element Q1 to a control period Tcyc. d = Tl / Tcyc (= VL / VH) (7)

[0060] Since the frequency fcr of the carrier signal CR, which is generated by the carrier generation section 64 in the control unit50 Since the output has a known value that is stored in advance, a control period Tcyc can be described by a ripple R of the choke current IL by 1 / fcr. Furthermore, the duty cycle command value d, which is determined by the current control calculation section, is 60 in the control unit 50 A known value is also output. Accordingly, the ON period T1 of switching element Q1 and the ON period T2 of switching element Q2 can be calculated using the following equations (8) and (9). T1 = d·Tcyc = d / fcr (8) T2 = Tcyc – T1 (9)

[0061] The inductor current IL is usually measured by the IL sensor. 21 each tax period Tcyc. In the Fig. Figure 5 shows the normal sampling times ST1, ST2, and ST3 with respect to the three ripple R, indicated by O-markings. In this way, when the choke coil current IL is sampled each control period, the sensing values ​​IL are taken by the IL sensor. 21 , which works normally, essentially gives the same value.

[0062] In contrast, at the time of processing to detect the abnormality of the IL sensor, 21 , with respect to the ripple R, which forms a triangular shape in the waveform of the choke current IL, a detection value ILA of the IL sensor 21 at a sampling time STA corresponding to a peak section and a detection value ILB of the IL sensor 21At a sampling time STB, a bottom section is obtained. The sampling times STA and STB can be determined by the input times T1 and T2, which are obtained by equations (8) and (9) described above. Subsequently, by calculating a difference by subtracting the acquisition value ILB from the acquisition value ILA, a maximum current fluctuation range ΔILmax of the ripple R can be obtained; i.e., the amplitude of the ripple R. Here, the "peak section" is defined as containing an inflection point on the upper side of the triangular ripple R and a high-current section in its immediate vicinity, and the "bottom section" is defined as containing an inflection point on the lower side of the triangular ripple R and a low-current section in its immediate vicinity.

[0063] Furthermore, the sampling times STA and STB of the choke current IL during the abnormality detection processing of the IL sensor 21 can be added to the normal sampling time ST2. Alternatively, the choke current IL at sampling times STA and STB can be obtained in a single period during the abnormality detection processing instead of the normal sampling time ST2. In addition, the choke current IL can be, as in Fig. Figure 5 shows that the current can be sampled at a sampling time STC, where a current increase of the ripple R begins, in addition to (or instead of) the sampling time STB. In this case, the current value ILB of the bottom section of the ripple R can be determined by averaging the acquisition values ​​at the two sampling times STB and STC.

[0064] By calculating and monitoring the maximum current fluctuation range ΔILmax with respect to the ripple R of the choke current IL, the abnormality of the IL sensor can be detected. 21 , as described above, are appropriately detected. In the IL sensor 21 For example, an abnormality sometimes occurs in which a state is created according to which a detection value is fixed at an essentially constant value. This abnormality is referred to below as a "fixing error." In this case, the detection value of the IL sensor fluctuates. 21 , as in Fig. Figure 5 shows the current fluctuation range only within a predetermined narrow range ΔILth. The current fluctuation range ΔILth is a specific value corresponding to the type or similar of the IL sensor. 21 and is stored in advance in the memory of the control unit 50 saved.

[0065] If the maximum current fluctuation range ΔILmax of the choke coil current IL, obtained according to the above description, is greater than the current fluctuation range ΔILth, it can be determined that the fixing error in the IL sensor 21 This has not occurred. However, there is a case where the maximum current fluctuation range ΔILmax of the ripple R of the choke current IL is smaller than the current fluctuation range ΔILth, and in this case, there is a possibility that the fixing error of the IL sensor 21 cannot be captured in a suitable manner.

[0066] Consequently, in the upscaling device 10 in this embodiment, the fixing error of the IL sensor 21 detected by temporarily increasing a fluctuation range of the ripple R and then performing the sampling of the choke current IL as described above.

[0067] Fig. Figure 6 shows a flowchart to illustrate a process in the control unit. 50 processing method implemented in this embodiment for detecting the fixing error of the IL sensor 21 This processing takes place at each predetermined time in the CPU of the control unit. 50 .

[0068] The control unit 50 First, in step S10, it is determined whether or not a vehicle speed of a motor control device is determined. 100 The equipped electric vehicle exceeds a predetermined speed SPth. In this way, the processing serves to detect the abnormality of the IL sensor. 21 In the event that the vehicle speed exceeds the predetermined speed SPth, to suppress the influence on drivability, the processing is carried out within a vehicle speed range up to the extent that noise and vibration of the converter are minimized.20 Driving noises (such as engine noise, road noise, or the like) are not a cause for concern, since a carrier frequency used in the control of the converter is present. 20 The use of the device decreases the likelihood of noise or vibration increasing. If the determination in step S10 above is positive, the routine proceeds to step S12; otherwise, it proceeds to step S26.

[0069] In the event of a positive determination in step S10, the control unit reduces 50 In the subsequent step S12, the carrier frequency fcr changes from f1 to f2. This processing is carried out by the carrier generation section. 64 executed (see Fig. 4) Here, f1 describes a carrier frequency used in the gain control of the converter. 20 was used before an operation to detect the fixing error of the IL sensor 21This occurs, and f2 describes a carrier frequency for fixing error detection that is lower than f1.

[0070] The carrier frequency fcr, which is used in the control of the converter 20 The amount used is reduced in this way, so that, as in Fig. As shown in Figure 7, the control periods of the switching elements Q1 and Q2 increase, so that the ripple R of the choke current IL increases. In this state, in the subsequent step S14, the detection values ​​ILA and ILB, which are measured by the IL sensor, are 21 are recorded, sampled at the sampling times STA and STB, as with reference to the Fig. 5 described. The control unit then calculates 50 , in the following step S16, an actual measurement value of a current fluctuation range ΔIL (hereinafter referred to as a “ΔIL actual measurement value”) by subtracting the acquisition value ILB from the acquisition value ILA.

[0071] The control unit then determines 50, in the following step S18, whether or not the ΔIL actual measured value during the fixing error of the IL sensor 21 below the current fluctuation range ΔILth. Here, as in Fig. 5 shown, in the case that the IL sensor 21 subject to a setting error, the measured values ​​ILA and ILB, which are taken by the IL sensor 21 The values ​​are captured as a waveform that, during the settling error, fluctuates only within a narrow range of the current fluctuation range ΔILth. Consequently, if the ΔIL actual measured value, which describes the difference between the detected values ​​ILA and ILB, is smaller or narrower than the current fluctuation range during the settling error, the IL sensor 21 as being subject to an error in assessment.

[0072] In step S18 above, if the actual ΔIL reading during the locking error is not less than the current fluctuation range ΔILth (NO in step S18), the sensor locking error detection processing is terminated by processing in step S26. Conversely, if it is determined that the actual ΔIL reading during the locking error is less than the current fluctuation range ΔILth, in the following step S20, n (initial value = 0) is incremented to take on the value n + 1, and in the following step S22, it is determined whether or not n has taken on a predetermined value m (an integer greater than or equal to 2). Subsequently, the processing from step S14 to step S20 is repeated until it is determined that n equals m. The degree of accuracy of the locking error detection is improved by determining the locking error of the IL sensor. 21 This is repeated several times.

[0073] In the event that step S22 determines that n equals m; i.e., in the event that a state in which the ΔIL actual measured value during the fixing error is smaller than the current fluctuation range ΔILth is continuously detected several times, in the following step S24 it is detected that the IL sensor 21 is subject to a fix error. In this way, if the fix error of the IL sensor occurs, 21 The control unit is determined 50 switch to a fail-safe mode. According to the fail-safe mode, for example, instead of the IL sensor's measured value, 21 , which is from the IB sensor 13 Battery current IB measured during current control of the converter 20 can be used. Furthermore, the control unit provides information 50 preferably a driver via a lamp indicator, an acoustic signal or the like about the IL sensor's fixing error 21 .

[0074] The control unit then sets 50 , in the following step S26, n is set back to 0 and the carrier frequency fcr is set back from f2 to f1 (see Fig. 7) This terminates the processing for detecting the IL sensor's fix error.

[0075] According to the boost converter device 10 This embodiment, as described above, is used when the fixing error of the IL sensor occurs. 21 The detection error of the IL sensor is recorded. 21This is determined by temporarily increasing the ripple R of the choke current IL and then sampling the peak section, which describes the high-current portion of the ripple R, and the bottom section, which describes the low-current portion of the ripple R, with the IL sensor. In this way, the difference between current fluctuations during normal conditions and during the settling error with respect to the ripple R of the choke current IL becomes significant, and thus the settling error of the IL sensor can be determined. 21 be recorded in a suitable manner.

[0076] Furthermore, according to the booster device 10 In this embodiment, the carrier frequency is changed only in one period, for example to sample the two current values ​​ILA and ILB for the fixing error detection processing, and the influence on drivability can thus be reduced by keeping the sampling period as short as possible.

[0077] Furthermore, according to the upconverter device 10 in this embodiment, the processing for detecting the fixing error of the IL sensor 21 , if the vehicle speed is greater than or equal to a predetermined value, so that even if noise and vibration occur due to a reduction in the carrier frequency fcr in the converter 20 a driving condition is created in which the noise and vibration do not disturb the driver or the like, so that the impact on drivability can be reduced.

[0078] It should be noted that the upconverter device of the present invention is not limited to the embodiment and modification thereof described above, but can be modified or improved in various ways without abandoning its scope of protection as set out in the attached claims.

[0079] It has been described above that the ripple R of the choke current IL is increased, for example, by decreasing the carrier frequency fcr. However, the present invention is not limited to this, and the ripple R of the choke current IL can be, as shown in parentheses in step S12 of the Fig. As shown in Figure 6, the duty cycle command value d can be increased by using the duty cycle command value change section. 66 The voltage is temporarily reduced. This reduces the duty cycle, thereby decreasing the gain voltage of the converter. 20 is temporarily increased, so that, as in the Fig. Figure 8 shows a state in which the amplitude of the ripple R of the choke coil current IL is equally high. In this state, the detection of the fixing error of the IL sensor is possible. 21This is achieved by capturing the current value ILA at sampling time STA corresponding to the peak section of the ripple R and a current value ILC at sampling time STC corresponding to the bottom section of the ripple R. The same effects as in the embodiment described above can be produced in this way. Furthermore, a state in which the amplitude of the ripple R is high can be generated by performing processing to decrease such a duty cycle command value simultaneously with the reduction of the carrier frequency fcr. Furthermore, when a sampling for the choke current IL fix error detection processing is completed, the duty cycle command value d is reset to its original value in step S26. In this way, as described in Fig. Figure 8 shows the output voltage of the converter. 20 away.

[0080] Furthermore, in the embodiment described above, the processing is for detecting the fixing error of the IL sensor. 21 The invention is described as being achieved by reducing the carrier frequency (and / or the duty cycle command value) when the vehicle speed is greater than or equal to a predetermined value. However, the invention is not limited to this. In the event that the degradation of noise and vibration (NV) performance can be avoided by adjusting the carrier frequency f2 to a frequency at which noise and vibration are severe, or the like, the processing can be adapted to detect the IL sensor's fixing error. 21 This can be done regardless of the vehicle speed.

[0081] Furthermore, as in Fig. 9 shown, in the converter 20, an upper gain limit VHmax is set taking into account the compressive strength or similar properties of a switching element. However, it can also be in a state according to which the output voltage VH of the converter 20 Once the upper gain limit VHmax has been reached, the processing for detecting the fixing error of the IL sensor begins. 21 This occurs after the ripple of the choke current IL has been temporarily increased by a reduction in the carrier frequency or the like. This is because, usually, the upper gain limit VHmax (such as 600 V) is set with a margin, so that even if the gain voltage VH temporarily exceeds the designed value by ΔVH (such as 20 V), the switching element never fails. In this way, the detection of the IL sensor's locking fault can also occur in a state where the converter's output voltage is 20Once the upper gain limit has been reached, the fixing error of the IL sensor is detected in a suitable manner.

[0082] Furthermore, in the embodiment described above, the first step, S10, is the processing to detect the fixing error of the IL sensor. 21It is described as occurring when the vehicle speed is greater than or equal to a predetermined value. Before this, however, the maximum current fluctuation range ΔILmax of the ripple R of the choke current IL is estimated by calculation. It is then determined whether the estimated value during the settling error of the IL sensor is greater than the current fluctuation range ΔILth. For example, it may be the case that processing step S10 or later only occurs if it is determined that the estimated value is greater than the current fluctuation range ΔILth. In this case, the maximum current fluctuation range ΔILmax of the ripple R (i.e., the amplitude of the ripple) can be calculated based on the input voltage VL and the output voltage VH of the converter. 20and the carrier frequency f1 before sampling for fix error detection. More precisely, with respect to the ripple R of the choke current IL, the maximum current fluctuation range ΔILmax of the ripple can be calculated, as in Fig. As shown in Figure 5, the slope VL / L can be calculated by multiplying the slope VL / L by time T2 or by multiplying the slope (VL – VH) / L by time T1. This allows for more reliable detection of the IL sensor's locking error by performing the locking error detection processing only when it is estimated that the fluctuation range of the choke current IL exceeds the current fluctuation range during the sensor locking error.

[0083] The above describes a boost converter device.

[0084] A boost converter device comprises a converter, a current sensor that detects the inductor current flowing through an inductor, and a control unit that controls the converter by utilizing inductor current regulation. The switching element performs a reduction of a carrier frequency used in controlling the converter and / or a reduction of a duty cycle command value used in controlling the converter, detects the amplitude of a current ripple across the current sensor during an execution of the carrier frequency reduction or the duty cycle command value reduction, and detects that the current sensor is abnormal if the amplitude of the current ripple during the current sensor abnormality falls below a predetermined current ripple range. QUOTES INCLUDED IN THE DESCRIPTION

[0085] This list of documents cited by the applicant was automatically generated and is included solely for the reader's convenience. The list is not part of the German patent or utility model application. The DPMA accepts no liability for any errors or omissions. Cited patent literature

[0086] JP 2014-97799

[0001] JP 2006-311635 A [0003, 0004]

Claims

[1] Up-converter device with: – a boost converter which has a choke coil and a switching element and can amplify and output a voltage input from a battery; – a current sensor that detects the current flowing through the choke coil; and – a control unit that controls the boost converter by utilizing a regulation of the choke current, wherein – the control unit: – reduces a carrier frequency used in controlling the boost converter and / or reduces a duty cycle command value used in controlling the boost converter, – an amplitude of a current ripple detected via the current sensor when the carrier frequency is reduced or the duty cycle command value is reduced, and – the current sensor is detected as abnormal if the amplitude of the current ripple during a current sensor abnormality is below a predetermined current fluctuation range. [2] Up-converter device according to claim 1, characterized by that the reduction of the carrier frequency or the reduction of the duty cycle command value only occurs in one period in order to detect the amplitude of the current ripple via the current sensor. [3] Up-converter device according to claim 1 or 2, characterized by that the abnormality detection of the current sensor occurs when a vehicle speed is greater than or equal to a predetermined value.

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