Power conversion device

Abstract

Power conversion device comprising: an inverter (1) that converts a DC voltage into an AC voltage and drives a synchronous motor (2);and a magnetic pole position correction unit (46) that corrects an error in the magnetic pole position of a rotor from a rotation angle sensor (8) of the synchronous motor (2), wherein the magnetic pole position correction unit (46) comprises a calculation unit for the actual current phase, which calculates a current phase from a current when three-phase lines are short-circuited during the rotation of the synchronous motor (2), and a calculation unit (55) for the ideal current phase, which calculates an ideal current phase based on a received rotational speed of the rotor, a received temperature of a stator, and a received temperature of the rotor of the synchronous motor (2), and corrects the magnetic pole position from a difference between outputs of the calculation unit for the actual current phase and the calculation unit (55) for the ideal current phase.

Description

Technical field

[0001] The present invention relates to a power conversion device. State of the art

[0002] When a system in which a permanent magnet synchronous motor is driven by an inverter is controlled, information about the magnetic pole position of the rotor relative to the stator is required. In the case of an electric vehicle, this magnetic pole position information is acquired by a rotary angle sensor (rotation sensor or similar) attached to the rotor to control the motor's torque. However, if there is an angular error caused by a mounting fault in the rotary angle sensor, a torque that differs from the torque command value will be generated, resulting in a deterioration of the vehicle's drivability.

[0003] Therefore, it is necessary to estimate the actual rotor position of the permanent magnet synchronous motor and to detect any offset error of the rotation angle sensor based on this estimated actual rotor position. Such conventional techniques are described, for example, in PTL 1 and PTL 2. List of prior art patent literature PTL 1: JP 2014-050122 A PTL 2: JP 2017-212783 A Summary of the invention: Technical problem

[0004] In the invention described in PTL 1, only a d-axis voltage is applied in a stopped rotor state, and an angle at which a q-axis current becomes zero is sought. However, with this method, calibrating the amplitude of Vd and an excitation time, as well as determining the zero point with respect to the q-axis current, is difficult, and it is difficult to improve the accuracy sufficiently.

[0005] The invention described in PTL 2 calculates an offset error based on a difference between a phase current detected in a state where one motor end is short-circuited and a theoretical phase current value. However, if the rotational speed is relatively low (for example, 1000 to 2000 rpm) -1 ), is a large error caused by motor temperature, especially stator temperature, and it is difficult to improve the accuracy sufficiently.

[0006] One problem that the present invention aims to solve is to detect an offset error of a rotation angle sensor of a rotor with high accuracy. Solution to the problem

[0007] Therefore, a power conversion device according to the present invention comprises: an inverter that converts a DC voltage into an AC voltage and drives a synchronous motor; and a magnetic pole position correction unit that corrects an error in the magnetic pole position of a rotor from a rotation angle sensor of the synchronous motor, wherein the magnetic pole position correction unit comprises an actual current phase calculation unit, which calculates a current phase from a current when three-phase lines are short-circuited during rotation of the synchronous motor, and an ideal current phase calculation unit, which calculates an ideal current phase based on a rotational speed of the rotor and a temperature of a stator, and corrects the magnetic pole position from a difference between outputs of the actual current phase calculation unit and the ideal current phase calculation unit. Advantageous effects of the invention

[0008] The offset error of the rotor's rotation angle sensor can be detected with high accuracy. List of characters [ Fig. 1] Fig. Figure 1 is a diagram that represents a schematic circuit configuration of a power conversion device. [ Fig. 2] Fig. Figure 2 is a processing block diagram to describe the processing of a controller 4. [ Fig. 3] Fig. 3 is a flowchart that represents a procedure for calculating offset errors. [ Fig. 4] Fig. Figure 4 is a view showing a waveform of each phase current after the start of a three-phase short circuit. [ Fig. 5] Fig. Figure 5 is a block diagram representing a configuration of a magnetic pole position correction unit 46 according to a first embodiment. [ Fig. 6] Fig. Figure 6 is a graph that represents a relationship between rotational speed and an ideal current phase. [ Fig. 7] Fig. Figure 7 is a block diagram that represents a first modification of a calculation unit 55 for the ideal phase current. [ Fig. 8] Fig. Figure 8 is a block diagram that represents a second modification of the calculation unit 55 for the ideal phase current. [ Fig. 9] Fig. Figure 9 is a block diagram illustrating a configuration of the magnetic pole position correction unit 46 according to a second embodiment. [ Fig. 10] Fig. Figure 10 is a block diagram illustrating a modification of the magnetic pole position correction unit 46 according to the second embodiment. Description of embodiments

[0009] The following is a description of embodiments of a power conversion device according to the present invention with reference to the drawings. Furthermore, the same elements in the respective drawings are designated by the same reference numerals, and redundant descriptions thereof are omitted. First version

[0010] Fig. Figure 1 is a diagram representing a schematic circuit configuration of a power conversion device. Fig. 1 comprises the power conversion device of a motor 2, mainly an inverter 1, a battery 3, a controller 4, and the like. The inverter 1 is a three-phase, dual-level voltage inverter.

[0011] Motor 2 is connected to a drive mechanism of a motor vehicle system, and motor 2 rotates to propel a vehicle forward. In the present embodiment, motor 2 is an AC motor and is a three-phase internal permanent magnet synchronous motor (IPMSM). Motor 2 operates through an interaction between a magnetic flux generated by a permanent magnet provided in a rotor (not shown) and a magnetic field generated by currents iu, iv, and iw flowing through three-phase windings 5, 6, and 7 attached to an armature. Here, iu, iv, and iw represent a U-phase current, a V-phase current, and a W-phase current, respectively. It should be noted that a permanent magnet synchronous motor is described in the present embodiment, but the present invention is also applicable to other synchronous motors, such as a field-winding synchronous motor.

[0012] Motor 2 includes a magnetic pole position sensor 8. The magnetic pole position sensor 8 has a function for detecting the magnetic pole position of the rotor of motor 2 as an angle of rotation. The magnetic pole position sensor 8 outputs a magnetic pole position signal 9 (θ), which is to be input into the controller 4. The processing of the magnetic pole position signal 9 (θ) in the controller 4 will be described later with reference to Fig. 2 described. The magnetic pole position sensor 8 can be configured using a rotary encoder, a rotary encoder, an absolute encoder or the like.

[0013] The current sensor 10 detects currents flowing through windings 5, 6, and 7, outputs a U-phase current sensor signal 11 (signal indicating the current iu), a V-phase current sensor signal 12 (signal indicating the current iv), and a W-phase current sensor signal 13 (signal indicating the current iw) based on the detected currents, which are to be input into the controller 4. The processing of the current sensor signals 11, 12, and 13 in the controller 4 will also be described later with reference to Fig. 2 described.

[0014] The inverter 1 comprises switching elements 14, 15, 16, 17, 18, and 19 and freewheeling diodes 20, 21, 22, 23, 24, and 25. Each of the switching elements 14 to 19 of the present embodiment is a Si IGBT and comprises a gate terminal, a collector terminal, and an emitter terminal. Each of the freewheeling diodes 20 to 25 is connected between the collector terminal and the emitter terminal of each of the switching elements 14 to 19. If the collector terminal has a higher potential than the emitter terminal in each of the switching elements 14 to 19, each of the freewheeling diodes 20 to 25 prevents current from flowing through the freewheeling diodes 20 to 25 and prevents a high reverse voltage from being applied to the switching elements 14 to 19. However, the present invention is not limited to the combination of the Si-IGBT and the freewheeling diode, and an inverter circuit can be configured using other semiconductor elements.

[0015] The switching on and off of each of the switching elements 14 to 19 is performed by each of the gate drive signals 26, 27, 28, 29, 30, and 31, which are connected to the gate terminals of the respective switching elements 14 to 19. Six gate signals 32, which form the basis of the gate drive signals 26 to 31, are generated by the controller 4 and output to a gate drive circuit 35. The gate drive circuit 35 converts the gate signals 32 into potentials required to switch the switching elements 14, 15, 16, 17, 18, and 19 between on and off and outputs the gate drive signals 26, 27, 28, 29, 30, and 31. The generation of the gate signal 32 in the controller 4 will be described later with reference to Fig. 2 described.

[0016] The emitter terminal of switching element 14 and the collector terminal of switching element 15 are connected to each other, and their connection point is connected to winding 5 to allow the flow of current iu. The emitter terminal of switching element 16 and the collector terminal of switching element 17 are connected to each other, and their connection point is connected to winding 6 to allow the flow of current iv. The emitter terminal of switching element 18 and the collector terminal of switching element 19 are connected to each other, and their connection point is connected to winding 7 to allow the flow of current iw. The collector terminals of switching elements 14, 16, and 18 are connected to each other and to a high-potential DC wiring 33.Furthermore, the emitter terminals of the switching elements 15, 17 and 19 are connected to each other and to a low-potential DC wiring 34.

[0017] Consequently, the controller 4 switches the switching elements 14, 15, 16, 17, 18, and 19 on and off at suitable times based on the generated gate signals 32 and controls the currents iu, iv, and iw flowing through the windings 5, 6, and 7 to implement rotational control of the motor 2. The gate signal 32 is in the form of a pulse-width modulation (PWM) signal, so that the currents iu, iv, and iw are sinusoidal signals with phase differences of 120 degrees.

[0018] A voltage sensor 36 is connected to the high-potential DC wiring 33 and the low-potential DC wiring 34 and detects a potential difference between them. Since the potential difference between the high-potential DC wiring 33 and the low-potential DC wiring 34 is typically a high voltage of, for example, 100 V or more, the voltage sensor 36 generates a DC voltage sensor signal 37 (Vdc), which is converted into a low voltage detectable by the controller 4, and inputs the DC voltage sensor signal into the controller 4.

[0019] A smoothing capacitor 38, contained in the inverter 1, is connected between the high-potential DC wiring 33 and the low-potential DC wiring 34. The smoothing capacitor 38 has a function of suppressing a DC voltage pulsation generated by each switching operation of the switching elements 14 to 19.

[0020] In battery 3, a terminal on the high-potential side is connected to the high-potential DC wiring 33, and a terminal on the low-potential side is connected to the low-potential DC wiring 34. Therefore, battery 3 serves as a DC power supply, providing power to inverter 1 and motor 2.

[0021] A torque command 39 (T*), issued by a main controller of a vehicle, such as an electronic control unit (ECU), is input into the controller 4. The controller 4 performs torque control of the motor 2 based on the torque command 39 (T*). Furthermore, the controller 4 receives inputs of the temperatures of a stator and the rotor of the motor 2, although in Fig. 1 not shown. Although it is desirable that the temperatures of the stator and rotor be detected by a temperature sensor, it can be configured to use temperatures estimated by a known method.

[0022] Fig. Figure 2 is a processing block diagram to describe the processing of controller 4.

[0023] In Fig. 2. Processing blocks of the controller 4 include a current instruction calculation unit 40, a three-phase / two-phase conversion unit 41, a current control unit 42, a two-phase / three-phase conversion unit 43, a velocity calculation unit 44, a PWM gate control signal generation unit (control signal generation unit) 45, a magnetic pole position correction unit 46, a deviation calculator 47, a deviation calculator 48, and an adder 49. The current instruction calculation unit 40, the three-phase / two-phase conversion unit 41, the current control unit 42, the two-phase / three-phase conversion unit 43, the deviation calculator 47, and the deviation calculator 48 form a three-phase voltage instruction value generation unit. Normally, a three-phase voltage command value is generated based on the magnetic pole position signal 9 (θ) detected by the magnetic pole position sensor 8.In the present invention, however, a corrected magnetic pole position signal 50 (θ'), which is corrected by a method to be described later, is used instead of the magnetic pole position signal 9 (θ).

[0024] Controller 4 performs the rotation control by causing the three-phase currents iu, iv, and iw to flow through motor 2. Controller 4 employs a current vector control method, where processing is carried out in a coordinate system obtained by converting a fixed three-phase coordinate into a two-phase rotation coordinate represented by a d-axis and a q-axis.

[0025] The current command calculation unit 40 calculates a d-axis current command value id* and a q-axis current command value iq*. The torque command 39 (T*), an angular velocity ω, and the DC voltage sensor signal 37 (Vdc) are input to the current command calculation unit 40, and the d-axis current command value id* and the q-axis current command value iq* are calculated from these inputs. The angular velocity ω is calculated by the velocity calculation unit 44 based on the corrected magnetic pole position signal 50 (θ').

[0026] In a synchronous motor with a magnetic characteristic such as IPMSM, the torque T is expressed by the following formula (1) and depends on a current. T=Pp⋅{Φ+(Ld−Lq)id}⋅iq

[0027] In formula (1) id is a d-axis current, iq is a q-axis current, Pp is the number of pole pairs, Ld is a d-axis inductance, Lq is a q-axis inductance and Φ is a magnetic flux.

[0028] The corrected magnetic pole position signal 50 (θ') is input into the three-phase / two-phase conversion unit 41 and the two-phase / three-phase conversion unit 43 and is used for the conversion between a two-phase coordinate and a three-phase coordinate of the d-axis and the q-axis.

[0029] The three-phase / two-phase conversion unit 41 performs a coordinate conversion on the current sensor signals 11 (iu), 12 (iv) and 13 (iw) into the d-axis and q-axis on the basis of information about the corrected magnetic pole position signal 50 (θ') and outputs a d-axis detection current id and a q-axis detection current iq.

[0030] The deviation calculator 47 calculates a deviation between the d-axis current command value id*, output by the current command calculation unit 40, and the d-axis detection current id, output by the three-phase / two-phase conversion unit 41, and outputs a d-axis current deviation Δid to the current control unit 42. The deviation calculator 48 calculates a deviation between the q-axis current command value iq*, output by the current command calculation unit 40, and the q-axis detection current iq, output by the three-phase / two-phase conversion unit 41, and outputs a q-axis current deviation Δiq to the current control unit 42.

[0031] The current control unit 42 performs feedback control so that the d-axis differential current Δid and the q-axis current deviation Δiq, which each represent a deviation between a command value (a target value) and a measured value (an output value), become zero. It then calculates and outputs a d-axis voltage command Vd* and a q-axis voltage command Vq* as voltage commands to update the output values. The feedback control in the current control unit 42 is implemented, for example, by PI control. The d-axis voltage command value Vd* and the q-axis voltage command value Vq*, which are output from the current control unit 42, are input into the two-phase / three-phase conversion unit 43, whereby three-phase voltage command values ​​Vu*, Vv* and Vw* are calculated and output on the basis of the corrected magnetic pole position signal 50 (θ').

[0032] The PWM gate control signal generation unit (control signal generation unit) 45 compares the three-phase voltage command values ​​Vu*, Vv* and Vw* with a carrier wave (not shown) to generate the six gate signals 32, which are PWM signals, and outputs the gate signals to the gate drive circuit 35.

[0033] The magnetic pole position correction unit 46 is a characteristic section of the present invention. The magnetic pole position correction unit 46 calculates an offset error Δθ caused by a mounting error of the magnetic pole position sensor 8. The calculated offset error Δθ is added to the magnetic pole position signal 9 (θ) detected by the magnetic pole position sensor 8 and is used to generate the three-phase voltage command value as a corrected magnetic pole position signal 50 (θ').

[0034] Fig. 3 is a special procedure for calculating offset errors.

[0035] First, it is determined whether a requirement for angle error correction exists from the main controller. As an embodiment, for example, the angle error correction is performed before product shipment, after the magnetic pole position sensor 8 is mounted on the motor 2. Furthermore, the processing can be performed when the inverter 1 or the motor 2 is replaced, or when a decrease in the output or torque of the motor 2 is detected. The main controller mounted on the vehicle issues the angle error correction request when, for example, information about the rotational speed of the motor 2 and the like indicates that the angle error correction can be performed according to the present invention.

[0036] When the angle error correction request is executed by the main controller, a three-phase short circuit is initiated. The three-phase short circuit is a state in which switching elements 14, 16, and 18, forming an upper branch of inverter 1, are simultaneously switched on and switching elements 15, 17, and 19, forming a lower branch, are simultaneously switched off, or a state in which switching elements 14, 16, and 18, forming the upper branch, are simultaneously switched off and switching elements 15, 17, and 19, forming the lower branch, are simultaneously switched on.

[0037] After the three-phase short circuit is initiated, processing is ready until a predetermined threshold time expires. The reason for this is explained with reference to... Fig. 4 described. Fig. Figure 4 represents current waveforms of the respective phases from U to W when the three-phase short circuit is initiated at a specific time during the rotation of motor 2. As in Fig. As shown in Figure 4, the current waveforms vary in a transitional state immediately after the start of the three-phase short circuit. In this state, the d-axis and q-axis currents vibrate, making it difficult to accurately perform the magnetic pole position error correction. Consequently, the magnetic pole position correction operation is performed after the transition to a steady state, where the variations of each phase current are calmed.

[0038] The threshold time until the transition to steady state can be calculated from the properties of motor 2. The time constants of the d-axis inductance Ld, the q-axis inductance Lq, and a stator resistance R in the motor can be expressed by 2Ld · Lq / {R · (Ld + Lq)}.

[0039] It should be noted that in the present embodiment, the determination up to the transition to the steady state is carried out by comparison with the threshold time, but the determination can be carried out by another method, for example, a case in which the amplitude of each phase current falls within a predetermined range of values, a case in which the vibration of the d-axis current id and the q-axis current iq falls below a predetermined threshold, or the like.

[0040] In Fig. 3. If a three-phase short-circuit time exceeds the threshold, the magnetic pole position correction operation is executed. Details of the magnetic pole position correction operation will be described later with reference to Fig. 5 described. After completion of the magnetic pole position correction operation, the three-phase short circuit is terminated and the processing returns to a normal PWM mode.

[0041] Fig. Figure 5 is a block diagram illustrating a configuration of the magnetic pole position correction unit 46 in the present embodiment. A procedure for the magnetic pole position correction operation in Fig. 3 refers to the block diagram of Fig. 5 described.

[0042] As in Fig. As described in section 3, the magnetic pole position correction operation is performed at the time of the three-phase short circuit. First, a dq conversion is performed on each phase current detected at the time of the three-phase short circuit to calculate an actual current phase βact. The actual current phase βact at the time of the three-phase short circuit is expressed by formula (2). The current phase has the q-axis at 0 degrees as its reference, and its counterclockwise direction is positive. βact=tan−1(−id / iq)

[0043] Furthermore, the calculation unit 55 calculates an ideal current phase βideal based on the rotational speed (angular velocity ω) of motor 2 and a stator temperature Ts. The ideal current phase βideal at the time of the three-phase short circuit is expressed by formula (3). Here, ω is an electric angular velocity [rad / s], Lq is the q-axis inductance [H], and R is the stator resistance [Ω]. βideal=tan−1(−ω⋅Lq / R)

[0044] Here, the ideal current phase βideal is a current phase in a case where there is no error in the magnetic pole position. The q-axis inductance Lq changes depending on the q-axis current, and the stator resistance R changes depending on the stator temperature. Meanwhile, the q-axis current hardly changes as long as it is equal to or greater than a predetermined rotational speed (for example, 1000 rpm). -1), and consequently the q-axis current can be considered essentially constant. However, the stator resistance R depends on the temperature, and a resistance value also increases as the temperature increases. Therefore, the correction operation in the present invention is performed using the stator temperature Ts.

[0045] Fig. Figure 6 is a graph that illustrates the relationship between rotational speed and an ideal current phase. The current phase on the vertical axis is given by a numerical value when the q-axis is set to 0 degrees as a reference. As shown in Fig. As shown in Figure 6, the variation due to temperature is large in a range where the rotational speed is relatively low, and the error is large in a case where the temperature is not corrected.

[0046] After the ideal current phase βideal is calculated taking into account the influence of the stator temperature Ts, as described above, a difference from the actual current phase βact is obtained. This difference corresponds to the offset error Δθ of the magnetic pole position sensor 8.

[0047] As described above, the power conversion device according to the present embodiment comprises: the inverter, which converts the DC voltage into the AC voltage and drives the synchronous motor; and the magnetic pole position correction unit 46, which corrects an error in the rotor magnetic pole position from the magnetic pole position sensor 8 of the motor 2. The magnetic pole position correction unit 46 comprises an actual current phase calculation unit, which calculates the current phase βact from the current when three-phase lines are short-circuited during rotation of the motor 2, and an ideal current phase calculation unit, which calculates the ideal current phase βideal based on the rotational speed of the rotor and the temperature of the stator. The magnetic pole position is then corrected by the difference Δθ between the outputs of the actual current phase calculation unit and the ideal current phase calculation unit.

[0048] Consequently, the magnetic pole position can be obtained with high accuracy even at low rotational speeds. In other words, the offset error can be corrected by a simple method according to the present invention, thus simplifying tolerance management during the assembly of the magnetic pole position sensor. Furthermore, in the present embodiment, the current phase is obtained directly from the current value at the rotational coordinate during magnetic pole position error correction. Consequently, the correction can be performed with a simpler configuration compared to a method for calculating an ideal current as three-phase currents at the fixed coordinates. Additionally, the offset angle is obtained from the three-phase short-circuit currents, and consequently, it is possible to perform the correction with high accuracy without being affected by errors caused by dead time due to switching.

[0049] It should be noted that the present embodiment is configured such that a very accurate offset correction can be performed with fewer parameters using the stator temperature Ts to calculate the ideal current phase. However, the use of other parameters is not excluded. For example, the ideal current phase can be calculated taking into account not only the stator temperature but also the influence of a rotor temperature.

[0050] Next, modifications to the calculation unit for the ideal current phase will be discussed. Fig. 5 described. A modification in Fig. Figure 7 is an example using a one-dimensional map to calculate the stator resistance R from the stator temperature Ts. The stator resistance R obtained from the one-dimensional map, the q-axis inductance, which is considered a constant value, and the rotor speed (rotational speed ω) are used to calculate βideal = tan -1 to calculate (-ω · Lq / R). Furthermore, in a modification of Fig. 8. A two-dimensional map or table for the rotational speed and stator temperature is stored, and the ideal current phase βideal is calculated with reference to the two-dimensional map. The two-dimensional map is created in advance by changing the stator temperature and rotational speed and capturing the current phases with an actual machine. Second embodiment

[0051] In the present embodiment, a noise reduction unit is also included in addition to the configuration of the Fig. The magnetic pole position correction unit 46 shown in Figure 5 is provided. The other points are the same as those of the first embodiment. Ideally, three-phase short-circuit currents are symmetrical three-phase sinusoidal currents, as in the steady state of Fig. Figure 4 illustrates this. In practice, however, the current becomes a current in which noise is superimposed due to variations in the impedance of each phase, a spatial harmonic of an induced voltage, a current sensor error, or the like. Furthermore, the dq-axis current itself vibrates even when a completely steady state is not established. Therefore, the noise removal unit in the present embodiment is provided to eliminate this influence, and a more accurate angle correction can be implemented. The noise removal unit removes noise by means of a first-order delay filter or an averaging process.

[0052] Fig. Figure 9 is a block diagram in a case where a noise removal unit 56 is provided in a subsequent stage of a deviation calculation unit that maintains a deviation between the actual current phase βact and the ideal current phase βideal. Alternatively, noise removal units 57 and 58 can be provided in subsequent stages of an actual phase calculation unit and an ideal current phase calculation unit, respectively, as shown in Figure 9. Fig. 10 shown. Reference symbol list 1 inverter 2 Engine 3 batteries 4 Controller 5 to 7 three-phase winding 8 Magnetic pole position sensor 9 Magnetic pole position signal 10 Current sensor 11 U-phase current sensor signal 12V phase current sensor signal 13 W phase current sensor signal 14 to 19 switching element 20 to 25 IGBTs 26 to 31 Gate control signal 32 Gate signal 33 High-potential DC wiring 34 Low-potential DC wiring 35 Gate drive circuit 36 Voltage sensor 37 DC voltage sensor signal 38 Smoothing capacitor 39 Torque command 40 Current command calculation unit 41 Three-phase / two-phase implementation unit 42 Power control unit 43 Two-phase / three-phase conversion unit 44 Speed ​​calculation unit 45 PWM gate control signal generation unit (control signal generation unit) 46 Magnetic pole position correction unit 47 Deviation Calculator 48 deviation calculators 49 Adders 50 corrected magnetic pole position signal (θ') 55 Calculation unit for the ideal current phase 56 to 58 noise reduction units 65 Motor drive device (motor control device) QUOTES INCLUDED IN THE DESCRIPTION

[0000] This list of documents cited by the applicant was automatically generated and is included solely for the reader's convenience. The list is not part of the German patent or utility model application. The DPMA accepts no liability for any errors or omissions. Cited patent literature

[0000] JP 2014050122 A

[0003] JP 2017212783 A

[0003]

Claims

[1] Power conversion device comprising: an inverter that converts a direct current into an alternating current and drives a synchronous motor; and a magnetic pole position correction unit that corrects an error in the magnetic pole position of a rotor from a rotation angle sensor of the synchronous motor, wherein the magnetic pole position correction unit a calculation unit for the actual current phase, which calculates a current phase from a current when three-phase lines are short-circuited during the rotation of the synchronous motor, and a calculation unit for the ideal current phase, which calculates an ideal current phase based on a rotor speed and a stator temperature, and The magnetic pole position is corrected from a difference between the outputs of the calculation unit for the actual current phase and the calculation unit for the ideal current phase. [2] Power conversion device according to claim 1, wherein the magnetic pole position correction unit begins to operate after three-phase currents transition to a steady state following the short circuit under the three-phase lines. [3] Power conversion device according to claim 1, further comprising a noise removal unit configured for a difference between the outputs of the computation unit for the actual current phase and the computation unit for the ideal current phase or the calculated actual phase and ideal phase. [4] Power conversion device according to claim 1, wherein the calculation unit for the ideal current phase calculates an ideal current phase based on the rotational speed of the rotor, the temperature of the stator and a temperature of the rotor of the synchronous motor.