Radar detection with doppler-based delay correction

Abstract

Embodiments include methods for a radar receiver. Such methods include receiving a composite signal corresponding to a transmitted radar signal. The received composite signal comprises a plurality (M) of signal streams corresponding to a respective plurality (M) of signal pulses in the transmitted radar signal. Each signal stream includes delayed samples corresponding to a plurality of propagation delays of the corresponding signal pulse. Such methods include processing the received composite signal to generate a delay-Doppler plane representation, detecting an object at a first delay value and at a first Doppler value in the delay-Doppler plane representation, and correcting the first delay value based on a delay correction that is a function of the following: the first Doppler value, and a sequence of values used to generate the plurality of signal pulses in the transmitted radar signal. Other embodiments include radar receivers and radar transceivers configured to perform such methods.

Description

[0001] RADAR DETECTION WITH DOPPLER-BASED DELAY CORRECTION

[0002] TECHNICAL FIELD

[0003] The present disclosure relates generally to detection of objects using reflected radar signals, and more specifically to techniques for correcting an estimated delay of reflections from an object based on an estimated Doppler shift for the object.

[0004] BACKGROUND

[0005] Radar is a detection system that uses radio waves to determine the distance, angle, and / or radial velocity of objects relative to a transmitter of the radio waves. Radar can be used to detect various stationary and moving objects such as aircraft, ships, spacecraft, missiles, cars and other motor vehicles, weather formations, and terrain. A radar system in general consists of a transmitter that generates the radar signal, one or more transmitting antennas, one or more receiving antennas (which may or may not be the transmitting antennas), and a receiver that processes the signals from the receiving antennas to detect and / or determine properties of objects. In particular, the transmitted signal (which typically consists of multiple pulses) reflect off objects in its paths and returns to the receiver, which can detect properties of the objects based on transmitreceive round-trip delay and other characteristics of the received signals. For example, the delay between a transmitted pulse and the received echo (referred to as “round trip time” or RTT) determines range to the target.

[0006] Figure 1 illustrates an exemplary transmitted radar signal comprising a series of M pulses, indexed 0 to M-l . In the time domain, the M pulses are generally uniform in shape and are spaced apart by a period T (e.g., between the starts of successive pulses). The signal shown in Figure 1, reflects from an object (target), with the reflected signal (“echo”) being received by a radar receiver that may be (but is not necessarily) co-located with the transmitter. The pulsed arrangement of the signal in Figure 1 facilitates detection of range or distance to the target and target velocity.

[0007] The frequency bandwidth of each pulse determines the resolution of the range determination. A moving object introduces a Doppler shift in the echo, which manifests itself as a phase change between successive received pulses. The amount of phase change can be used to determine target velocity. The pulse period T establishes the maximum velocity that can be detected without ambiguity, and the number of pulses (M) establishes the resolution of the velocity determination.

[0008] Figure 2 illustrates exemplary processing of echoes of the radar signal shown in Figure 1 by a radar receiver. The receiver correlates each received (echo) pulse with a copy of the transmitted pulse over a range of delays using a time-domain matched filter. As shown in Figure i 2, the receiver uses M matched filters corresponding to the M pulses. The 2D-representation of the signals after matched filtering is referred to as “delay -time plane”. The receiver then applies multiple fast Fourier transforms (FFTs) to the matched filter outputs, specifically each FFT across M samples from the matched filter outputs corresponding to a single delay value or hypothesis. In other words, each FFT corresponds to a different delay. The 2D representation of the FFT outputs is referred to as “delay-Doppler plane”.

[0009] A magnitude peak in the delay-Doppler plane corresponds to a detected object, with the delay and Doppler coordinates of the peak corresponding to the detected object’s distance and velocity, respectively. For example, echoes from a moving (or non-stationary) object will generate a peak at some non-zero Doppler while echoes from a stationary object will generate a peak at zero Doppler. In the example shown in Figure 2, three objects are identified in the delay- Doppler plane - one stationary and two moving.

[0010] SUMMARY

[0011] Even so, there are some problems, issues, and / or difficulties with detecting moving objects using radar. For example, when a moving object introduces a Doppler shift in a reflected radar pulse, the output of the receiver’s matched filer for that pulse is distorted. This distortion can be used to detect movement and to discriminate between different objects depending on the respective speeds of the objects. However, this distortion may also affect the receiver’s measurements of range or distance for these object. For example, even if the receiver detects a moving object, its measurement of range or distance of that object may be incorrect due to the radar pulse distortion.

[0012] An object of embodiments of the present disclosure is to improve radar-based detection of objects (e.g., moving objects), such as by providing, enabling, and / or facilitating solutions to overcome exemplary problems summarized above and described in more detail below.

[0013] Some embodiments include methods (e.g., procedures) for a radar receiver.

[0014] These exemplary methods include receiving a composite signal corresponding to a transmitted radar signal. The composite signal comprises a plurality (M) of signal streams corresponding to a respective plurality (M) of signal pulses in the transmitted radar signal. Each signal stream includes delayed samples corresponding to a plurality of propagation delays of the corresponding signal pulse. These exemplary methods also include processing the received composite signal to generate a delay-Doppler plane representation, and detecting an object at a first delay value and a first Doppler value in the delay-Doppler plane representation. These exemplary methods also include correcting the first delay value based on a delay correction that is a function of the following: the first Doppler value, and a sequence of values used to generate the plurality (M) of signal pulses in the transmitted radar signal.

[0015] In some embodiments, for a first sequence of values used to generate the plurality (M) of signal pulses in the transmitted radar signal, the delay correction increases approximately linearly in relation to one of the following: velocity of an object that reflected the transmitted radar signal, or Doppler shift in the received signal due to the velocity of the object that reflected the transmitted radar signal.

[0016] In some embodiments, for a second sequence of values used to generate the plurality (M) of signal pulses in the transmitted radar signal, the delay correction has one or more approximate step changes in relation to one of the following: velocity of an object that reflected the transmitted radar signal, or Doppler shift in the received signal due to the velocity of the object that reflected the transmitted radar signal.

[0017] In some embodiments, each signal pulse of the plurality (M) of signal pulses in the transmitted radar signal includes a main pulse, and the sequence of values is used to generate the main pulse. Moreover, correcting the first delay value is based on a function of the sequence of values used to generate the main pulse of each signal pulse.

[0018] In some of these embodiments, the sequence of values is a frequency domain sequence, and the main pulse is generated based on an Inverse Discrete Fourier Transform (IDFT) of the frequency domain sequence. In some variants of these embodiments, the frequency domain sequence is one of the following: a quadrature phase shift keying (QPSK) sequence, a binary phase shift keying (BPSK) sequence, a Zadoff-Chu sequence, or any cyclic shift of a Zadoff-Chu sequence. For example, when the frequency-domain sequence is a Zadoff-Chu sequence (or a cyclic shift thereof), the delay correction is a function of a root index, u, of the Zadoff-Chu sequence.

[0019] In some of these embodiments, each signal pulse of the plurality (M) of signal pulses in the transmitted signal also includes a cyclic extension (CE) of the main pulse. In some variants of these embodiments, processing the received composite signal to generate the delay -Doppler plane representation includes the following operations:

[0020] • performing a cyclic correlation of each signal stream with a replica of the main pulse of the corresponding signal pulse, thereby generating a delay-time plane representation comprising a plurality (M) of correlation streams including samples corresponding to the plurality of propagation delays;

[0021] • performing a plurality of discrete Fourier transforms (DFTs) on the correlation streams to generate the delay -Doppler plane representation, wherein the respective DFTs are performed on samples from all correlation streams corresponding to respective propagation delays.

[0022] Other embodiments and variants of the exemplary methods summarized above are described herein. Other embodiments include radar receivers and radar transceivers configured to perform operations corresponding to any of the exemplary methods described herein. Other embodiments include non-transitory, computer-readable media storing program instructions that, when executed by processing circuitry, configure such radar receivers and radar transceivers to perform operations corresponding to any of the exemplary methods described herein.

[0023] These and other embodiments described herein may mitigate and / or remove growth of sidelobes and / or the loss in peak power in the radar receiver’s matched filter output due to increasing Doppler shift. Also, embodiments may mitigate and / or remove potential peak delay shifts that can vary with Doppler shift. More generally, embodiments may significantly improve fidelity of the radar processing for range estimation in the presence of Doppler shift and phase error with only minor impacts to receiver complexity.

[0024] These and other objects, features, and advantages of embodiments of the present disclosure will become apparent upon reading the following Detailed Description in view of the Drawings briefly described below.

[0025] BRIEF DESCRIPTION OF THE DRAWINGS

[0026] Figure 1 illustrates an exemplary transmitted radar signal comprising a series of M pulses.

[0027] Figure 2 illustrates exemplary processing of echoes of the radar signal shown in Figure 1 by a radar receiver.

[0028] Figure 3 illustrates monostatic sensing, where the radar transmitter and radar receiver are collocated.

[0029] Figure 4 illustrates bistatic sensing, where the radar transmitter and radar receiver are geographically separated.

[0030] Figure 5 shows an ambiguity function when quadrature phase shift keyed (QPSK) pulses are used to modulate a CP-ODFM radar signal in the frequency domain.

[0031] Figure 6 further illustrates bistatic sensing involving a transmitter, receiver, and target.

[0032] Figure 7 illustrates an exemplary transmitted radar signal comprising a series of M pulses with cyclic extension (CE).

[0033] Figure 8 illustrates exemplary radar receiver processing of a received composite signal comprising echoes of the M-pulse radar signal shown in Figure 7.

[0034] Figure 9 illustrates operation of an exemplary radar receiver including a cyclic correlator. Figure 10 shows an example of delay error and matched filter output loss for QPSK- modulated radar pulses subjected to Doppler shift.

[0035] Figure 11 shows an exemplary radar transmitter.

[0036] Figure 12 shows an exemplary radar receiver, according to some embodiments of the present disclosure.

[0037] Figure 13 shows an example of delay error and matched filter output loss for Zadoff-Chu- modulated radar pulses subjected to Doppler shift.

[0038] Figure 14 shows a flow diagram of an exemplary method (e.g., procedure) for a radar receiver, according to various embodiments of the present disclosure.

[0039] Figure 15 shows an exemplary device configurable as a radar receiver and / or radar transmitter, according to various embodiments of the present disclosure.

[0040] DETAILED DESCRIPTION

[0041] Embodiments briefly summarized above will now be described more fully with reference to the accompanying drawings. These descriptions are provided by way of example to explain the subject matter to those skilled in the art and should not be construed as limiting the scope of the subject matter to only the embodiments described herein. More specifically, examples are provided below that illustrate the operation of various embodiments according to the advantages discussed above.

[0042] In general, all terms used herein are to be interpreted according to their ordinary meaning in the relevant technical field, unless a different meaning is clearly given and / or is implied from the context in which it is used. All references to a / an / the element, apparatus, component, means, step, etc. are to be interpreted openly as referring to at least one instance of the element, apparatus, component, means, step, etc., unless explicitly stated otherwise. The operations of any methods and / or procedures disclosed herein do not have to be performed in the exact order disclosed, unless an operation is explicitly described as following or preceding another operation and / or where it is implicit that an operation must follow or precede another operation. Any feature of any embodiment disclosed herein can apply to any other disclosed embodiment, as appropriate. Likewise, any advantage of any embodiment described herein can apply to any other disclosed embodiment, as appropriate.

[0043] Currently the fifth generation (5G) of cellular systems, also referred to as New Radio (NR), is being standardized within the Third-Generation Partnership Project (3GPP). NR is developed for maximum flexibility to support multiple and substantially different use cases. These include enhanced mobile broadband (eMBB), machine type communications (MTC), ultra-reliable low latency communications (URLLC), side-link device-to-device (D2D), and several other use cases.

[0044] 5G / NR technology shares many similarities with fourth-generation LTE. For example, NR uses Cyclic Prefix Orthogonal Frequency Division Multiplexing (CP-OFDM) in the DL and CP- OFDM or DFT-spread OFDM (DFT-S-OFDM) in the UL. As another example, in the time domain, NR DL and UL physical resources are organized into equal-sized 1-ms subframes. A subframe is further divided into multiple slots of equal duration, with each slot including multiple OFDM-based symbols. An NR slot can include 14 OFDM symbols for normal cyclic prefix (CP) and 12 symbols for extended CP. A resource block (RB) consists of a group of 12 contiguous OFDM subcarriers for a duration of a 12- or 14-symbol slot. A resource element (RE) corresponds to one OFDM subcarrier during one OFDM symbol interval.

[0045] NR supports various OFDM subcarrier spacing (SCS) values A = (15 X 2R) kHz, where p G (0,1, 2, 3, 4) are referred to as “numerologies.” Numerology p = 0 ( / .< ., A = 15kHz) provides the basic (or reference) SCS that is also used in LTE. The symbol duration, CP duration, and slot duration are inversely related to SCS or numerology. For example, there is one (1-ms) slot per subframe for A = 15kHz, two 0.5-ms slots per subframe for A = 30kHz, etc. In addition, the maximum carrier bandwidth is related to numerology according to 2 * 50MHz. Different DL and UL numerologies can be configured by the network.

[0046] Joint Communication and Sensing (JCAS) is an emerging technology standardized by IEEE 802.11, discussed in Third Generation Partnership Project (3GPP), and expected to be a fundamental element of sixth generation (6G) communication networks. In 3GPP context, JCAS is often referred to as Integrated Sensing and Communicatoins (ISAC). Sensing in an ISAC or JCAS system may be done using radar signal or Channel State Information (CSI) reference signals (RS) transmitted by a radio access network (RAN) node (e.g., base station).

[0047] For radar-based sensing, as briefly discussed above, one or more pulses are transmitted and their received echoes are used for range and Doppler measurements. Even though the use of radar in line-of-sight (LOS) propagation environments is highly effective, its deployment in environments with strong non-LOS (NLOS) characteristics becomes challenging.

[0048] In contrast, CSLbased sensing has the potential to overcome some of these limitations by undertaking measurements of the underlying channel (or environment) between a transmitter and a receiver. In this approach, inference regarding the presence of a non-device target and / or any other quantity of interest is done through the change of CSI at multiple time instances or with the appropriate processing of the CSI of a single time instance. This form of sensing may be referred to as device-free sensing (DFS), with the word “free” indicating that the target is the physical environment rather than any particular device that may (or may not) be present in it. For ISAC in 6G networks, the goal is to have RAN nodes (e.g., base stations, gNBs) sense the physical environment using either signals also used for communication or dedicated sensing signals, thereby obtaining information such as location, shape, and speed of the objects in the environment. Some applications of ISAC sensing include traffic monitoring, drone detection, gesture / motion detection, presence detection of objects or persons, vital sign detection, environment mapping, particle / pollution detection, etc.

[0049] Two types of radar-based sensing are monostatic and bistatic. Figure 3 illustrates monostatic sensing, where the radar transmitter (Tx) and radar receiver (Rx) are collocated. In this case, the radar Tx and radar Rx are collocated in a RAN node. The radar Tx sends a waveform that is reflected or echoed by the environment and any targets present. The reflections are received and processed by the radar Rx to obtain information about the environment and / or targets. In some instances, the receiver processing may be done by a network element other than the RAN node.

[0050] A radar pulse that spans one or multiple OFDM symbols would allow an easy integration of radar into a 5G / NR network. For example, for NR 30 kHz SCS, the OFDM symbol duration (including CP) is approximately 36 / s. A radar pulse spanning a single OFDM symbol duration would have the same length. A round trip time of 36 / s corresponds to one-way distance of 5.4 km. For objects closer than 5.4 km, the echo would arrive before the radar Tx has finished transmitting that radar pulse.

[0051] Thus, the radar Rx must be capable of receiving the reflected signal in the presence of strong self-interference due to the transmitted signal leaking into the radar Rx. A receiver capable of this is often called a full-duplex (FD) capable receiver. This capability places high requirements (e.g., for self-interference cancellation) on the receiver and possibly other requirements (e.g., linearity) on the transmitter, especially when high transmit power is used.

[0052] In a CP-OFDM-based radar system, the pulses may be generated by applying a sequence in the frequency domain to the allocated subcarriers of the OFDM modulator and by adding a CP. In principle every sequence that has constant magnitude would result in a time-domain signal with a perfectly periodic autocorrelation function (ACF) and could be used as radar sequence.

[0053] Due to technical challenges and complexity of FD capability in monostatic sensing, bistatic sensing has been considered as an alternative option. Figure 4 illustrates bistatic sensing, where the radar Tx (in BS1) and radar Rx (in BS2) are geographically separated but operating in the same time-frequency radio resources.

[0054] In radar systems, distance measurements are based on measuring the Time of Flight (ToF) of the signal from radar Tx via the target to radar Rx. All possible target positions for a measured ToF value are located on an ellipsis (in 2D, or ellipsoid in 3D) with focal points given by radar Tx and radar Rx locations. To determine a target position, radar Tx and radar Rx locations, ToF, and either Angle of Departure (AoD, from Tx) or Angle of Arrival (AoA, at Rx) must be known. Specifically, AoD or AoA is needed to determine the target location on the ellipse given by radar Tx and radar Rx locations and ToF. Also, to accurately determine ToF, radar Tx and radar Rx need to be time synchronized, which is one of the main challenges for bi- and multi-static radar.

[0055] A further extension is multi-static sensing, where one or more geographically separated radar Tx’s transmit signals whose echoes or reflections are received by one or more geographically separated radar Rx’s. For example, with one radar Tx and three radar Rx’s, three ellipses can be determined and the target is located where the three ellipses intersect. To obtain a 3D position of a target, this method requires ToF estimates from at least three receivers but removes the need for AoD / AoA information, which nevertheless can be used to improve performance.

[0056] For radar processing using CP-OFDM signals, the receiver typically has a frequencydomain matched filter. For example, the receiver’s frequency-domain matched filter can be a complex conjugate of the discrete Fourier transform (DFT, e.g., FFT) of the transmitted pulse sequence, which the receiver multiplies element-by-element with a FFT of the received signal. The estimation of ToF, AoA, and AoD is then based on evaluating the matched filter output for different ToF, AoA, and AoD. As illustrated in Figure 2, this evaluation may be performed in a delay -Doppler plane, which is created by applying further FFTs to the matched filter outputs.

[0057] Typically the absolute value of the time-domain output of the matched filter is optimized over ToF, AoA and AoD and if the maximum absolute value of the time-domain output of the matched filter is larger than a threshold it is concluded that there is a likelihood that a target is found in a direction given by AoA and / or AoD and that the radar pulse has travelled a distance proportional to ToF from radar Tx to radar Rx.

[0058] If the physical environment of interest includes moving objects, then the radar signal may be affected by Doppler shifts. In other words, the radar signal is not only delayed due to propagation but also distorted in shape due to a frequency shift. As such, the radar Rx’s matched filter will be mismatched, causing reduced detection performance due to lower peaks and higher sidelobes.

[0059] Figure 5 shows an ambiguity function when quadrature phase shift keyed (QPSK) pulses are used to modulate a CP-ODFM radar signal in the frequency domain. With no Doppler shift the peak of the received pulse is high and the side lobes are low. With high Doppler shift the peak is reduced and the side lobes are much higher. In such case, the pulse needs to be transmitted with higher energy to achieve the same detection performance as with no Doppler shift.

[0060] In a monostatic radar the distance between radar node and target is determined based on the ToF, i.e., / ? = c0T / 2 with T the time duration between pulse transmission and reception and c0the speed of light. Two targets that are further apart than the range resolution Rr= c0 / 2BW) can still be distinguished as separate objects. In other words, the range resolution is inverse proportional to the radar signal bandwidth BW .

[0061] Figure 6 further illustrates bistatic sensing involving a transmitter (TX), receiver (RX), and target. The TX emits a sensing signal, x(t), which propagates through the physical environment and is collected by the RX. To increase the probability of detection of targets, it is desirable that the Tx transmits the signal in a wide angular ranges, preferably using an isotropic antenna. Assuming that there is only one target and that TX and RX are synchronized in time, the RX can measure ToF at the output of its matched filter and determine from measure ToF the bistatic range, i.e., distance between TX and target plus distance between target and RX.

[0062] As mentioned above, without knowledge of TX AoD and / or RX AoA for the signal, the bistatic range only indicates that the target is placed somewhere on the surface of an ellipsoid with focal points at TX and RX. If the AoD and / or AoA are available, a more accurate estimate of a position of the target on the ellipsoid can be extracted. In any event, the actual distance between the TX and RX is assumed to be known.

[0063] In some cases, RX AoA can be used to obtain a more accurate estimate of a position of the target on the ellipsoid. Assume the TX uses an isotropic antenna while the RX is equipped with an antenna array capable of forming directional beams to determine AoA (or equipped with other functionality to determine AoA). Furthermore, in this arrangement, the operating principle for estimating the target-to-RX distance Dr, the TX-to-Target distance Dt, and the position of the target r rely on the measurement of ToF of the propagation path TX-Target-RX and the knowledge of AoA in the RX, ipr. From the measured ToF, the distance of flight (DoF), S =

[0064] Dt+ Dr, can be obtained. Also, the TX-to-Target distance can be determined by: where, Dbis the TX-to-RX distance. Likewise, target-to-RX distance Drcan be determined by: while the position of the target is expressed as:

[0065] Here, 6 and (p are the zenith and azimuth RX AoAs, respectively. Note that RX AoA iprcan be expressed in terms of angles 0 and <p, TX-RX distance Db, and 3D TX position t = [t1(t2, t3]Tas: tj sin 0 cos (p + t2sin 0 sin (p + t3cos 0 cos ipr= - ,

[0066] Dbj (sin 0 cos )2+ (sin 0 sin )2+ (cos 0)2 where the zenith and azimuth RX AoAs 6 and (p, and consequently RX AoA ir, need to be estimated using the appropriate procedure.

[0067] As discussed above, velocity of a target can be estimated based on measured Doppler- induced frequency shift on a radar signal reflected from the target. Measuring Doppler shift can also aid in target classification. For example, a pedestrian has a specific Doppler profile due to movements of legs and arms and is typically moving slower than surrounding vehicles. Measuring Doppler shift can also aid in differentiating between stationary clutter and moving targets.

[0068] Figure 7 illustrates an exemplary transmitted radar signal comprising a series of M pulses with cyclic extension (CE), which may be referred to alternately as CP. Each pulse CE is identical and preferably long enough to capture echoes from objects at a maximum detection range (i.e., echoes with highest expected RTT). A CE that is too short to capture echoes of the transmitted signal will leads to performance degradation. The pulses are periodic with a period, T.

[0069] The Doppler-induced frequency shift on the received signal reflected by a moving target can be determined by measuring the phase difference between received pulses of a pulse train. For a stationary target and perfect phase and frequency synchronization between stationary TX / RX, the TX-target-RX distance is constant and each received pulse has the same phase shift relative to its transmitted copy, i.e., there are no phase differences between successive received pulses. In contrast, the TX-target-RX distance for a moving target changes over time, which manifests itself as a phase difference (or change) between successive received pulses and can be used to determine Doppler shift. For example, the phase change can be determined as A<p = fd■ T where T is the pulse period and fdis Doppler shift induced by the moving target.

[0070] Doppler shift depends on the velocity of the target as well as the target movement direction relative to TX and RX. For example, a target moving with velocity v towards a monostatic radar leads to a Doppler shift of fd= 2v / c / c0with c0andcspeed of light and carrier frequency, respectively. In a bistatic and multi-static radar, Doppler shift depends on geometry between TX, Rx, and target but is typically less than for monostatic radar.

[0071] Figure 8 illustrates exemplary radar receiver processing of a received composite signal comprising echoes of the M-pulse radar signal shown in Figure 7. The receiver includes M cyclic correlators, each of which is applied to a signal stream with samples covering a range of delays of the corresponding pulse. The M cyclic correlators may be implemented as separate hardware units, a common hardware unit that performs all cyclic correlations, or a combination thereof.

[0072] The output of a cyclic correlation of a signal stream with a transmitted pulse (without CE) corresponds to a cyclic convolution between a periodic autocorrelation function (ACF) of the transmitted pulse (without CE) and the impulse response of the channel between transmitter and receiver (i.e., response of the objects). The output of a cyclic correlation may be referred to as a “correlation stream.” As explained in more detail below, pulses may be constructed to have very desirable (e.g., only one non-zero value) and identical periodic ACFs, even pulses that are different.

[0073] In some embodiments, the cyclic correlators may be realized as frequency-domain matched filters. Figure 9 illustrates operation of an exemplary cyclic correlator 910. In this arrangement, each signal stream (per pulse) is converted to a frequency-domain representation using an FFT. This representation is then element-by-element multiplied by the complex conjugate of a frequency-domain representation (e.g., FFT) of the transmitted pulse (without cyclic extension). The product of these terms is then converted back into time-domain using an inverse FFT (IFFT). The size of the FFT is determined by the pulse length (without cyclic extension) and the sampling rate, while the IFFT size may be selected based on the desired range resolution. The output of the cyclic correlators is the 2D delay-time plane.

[0074] Subsequently, multiple FFTs are applied to the cyclic correlator outputs, with each FFT being applied to outputs from all M cyclic correlators corresponding to the same delay value or hypothesis. The output of the FFTs (i.e., the M correlation streams) is the 2D delay-Doppler plane. In the example shown in Figure 8, three targets are detected in the delay-Doppler plane - one stationary and two moving.

[0075] As mentioned above, the output of a cyclic correlation of a received signal with a transmitted pulse (without CE) corresponds to a cyclic convolution between a periodic ACF of the transmitted pulse (without CE) and the impulse response of the objects. For stationary objects, the impulse response is constant and the periodic ACF does not vary across the M pulses. Thus, the output of each cyclic correlator is the same for a stationary object and each FFT input (corresponding to a particular delay value or hypothesis) consists of identical (or constant) values for all M pulses. The FFT then generates a corresponding output having a large value at zero Doppler and small or zero values at non-zero Dopplers, which reduce or eliminate masking of responses from moving objects at any of the non-zero Dopplers.

[0076] The pulses in Figure 7 would be typically generated by applying a sequence in frequencydomain to the allocated subcarriers of the OFDM modulator and adding a CP. In principle every sequence that has constant magnitude would result in a time-domain signal with a perfectly periodic ACF and could be used as radar sequence. One type of sequence with constant magnitude is Zadoff-Chu, which is defined as: where 0<u<Nzc and greatest common denominator (gcd) of u and Nzcis 1. All complex values in the Zadoff-Chu sequence have the same magnitude (i.e., distance from origin in complex plane). Additionally, the autocorrelation of a DFT of a Zadoff-Chu sequence with a DFT of a cyclically shifted version is non-zero at one offset that corresponds to the cyclic shift and zero at other offsets. Furthermore, if Nzc is a prime number, the DFT of a Zadoff-Chu sequence is a complex-conjugated and time-scaled version of another Zadoff-Chu sequence. IfNzc is not a prime number, the Zadoff-Chu sequence still has the desirable autocorrel tion properties but its DFT may not necessarily have constant amplitude.

[0077] Even so, there are some problems, issues, and / or difficulties with detecting moving objects using radar. As mentioned above, when a moving object introduces a Doppler shift in a reflected radar pulse, the output of the receiver’s matched filter for that pulse is distorted. This distortion can be used to detect movement and to discriminate between different objects depending on the respective speeds of the objects. However, this distortion may also affect the receiver’s measurements of range or distance for these objects.

[0078] For example, even if the receiver detects a moving object, its measurement of range or distance of that object may be incorrect due to the radar pulse distortion. More specifically, the distortion can shift the peak of the matched filter output and causes an error in the ToF estimation. Furthermore, as discussed above, the peak of the matched filter output is typically lower and side-lobes of the autocorrelation are higher for a Doppler-shifted signal, which can reduce sensitivity of the radar receiver.

[0079] Figure 10 shows an example of delay error and matched filter output loss of radar pulses subjected to Doppler shift, particularly for radar pulses in which pseudo-random QPSK is used to modulate a CP-ODFM signal in the frequency domain. Output power of the matched filter decreases with increased vehicular speed and delay error varies between zero and non-zero values according to vehicular speed. However, the actual impact depends on signal shape, such that different transmit signals will generate different delay errors and peak power loss versus vehicular speed. Even so, these errors and losses may cause the receiver to fail to detect a target or, if detected, estimate an incorrect range for the target.

[0080] Embodiments of the present disclosure address these and other problems, issues, and / or difficulties by a radar receiver estimating a time-of-flight (ToF) and Doppler shift for a received signal based on an output of its matched filter, and then determining a ToF compensation factor based on the shape of the transmitted signal and the estimated Doppler shift. The radar receiver then applies the ToF compensation factor to the estimated ToF to obtain a more accurate ToF estimate. In some embodiments, Zadoff-Chu sequences may be used advantageously for the transmitted signal pulses. Embodiments may provide various benefits and / or advantages. For example, embodiments may mitigate and / or remove growth of sidelobes and / or loss in peak power in receiver matched filter output due to increasing Doppler shift. Also, embodiments may mitigate and / or remove potential peak delay shifts that can vary with Doppler shift. More generally, embodiments may significantly improve fidelity of the radar processing for range estimation in the presence of Doppler shift and phase error with only minor impacts to receiver complexity.

[0081] Figure 11 shows an exemplary radar transmitter 1100, which includes processing circuitry 1110 configured to generate a time-domain radar signal (e.g., M pulses) and transmitter circuitry 1120 configured to transmit the time-domain radar signal. For example, the radar transmitter may be part of, or attached to, a RAN node.

[0082] The input sequence X(k) is specified in the frequency domain. A time-domain signal x(r) is then obtained from an inverse DFT (e.g., IFFT) on X(k), optionally performing frequencydomain windowing on X(k) prior to the IFFT. By applying the windowing in the frequencydomain, the energy in each pulse may be maintained by scaling the window function, e.g., to amplify certain values to which it is applied. Next, a cyclic extension (CE) is added to x(n) by prepending the last L samples of x(n), n=l . . .N, to create the following time-domain sequence for the mthsignal pulse,

[0083] Sm(n) = [x(N-L+l), x(N-L+2),. . ., x(N), x(l), x(2),. . ., x(N)), which is transmitted by the transmitter circuitry. These operations take place for all pulses with index m=0. . .M-l, as illustrated in Figure 7. For example, the radar transmitter in Figure 11 may generate and transmit the M-pulse signal shown in Figure 7.

[0084] Figure 12 shows an exemplary radar receiver 1200, according to some embodiments of the present disclosure. The exemplary receiver includes various functional blocks, which may be implemented by various combinations of processing circuitry and software.

[0085] The composite signal input to the receiver from the channel is rm(n), m=0...M-l, where the mth received signal stream includes a delayed, attenuated, and Doppler-shifted version of corresponding signal pulse sm(n), plus interference and noise. Put differently, the mth signal stream includes samples corresponding to delayed copies (i.e., at multiple propagation delays) of the mth signal pulse generated and transmitted by the radar transmitter.

[0086] For illustration, consider a scenario with mono-static radar and a single target moving toward the receiver with speed v m / s. In this scenario, a continuous time version of the mth received signal stream can be expressed as: where fcis the signal carrier frequency (in Hz) and c is the speed of light (in m / s). Although not shown explicitly, the receiver may include receiver circuitry (e.g., antennas, amplifiers, mixers, analog-to-digital converters, etc.) configured to receive continuous time radar signals from the channel and to convert the received signals into discrete-time samples for processing.

[0087] The larger dashed box indicates the per-pulse processing performed by the receiver, i.e., for each of pulses m=0. . .M-l . After synchronization, the receiver first removes the CE from each signal stream and then converts the result to a frequency-domain representation using an FFT. This representation is then element-by-element multiplied by the complex conjugate of the original (without CE) pulse X(k), which is known by the receiver. The product of these terms is then converted back into time-domain using an inverse FFT (IFFT), optionally applying a frequency-domain window W prior to the IFFT. The size of the FFT is determined by the pulse length (without cyclic extension) and the sampling rate, while the IFFT size may be selected based on the desired range resolution. The IFFT output for each pulse index m is the matched filter (i.e., cyclic correlator) output for that pulse, rnfm(n), n=l ...N, which may also be referred to as a “correlation stream”. The collection of FFT outputs (i.e., the correlation streams) for all pulses m=0. . .M-l constitutes the delay -time plane.

[0088] Subsequently, multiple FFTs are applied to the cyclic correlator outputs (i.e., the correlation streams), with each FFT being applied to outputs from all M cyclic correlators corresponding to the same delay value or hypothesis. The output of the FFTs (i.e., the M correlation streams) constitutes the delay -Doppler plane.

[0089] Next, the receiver estimates the position [d, D] of the peak power in the delay-Doppler plane. Although the Doppler dimension D of the peak may be sufficiently accurate, the delay dimension d of the peak may be time-shifted due to the Doppler shift, as illustrated above by examples. According to embodiments of the present disclosure, the estimated Doppler shift D is used to determine a compensation value for the estimated delay d. In other words, the receiver estimates the actual delay, d’, based on d-f(X(k), where f(X(k),D) is a function that depends on the signal pulse X(k) and the Doppler shift D. For example, the values of f(X(k),D) can be precalculated for all relevant signal pulses X(k) and all relevant Doppler values D.

[0090] Figure 10 discussed above shows an example of delay error when X(k) is a QPSK sequence, for which f(X(k), D) can be designed to compensate. This example is challenging since the delay error changes rapidly (e.g., via step change or discontinuity, as illustrated) at some speeds (or Dopplers), at which f(X(k), D) must also change rapidly. Also, the there is a power loss that may impact delay and Doppler estimate accuracy due to reduced SNR. As such, the Doppler estimate may be inaccurate in the regions of rapid change but may be sufficiently accurate in other regions to be used for compensating the delay estimate via f(X(k), D) for inaccuracies introduced by actual Doppler shifts. In some embodiments, Zadoff-Chu sequences can be used instead of random QPSK sequences. For instance, using a Zadoff-Chu sequence with, Nzc=2048< u = 1 and q = 0, the variation of f(X(k), D) is slow and there is no impact on the SNR. Figure 13 shows an example of delay error and matched filter output loss when CP-OFDM radar signals modulated by such Zadoff-Chu sequences are subjected to Doppler shift. In this example, the delay error increases approximately linearly with Doppler shift. This linear relation is a function of the particular root index, u, of the Zadoff-Chu sequence used to generate the CP-OFDM signal.

[0091] In other words, if d is the sum of actual delay (dact) and delay error (Ad), then Ad increases approximately linearly with target velocity, which is related to Doppler by carrier frequency of the radar signal and direction of the target relative the transmitter and the receiver. Thus, f(X(k), D) can be designed to also increase approximately linearly with target velocity, such that d’=d- f(X(k), D) gives an accurate estimate of the actual delay, dact. Alternately, f(X(k), D) can be designed to increase approximately linearly with measured Doppler shift D, which avoids dependency on carrier frequency of the radar signal and direction of the target relative the transmitter and the receiver.

[0092] Performance of embodiments of the present disclosure will depend on the chosen sequence X(k). Using a sequence with constant amplitude in the frequency domain is beneficial since it provides desirable autocorrelation properties for zero Doppler shift. Additionally, a Zadoff-Chu sequence with length equal to a prime number has an advantage that the transmitted signal Sm(n) has constant amplitude.

[0093] Frequency-domain windowing in transmitter and / or receiver is beneficial to reduce the impact of autocorrelation side lobes. A wide variety of window shapes with may be used, including Hamming, Hanning, Kaiser, etc. The choice of window shape may be based on balancing requirements for sidelobe suppression and width of the peak / main lobe.

[0094] In some embodiments, both transmitter and receiver may apply windowing. For example, a first part of the windowing (e.g., with frequency response can be applied in the transmitter and a second part of the windowing (e.g., with frequency response wrx( / )) can be applied in the receiver, with the total windowing being the product of the two parts w( ) = wtx( / ) * conj(wrx(j ), where “conj” represents complex conjugate. In some variants,

[0095] Various features of the embodiments described above correspond to various operations illustrated in Figure 14, which show an exemplary method (e.g., procedures) for a radar receiver. In other words, various features of the operations described below correspond to various embodiments described above. Although Figure 14 shows specific blocks in a particular order, the operations of the exemplary method may be performed in different orders than shown and may be combined and / or divided into blocks having different functionality than shown. Optional blocks or operations are indicated by dashed lines. The exemplary method can or may be performed by any device or apparatus (e.g., wireless device, etc.) capable of receiving radar signals, such as described elsewhere herein.

[0096] The exemplary method includes the operations of block 1410, where the radar receiver receives a composite signal corresponding to a transmitted radar signal. The composite signal comprises a plurality (M) of signal streams corresponding to a respective plurality (M) of signal pulses in the transmitted radar signal. Each signal stream includes delayed samples corresponding to a plurality of propagation delays of the corresponding signal pulse. In other words, each signal stream includes samples covering delayed copies of the corresponding signal pulse.

[0097] The exemplary method also includes the operations of block 1420, where the radar receiver processes the received composite signal to generate a delay -Doppler plane representation. The exemplary method also includes the operations of block 1430, where the radar receiver detects an object at a first delay value and a first Doppler value in the delay -Doppler plane representation. The exemplary method also includes the operations of block 1440, where the radar receiver corrects the first delay value based on a delay correction that is a function of the following: the first Doppler value, and a sequence of values used to generate the plurality (M) of signal pulses in the transmitted radar signal.

[0098] In some embodiments, for a first sequence of values used to generate the plurality (M) of signal pulses in the transmitted radar signal, the delay correction increases approximately linearly in relation to one of the following: velocity of an object that reflected the transmitted radar signal, or Doppler shift in the received signal due to the velocity of the obj ect that reflected the transmitted radar signal. Figure 13 shows an example of delay error for which these embodiments of delay correction would be beneficial.

[0099] In some embodiments, for a second sequence of values used to generate the plurality (M) of signal pulses in the transmitted radar signal, the delay correction has one or more approximate step changes in relation to one of the following: velocity of an object that reflected the transmitted radar signal, or Doppler shift in the received signal due to the velocity of the object that reflected the transmitted radar signal. Figure 10 shows an example of delay error for which these embodiments of delay correction would be beneficial.

[0100] In some embodiments, in the samples of each signal stream, the magnitude of the corresponding signal pulse is inversely proportional to one of the following: velocity of an object that reflected the transmitted radar signal, or Doppler shift in the received signal due to the velocity of the object that reflected the transmitted radar signal. In some embodiments, each signal pulse of the plurality (M) of signal pulses in the transmitted radar signal includes a main pulse, and the sequence of values is used to generate the main pulse. Moreover, correcting the first delay value in block 1440 is based on a function of the sequence of values used to generate the main pulse of each signal pulse. In some of these embodiments, the sequence of values is a frequency domain sequence, and the main pulse is generated based on an Inverse Discrete Fourier Transform (IDFT) of the frequency domain sequence. In some variants of these embodiments, the frequency domain sequence is one of the following: a quadrature phase shift keying (QPSK) sequence, a binary phase shift keying (BPSK) sequence, a Zadoff-Chu sequence, or any cyclic shift of a Zadoff-Chu sequence.

[0101] In some further variants, the Zadoff-Chu sequence is defined as: where q is an integer in the range 0. . .Nzc-1, and u is an integer in the range 1 . . . Nzc-1. Each of these ranges is inclusive of the two endpoints. For example, u may be a root index. In some further variants, the delay correction is a function of the root index, u, of the Zadoff-Chu sequence. In some further variants, the Zadoff-Chu sequence has one or more of the following characteristics:

[0102] • a first characteristic, in which u=l or u= Nzc-1; and

[0103] • a second characteristic, in which Nzcis a prime number.

[0104] In some of these embodiments, the sequence of values used to generate the main pulse has a periodic autocorrelation function (ACF) with a single non-zero value in each period. Alternatively or additionally, the values of the sequence have a constant modulus or magnitude.

[0105] In some of these embodiments, each signal pulse of the plurality (M) of signal pulses in the transmitted signal also includes a cyclic extension (CE) of the main pulse. As noted above, a CP is a type of CE. In some variants of these embodiments, the CEs of the plurality (M) of signal pulses have a common length, wherein the common length is sufficient to capture echoes from objects at a maximum detection range. In other variants of these embodiments, at least one of the CEs of the plurality (M) of signal pulses has a different length than at least one other of the CEs of the plurality (M) of signal pulses.

[0106] In some variants of these embodiments, processing the received composite signal to generate the delay -Doppler plane representation in block 1420 includes the following operations, labelled with corresponding sub-block numbers:

[0107] • (1421) performing a cyclic correlation of each signal stream with a replica of the main pulse of the corresponding signal pulse, thereby generating a delay-time plane representation comprising a plurality (M) of correlation streams including samples corresponding to the plurality of propagation delays;

[0108] • (1422) performing a plurality of discrete Fourier transforms (DFTs) on the correlation streams to generate the delay -Doppler plane representation, wherein the respective DFTs are performed on samples from all correlation streams corresponding to respective propagation delays.

[0109] In some further variants, performing the cyclic correlation of each signal stream with the replica of the main pulse of the corresponding signal pulse in sub-block 1421 comprises the following operations, for each signal stream:

[0110] • performing a DFT of the signal stream;

[0111] • computing an element-by-element product between the DFT of the signal stream and a complex conjugate of the sequence of values used to generate the main pulse; and

[0112] • performing an inverse DFT of the element-by-element product, thereby generating the cyclic correlation of the signal stream.

[0113] In some further variants, performing the cyclic correlation of each signal stream with the replica of the main pulse of the corresponding signal pulse in sub-block 1421 further comprises the radar receiver applying a first window function to the element-by-element product for each signal stream, i.e., prior to performing the inverse DFT. For example, the first window function may be configured to improve localization of objects in the delay dimension of the delay -Doppler plane representation, relative to when the first window function is not applied. As another example, the first window function may be configured to be complementary to a second window function configured to be applied by a radar transmitter to each main pulse of the plurality (M) of signal pulses included in the transmitted radar signal.

[0114] Although various embodiments are described above in terms of methods, techniques, and / or procedures, the person of ordinary skill will readily comprehend that such methods, techniques, and / or procedures can be embodied by various combinations of hardware and software in various systems, communication devices, computing devices, control devices, apparatuses, non-transitory computer-readable media, computer program products, etc. For example, a radar receiver may comprise receiver circuitry configured to receive echoes of transmitted radar signals and processing circuitry operatively coupled to the receiver circuitry. The processing circuitry (and optionally the receiver circuitry) is configured to perform operations corresponding to methods described above.

[0115] Figure 15 shows a block diagram of an exemplary device configurable according to various embodiments of the present disclosure, including by execution of instructions on a computer- readable medium that correspond to, or comprise, any of the exemplary methods and / or procedures described above. For example, device 1500 can or may be configured as a radar receiver, a radar transmitter, or as a radar transmitter / receiver (“transceiver”).

[0116] Device 1500 comprises processing circuitry 1510 that may be operably connected to a program memory 1520 and / or a data memory 1530 via a bus 1570 that may comprise parallel address and data buses, serial ports, or other methods and / or structures known to those of ordinary skill in the art. Program memory 1520 may store software code, programs, and / or instructions (collectively shown as computer program product 1521 in Figure 15) executed by processing circuitry 1510 that may configure and / or facilitate device 1500 to perform operations corresponding to various methods or procedures described herein. For example, execution of computer program product 1521 may configure device 1500 to perform operations attributed to a radar receiver and / or to a radar transmitter in the above descriptions related to other figures.

[0117] In some embodiments, program memory 1520 may store software code, programs, and / or instructions that may facilitate device 1500 to communicate with various Fifth Generation / New Radio (5G / NR), Fourth Generation / Long Term Evolution (4G / LTE), Third Generation / General Universal Mobile Telecommunications System (3G / UMTS), Second Generation / Global System for Mobile Communications / General Packet Radio Service (2G / GSM / GPRS), etc. networks according to standards promulgated by 3rd Generation Partnership Project (3GPP). In some embodiments, program memory 1520 may store software code, programs, and / or instructions that may facilitate device 1500 to communicate with various Wireless Fidelity (WiFi) networks according to standards promulgated by Institute of Electrical and Electronics Engineers (IEEE). In some embodiments, program memory 1520 may store software code, programs, and / or instructions that may facilitate device 1500 to communicate with other devices or networks via radio-frequency technologies such as Bluetooth, Zigbee, Global Navigation Satellite Systems (GNSS), such as GPS, etc.

[0118] Program memory 1520 may also include software code executed by processing circuitry 1510 to control the function of device 1500, including configuring and controlling various components such as transceiver 1540, user interface 1550 (optional), and / or control interface 1560 (optional). Program memory 1520 may also comprise one or more application programs and / or modules comprising computer-executable instructions embodying any of the exemplary methods and / or procedures described herein. Such software code may be specified or written using any known or future developed programming language, such as e.g., Java, C++, C, Objective C, HTML, XHTML, machine code, and Assembler, as long as the desired functionality, e.g., as defined by the implemented method steps, is preserved. In addition, or as an alternative, program memory 1520 may comprise an external storage arrangement (not shown) remote from device 1500, from which the instructions may be downloaded into program memory 1520 located within or removably coupled to device 1500, so as to enable execution of such instructions.

[0119] Data memory 1530 may comprise memory area for processing circuitry 1510 to store variables used in protocols, configuration, control, and other functions of device 1500, including operations corresponding to, or comprising, any of the exemplary methods and / or procedures described herein. Moreover, program memory 1520 and / or data memory 1530 may comprise nonvolatile memory (e.g., flash memory), volatile memory (e.g., static or dynamic RAM), or a combination thereof. Furthermore, data memory 1530 may comprise a memory slot by which removable memory cards in one or more formats (e.g., SD Card, Memory Stick, Compact Flash, etc.) may be inserted and removed.

[0120] Processing circuitry 1510 may include one or more individual processors (including, e.g., multi-core processors), each of which implements a portion of the functionality described above. Multiple individual processors may be commonly connected to program memory 1520 and data memory 1530 or individually connected to multiple individual program memories and or data memories. More generally, persons of ordinary skill in the art will recognize that device 1500 may comprise various computing arrangements having different combinations of processing circuitry and software including, but not limited to, application processors, signal processors, general- purpose processors, multi-core processors, ASICs, fixed and / or programmable digital circuitry, analog baseband circuitry, radio-frequency circuitry, software, firmware, and middleware.

[0121] Transceiver 1540 may or may include transmitter circuitry 1541 and / or receiver circuitry 1542 that facilitates device 1500 to transmit and / or receive radar signals according to various embodiments described above. For example, the receiver circuitry may include one or more antennas, low-noise amplifiers, filters, mixers, analog-to-digital converters (ADCs), etc. that are arranged to receive echoes of transmitted radar signals and to produce samples for processing by processing circuitry 1510. Similarly, the transmitter circuitry may include one or more antennas, power amplifiers, mixers, filters, digital-to-analog converters (DACs), etc. arranged to generate and transmit radar signals based on samples provided by processing circuitry 1510. In some embodiments, receiver circuitry 1542 and transmitter circuitry 1541 may share one or more common components, such as antenna(s), mixer(s), etc.

[0122] In some embodiments, transceiver 1540 may include radio-frequency transmitter and / or receiver circuitry that may facilitate device 1500 to communicate with various 5G / NR,4G / LTE, 3G / UMTS, 2G / GSM / GPRS, etc. networks according to standards promulgated by 3GPP and / or with WiFi networks according to standards promulgated by IEEE. In different variants, this radiofrequency transmitter and / or receiver circuitry may be the same as, include parts of, or be different than transmitter circuitry 1541 and / or receiver circuitry 1542. In some embodiments, transceiver 1540 may include radio-frequency transmitter and / or receiver circuitry that may facilitate device 1500 to communicate with other devices or networks via radio-frequency technologies such as Bluetooth, Zigbee, GNSS (e.g., GPS), etc. In different variants, this radio-frequency transmitter and / or receiver circuitry may be the same as, include parts of, or be different than transmitter circuitry 1541 and / or receiver circuitry 1542.

[0123] User interface 1550 may take various forms depending on the embodiment of device 1500, or may be absent from device 1500 entirely. In some embodiments, user interface 1550 may comprise a microphone, a loudspeaker, slidable buttons, depressible buttons, a display, a touchscreen display, a mechanical or virtual keypad, a mechanical or virtual keyboard, and / or any other user-interface features commonly found on mobile phones. In other embodiments, device 1500 may comprise a tablet computing device including a larger touchscreen display. In such embodiments, one or more of the mechanical features of user interface 1550 may be replaced by comparable or functionally equivalent virtual user interface features (e.g., virtual keypad, virtual buttons, etc.) implemented using the touchscreen display, as familiar to persons of ordinary skill in the art. In other embodiments, device 1500 may be a digital computing device, such as a laptop computer, desktop computer, workstation, etc. that comprises a mechanical keyboard that may be integrated, detached, or detachable depending on the particular embodiment. Such a digital computing device may also comprise a touch screen display. Embodiments of device 1500 having a touch screen display may receive user inputs, such as inputs related to exemplary methods and / or procedures described herein or otherwise known to persons of ordinary skill in the art.

[0124] In some embodiments, device 1500 may include an orientation sensor, which may be used in various ways by features and functions of device 1500. For example, device 1500 may use outputs of the orientation sensor to determine when a user has changed the physical orientation of device 1500’s touch screen display. An indication signal from the orientation sensor may be available to any application program executing on device 1500, such that an application program may change the orientation of a screen display (e.g., from portrait to landscape) automatically when the indication signal indicates an approximate 90-degree change in physical orientation of the device. In this manner, the application program may maintain the screen display in a manner that is readable by the user, regardless of the physical orientation of the device. In addition, the output of the orientation sensor may be used in conjunction with various embodiments of the present disclosure.

[0125] When present in device 1500, control interface 1560 may take various forms depending on the particular embodiment of device 1500 and the particular interface requirements of other devices that device 1500 is intended to communicate with and / or control. For example, the control interface 1560 may comprise an RS-232 interface, an RS-485 interface, a USB interface, an HDMI interface, a Bluetooth interface, an IEEE (“Firewire”) interface, an I2C interface, a PCMCIA interface, or the like. In some embodiments of the present disclosure, control interface 1560 may comprise an IEEE 802.3 Ethernet interface such as described above. In some embodiments of the present disclosure, the control interface 1560 may comprise analog interface circuitry including, for example, one or more digital-to-analog (D / A) and / or analog-to-digital (A / D) converters.

[0126] Persons of ordinary skill in the art may recognize that the above list of features, interfaces, and radio-frequency communication standards is merely exemplary, and not limiting to the scope of the present disclosure. In other words, device 1500 may comprise more functionality than is shown in Figure 15 including, for example, a video and / or still -image camera, microphone, media player and / or recorder, etc. Moreover, the processing circuitry 1510 may execute software code stored in the program memory 1520 to control such additional functionality. For example, directional velocity and / or position estimates output from a GPS receiver may be available to any application program executing on device 1500. As another example, sensing outputs from receiver 1541 may be available to any application program executing on device 1500.

[0127] The foregoing merely illustrates the principles of the disclosure. Various modifications and alterations to the described embodiments will be apparent to those skilled in the art in view of the teachings herein. It will thus be appreciated that those skilled in the art will be able to devise numerous systems, arrangements, and procedures that, although not explicitly shown or described herein, embody the principles of the disclosure and can be thus within the spirit and scope of the disclosure. Various embodiments can be used together with one another, as well as interchangeably therewith, as should be understood by those having ordinary skill in the art.

[0128] The term unit, as used herein, can have conventional meaning in the field of electronics, electrical devices and / or electronic devices and can include, for example, electrical and / or electronic circuitry, devices, modules, processors, memories, logic solid state and / or discrete devices, computer programs or instructions for carrying out respective tasks, procedures, computations, outputs, and / or displaying functions, and so on, such as those that are described herein.

[0129] Any appropriate steps, methods, features, functions, or benefits disclosed herein may be performed through one or more functional units or modules of one or more virtual apparatuses. Each virtual apparatus may comprise a number of these functional units. These functional units may be implemented via processing circuitry, which may include one or more microprocessor or microcontrollers, as well as other digital hardware, which may include Digital Signal Processor (DSPs), special-purpose digital logic, and the like. The processing circuitry may be configured to execute program code stored in memory, which may include one or several types of memory such as Read Only Memory (ROM), Random Access Memory (RAM), cache memory, flash memory devices, optical storage devices, etc. Program code stored in memory includes program instructions for executing one or more telecommunications and / or data communications protocols as well as instructions for performing one or more of the techniques described herein. In some implementations, the processing circuitry may be used to cause the respective functional unit to perform corresponding functions according to one or more embodiments of the present disclosure.

[0130] As described herein, device and / or apparatus can be represented by a semiconductor chip, a chipset, or a (hardware) module comprising such chip or chipset; this, however, does not exclude the possibility that a functionality of a device or apparatus, instead of being hardware implemented, be implemented as a software module such as a computer program or a computer program product comprising executable software code portions for execution or being run on a processor. Furthermore, functionality of a device or apparatus can be implemented by any combination of hardware and software. A device or apparatus can also be regarded as an assembly of multiple devices and / or apparatuses, whether functionally in cooperation with or independently of each other. Moreover, devices and apparatuses can be implemented in a distributed fashion throughout a system, so long as the functionality of the device or apparatus is preserved. Such and similar principles are considered known to a skilled person.

[0131] Unless otherwise defined, all terms (including technical and scientific terms) used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this disclosure belongs. It will be further understood that terms used herein should be interpreted as having a meaning that is consistent with their meaning in the context of this specification and the relevant art and will not be interpreted in an idealized or overly formal sense unless expressly so defined herein.

[0132] In addition, certain terms used in the present disclosure, including the specification and drawings, can be used synonymously in certain instances (e.g., “data” and “information”). It should be understood that although these terms (and / or other terms that can be synonymous to one another) can be used synonymously herein, there may be instances where such words are not intended to be used synonymously.

Claims

CLAIMS1. A method for a radar receiver, the method comprising: receiving (1410) a composite signal corresponding to a transmitted radar signal, wherein: the received composite signal comprises a plurality, M, of signal streams corresponding to a respective plurality, M, of signal pulses in the transmitted radar signal, and each signal stream includes delayed samples corresponding to a plurality of propagation delays of the corresponding signal pulse; processing (1420) the received composite signal to generate a delay -Doppler plane representation; detecting (1430) an object at a first delay value and at a first Doppler value in the delay- Doppler plane representation; and correcting (1440) the first delay value based on a delay correction that is a function of the following: the first Doppler value, and a sequence of values used to generate the plurality, M, of signal pulses in the transmitted radar signal.

2. The method of claim 1, wherein for a first sequence of values used to generate the plurality, M, of signal pulses in the transmitted radar signal, the delay correction increases approximately linearly in relation to one of the following: velocity of an object that reflected the transmitted radar signal, orDoppler shift in the received signal due to the velocity of the object that reflected the transmitted radar signal.

3. The method of any one of claims 1-2, wherein for a second sequence of values used to generate the plurality, M, of signal pulses in the transmitted radar signal, the delay correction has one or more approximate step changes in relation to one of the following: velocity of an object that reflected the transmitted radar signal, orDoppler shift in the received signal due to the velocity of the object that reflected the transmitted radar signal.

4. The method of any one of claims 1-3, wherein in the samples of each signal stream, the magnitude of the corresponding signal pulse is inversely proportional to one of the following: velocity of an object that reflected the transmitted radar signal, orDoppler shift in the received signal due to the velocity of the object that reflected the transmitted radar signal.

5. The method of any one of claims 1-4, wherein: each signal pulse of the plurality, M, of signal pulses in the transmitted radar signal includes a main pulse; the sequence of values is used to generate the main pulse; and correcting (1440) the first delay value is based on a function of the sequence of values used to generate the main pulse of each signal pulse.

6. The method of claim 5, wherein the sequence of values is a frequency domain sequence and the main pulse is generated based on an Inverse Discrete Fourier Transform, IDFT, of the frequency domain sequence.

7. The method of claim 6, wherein the frequency domain sequence is one of the following: a quadrature phase shift keying, QPSK, sequence; a binary phase shift keying, BPSK, sequence; a Zadoff-Chu sequence; or any cyclic shift of a Zadoff-Chu sequence.

8. The method of claim 7, wherein the Zadoff-Chu sequence is defined as:q is an integer in the range 0. . .Nzc-1; and u is an integer root index in the range 1 . . .Nzc-1.

9. The method of claim 8, wherein the Zadoff-Chu sequence has one or more of the following characteristics: a first characteristic, in which u=l or u=Nzc-l; and a second characteristic, in which Nzcis a prime number.

10. The method of any one of claims 8-9, wherein the delay correction is a function of the root index, u, of the Zadoff-Chu sequence.

11. The method of any one of claims 5-10, wherein one or more of the following applies: the sequence of values used to generate the main pulse has a periodic autocorrelation function, ACF, with a single non-zero value in each period; andthe values of the sequence have a constant modulus or magnitude.

12. The method of any one of claims 5-11, wherein each signal pulse of the plurality, M, of signal pulses in the transmitted signal also includes a cyclic extension, CE, of the main pulse.

13. The method of claim 12, wherein one of the following applies: the CEs of the plurality, M, of signal pulses have a common length, wherein the common length is sufficient to capture echoes from objects at a maximum detection range; or at least one of the CEs of the plurality, M, of signal pulses has a different length than at least one other of the CEs of the plurality, M, of signal pulses.

14. The method of any one of claims 12-13, wherein processing (1420) the received composite signal to generate the delay-Doppler plane representation comprises: performing (1421) a cyclic correlation of each signal stream with a replica of the main pulse of the corresponding signal pulse, thereby generating a delay-time plane representation comprising a plurality, M, of correlation streams including samples corresponding to the plurality of propagation delays; performing (1422) a plurality of discrete Fourier transforms, DFTs, on the correlation streams to generate the delay-Doppler plane representation, wherein the respective DFTs are performed on samples from all correlation streams corresponding to respective propagation delays.

15. The method of claim 14, wherein performing (1421) the cyclic correlation of each signal stream with the replica of the main pulse of the corresponding signal pulse comprises, for each signal stream: performing a DFT of the signal stream; computing an element-by-element product between the DFT of the signal stream and a complex conjugate of the sequence of values used to generate the main pulse; and performing an inverse DFT of the element-by-element product, thereby generating the cyclic correlation of the signal stream.

16. The method of claim 15, wherein performing (1421) the cyclic correlation of each signal stream with the replica of the main pulse of the corresponding signal pulse further comprises, foreach signal stream, applying a first window function to the element-by-element product prior to performing the inverse DFT.

17. The method of claim 16, wherein the first window function is configured to improve localization of objects in the delay dimension of the delay -Doppler plane representation, relative to when the first window function is not applied.

18. The method of any one of claims 16-17, wherein the first window function is configured to be complementary to a second window function configured to be applied by a radar transmitter to each main pulse of the plurality, M, of signal pulses included in the transmitted radar signal.

19. A radar receiver (1200, 1500) comprising processing circuitry (1510) configured to: receive a composite signal corresponding to a transmitted radar signal, wherein: the received composite signal comprises a plurality, M, of signal streams corresponding to a respective plurality, M, of signal pulses in the transmitted radar signal, and each signal stream includes delayed samples corresponding to a plurality of propagation delays of the corresponding signal pulse; process the received composite signal to generate a delay -Doppler plane representation; detect an object at a first delay value and at a first Doppler value in the delay -Doppler plane representation; and correct the first delay value based on a delay correction that is a function of the following: the first Doppler value, and a sequence of values used to generate the plurality, M, of signal pulses in the transmitted radar signal.

20. The radar receiver (1200, 1500) of claim 19, wherein the processing circuitry (1510) is further configured to perform operations corresponding to any one of the methods of claims 2- 18.

21. A radar receiver (1200, 1500) configured to: receive a composite signal corresponding to a transmitted radar signal, wherein: the received composite signal comprises a plurality, M, of signal streams corresponding to a respective plurality, M, of signal pulses in the transmitted radar signal, andeach signal stream includes delayed samples corresponding to a plurality of propagation delays of the corresponding signal pulse; process the received composite signal to generate a delay -Doppler plane representation; detect an object at a first delay value and at a first Doppler value in the delay -Doppler plane representation; and correct the first delay value based on a delay correction that is a function of the following: the first Doppler value, and a sequence of values used to generate the plurality, M, of signal pulses in the transmitted radar signal.

22. The radar receiver (1200, 1500) of claim 21, being further configured to perform operations corresponding to any one of the methods of claims 2-18.

23. A radar transceiver (1500) compri sing : the radar receiver of any one of claims 19-22; and a radar transmitter (1100, 1510, 1541) configured to generate and transmit radar signals, wherein the radar receiver is configured to receive echoes of the radar signals transmitted by the radar transmitter.

24. A non-transitory, computer-readable medium (1520) storing computer-executable instructions that, when executed by processing circuitry (1510) of a radar receiver (1200, 1500), configure the radar receiver to perform operations corresponding to any one of the methods of claims 1-18.

25. A computer program product (1521) comprising computer-executable instructions that, when executed by processing circuitry (1510) of a radar receiver (1200, 1500), configure the radar receiver to perform operations corresponding to any one of the methods of claims 1-18.

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