Antenna array element for beamforming antenna arrays

Abstract

An antenna array element (1) for beamforming antenna arrays, comprising a dielectric enclosure (16), two bowtie radiating elements (2a, 2b) orthogonally arranged in parallel planes (7, 8), and a feeding network comprising four T-shaped monopoles (10a, 10b, 10c, 10d), the ends (11a, 11b, 11c, 11d) of which are arranged respectively parallel to the extremities (9a, 9b, 9c, 9d) of the bowtie radiating elements (2a, 2b). The antenna array element (1) provides wide-angle beam steering with minimal gain degradation, and allows achieving a compact design with reduced mutual coupling in antenna arrays, making it suitable for satellite systems.

Description

FIELD

[0001] The present invention generally relates to the field of radio frequency (RF) communications, and more specifically to an antenna array element designed to enable the development of high-performance beamforming antenna arrays, particularly suitable for satellite communication systems.BACKGROUND

[0002] Beam steering in satellite communication applications provides an additional degree of freedom for satellite pointing and energy consumption. Beam steering implies that the direction of maximum gain of the antenna can be modified in the angular domain, or the gain can be maximized in a certain direction. This ability simplifies the satellite operation by enabling the antenna to be directed toward a target without requiring physical reorientation of the satellite, which would be otherwise required to maintain an optimal communication link. It is always desirable to point the antenna with the maximum gain in the direction of desired communication. In addition, beam steering ensures that the SNR (signal to noise ratio) is optimized for peak performance of the communication protocol.

[0003] There are two main types of beam-steerable antennas: mechanical beam-steering antennas and electronic beam-steering antennas. Certain communication protocols, such as Narrowband Internet of Things (NB-loT), involve short transmission windows, for example with frame duration of 10 milliseconds. In such cases, mechanical beam steering may be unsuitable due to the required beam-pointing agility. Furthermore, from a structural standpoint, satellite designs with flat, low-profile surfaces are preferable, which can further limit the feasibility of mechanically steerable solutions.

[0004] The topic of electronically steerable antennas has been addressed extensively in literature. The design principles are generally applicable across different frequency bands. However, some practical limitations exist due to the physical size constraints and the limiting nature of satellite platforms.

[0005] One example of a typical electronically steerable antenna array application is disclosed in [1]. The antenna array operates at a frequency of 11.95 GHz, resulting in a wavelength (λ) of 25.1 mm. The array features a triangular lattice of 94 elements, with 13.5 mm (0.53λ) spacing between individual elements. The authors report that a beam steering angle of 60 degrees and a gain degradation of 3 dB (from 24.8 to 21.9 dB) is achieved.

[0006] Another example of an electronically steerable antenna design is presented in [2], which discloses the use of a staked patch antenna element. The configuration is reported to provide high gain at nadir. The antenna element has a peak directivity of 8.2 dBi (realized gain estimated to be around 6.2 dBi) and a directivity of 0 dBi (realized gain estimated to be around -2dBi). There is a degradation in gain due to beam steering (scanning loss) of 3 dB at 50 degrees beam steering angle from nadir.

[0007] Document [3] provides a comprehensive literature review on electronically steerable antennas. It identifies and categorizes the principal known approaches to solve beam steerable array design challenges available in literature. The different design methodologies are summarized in Tables 2 and 3 of said document. As can be seen, most array topologies have a scanning loss that is at least 2.5 dB for a dual axis scanning of ± 60 degrees.

[0008] In [4] a 2.5 dB scanning loss has been obtained for an array comprising large elements when compared to classical arrays (approximately eight times the size of a classical element). The reduced scanning loss is attributed to the use of an antenna element with a very wide beam.

[0009] Document [5] presents another well-known method for beam-steering antenna arrays operating at frequencies in the order of tens of GHz. The method is called a lens array. In this approach, a superstrate or a reflector changes the phase of the propagated wave to sum coherently in a desired direction. This method presents limitations when applied to lower frequency bands such as the S-band. At these frequencies, the associated wavelengths result in electrically and physically large structures, which may not be practical for size- and mass-constrained platforms like satellites. Furthermore, such architectures typically have limited aperture efficiency and a narrow beam-scanning range, with noticeable gain degradation at larger steering angles (scanning loss).

[0010] Document [6] discloses an antenna system for space communication comprising a compact patch antenna configured for S-band operation on satellite platforms. The design included a feeding network intended to address bandwidth constraints inherent to miniaturized antenna structures. This feeding network comprises a capacitive coupling element disposed beneath the radiating patch and an L-type matching network configured to achieve impedance matching. Although its compact form factor, the antenna element presents limited radiation efficiency, primarily due to impedance mismatch and dielectric losses. As shown in Figure 4 of [6], the proposed design achieves high efficiency only within a very narrow bandwidth of the target operational frequency range, thereby limiting its applicability.

[0011] There are some well-known limitations of beam-steerable antennas when it comes to satellite platforms. One important limitation in antenna arrays is mutual coupling, which refers to the interaction between closely spaced antenna elements. When elements are placed near each other, the electromagnetic fields radiated by one element can induce currents in neighboring elements. This interaction modifies the input impedance and radiation pattern of the elements, potentially reducing the overall efficiency of the array and distorting its intended beam shape.

[0012] Another key issue is the occurrence of grating lobes, which are unwanted secondary lobes that appear when the spacing between array elements exceeds a certain threshold. According to the established antenna theory, to suppress grating lobes across the intended beam steering range, the spacing between elements in a uniform planar array must not exceed half the wavelength (λ / 2).

[0013] However, achieving high radiation efficiency and wide operational bandwidth typically requires antenna elements with physical dimensions larger than λ / 2. This introduces a tradeoff: reducing inter-element spacing to avoid grating lobes increases mutual coupling, which in turn degrades element efficiency due to undesired energy transfer between elements. Thus, an inherent design compromise must be made, maximizing element size and performance while minimizing element spacing within the constraints imposed by the application.

[0014] Additionally, beamwidth is also a critical parameter for beam-steerable antenna arrays. The overall array gain in a given direction is determined by the product of the element gain in that direction and the array factor, which is dependent on the array geometry. To enable wide-angle beam steering with minimal scanning loss, it is advantageous for each antenna element to have a wide beamwidth, thereby maintaining sufficient gain across a broad range of scan angles.

[0015] The prior art fails to provide a solution that simultaneously satisfies requirements of wide-angle beam steering with minimal scanning loss, high radiation efficiency, wide operational bandwidth and compactness, particularly in the operational frequency of satellite platforms. Known array architectures, such as the ones disclosed in [1]-[6], typically present drawbacks of limited element efficiency, high scanning loss, and / or impractically large structural dimensions for integration on flat, low-profile satellite surfaces.

[0016] Therefore, there is a need for an antenna element that enables wide-angle beam steering with minimal gain degradation, reduced mutual coupling, and compatibility with the physical and functional constraints of satellite systems. The present invention solves these problems.References

[0017] [1] Kaplan, I. et al., "Electronically Beam Steerable Antennas for Broadband Satellite Communications", the 8th European Conference on Antennas and Propagation (EuCAP 2014), Apr. 2014, pp. 2450-2454. [2] Expósito-Domínguez, G. et al., "Electronic Steering Antenna Onboard for Satellite Communications in X Band", Proceedings of the 5th European Conference on Antennas and Propagation (EUCAP), Rome, Italy, 2011, pp. 2120-2123. [3] Chaloun, T. et al., "Electronically Steerable Antennas for Future Heterogeneous Communication Networks: Review and Perspectives," IEEE Journal of Microwaves, vol. 2, no. 4, pp. 545-581, Oct. 2022. [4] Logan, J. T. et al., "A New Class of Planar Ultrawideband Modular Antenna Arrays With Improved Bandwidth," IEEE Transactions on Antennas and Propagation, vol. 66, no. 2, pp. 692-701, Feb. 2018. [5] Mei, P. et al., "A Low-Profile and Beam-Steerable Transmitarray Antenna: Design, Fabrication, and Measurement", IEEE Antennas and Propagation Magazine, vol. 63, no. 5, pp. 88-101, Oct. 2021. [6] Tatomirescu, A. et al., "Compact S Band Antenna for CubeSat", 2018 International Conference on Communications (COMM), Jun. 2018, pp. 231-234. SUMMARY

[0018] The present invention relates to an antenna array element for beamforming antenna arrays that overcomes the aforementioned problems using a particular geometrical configuration of the radiating element (based on a bowtie design) combined with a unique feeding mechanism.

[0019] In particular, the antenna array element comprises a dielectric enclosure, two bowtie radiating elements orthogonally arranged in parallel planes, and a feeding network comprising four T-shaped monopole feeds, the ends of which are arranged respectively parallel to the extremities of the bowtie radiating elements.

[0020] Another aspect of the present invention refers to a beamforming antenna array or an electronically steerable antenna system comprising a plurality of said antenna array elements arranged in a planar or conformal array configuration.

[0021] The antenna array element offers key advantages including wide operational bandwidth, high element gain, low scanning loss, reduced weight, and a structurally simple design, thus making it particularly well-suited for beamforming antenna arrays on satellite platforms. Although the invention is optimized for satellite-based beamforming antenna arrays, its benefits are applicable to any electronically steerable antenna system requiring high performance and compact integration.

[0022] The present invention overcomes the limitations in the prior art by introducing an antenna array element suitable for use in various array configurations, including but not limited to planar and conformal arrays, providing wide beam steering capabilities and without the occurrence of grating lobes. In a uniform planar array configuration according to an embodiment of the invention, the antenna array element enables beam steering up to ±60 degrees from boresight (i.e., normal to the array surface) without grating lobes and avoiding significant gain degradation (scanning loss). This design is particularly well-suited for compact, low-profile array implementations, such as those required in satellite platforms where space and weight constraints are critical.

[0023] A key performance metric of the antenna array element is the scanning loss, which is approximately 2 dB at a steering angle of 50 degrees. This represents a significant improvement over conventional solutions, which typically have scanning losses of around 3 dB for realistic array configurations in satellite applications. Furthermore, the physical design of the antenna array element is characterized by a compact size, lightweight structure, and a simplified layout, facilitating low-cost manufacturing and ensuring compatibility with the structural and functional constraints commonly associated with satellite-based systems.BRIEF DESCRIPTION OF THE DRAWINGS

[0024] A series of drawings which aid in better understanding the invention and which are expressly related with an embodiment of said invention, presented as a non-limiting example thereof, are very briefly described below. Figure 1 shows an antenna array element according to an embodiment. Figure 2 shows the current distribution of one of the polarizations for the embodiment of Figure 1. Figure 3 shows the broadside radiation pattern of the antenna array element in a spherical coordinate system. Figures 4A and 4B show exemplary gain patterns of the antenna array element at different operational frequencies. Figure 5 shows a beamforming antenna array formed by multiple antenna array elements arranged in a planar array configuration. Figure 6 is a graph depicting the mutual coupling for the beamforming antenna array of Figure 5. Figure 7 is a graph depicting the realized gain of an 8x8 array of antenna array elements at different tilt angles. Figure 8 is a schematic representation of the connection of the antenna array element with a transceiver. DETAILED DESCRIPTION

[0025] The antenna array element of the present invention is based on a cross polarized bowtie antenna. Figure 1 illustrates an antenna array element 1 according to an embodiment of the present invention, where two bowtie radiating elements (a first bowtie radiating element 2a and a second bowtie radiating element 2b) are positioned in an orthogonal layout (i.e., their corresponding longitudinal axes 3a and 3b being orthogonal). Each bowtie radiating element (2a, 2b) is formed by two opposing triangular elements (first bowtie radiating element 2a formed by first triangular element 4a and second triangular element 5a, and second bowtie radiating element 2b formed by first triangular element 4b and second triangular element 5b) contacting at a central region 6, the extremities of each bowtie radiating element (first extremity 9a and second extremity 9c for first bowtie radiating element 2a, and first extremity 9b and second extremity 9d for second bowtie radiating element 2b) being arranged at opposing sides of the central region 6.

[0026] The two bowtie radiating elements (2a, 2b) are arranged in parallel planes separated by an offset e and configured to radiate orthogonal current distributions. In the embodiment depicted in Figure 1 the top of the first bowtie radiating element 2a is disposed on a first plane 7 and the top of the second bowtie radiating element 2b is disposed on a second plane 8, parallel and arranged on or below the first plane 7. In an embodiment, the bowtie radiating elements (2a, 2b) are arranged in parallel planes (e.g., horizontal planes XY) separated by a distance (offset e) lower than 2 mm. First 7 and second 8 planes may be coincident, such that in this embodiment the offset e is zero and the bowtie radiating elements are disposed on the same plane, thereby forming a cross-shaped element.

[0027] A challenge associated with the use of bowtie antennas lies in the feeding method. The bowtie antenna is inherently a balanced radiator, whereas typical feed lines are unbalanced. In conventional patch antenna designs, unbalanced feeding can be applied directly without requiring special adaptation. However, to properly feed a balanced antenna like the bowtie, a balun (i.e., a balanced-to-unbalanced transition circuit) is traditionally required. The inclusion of a balun introduces additional insertion losses and imposes constraints on the operational bandwidth, thereby limiting overall antenna performance.

[0028] The present invention addresses this issue by introducing a coupled feeding network that simultaneously functions as an impedance matching circuit, thereby enhancing the bandwidth of the antenna array element 1. The coupled feeding network of the antenna array element 1 comprises a set of miniaturized monopoles T. The set of T-shaped monopoles are placed to introduce a second resonance, complementing the primary resonance provided by the bowtie elements.

[0029] Figure 1 illustrates the structure of the feeding network, where four T-shaped monopoles (first T-shaped monopole 10a, second T-shaped monopole 10b, third T-shaped monopole 10c and fourth T-shaped monopole 10d) are positioned such that their respective ends (11a, 11b, 11c, 11d), i.e. the horizontal element or top hat of the T, are parallel to the corresponding extremities of the bowtie radiating elements (2a, 2b): The end 11a of the first T-shaped monopole 10a is parallel to the first extremity 9a of the first bowtie radiating element 2a. The end 11b of the second T-shaped monopole 10b is parallel to the first extremity 9b of the second bowtie radiating element 2b. The end 11c of the third T-shaped monopole 10c is parallel to the second extremity 9c of the first bowtie radiating element 2a. The end 11d of the fourth T-shaped monopole 10d is parallel to the second extremity 9d of the second bowtie radiating element 2b.

[0030] Therefore, the upper part of each T-shaped monopole is positioned at the extremities of the bowtie elements, specifically in regions where the electric field is strongest. This arrangement generates a strong capacitive coupling between the T-shaped monopole and the bowtie radiating elements. This added capacitance creates a second resonant mode. This effect broadens the frequency range over which the overall antenna element presents a favourable impedance to the transceiver, thereby enhancing the operational bandwidth of the antenna array.

[0031] The monopoles also contribute to the overall electromagnetic radiation, emitting energy in directions where the bowtie elements might have reduced radiation. This dual function of the T-shaped monopoles, providing both a second resonance for impedance matching and contributing to the radiation pattern, effectively widens the beam of the antenna element and enhances its operational bandwidth.

[0032] The T-shaped monopoles (10a, 10b, 10c, 10d) are preferably arranged in planes (13a, 13b, 13c, 13d) perpendicular to the bowtie radiating elements (i.e., vertical planes perpendicular to the first 7 and second 8 horizontal planes) and containing the corresponding extremity (9a, 9b, 9c, 9d). Positioning the T-shaped monopoles (10a, 10b, 10c, 10d) at the extremities (9a, 9b, 9c, 9d) of the bowtie elements (2a, 2b) maximizes capacitive coupling between the monopoles and the bowtie elements. The strong coupling is essential for efficient impedance matching, and displacing the monopoles from the extremities (9a, 9b, 9c, 9d) would negatively impact coupling, degrading impedance matching and reducing operational bandwidth.

[0033] Furthermore, the vertical element (12a, 12b, 12c, 12d) of each T-shaped monople (10a, 10b, 10c, 10d), which extends upwards from a corresponding feed point (14a, 14b, 14c, 14d), is aligned with the center of the corresponding extremity (9a, 9b, 9c, 9d). In the embodiment depicted in Figure 1, the feed points 14a and 14d function as input interfaces, while the feed points 14b and 14c are connected to ground. Misalignments may impact the radiation efficiency of the antenna array element 1. Specifically, displacing the vertical element (12a, 12b, 12c, 12d) from the center of the corresponding extremity (9a, 9b, 9c, 9d) results in a degraded cross-polarization ratio, leading to reduced axial ratio and radiation efficiency.

[0034] In this arrangement, the ends (11a, 11b, 11c, 11d) of the T-shaped monopoles (10a, 10b, 10c, 10d) are separated by a distance d (in the Z axis) from the corresponding extremity (9a, 9b, 9c, 9d). The distance d is dependent on the frequency of operation and the specific dielectric material (dielectric enclosure 16) used as a support for the structure and can range between 0.01λ and 0.25λ at the frequency of operation. In one embodiment of the present invention, the dielectric enclosure 16 encasing the antenna array element 1 possesses a dielectric constant of approximately 3.

[0035] The vertical planes (13a, 13b, 13c, 13d) define, together with an upper horizontal plane (first plane 7) and a lower horizontal plane 15 that pass through the feed points (14a, 14b, 14c, 14d), a volume (in particular, a parallelepiped) of the antenna array element 1. In the embodiment of Figure 1 the offset e is 0.2 mm in the Z axis, and the dimensions of the volume are defined by the following distances between opposite end points (P1, P2) of the parallelepiped: 32 mm in the X axis, 32 mm in the Y axis and 18 mm in the Z axis. However, depending on the particular application, the value of the offset e and the dimensions of the volume may vary according to the wavelength (λ) at the operation frequency. Preferred embodiments of the present invention define the dimensions of the parallelepiped along the X, Y, and Z axes to range between 0.05λ and 0.35λ.

[0036] Figure 2 shows the current distribution of one of the polarizations, wherein the active monopole is the fourth T-shaped monopole 10d which is connected to feed point 14d, serving as an input interface for the antenna array element 1, and a parasitic opposite monopole for balancing the radiation pattern is the second T-shaped monopole 10b, which is connected to ground. The radiation pattern balancing works by designing and placing the parasitic element (second T-shaped monopole 10b) to counteract undesirable characteristics of the active element's radiation (fourth T-shaped monopole 10d). For example, if an active monopole inherently radiates in an undesired direction, the induced currents on the parasitic element are designed to radiate in a way that cancels out or reduces these unwanted lobes, improving the radiation pattern.

[0037] Figure 3 illustrates an exemplary radiation pattern of the antenna array element 1. The main lobe 17 of the radiation pattern is oriented in the broadside direction, specifically aligned with the Z-axis, which is perpendicular to the plane of the antenna array element 1 (lower horizontal plane 15). The spherical coordinate system employed for illustrating the radiation pattern is defined such that the Theta (θ) angle ranges from 0° (along the positive Z-direction) to 180° (along the negative Z-direction), and the Phi (Φ) angle sweeps from 0° to 360° around the Z-axis, consistent with conventional antenna theory.

[0038] The bowtie element has superior beamwidth compared to a square or circular patch antenna. As depicted in Figures 4A and 4B, respectively showing the gain pattern of frequencies 1.98 GHz and 2.2 GHz, for different φ and θ angles, the beamwidth at 0 dBi realized gain of the antenna array element 1 is around 120 degrees. This means a beam steering up to ±60 degrees from broadside without grating lobes. Electronically changing the tilt angle in these ranges is accomplished by applying a progressive phase shift to the signals fed to each individual radiating element in the array.

[0039] Thus, the antenna array element 1 is superior in terms of beamwidth. For example, the 3dB beamwidth is around 85-90 degrees, whereas the typical 3 dB beamwidth for normal patches is around 70 degrees. In Figure 4A, which refers to the frequency of 1.98 GHz, the total efficiency of the element is -0.7451 dBi, and the realized gain is 6.551 dBi. In Figure 4B, which refers to the frequency of 2.2 GHz, the total efficiency is -0.6240 dBi and the realized gain is 7.280 dBi. The realized gain is defined as the product of the antenna element's directivity and its total efficiency. To achieve a high gain, an efficiency as close as possible to 1 (i.e., 0 dB) is desired.

[0040] Figure 5 shows a beamforming antenna array 20 comprising a plurality of antenna array elements (1a, 1b, 1c, 1d) arranged in a planar array configuration. In this embodiment, the antenna array elements are separated by 72 mm in the X-axis (dx spacing) and Y-axis (dy spacing). In other embodiments, the values of dx and dy may vary from one another and may be comprised between 0.4λ and 0.6λ at the operational frequency.

[0041] Figure 6 illustrates an exemplary performance of the antenna array 20 by depicting the magnitude of the scattering parameters (S-parameters), expressed in decibels (dB). For example, the curves labelled S1,1 and S2,2 represent the reflection coefficients of the respective antenna ports. A reflection coefficient value below approximately -10 dB indicates that the antenna element is well-matched to its feeding network at the corresponding frequency. As depicted, the antenna array exhibits a good impedance match for frequencies ranging from approximately 2.0 GHz to 2.2 GHz. This matching corresponds to a wide operational bandwidth of approximately 200 MHz.

[0042] For example, in the S-band, for a central frequency around 2.2 GHz, the bandwidth achieved by the antenna array element 1 is around 13% of the central operating frequency, whereas a typical patch antenna supports approximately 5-6%. For a service such as NB-loT over satellite, this increased bandwidth effectively spans the S-band allocated for the service link. Another advantage achieved is that the enhanced bandwidth allows for a miniaturization of the antenna array element.

[0043] Furthermore, the mutual coupling between adjacent and non-adjacent antenna array elements is characterized by S-parameters such as S2,1, S3,1, and S4,1. Notably, high performance is maintained even when spacing the antenna array elements (1a, 1b, 1c, 1d) only 72 mm from each other. This can be observed as the mutual coupling remain generally below -15 dB across the operational bandwidth, indicating an effective isolation between antenna array elements. This close spacing between antenna array elements is enabled by the miniaturization of each antenna array element, which results in concentration of the fields around it and limited mutual coupling even at 72 mm (0.53λ at 2.2 Ghz) spacing.

[0044] Figure 7, a graph depicting the realized gain of an 8x8 array of antenna array elements 1 at 2170 MHz for different beam tilt angles θ. Advantageously, it can be observed that the sidelobe level remains low even at 50 degrees beam tilt.

[0045] Figure 8 illustrates a high-level electronic scheme for the connection of the antenna array element 1 comprising the proposed feeding network according to the present invention, depicting the signal path from the Radio Frequency Front-End (RF FE) of a transceiver 30 to the antenna array element 1. The RF FE transceiver 3 is configured to process both Right-Hand Circular Polarization (RHCP) signals and Left-Hand Circular Polarization (LHCP) signals. To achieve circular polarization from the two orthogonal linear polarizations inherent to the antenna array element 1, accessible via Port 1 and Port 2 (for the embodiment of Figure 1, feed points 14a and 14d, respectively). A 90-degree hybrid coupler 40 is operatively connected. This 90-degree hybrid coupler 40 is configured to facilitate the excitation of the circular polarizations, thereby ensuring proper feeding of the antenna array element 1 for various applications.

[0046] The antenna array element 1, while primarily described in planar configurations, is equally applicable to conformal and other non-planar antenna arrangements. Its principles adapt to various curved surfaces, including cylindrical, spherical, conical, and arbitrarily curved geometries. This enables integration into diverse platforms requiring high-performance, compact, electronically steerable antenna designs.

Claims

1. An antenna array element for beamforming antenna arrays, comprising: a dielectric enclosure (16); two bowtie radiating elements (2a, 2b) orthogonally arranged in parallel planes (7, 8); and a feeding network comprising four T-shaped monopoles (10a, 10b, 10c, 10d), the ends (11a, 11b, 11c, 11d) of which are arranged respectively parallel to the extremities (9a, 9b, 9c, 9d) of the bowtie radiating elements (2a, 2b).

2. The antenna array element of claim 1, wherein the parallel planes (7, 8) of the bowtie radiating elements (2a, 2b) are separated by an offset (e) lower than 2 mm.

3. The antenna array element of any preceding claim, wherein the T-shaped monopoles (10a, 10b, 10c, 10d) are arranged in vertical planes (13a, 13b, 13c, 13d) perpendicular to the bowtie radiating elements (2a, 2b) and containing the corresponding extremity (9a, 9b, 9c, 9d).

4. The antenna array element of claim 3, wherein the ends (11a, 11b, 11c, 11d) of the T-shaped monopoles (10a, 10b, 10c, 10d) are arranged at a distance (d) lower than 0.25λ at the operational frequency from the corresponding extremity (9a, 9b, 9c, 9d) of the bowtie radiating elements (2a, 2b).

5. The antenna array element of any preceding claim, wherein the dimensions of the antenna array element (1) in the X, Y and Z axes range between 0.05λ and 0.35λ at the operational frequency.

6. The antenna array element of any preceding claim, wherein the operational frequency is in the range of 2.0 GHz to 2.2 GHz.

7. A beamforming antenna array, comprising a plurality of antenna array elements (1a, 1b, 1c, 1d) of any preceding claim.

8. The beamforming antenna array of claim 7, wherein the antenna array elements (1a, 1b, 1c, 1d) are arranged in a planar array configuration.

9. The beamforming antenna array of claim 7, wherein the antenna array elements (1a, 1b, 1c, 1d) are arranged in a conformal array configuration on a non-planar surface.

10. The beamforming antenna array of any preceding claim, wherein the antenna array elements (1a, 1b, 1c, 1d) are separated by a distance (dx, dy) in each axis (X, Y) of the planar configuration comprised between 0.4λ and 0.6λ at the operational frequency.

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