Insulation test apparatus

Abstract

Insulation test apparatus comprises a flyback transformer circuit comprising an input circuit arranged to switch a current to a primary winding of a transformer 8 comprising a bobbin 15 around which t

Description

Technical Field The present invention relates generally to insulation test apparatus, and in particular, but not exclusively, to a battery powered insulation test meter comprising a flyback transformer circuit. Background Measurements of electrical resistance of an electrical insulator may be performed by insulation test apparatus, such as a portable battery powered test meter, by applying a test voltage to conductors separated by the insulator and measuring the resulting current flowing in the insulator. The insulation resistance may be calculated in the meter on the basis of the applied test voltage and the resulting current. Typically, an insulation test meter may be adjustable to produce a test voltage that may be set to values over a wide range, to suit the specific test performed, for example generating a test voltage that may be in a range from 50V to 5kV, lOkV or higher. In order to generate the test voltage from a lower supply voltage, for example a DC voltage supplied by a battery, a flyback transformer circuit may be used. A flyback transformer circuit has a switch, such as a transistor, arranged to interrupt an input current to the primary winding of a flyback transformer. A flyback transformer is a transformer specifically designed to have an air gap in its magnetic core, which stores magnetic energy for transfer between the primary and secondary windings when the input current is interrupted. A diode is connected in series with the secondary winding of the flyback transformer to charge up a capacitor to the required voltage to provide the test voltage. The advantage of the flyback circuit over a conventional transformer circuit is that the voltage may be stepped up by more than the turns ratio between the secondary and primary winding. However, the construction of the flyback transformer for operation at the high voltages required for insulation testing is typically complex and expensive, and the efficiency of the flyback transformer circuit may be low, limiting battery life. It would be advantageous to provide insulation test apparatus that is capable of generating a test voltage with greater efficiency and at lower cost. Summary In accordance with a first aspect, there is provided insulation test apparatus comprising a flyback transformer circuit, the flyback transformer circuit comprising an input circuit, a transformer and an output circuit, the input circuit being arranged to switch a current to a primary winding of the transformer, the transformer comprising a bobbin around which the primary winding and a secondary winding are wound and a magnetic core, and the output circuit comprising a diode connected in series with the secondary winding of the transformer and at least one capacitor connected in parallel with the series combination of the secondary winding and the diode, wherein: the magnetic core comprises two parts connected to provide a central member disposed within the bobbin and to provide two outer members outside the bobbin, each outer member connecting the two parts, and the central member having a gap disposed within the bobbin; the secondary winding is wound as a single layer of turns; a spacing layer composed of insulating material is provided between the secondary winding and the primary winding; and the primary winding is wound around the spacing layer, further from the bobbin than is the secondary winding. The spacing layer allows the secondary winding to be wound from fine wire, thereby allowing a greater number of turns to be wound as a single layer, by protecting the secondary winding mechanically from the forces that would be applied to it when winding the primary winding on top of it. The primary winding requires thicker wire than the secondary, which would potentially damage the secondary due to the forces involved in bending the wire, if it were not protected by the spacing layer. Positioning the primary winding and the secondary winding on opposite sides of the spacing layer allows efficient operation, by allowing the primary winding to overlay the secondary winding, while the spacing layer reduces the capacitance between the primary and secondary windings. Providing the secondary winding as a single layer of turns allows high voltage to be generated using an economical construction. In an example, the spacing layer has a thickness of at least 0.5 mm. In other examples, the spacing layer has a thickness of at least 1 mm or more. The thickness of the spacing layer reduces the capacitance between the primary and secondary windings, increasing the efficiency of the flyback circuit. The spacing layer may comprise insulating tape wound around the secondary winding, providing a firm basis for the primary winding, protecting the secondary winding. In addition, or instead of the insulating tape, the spacing layer may comprise an insulating mesh, for example a cross-linked polymer mesh. The cross-linked polymer mesh resists compression, and the air content of the mesh further reduces the capacitance between the primary and secondary windings. In an example, the secondary winding covers at least 80% of the width of the bobbin, typically substantially the whole width of the bobbin. This feature allows the maximum number of turns to be produced as a single layer, increasing the voltage that may be generated by the circuit. The secondary winding may be wound so that each turn is disposed further from a given end of the bobbin than the preceding turn and no turn crosses another turn, so that the chance of insulation breakdown between the turns is reduced. In an example, the primary winding is disposed such that the midpoint of the central member of the magnetic core is within the primary winding, and in an example, the gap in the central member of the magnetic core is within the primary winding, providing for efficient operation of the transformer. In an example, the end of the primary winding that is nearest to the higher voltage end of the secondary winding is at least 3mm from the higher voltage end of the secondary winding. This reduces the chance of insulation breakdown between the primary and secondary windings. In an example, the transformer is not set in a potting compound, providing a more economical implementation. In an example, the combined cross-sectional area of the primary winding, the secondary winding and the spacing layer occupies less than 25% of the cross sectional area of a window in the core between the central member and an outer member. This provides efficient cooling for the flyback transformer. In an example, the secondary to primary voltage ratio divided by the secondary to primary turns ratio is at least 3, and typically 8 or more, due to the operation of the flyback transformer circuit. In an example, the output circuit comprises a voltage multiplier circuit. The use of a voltage multiplier circuit, for example providing a test voltage that is greater than the voltage on the secondary winding by a factor of 2, 3 or more, allows a lower voltage to be generated by the flyback transformer for a given test voltage. This allows the flyback transformer to have a reduced turns ratio, so that the turns of the secondary can be accommodated in a single layer, while producing a high output voltage. In an example, the insulation test apparatus comprises a feedback circuit configured to control an output voltage of the voltage multiplier by pulse width modulation of a current applied to the primary winding by the input circuit. The feedback circuit allows the test voltage to be accurately generated despite the inherent lack of precision of the use of a voltage multiplier. The feedback circuit may be configured to set an output voltage of the output circuit to a voltage within a controllable range from 2.5 kV or less to 5 kV or more, or in another example, the feedback circuit is configured to set an output voltage of the output circuit to a voltage within a controllable range from 50 V or less to 15 kV or more, providing a wide range of test voltages. In an example, the input circuit is configured to provide a turn-on waveform that is more gradual than a turn-off waveform. This allows the feedback circuit to control the voltage over a greater range, to a lower voltage than would be possible using a symmetrical waveform. To provide a turn-on waveform that is more gradual than a turn-off waveform, the input circuit may comprise a parallel combination of a resistor and a diode in series with a gate of a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) transistor, the MOSFET transistor being arranged to switch the input current to the primary winding. In an example, the feedback circuit is configured to provide feedback of the output voltage of the output circuit and a voltage representing the current in the primary winding to control the pulse width modulation of the current applied to the primary winding. This arrangement provides greater stability in the generation of the output voltage. In an example, the feedback to control the pulse width modulation comprises a signal combining the signal used to switch the current in the primary winding and the voltage representing the current in the primary winding. This arrangement provides more stable voltage generation to mitigate generation of current spikes when a flyback transfer is used with a voltage multiplier. Further features and advantages of the will be apparent from the following description of exemplary embodiments, which are given by way of example only. Brief Description of the Drawings Figure 1 is a schematic diagram illustrating insulation test apparatus having a voltage generation circuit comprising an improved flyback transformer; Figure 2 is a schematic diagram illustrating insulation test apparatus having a voltage generation circuit comprising an improved flyback transformer used in conjunction with a voltage multiplier circuit; Figure 3 is an exploded view of an improved flyback transformer; Figure 4 shows the secondary winding of the improved flyback transformer (spacing layer and primary winding removed for clarity); Figure 5 shows the spacing layer between the primary and secondary windings comprising insulating tape and a cross-linked polymer mesh (primary winding not shown for clarity); Figure 6 shows the magnetic core comprising two E-shaped components and having a gap in the central member; Figure 7 is a schematic diagram showing a cross section of the primary winding, the spacing layer, and the secondary winding (not to scale); Figure 8 is a circuit diagram showing a flyback transformer circuit; Figure 9 is a circuit diagram showing a flyback transformer circuit having a voltage multiplier; Figure 10 illustrates the input circuit in which a signal from the gate driver output is added using resistor RPWM to a feedback signal representing the current in the primary winding and fed to the current feedback input of the pulse width modulation controller; Figure 11 shows the signal fed back to the input of the pulse width modulation controller, for the case that the flyback transformer is connected to a voltage multiplier, without adding the signal from the gate driver output; and Figure 12 shows the signal fed back to the input of the pulse width modulation controller, for the case that the flyback transformer is connected to a voltage multiplier, with the addition of the signal from the gate driver output. Detailed Description By way of example, embodiments will now be described in the context of a portable battery powered meter for measuring the insulation resistance of an insulator, which is configured to be capable of generating a stable high DC voltage (1 kV and higher) using a flyback transformer circuit. In an example, a DC voltage may be generated that is controllable in a range from 40V to IkV and higher. However, it will be understood that embodiments may also apply to other types of electrical testing, and the generation of a non-DC voltage, for example by pulse-width modulation of an input current, to give a time-varying output voltage, for example in the form of a ramp, and the meter may not necessarily be battery-powered. Figure lisa schematic diagram illustrating insulation test apparatus in the form of a meter 1 having a voltage generation circuit 7 comprising an improved flyback transformer 8. As shown in Figure 2, the voltage generation circuit may comprise a voltage multiplier circuit 11 used in conjunction with the improved flyback transformer as described herein. The inventors have found that the improved flyback transformer allows a more efficient voltage generation circuit to be provided, which can extend the battery life of the meter. The production cost of the flyback transformer 8 is also reduced compared to conventional flyback transformers, by its simplicity of design. In addition, a single universal design of flyback transformer 8 can be used for a variety of different models of the test meter, which may be specified to produce different maximum test voltages. The use of a voltage multiplier following the flyback transformer allows a test voltage to be generated by the meter that is higher than the voltage specified for the flyback transformer. For example, limiting the voltage specified for the flyback transformer to 5 kV allows the more efficient and simplified design of flyback transformer to be used, and the use of a voltage multiplier, such as, for example, a voltage doubler or tripler, allows the same flyback transformer to be used in a variety of models of test meter, for example specified to produce a test voltage of 10 kV or 15 kV. It is not conventional to use a voltage multiplier in conjunction with a flyback transformer circuit in a test meter to produce a well-controlled and stable test voltage, because a voltage multiplier is known to reduce its voltage output and to increase voltage ripple under heavy load. In an insulation test meter, the output load can be significant, for example 30 W for 15 kV at 2 mA, but the stability of the output voltage must be very high, for example better than 100 mV in 1 kV or better than 100 mV in 10 kV. Furthermore, the connection of a voltage multiplier at the output of the flyback transformer circuit interferes with the operation of flyback transformer circuit, causing some energy to be transferred to the secondary winding as soon as the primary current begins to flow. However, the inventors have found that careful design of the feedback loop controlling the voltage generation circuit can mitigate the shortcomings of the combination of a flyback transformer circuit in combination with a voltage multiplier. Conventionally, a flyback transformer circuit to produce voltages of 5 kV and above uses multiple stages of secondary turns, each secondary turn being separated by an insulating layer, instead of using a voltage multiplier. This results in a complex design of flyback transformer. As shown in Figure 1, under control of a processor 9, the voltage generation circuit 7, provides a test voltage to terminals 3, 4 of the meter which are connected to a device under test 2. The device under test 2 may be, for example, a part of an electrical machine such as a motor or a generator, in which metallic parts, such as windings, are separated by an insulator. To test the condition of the insulator, the test voltage is applied and the resulting leakage current is measured using a current measurement circuit 5. The applied voltage is measured using a voltage measurement circuit 6, and the insulation resistance is calculated from the measured current and voltage by the processor 9, and the resulting insulation value is shown on the display 10. As shown in Figure 2, the test meter may have a voltage generation circuit 7 comprising a voltage multiplier circuit 11. Figure 3 is an exploded view of an example of a flyback transformer in an embodiment of the invention. It can be seen that the transformer 8 comprises a bobbin 15 around which the primary winding 14 and a secondary winding are wound and a magnetic core 20, 18. The bobbin is a former made of rigid and typically non-conducting material, such as a composite or polymer, and comprises a tubular section, into which a central part of the magnetic core is inserted, and typically also comprises two opposite edge sections which may support the wires at the ends of the primary and secondary windings. In the example shown in the exploded view of Figure 3, the magnetic core comprises two E-shaped components 20, 18, such that the magnetic core, when assembled, comprises a central member, passing through the bobbin 15 and two outer members, passing outside the bobbin 15. The central member has a gap, typically an air gap, when assembled. The Clips 13, 19, are provided to hold the two parts of the magnetic core 20, 18, together. Figure 4 shows the secondary winding 17 of flyback transformer 8, which is wound as a single layer of turns around the bobbin 15. The spacing layer and primary winding are not shown, for clarity. The secondary winding covers at least 80% of the width of the bobbin. In the case illustrated the secondary winding covers substantially 100% of the width of the bobbin, that is to say substantially the whole width of the bobbin provided for the secondary winding. The secondary winding is wound so that each turn is disposed further from a given end of the bobbin than the preceding turn and no turn crosses another turn, reducing the risk of insulation breakdown. Providing the secondary winding as a single layer of turns allows high voltage to be generated using an economical construction. Figure 5 shows the spacing layer between the primary and secondary windings comprising insulating tape 21 and a cross-linked polymer mesh 22. The primary winding is not shown, for clarity. In this example, the cross-linked polymer mesh is wrapped around the insulating tape layer. In an example, the mesh may be attached to the insulating tape layer by adhesive tape 23. The thickness of the spacing layer reduces the capacitance between the primary 14 and secondary 17 windings, increasing the efficiency of the flyback circuit. The thickness of the spacing layer may be greater than 0.5 mm, greater than 1 mm, and greater than 1.5 mm, in examples. The spacing layer allows the secondary winding to be wound from fine wire, thereby allowing a greater number of turns to be wound as a single layer, by protecting the secondary winding mechanically from the forces that would be applied to it when winding the primary winding on top of it. The primary winding requires thicker wire than the secondary, which would potentially damage the secondary due to the forces involved in bending the wire, if it were not protected by the spacing layer. Positioning the primary winding and the secondary winding on opposite sides of the spacing layer allows efficient operation, by allowing the primary winding to overlay the secondary winding and both windings to overlay the gap in the magnetic core. Figure 6 shows that the magnetic core 26 comprises two E-shaped components 20, 18, such that the magnetic core comprises a central member 28 passing through the bobbin 15 and two outer members 24,27 passing outside the bobbin 15, the central member 28 having a gap 25 disposed within the bobbin 15 when the flyback transformer is assembled. As illustrated by Figure 3 and Figure 6, in this example, when assembled, the combined cross-sectional area of the primary winding 14, the secondary winding 17 and the spacing layer 29 occupies less than 25% of the cross-sectional area of a window in the core 26 between the central member 28 and an outer member 27. It can be seen that in the example illustrated, the magnetic core 26 comprises two parts 20, 18, which may each be described as E-shaped, connected (when clipped together, for example, with the clips 13, 19 shown in Figure 3) to provide a central member 28 disposed within the bobbin, and within the primary and secondary windings 14, 17, and to provide two outer members 24, 27, outside the bobbin, and outside the primary and secondary windings. Each outer member connects the two parts of the magnetic core 20, 18, in the sense that the two parts touch each other, but are not necessarily otherwise mechanically attached by the outer members. The central member 28 has a gap 25 disposed within the bobbin, and within the primary and secondary windings. The central member 27 accordingly comprises, in this example, protrusions from both parts 20, 18 of the core, and the outer members 24, 27 also comprise protrusions from both parts of the core. In other examples, the magnetic core may be formed of other arrangements that provide a central member, passing through the bobbin and the windings and having a gap within the bobbin and the windings, and outer members passing outside the bobbin and the windings. The magnetic core may comprise any number of parts. The parts may be sections of the core that are not necessarily separable parts. The magnetic core is composed of a magnetic material, for example ferrite. The “bobbin” may be any former that provides a base to support the secondary winding. Figure 7 is a schematic diagram showing a cross section of the primary winding 14, the spacing layer 29, and the secondary winding 17, wound onto the bobbin 15. The diagram is not to scale; in particular, the secondary winding 14 typically has more turns than illustrated, for example more than 400 turns, for example 410 turns, and the primary winding 14 typically has 10-20 turns, for example 14 turns. The secondary winding is wound from finer wire than the primary, typically 0.05 mm - 0.1 mm diameter. The primary winding is made of thicker wire, typically 0.5 - 1.5 mm in diameter. A spacing layer 29 composed of insulating material is provided between the secondary winding 17 and the primary winding 14. The spacing layer has a thickness of at least 0.5 mm, and in an example 1 mm. In the example shown, the spacing layer 29 comprises insulating tape 21 wound around the secondary winding 17, and the spacing layer 29 also comprises an insulating mesh 22. In this example, the insulating mesh 22 is composed of cross-linked polymer, such as nylon. In this example, the insulating mesh is a flattened tube, wound on top of the insulating tape 21, so that there are two layers of mesh between the primary 14 and secondary 17 windings. As a result, the spacing layer comprises air, in the gaps of the mesh, which reduces the capacitance between the primary and secondary windings, increasing the efficiency of the transformer. As can be seen from Figure 3, the primary winding 14 is wound as a single layer of turns around the spacing layer 29, further from the bobbin 15 than is the secondary winding 17. It can be seen from Figures 3 and 6 in particular, that the primary winding 14 is disposed such that the midpoint of the central member 28 of the magnetic core 26 and the gap 25 in the central member 28 of the magnetic core are within the primary winding. This improves the efficiency of the operation of the flyback transformer. Figure 7 illustrates that there is a gap 30 between the end of the primary winding 14 that is nearest to the higher voltage end of the secondary winding 17. The gap 30 is at least 3mm in an example. The transformer is not set in a potting compound. It can be seen from Figures 3 to 7 that the primary winding is positioned at the outer-most layer, which facilitates much better cooling, because most losses are dissipated in the primary winding. Most of the available core window is empty, i.e. not occupied by the cross-sectional area of the copper of the wire or the insulating material. Therefore, only less than 10 % or less than 25 % of the core window is occupied by the combined cross-sectional area of copper, and over 75-90 % is just air. This is contrary to the normal way the transformers are designed in which the window utilisation factor as high as possible. Additionally, the major part of the bobbin height is available as an insulating distance, which is beneficial for reducing partial discharge and other high voltage related problems, because the magnetic core will be typically at the potential of HV / 2, but certainly at a potential much lower than the HV end of the bobbin. Figure 8 is a circuit diagram showing an example of a flyback transformer circuit comprising the improved flyback transformer 8. The input voltage Vin is supplied from the power source (VI) to the primary winding LI, which is switched by Ml (MOSFET or any other type of transistor). The switching is controlled by a gate driver circuit, which controls the duty cycle, so that appropriate output voltage HVout or output power is delivered to the load Rload (or Zload), which is the device under test 2. The secondary winding L2 is connected in such a way that is blocked with a diode DI when Ml is ON. LI and L2 have reverse polarity, indicated by the dots. At the instant when Ml is turned OFF, the voltage across L2 reverses, and diode DI conducts the current into the output capacitance Cl. If the turn ratio of the transformer, duty cycle, and output load have the correct parameters then very high voltage can be generated at HVout. In examples, Vin can be between 5-25 V and HV out can be up to 16 kV or higher. The output voltage is also fed to a high-impedance voltage divider Rl-R2 (typically between MQ and hundreds of MQ), which provides information to the Voltage Feedback (VF) circuit, which affects the duty cycle in order to maintain the required level of output voltage. In addition, in this example, feedback is also provided of the current in the primary winding, by using a low-value (e.g. 0.05 Q) shunt resistor Rshunt connected between Ml and local ground. The signal on this resistor is used for the current feedback (IF). There are therefore two feedback loops working in parallel: the slower VF (voltage feedback) regulates the overall voltage level, and the faster IF (current feedback) helps with suppressing the faster deviations on pulse-to-pulse basis, improving instantaneous stability of the output voltage. The circuit works such that at the instant after Ml is turned OFF the output capacitance is being charged by the energy stored in the flyback transformer, and once the pulse energy is transferred, then the output capacitance Cl is the energy reservoir which sustains HVout until the next switching cycle (and also for the interval when Ml is ON). Switching frequency is typically in excess of 10 kHz, but can be hundreds of kHz. The circuit of Figure 8 may be used for lower voltages, for example if 1 kV or 2.5 kV is to be generated. If higher voltages are to be generated (5-15 kV) then conventionally the output voltage can be built up from several winding sections, however this results in a complex transformer structure having several secondary winding layers and insulation layers between the windings. Additionally, the windings can be wound only in the middle of the bobbin, because certain spacing is required to withstand the voltage between the stages. A flyback transformer behaves more like a coupled inductor than an ordinary transformer that does not have a gap in the core. The energy is first stored magnetically in the air gap of the core, and only after the primary current is suddenly interrupted it is delivered to the output, so it is a two-step process. By contrast, in a normal transformer there is no energy storage, and the secondary current flows always at the same time as the primary current. After turning the transistor Ml on, the primary current ramps up, only after the primary current is switched off at the peak of the ramp, the stored magnetic energy is transferred to the secondary winding to be discharged through the output diodes. The whole cycle repeats at a kHz rate (usually somewhere between 5-500 kHz). It should be noted that the output voltage level does not depend directly on the turn ratio of the transformer. This is contrary to any other transformer in which the voltage ratio (secondary to primary, V2 / V1) is directly proportional to the turns ratio (N2 / N1) so that V2 / V1 ~ N2 / N1 (when losses are neglected, etc.). This is not the case for the flyback transformer in which for producing the output voltage V2 / V1 N2 / N1. In the flyback transformer the energy is stored in the airgap in the first step, and it is responsible for all the energy transferred which happens only in the second step. The secondary current flow is sustained by the energy stored in the gapped core, because during this time the instantaneous primary current is zero. This is by contrast to conventional switched mode power supply topologies, which use non-gapped cores, because for such circuits any energy storing in the core is an unwanted parasitic. Figure 9 is a circuit diagram showing a flyback transformer circuit having a voltage multiplier, in this case a voltage doubler. The voltage doubler comprising D2, C2, D3 and D3, and shares DI, Cl with the flyback transformer circuit. The voltage doubler is an example of a voltage multiplier circuit that may be referred to as a Cockcroft-Walton multiplier. If such diode-capacitor circuit is to be fed with a bipolar or bi-state voltage, then during the positive half-cycle Cl charges through DI to the positive peak of the waveform. During the negative half cycle, Cl is already charged to the peak positive voltage, but left side of C2 goes to peak negative voltage. As a result, if C1=C2, during the negative half-cycle C2 charges also to the same voltage as Cl namely to have a voltage difference of peak positive across itself. During the next positive half-cycle the voltage across C2 is effectively added in series to the input voltage, and as a result the voltage at the output is twice the peak of the input. The final output diode D3 rectifies this voltage to produce DC. Higher voltage may be generated by having more diode-capacitor stages in the voltage multiplier. The flyback transformer shown in Figure 9 is the flyback transformer as illustrated by Figures 3 to 7. Compared to a conventional flyback transformer having several secondary windings, operation of the voltage generation circuit is improved, because the parasitic self-capacitance of the secondary winding as well as the parasitic capacitance from secondary to primary is reduced. This is because only one secondary section is present, and the spacing between the windings is controlled. Each time a high voltage pulse is generated on the secondary winding, the parasitic capacitance represented by Cp in Figure 9 has to be charged up and this extra energy is dissipated eventually as loss. The parasitic capacitance is the capacitance that results from the construction of the transformer rather than being a separate component. With smaller parasitic capacitance less energy is wasted for each HV pulse, but also at lower voltage less energy is stored in this capacitance. It is therefore possible to achieve significant efficiency improvement which allows for longer battery life either by reducing this capacitance or the voltage to which it is charged. Previous solutions can have efficiency of around 50% whereas with the new approach over 80% is possible. Because of the safety requirements, in some instruments the HV generator has to be powered up by an isolated power supply with additional inefficiency which increases with the level of power. By reducing the amount of power needed by the HV circuit also the demands on the isolating power supply are reduced. As a result, the battery life can be almost doubled, with the same power delivered to the HV output, from the same battery. The primary winding is kept at low potential (e.g. supplied from 24 V), which means that towards the HV end of the bobbin there would be a substantial electric field between 5 kV of the secondary and 24 V of the primary. This electric field intensity can be so high that it could generate partial discharge at the surface of the wires (primary or secondary). In order to reduce the intensity of the field a physical spacing is added, so that, as shown in Figure 7, the thickness of at least “c” is provided by that layer. This can be achieved for example by a cross-linked polymer braid mesh, which is flexible, but has walls strong enough not to be squashed by the tension of the primary winding. The value of “c” is to be not less than 0.5 mm. If “c” is too small the partial discharge can be generated, which would eventually damage the insulation of the wires and thus will lead to a premature failure. The strong mechanical base of the bobbin plus the secondary winding plus the insulation tape plus a mesh layer, for example allows the primary winding N1 to be be safely wound on top of the secondary, which is contrary to previous designs. However, the primary winding N1 being in proximity to the secondary winding N2 results in a certain amount of parasitic capacitance between these two conductors. This capacitance needs to be minimised to be very low, e.g. below 100 pF or 50 pF or 25 pF for the transformer to be efficient. This is why having several turns of the insulation tape and a spacer, such as the mesh, is further beneficial, not only mechanical and electrical protection, but also for increased mechanical thickness so that the mechanical spacing between the conductor surfaces of the primary winding and the secondary winding is increased and thus the resulting capacitance is lowered. The required number of turns of the tape / spacer will be different depending on its thickness, but in an example, at least 1 full turn is needed to fully cover and protect the secondary winding from the forces due the primary winding. The voltage distribution over the bobbin (especially over the secondary winding) is such that on one end of the bobbin the voltages are low (close to circuit ground). The other end of the bobbin becomes the high-voltage side. As can be seen in Figure 7, for a 5 kV transformer the middle of the bobbin would have only HV / 2 = 2.5 kV. For the critical field of 3 kV / mm (partial discharge threshold) not to be exceeded it is therefore theoretically enough to have just 0.8 mm spacing between the middle of the bobbin and the primary winding. Similar spacing considerations need to be made vertically (distance “c”) and horizontally (distance “b”). Therefore, provided that the distance “b” in Figure 7 (approximately the same as diagonal distance 30) is longer than 3 mm, or 6 mm to provide a safety margin, and “c” is greater than 1 mm this arrangement does not require any additional insulation to be applied between the primary winding N1 and the high voltage (HV) end of the secondary winding (N2). The secondary winding having a single layer and no wire crossing has the effect that if the winding produces 5000 V and this is divided equally into 410 turns, for example, then between each neighbouring turn the voltage is only 12.2 V, and even extremely thin insulation in the form of enamel with thickness of 0.0025 mm (typical for such 0.071 mm wire) can easily withstand the voltage applied, because the insulation is typically rated for a breakdown voltage of up to 300 V. However, by contrast, if just 25 turns were to be crossed by a loose or crossing wire, then the resulting potential would be 25 x 12.2 V = 305 V and it would make the wire insulation fail. This illustrates the advantage of a single electrical layer of secondary winding. As shown in Figure 7, the secondary is overlapped by the full length of the primary winding. This greatly improves the efficiency due to increased magnetic coupling. It is also important for the efficiency that the primary winding is located over the air gap in the magnetic core, which is typically at the centre of the bobbin. With the primary winding having relatively few turns (e.g. N1 = 14 the full width of the bobbin is not utilised and there can be a separation distance “a”. This is because it is more beneficial to wind the wires such that they form a tight solenoid, so that the electrically active surface area is minimised to keep the parasitic capacitance to a minimum. The distance “a” is not critical and it can be that a = 0, whereas “b” is critical and is kept greater than a certain minimum such that, for example, for a 5 kV output voltage, b >3 mm, or b >6 mm, so the primary winding can be positioned asymmetrically on the bobbin, in any case c >0.5 mm or c >1 mm. These values are valid if the universal flyback transformer is used up to 5 kV. For higher values proportionally larger b, c values will apply. Returning to Figure 9, the main flyback switch, i.e. transistor Ml, is turned on to start the primary current ramp. This is achieved by applying a rectangular pulse to the gate of the transistor Ml. The output voltage and / or power are regulated by using Pulse-Width Modulation (PWM) of the gate driving pulses. However, the output power must be regulated in an extremely wide range of values, ranging from tens of watts at 15 kV (typically between 10-90 W), to microwatts at 50 V (30 microwatts and less), which is over 6 orders of magnitude. It is very difficult to maintain the stability of the output voltage over such a wide single range because of the limitations of the available duty cycle control in the PWM. To improve the output regulation and stability, especially for the light load, the gate driver circuit is provided with an additional dedicated resistor Rg and diode Dg as shown in Figure 9, which allow obtaining an asymmetric gate drive signal, rather than substantially a rectangular signal, with a sharp on and sharp off. For turning the Ml on, the gate signal is applied. However, as the signal is applied the gate capacitance is discharged so the voltage across it is zero. This capacitance must be charged through the resistor Rg and therefore the speed of charging is slowed down and produces a “rounded off’ waveform. As a result, Ml turns on more slowly and in the initial part behaves more like a variable resistor, because it operates in its linear mode. This causes a large voltage drop across Ml and thus less voltage being available to drive the primary winding. Therefore, low output voltages from the transformer are easily achievable. The value of Rg to achieve this soft edge is greater than 50 Q or 100 Q or 500 Q, which is counterintuitive because typically low-resistance connection to gate is used, for example 10 Q or less. It is possible to obtain output voltages much lower than is dictated just by the transformer turn ratio, because most of the voltage can be made to be developed across the high resistance of the partially-turned-on transistor. Without the soft drive and with a voltage multiplier this is not possible, and the smallest output voltage would be dictated by at least the turn ratio, because the output pulse is not fully blocked with Ml turns on, as it is the case with direct flyback without a multiplier. So in this case the output would become around 24 V * 410 turns / 16 turns = 615 V, and it would be impossible to reduce this to a lower value, regardless the pulse width (ignoring parasitics). This is another reason why the physical turns ratio in the transformer is kept deliberately much lower than the required nominal voltage ratio. In an example, the turns ratio TR = N2 / N1 = 410 / 16 = 25.6, whereas voltage ratio VR = 5000 V / 24 V = 208, and so VR / TR = 8.1, with the minimum expected value to be at least 5 or 3, but it can be even higher. The soft edge also has the effect that it takes longer for Ml to become fully turned on and therefore the duty cycle of PWM becomes significantly longer for the shortest pulses, when the output running at low voltage is loaded with very high resistance, so this becomes easier to control in a stable way by the PWM controller. After Ml is turned on it must be also turned off. When the gate signal is pulled from high to low, e.g. from 10 V to ground potential, then the gate capacitance must be discharged. It is beneficial that turning off is achieved as quickly as possible to minimise switching losses in Ml. This is because the primary current as at the peak of the ramp and therefore the switching loss will be at its maximum (given by highest voltage multiplied by highest current). Therefore, the role of the diode Dg is to facilitate fast discharge of gate capacitance because Rg is bypassed by the diode and thus provides fast turning off for Ml. As a result of the Rg / Dg pair, the voltage on the gate of Ml is asymmetric, with soft rising, and sharp falling. Figure 9 shows the flyback circuit with a voltage doubler and asymmetric gate driver. AUX represents additional circuitry typically held at voltage close to ground. VF represents voltage feedback used in the control loop of the output voltage level. The AUX circuitry is used for providing further functions for example for current limiting on the secondary side of the transformer, for example for overcurrent protection. The output current can be measured / detected and this information can be provided to the PWM controller. If the output current exceeds the allowed maximum then the PWM generator can reduce the pulse width or even shutdown the PWM signal. The circuit of Figure 9 shows the version with a voltage doubler (Cl, DI, C2, D2, C3, D3). An analogous circuit can be used with a voltage tripler (3 stages) or any other number of sections in the voltage multiplier part. This circuit can also be used without the multiplier at all, with just one diode and capacitor, similar to Figure 8. As previously mentioned, using a voltage multiplier with a flyback transformer circuit can interfere with the operation of the flyback transformer circuit. A direct flyback transformer without a voltage multiplier transfers energy to the secondary winding only after the primary current is interrupted, because when the primary current is flowing the secondary diode is biased in the reverse direction. However, once the voltage multiplier is added this is no longer the case and some energy is transferred as soon as the primary current begins to flow. This is because Cl-D2-C2 (Figure 9) form an alternative path and allow energy transfer when Ml is on. As a result, it has been found that, when the voltage multiplier is present, there is an additional spike at the beginning of the current ramp, as shown in Fig. 11. This spike can have a very high amplitude which can even exceed the final peak of the ramp. The information about the current ramp is fed to the PWM controller, to its current feedback pin IF input (Figure 9). The PWM controller uses the peak of the signal supplied to the IF input for cycle-to-cycle control of the width of PWM. The peak current should be satisfied at all times to store the same energy, but if the supply voltage is somewhat lower (e.g. due to noise or higher load on the battery) then the same peak current will be reached only by making the pulse proportionally longer. So the peak current information is added to the PWM control loop. The peak current is also used for power limiting or to be fault tolerant, because the PWM controller will not allow the peak current to exceed the specified value. In conventional flyback converters the V(IF) signal (voltage representing current in the primary winding) is low-pass filtered to remove any fast glitches of noise, and this is achieved by the RFI and CFI low-pass filter in the circuit of Figure 10. However, with the voltage multiplier, this initial spike is much more significant, and it is not possible to completely filter it out. The filtered version of V(IF) is no longer a clean ramp, but it has a distortion at its start due to the energy contained in the large initial spike, as shown in trace 36 in Figure 11. In Figure 11 and Figure 12, the trace 37 is voltage across Rs, that is to say the voltage representing the current through the primary winding of the flyback transformer LI before filtering. For very short duty cycles the energy of this distortion can be higher than in the main ramp, and therefore the information is of little use for the cycle-to-cycle control. In order to further improve stability of the PWM control, in an embodiment, a part of the rectangular pulse from the output of the gate driver is connected to the same low-pass filter, by means of the resistor RPWM as shown in Figure 10. It is counterintuitive to connect the PWM output directly back the feedback input. With RPWM present, the CFI needs to have a proportionally larger value, because now the role of CFI is to form simultaneously one low-pass filter with RFI (for the ramp with leading impulse signal) and another low-pass filter with RPWM (for the rectangular signal). Because the RPWM + CFI forms a relatively low-frequency low pass filter then the rectangular waveform of the gate driver becomes in essence triangular and is added to the ramp signal. This is shown in Figure 12, and as can be seen the shape of trace 38 the filtered signal with gate signal and current feedback signal added, forms a much cleaner triangular ramp than the shape of the signal 36 which includes only the filtered current ramp signal. In an example, the proportion of information from the two sources can be adjusted so that it is roughly half of each (note the amplitude of trace 38 is twice the amplitude of trace 36), so each component is substantial, and typically would not be less than 25 % of the total. But in any case, the actual peak of such combined filtered signal must be scaled accordingly as dictated by the design specification of the PWM controller. If an off-the-shelf integrated circuit (a chip) is used then this can have a pre-set maximum value (for peak current limit) such as 1.25 V. This mixing scheme described above gives the best compromise between all the required aspects: a) clean signal because of good filtering, b) good quality ramp-like signal even for extremely short pulse widths, c) reasonable information about current peak in case of failure or over-current conditions. In examples, the flyback transformer circuit comprises an input circuit, a transformer and an output circuit, the input circuit being arranged to switch a current to a primary winding of the transformer, the transformer comprising the primary winding and a secondary winding and a magnetic core and the output circuit comprises a diode connected in series with the secondary winding of the transformer and at least one capacitor connected in parallel with the series combination of the secondary winding and the diode, wherein: the magnetic core comprises two parts connected to provide a central member disposed within the primary and secondary windings and to provide two outer members outside the primary and secondary windings, each outer member connecting the two parts, and the central member having a gap disposed within the primary and secondary windings; the secondary winding is wound as a single layer of turns; a spacing layer composed of insulating material is provided between the secondary winding and the primary winding; and the primary winding is wound around the spacing layer, further from the central member than is the secondary winding. In the examples described above, the primary and secondary windings are separated by a spacing layer. In alternative examples, the spacing may be provided by thick insulation on the primary winding, sufficient to provide high voltage isolation and to reduce inter-winding capacitance. The above embodiments are to be understood as illustrative examples of the invention. It is to be understood that any feature described in relation to any one embodiment may be used alone, or in combination with other features described, and may also be used in combination with one or more features of any other of the embodiments, or any combination of any other of the embodiments. Furthermore, equivalents and modifications not described above may also be employed without departing from the scope of the invention, which is defined in the accompanying claims.

Claims

1. Insulation test apparatus comprising a flyback transformer circuit, the flyback transformer circuit comprising an input circuit, a transformer and an output circuit, the input circuit being arranged to switch a current to a primary winding of the transformer, the transformer comprising a bobbin around which the primary winding and a secondary winding are wound and a magnetic core, and the output circuit comprising a diode connected in series with the secondary winding of the transformer and at least one capacitor connected in parallel with the series combination of the secondary winding and the diode, wherein:the magnetic core comprises two parts connected to provide a central member disposed within the bobbin and to provide two outer members, outside the bobbin, each outer member connecting the two parts, and the central member having a gap disposed within the bobbin;the secondary winding is wound as a single layer of turns;a spacing layer composed of insulating material is provided between the secondary winding and the primary winding; andthe primary winding is wound around the spacing layer, further from the bobbin than is the secondary winding.

2. Insulation test apparatus according to claim 1, wherein the spacing layer has a thickness of at least 0.5 mm.

3. Insulation test apparatus according to claim 2, wherein the spacing layer comprises insulating tape wound around the secondary winding.

4. Insulation test apparatus according to claim 2 or claim 3, wherein the spacing layer comprises an insulating mesh.

5. Insulation test apparatus according to claim 4, wherein the insulating mesh is composed of cross-linked polymer.

6. Insulation test apparatus according to any preceding claim, wherein the spacing layer comprises air.

7. Insulation test apparatus according to any preceding claim,wherein the secondary winding covers at least 80% of the width of the bobbin.

8. Insulation test apparatus according to any preceding claim,wherein the secondary winding is wound so that each turn is disposed further from a given end of the bobbin that the preceding turn and no turn crosses another turn.

9. Insulation test apparatus according to any preceding claim, wherein the primary winding is disposed such that the midpoint of the central member of the magnetic core is within the primary winding.

10. Insulation test apparatus according to any preceding claim, wherein the primary winding is disposed such that the gap in the central member of the magnetic core is within the primary winding.

11. Insulation test apparatus according to any preceding claim, wherein the end of the primary winding that is nearest to the higher voltage end of the secondary winding is at least 3mm from the higher voltage end of the secondary winding.

12. Insulation test apparatus according to any preceding claim,wherein the transformer is not set in a potting compound.

13. Insulation test apparatus according to any preceding claim,wherein the combined cross-sectional area of the primary winding, the secondarywinding and the spacing layer occupies less than 25% of the cross-sectional area of a window in the core between the central member and an outer member.

14. Insulation test apparatus according to any preceding claim, wherein the secondary to primary voltage ratio divided by the secondary to primary turns ratio is at least 3.

15. Insulation test apparatus according to any preceding claim, wherein the output circuit comprises a voltage multiplier circuit.

16. Insulation test apparatus according to claim 15, comprising a feedback circuit configured to control an output voltage of the voltage multiplier by pulse width modulation of a current applied to the primary winding by the input circuit.

17. Insulation test apparatus according to claim 16, wherein the feedback circuit is configured to set an output voltage of the output circuit to a voltage within a controllable range from 2.5 kV or less to 5 kV or more.

18. Insulation test apparatus according to claim 17, wherein the feedback circuit is configured to set an output voltage of the output circuit to a voltage within a controllable range from 50 V or less to 15 kV or more.

19. Insulation test apparatus according to any one of claims 15 to 18, wherein the input circuit is configured to provide a turn-on waveform that is more gradual than a turn-off waveform.

20. Insulation test apparatus according to claim 19, wherein the input circuit comprises a parallel combination of a resistor and a diode in series with a gate of a MOSFET transistor.

21. Insulation test apparatus according to any one of claims 15 to 20, wherein the feedback circuit is configured to provide feedback of the output voltage of the output circuit and a voltage representing the current in the primary winding to control the pulse width modulation of the current applied to the 5 primary winding.

22. Insulation test apparatus according to claim 21, wherein feedback to control the pulse width modulation comprises a signal combining the signal used to switch the current in the primary winding and the voltage representing the 10 current in the primary winding.IntellectualPropertyOfficeApplication GB2502161.9Search report under Section 17 of the Patents Act 1977Date search completed: 04 August 2025Claims searched: 1-22International classificationSubclass and subgroup Valid from g01r1 / 20 01 / 01 / 2006 g01r31 / 00 01 / 01 / 2006 g01r31 / 12 01 / 01 / 2020 h01f38 / 40 01 / 01 / 2006Field of searchWorldwide search of patent documents classified in the following areas of the IPC:G01R, H01FDatabases used in the preparation of this search report:SEARCH-PATENTDocuments considered to be relevantPatent literatureCategory Relevant Document of relevanceclaimsA - GB 2517015 A (MEGGER INSTR LTD), - A - CN 117524685 A (TONG FANG ELECT SCI &TECH CO LTD), -Non-patent literatureCategory Relevant claims Document of relevanceCategoriesLetter or symbol Description X Document indicating lack of novelty or inventive step.Y Document indicating lack of inventive step, if combined with anotherdocument of the same category.& Member of the same patent family. A Document indicating technological background. P Document published on or after the priority date but before the fling date of the present application. E Earlier application published on or after the filing date of the present application.

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