sensor

Abstract

To solve the problem that, in a contact detection type input method of a display screen built-in system discriminable at a multipoint simultaneous contact location applicable to a screen size over seven inches, a detection electrode and a control circuit connection terminal are connected by one-to-one, thus the number of control circuit connection terminals is increased significantly, resulting an increased cost due to an increase of the number of driver ICs used.SOLUTION: By locating a transistor constituting a multiplexer on a pixel, the number of control circuit connection terminals is decreased significantly without accompanying an increased frame width and a large reduction of an opening ratio. This method accompanies decreased detection sensitivity due to variation of a detection current, therefore canceling a fundamental current by a frame voltage impression method relaxes a restriction of the variation of the detection current. To realize it, a derivation of new design condition for a distribution arrangement of a transistor is conducted.SELECTED DRAWING: Figure 61

Description

Technical Field

[0001] The present invention relates to a technique for detecting the approach or contact of a detected object by detecting a current change caused by a capacitance formed between a detection electrode and the detected object via an insulating layer, and more particularly, to a technique for reducing the number of terminals connecting a large number of densely arranged detection electrodes and a control circuit.

Background Art

[0002] Currently, various methods have been proposed for sensors that detect the approach or contact of objects or humans, and they are widely used in society, from automatic door opening / closing devices to automotive collision accident prevention devices, depending on the application. Among these, a sensor element called a capacitive type that detects the approach or contact of a detected object by detecting a current change caused by a capacitance formed between a detection electrode and the detected object via an insulating layer has high sensitivity. A sensor device that miniaturizes and densely arranges this can individually identify and detect the simultaneous multi-point contact positions of the detected object. If the sensitivity of the capacitive sensor element can be improved, the detection of non-contact events, which has conventionally been performed based on an imaging element and image processing technology, can be achieved only with the capacitive sensor element. This not only enables significant cost reduction but also enables the creation of a wide range of new markets. Already, a sensor device of the detection capacitance timely formation type has been proposed as a highly sensitive capacitive sensor capable of non-contact detection. However, when performing simultaneous multi-point contact position identification with the conventional technology, the cost increases, so it has not been put into practical use, and the establishment of the technology is awaited.

[0003] For example, in recent years, medical institutions have been advancing the mechanization of reception and accounting operations. Many of these are operated by patients themselves visiting the medical institution by touching a touch panel installed on the display screen. Therefore, during the period when infectious diseases are prevalent, there is a hesitation in touching the screen with bare hands from a hygienic perspective. If it is possible to inexpensively realize non-contact operation of this, it can be expected to rapidly spread. Or, if a doctor during surgery can perform PC operations non-contact, it can also be expected to lead to the establishment of new medical technologies. Therefore, if a sensor device with a highly sensitive multi-point simultaneous contact position identification function that can be operated non-contact can be realized, the social contribution will be extremely high.

[0004] On the other hand, in recent years, it is undeniable that the introduction of a contact detection type input device in which a contact detection sensor made of a transparent electrode is arranged on a glass substrate has led to the explosive spread of today's mobile terminals. However, many of the conventional mobile terminals have a structure in which a contact detection type input device made of a glass substrate is attached to the surface of the display screen. For this reason, not only is there an increase in cost due to an increase in the number of parts compared to the case where the contact detection type input device is not attached, but also an increase in the weight and thickness of the mobile terminal, and an increase in power consumption to compensate for the decrease in the brightness of the display screen. Therefore, if the contact detection type input function can be provided inside the display screen, it is expected to simultaneously achieve cost reduction, weight reduction, thinning, and low power consumption of the mobile terminal, leading to further promotion of its spread.

[0005] Currently, a contact detection type input method with a built-in display screen that incorporates a contact detection type input function capable of identifying multi-point simultaneous contact positions inside the display screen has been put into practical use and introduced into some smartphones, and the technology has been disclosed (Patent Document 1).

[0006] However, this technology was only applicable to screen sizes of 7 inches or less. Therefore, there is a strong demand for the development of a technology that incorporates a contact detection input function inside the display screen for screen sizes exceeding 7 inches. The invention described in this specification meets this demand by providing a contact detection input function technology capable of highly sensitive multi-point simultaneous contact position identification that enables non-contact operation. Also, the technology that forms the basis of the present invention has already been disclosed (Patent Document 2).

Prior Art Documents

Patent Documents

[0007]

Patent Document 1

Patent Document 2

Summary of the Invention

Problems to be Solved by the Invention

[0008] In a mobile terminal, operations such as enlarging or reducing a displayed image are generally performed by multi-point simultaneous contact. Therefore, it is essential for a contact detection input device introduced into a mobile terminal to have a function of individually identifying and detecting multi-point simultaneous contact points on the screen. To enable this, a detection capacitance variation type has been used in the conventional contact detection input method incorporated in the display screen.

[0009] The outline of the detection principle of contact events in a capacitance-variable contact detection input method will be described below with reference to the drawings. FIG. 1 is a plan view showing the schematic structure of a capacitance-variable contact detection input device. On a glass substrate, square electrodes made of transparent electrodes are densely laid out with gaps. Adjacent electrodes whose diagonals form the vertical or horizontal direction are connected to each other and integrated, and each forms an independent electrode line. As a result, the electrode lines are arranged in a substantially lattice pattern, and at each intersection, they overlap via an insulating film. Here, for convenience, the numbers of the electrode lines integrated in the vertical direction are respectively X1, X2, ···, Xn, and the numbers of the electrode lines integrated in the horizontal direction are respectively Y1, Y2, ···, Ym.

[0010] Each electrode line Xj (j = 1, 2, ···, n) and Yi (i = 1, 2, ···, m) is independently connected to a control circuit, and the control circuit connected to each electrode line Xj (j = 1, 2, ···, n) constantly measures the current. In contrast, the control circuits connected to each electrode line Yi (i = 1, 2, ···, m) are selected in a time-division manner in the order of Y1, Y2, ···, Ym, and an alternating voltage is applied to the selected electrode line Yi (i = 1, 2, ···, m).

[0011] FIGS. 2 to 5 are conceptual diagrams showing the contact detection principle in a capacitance-variable contact detection input method. Here, as an example, a cross-section of a location where the respective electrode portions are arranged with gaps near the intersection of electrode line X1 and electrode line Y1 is schematically shown. Electrode line Y1 is connected to an AC power source, and for example, a ground potential is supplied during the non-selection period. Electrode line X1 is grounded via an ammeter, and the potential of electrode line X1 can be regarded as being always fixed at the ground potential. This ammeter is arranged for convenience and is not actually placed at this site. It schematically shows that the control circuit measures the current flowing through this site. Here, since the current flowing through this part changes due to the contact event, the contact event can be detected by measuring this current, so it will be called the contact event detection current or simply the detection current.

[0012] Figure 2 shows the case where the electrode line Y1 is non - selected and the finger is not in contact. At this time, the potentials of the electrode part of the electrode line X1 and the electrode part of the electrode line Y1 can both be regarded as equal to the ground potential, and no electric force lines are generated between the two electrode parts. Figure 3 shows the case where the electrode line Y1 is selected and an alternating voltage is applied, and the finger is not in contact with the glass substrate. At this time, a potential difference occurs between the electrode part of the electrode line X1 and the electrode part of the electrode line Y1, electric force lines are generated between the two electrode parts, and a detection capacitance is formed between the two electrode parts. Due to the alternating voltage applied to the electrode line Y1 through this detection capacitance, a current flows through the electrode line X1. Figure 4 shows the case where the electrode line Y1 is non - selected and the finger is in contact with the glass substrate. Since the human body can be regarded as grounded to the ground through shoes etc., the potential of the finger can be regarded as the ground potential. At this time, similar to the case of Figure 2, the potentials of the electrode part of the electrode line X1 and the electrode part of the electrode line Y1 can both be regarded as equal to the ground potential. That is, there is no potential difference between the two electrode parts and the finger, no force lines are generated, and no current flows through the electrode line X1. Figure 5 shows the case where the electrode line Y1 is selected and an alternating voltage is applied, and the finger is in contact. At this time, similar to the case of Figure 3, a potential difference occurs between the electrode part of the electrode line X1 and the electrode part of the electrode line Y1, and electric force lines are generated between the two electrode parts, but a part of the electric force lines is directed towards the finger. Therefore, compared with the case of Figure 3 where the finger is not in contact, the detection capacitance formed between the two electrode parts becomes smaller, and the current flowing through the electrode line X1 also decreases.

[0013] Thus, in the capacitance - variation - type contact - detection input method, even when the finger is not in contact, an alternating current flows through the electrode line Xj (j = 1, 2, ···, n) for the time - division selection of the electrode line Yi (i = 1, 2, ···, m), which is called the basic current. And for example, when a finger contacts near the area where the electrode part of the electrode line Yi (i = 1, 2, ···, m) and the electrode part of the electrode line Xj (j = 1, 2, ···, n) face each other through a gap, the current flowing through the electrode line Xj (j = 1, 2, ···, n) is displaced from the basic current only when the electrode line Yi (i = 1, 2, ···, m) is selected. Therefore, by separating and measuring this displaced current from the basic current, contact detection is performed. Thereby, even when the finger makes multi - point simultaneous contact, the contact position can be identified.

[0014] Here, regarding the basic principle of detecting current measurement flowing through the electrode line X1, that is, the method of separating the displacement component of the current due to a finger touching the glass substrate in the control circuit, the stages of the current change processed by the control circuit will be described below with reference to the drawings. First, consider the state where the finger is not in contact with the glass substrate. The current flowing through the electrode line X1 transmitted to the control circuit in this case, that is, the basic current, is shown in FIG. 6. Since this is a current caused by the AC voltage applied to the electrode line Y1, it is an alternating current. As the first stage of processing for this alternating current, the alternating current is converted into a voltage signal by current-voltage conversion. The converted voltage waveform is shown in FIG. 7. Subsequently, rectification processing is performed. The voltage waveform after the rectification processing is shown in FIG. 8. Next, smoothing processing is performed. The smoothed voltage becomes a direct current, and its voltage waveform is shown in FIG. 9. This voltage is obtained by converting the basic current into a voltage signal, so it will be called the basic voltage. When a finger touches the glass substrate, as described above, the detected current decreases, so the voltage waveform after the smoothing processing becomes as shown in FIG. 10. That is, in the non-contact period of FIG. 10, the voltage corresponding to the basic current caused by the capacitance formed by the electric field generated between the electrodes is the same as the voltage in FIG. 9. However, in the contact period, a part of the electric force lines emitted from the electrode ends reaches the finger and a new capacitance is generated between the finger and the electrode, so the capacitance formed between the electrodes decreases. As a result, the current flowing through the electrode line X1 decreases, and consequently, the voltage in the contact period shown in FIG. 10 decreases. Next, by subtracting the basic voltage from the smoothed voltage, only the voltage component displaced by the contact of the finger can be separated, and this is shown in FIG. 11. Subsequently, FIG. 12 shows the state where polarity inversion and amplification processing are performed on the separated displacement voltage component. Here, FIG. 11 shows the case where the basic current can be completely subtracted. In this case, even if amplified at an arbitrary magnification, as shown in FIG. 12, only the amplified signal of the displacement voltage component caused by the contact event is included. Therefore, even if there is variation in the current caused by the contact event, as long as it is not extremely large, it is possible to easily perform the subsequent detection signal processing, and it is possible to realize highly sensitive contact event detection at low cost without requiring a complex processing circuit. Also, when there is no variation in the basic current, since the same value of the basic current can be subtracted uniformly across the entire screen, an even lower cost can be achieved.

[0015] However, when there is variation in the basic current, it is not possible to uniformly subtract the voltage corresponding to the same value of the basic current. Therefore, subtraction is performed using a look-up table that records the voltage corresponding to the basic current for each electrode wire associated with each inter-electrode capacitance. In this case, in order to create the look-up table, it is necessary to accurately measure the variation in the basic current, convert this into voltage and store it in memory, accurately read this out in a timely manner within the control circuit, and then accurately reproduce the voltage for subtraction. In this process, it is difficult to accurately reproduce the voltage for subtraction due to characteristic variations caused by manufacturing variations of each element constituting the control circuit, and the subtracted difference remains in the detection signal. This state is shown in FIG. 13. Since the voltage of this subtracted difference varies for each individual inter-electrode capacitance, it is transmitted as a detection signal in a state including new variations to the detection signal processing circuit in the next step. FIG. 14 shows a state where a signal including the variation of the difference voltage obtained by subtracting the voltage corresponding to the basic current and the variation of the displacement voltage component in the case where there is variation in the current caused by the contact event is transmitted to the signal processing circuit in the next step after the processes of polarity inversion and amplification are performed. The thick line indicates the displacement voltage component caused by the contact event in the case where there is no variation. From FIG. 14, even when there is no variation in the current caused by the contact event but there is variation in the basic current, if the difference between the displacement voltage component and the differential voltage caused by the basic current is small, the displacement voltage component cannot be easily detected by the control circuit as the voltage change caused by the contact event, and it is necessary to perform more complex signal processing, increasing the cost of the control circuit. Also, when the difference between the two voltages is small, there may be a case where the contact event cannot be detected even if an expensive circuit performing complex processing is used. In particular, when the variation in the current itself caused by the contact event is large, this situation is exacerbated, the contact event cannot be detected, and a further decrease in detection sensitivity occurs.

[0016] The displacement voltage component separation method described here describes the basic principle and is not limited to the method described here, and various methods have been proposed. For example, the case where amplification processing is performed at the stage shown in FIG. 12 was described as an example. In this amplification processing, it was necessary to use an operational amplifier or the like. As an alternative, there is a method of simplifying the control circuit and reducing costs by using an integrating circuit. In this case, for example, as shown in FIG. 7, after converting to an AC voltage signal, first, rectification processing is performed as shown in FIG. 15, and then the voltage signal is divided into a positive polarity and a negative polarity, and the polarity of the negative polarity voltage signal is inverted. At this time, as shown in FIG. 16, the two divided voltage signals are shifted by a half cycle from each other. Next, when the two voltage signals divided by the integrating circuit are combined, the voltage is amplified stepwise with the passage of time during the selection period as shown in FIG. 17. In this case, the higher the frequency of the applied AC voltage applied to the detection capacitor, the higher the amplification factor. Since the amplified voltage signal also includes the basic current amplified at the same ratio as in the case described with reference to FIGS. 6 to 14, it is similarly necessary to subtract the voltage signal amplified during the non-contact period as the voltage caused by the basic current. That is, similar to the case described with reference to FIGS. 6 to 14, if the basic current can be completely removed, high detection sensitivity can be obtained even if there is some variation in the current caused by the contact event. Conversely, when the basic current cannot be completely removed, such as when there is a basic current variation, an increase in the differential voltage caused by the basic current leads to a decrease in detection sensitivity and an increase in the control circuit cost. Furthermore, when there is a variation in the current caused by the contact event, it promotes a decrease in detection sensitivity and an increase in the control circuit cost. The separation methods of these displacement voltage components described with reference to FIGS. 6 to 17 are not specific to the detection capacitance variation type, but can be widely used for all methods of detecting contact by current change. Furthermore, various signal processing methods for contact event detection performed after separating the displacement voltage component have also been proposed. However, since these signal processing methods do not affect the present invention, no further explanation will be given.

[0017] Also, in the capacitance-varying type contact detection input method described with reference to FIGS. 1 to 5, an alternating voltage is applied to the electrode portion of the electrode line Yi (i = 1, 2, ···, m), which is one of the electrodes constituting the detection capacitance, and the potential of the electrode line Xj (j = 1, 2, ···, n), which is the other electrode, and the wiring connected thereto is set to the ground potential, and the current flowing through the wiring at the ground potential is measured. This is called low-side current detection and can be implemented by a simple current measurement circuit, and the cost of the circuit can be suppressed.

[0018] By the way, since the electrode portions of the electrode lines Yi (i = 1, 2, ···, m) and the electrode portions of the electrode lines Xj (j = 1, 2, ···, n) are both made of thin-film electrodes, the area where the two electrode portions face each other is extremely small, and the capacitance formed between the two electrode portions is small. Also, when a finger touches the glass substrate, a part of the electric lines of force generated between the two electrodes reaches the finger through the glass substrate having a thickness on the order of sub-millimeters, which is larger than the distance between the two electrode portions on the order of microns. Therefore, the change in the capacitance formed between the electrode portions due to the contact of the finger is very small. For this reason, the amount of change in the current flowing through the electrode lines Xj (J = 1, 2, ···, n) caused by the contact of the finger with the glass substrate is also very small.

[0019] Generally, when an alternating voltage is applied to a capacitance and an alternating current flows through the wiring, it is widely known that the current flowing increases as the frequency of the applied voltage increases. Therefore, in order to obtain sufficient sensitivity for finger contact detection, it is necessary to set a high frequency for the alternating voltage applied to the selected electrode line Yi (i = 1, 2, ···, m).

[0020] Here, since FIGS. 2 to 5 illustrate the concept of the current detection principle, only the electrode portions of the electrode line X1 and the electrode line Y1 are shown. However, when actually applied to a mobile terminal or the like as a contact detection type input method with a built-in display screen, these electrode lines and electrode portions cross the wirings and electrodes for the display function via an insulating film, and capacitances are formed at these crossing portions. In particular, when the capacitance of the electrode line Xj (j = 1, 2, ···, n) on the side for measuring the current increases, the current caused by the capacitance formed at the crossing portion increases, that is, the basic current increases. As a result, the ratio of the displacement current to the basic current becomes even smaller and separation becomes difficult, and a further increase in frequency is required.

[0021] However, generally, as the frequency of the applied voltage is increased, the current will eventually turn to decrease, and when the CR time constant determined by the resistance and capacitance of the wiring increases, the frequency at which the current turns to decrease will decrease. That is, there is a limit to the increase in frequency for enhancing the detection sensitivity. For this reason, in the detection type input method of the detection capacitance variation type, there is an upper limit applicable to the screen size and the number of pixels due to the constraint by the CR time constant of the wiring. The upper limit of the screen size when applying the detection capacitance variation type as a contact detection type input method with a built-in display screen is approximately 7 inches for full-spec high-definition with a pixel number of 1920×1080. As a result, the contact detection type input method of the detection capacitance variation type with a built-in display screen is only implemented for some smartphones with a small screen size.

[0022] In order to apply the contact detection type input method with a built-in display screen to a screen size exceeding 7 inches, it is necessary to reduce the applied voltage frequency by obtaining a detection current larger than that of the detection capacitance variation type, thereby relaxing the constraint conditions due to the time constant of the electrode line through which the detection current flows. For this purpose, it is possible to introduce a detection capacitance timely formation type that can obtain a detection current large enough to enable non-contact contact event detection instead of the detection capacitance variation type.

[0023] The outline of the detection principle of contact events in the contact detection type input method with timely formation of detection capacitance will be described below with reference to the drawings. FIG. 18 is a plan view showing the schematic structure of a contact detection type input device with timely formation of detection capacitance. This is the same as the schematic structure plan view of the contact detection type input method with capacitance variation shown in FIG. 1, except that sequential time-division selection is not performed on the electrode lines Yi (i = 1, 2, ···, m), that is, only the control method is different for the same structure.

[0024] FIGS. 19 to 20 are conceptual diagrams showing the contact detection principle in the contact detection type input method with timely formation of detection capacitance. Here, as an example, a cross-section of a portion where the respective electrode portions are arranged with a gap in the vicinity of the intersection of the electrode line X1 and the electrode line Y1 is schematically shown. The electrode line X1 and the electrode line Y1 are connected to an AC power source synchronized with each other, and the potentials of the electrode portion of the electrode line X1 and the electrode portion of the electrode line Y1 are set to be always equal. As a result, no electric field lines are generated between the two electrodes, and no capacitance is formed. Also, although the electrode portion of the electrode line X1 and the electrode portion of the electrode line Y1 are shown as being connected to the AC power source via ammeters in the figure, these ammeters are arranged for convenience and do not actually measure the detection current at that location. Instead, it schematically shows that the control circuit constantly measures the detection current flowing through that part.

[0025] FIG. 19 shows the case where the finger is non-contact with the glass substrate. As described above, no electric field lines are generated between the two electrode portions, and no capacitance is formed. FIG. 20 shows the case where the finger is in contact with the glass substrate. At this time, since the potentials of the two electrode portions and the finger are different, electric field lines are generated from the two electrodes toward the finger, a detection capacitance is formed between the two electrodes and the finger, and a current flows through the electrode line X1 and the electrode line Y1. Since the detection capacitance of the contact detection type with timely formation is formed in a region approximately equal to the overlapping area between the electrode and the finger in this way, it becomes sufficiently large compared to the case of the capacitance variation type. As a result, the detection current flowing through the two electrodes also becomes sufficiently large, and a high detection sensitivity that can be detected even in a non-contact state is obtained. Therefore, the frequency of the AC power source can be reduced, and it becomes possible to greatly increase the upper limit of the applicable screen size due to the CR time constant of the electrode line.

[0026] However, in the contact detection method of the detection capacitance timely formation type described here, since a large detection capacitance is obtained by setting the potentials of both electrode portions to be always equal, it is not possible to perform driving for sequentially selecting each electrode line in a time-division manner, and it is not possible to identify the contact points in the case of multi-point simultaneous contact. In order to solve this problem, as shown in FIG. 21, there is a method of separating each electrode portion from the electrode line to form detection electrodes 120a, 120b, 120c, ···, and connecting each detection electrode and the connection terminal of the control circuit one-to-one. When the sensor device has functions other than the sensor, each of the detection electrodes 120a, 120b, 120c, ··· becomes an aggregate of fine sensor electrodes, and each sensor electrode is arranged in an active circuit region having other functions. That is, each of the detection electrodes 120a, 120b, 120c, ··· is an aggregate of sensor electrodes in an active unit that is an aggregate of active circuit regions, or each sensor electrode is electrically connected to each other to form an integrated electrode.

[0027] Here, since FIGS. 19 to 20 show the concept of the current detection principle, only the electrode portions of the electrode line X1 and the electrode line Y1 are illustrated. However, when actually applied as a contact detection type input method with a display screen built-in type in a mobile terminal or the like, these electrode lines and electrode portions cross the wirings and electrodes for the display function via an insulating film, and capacitances are formed at these crossing portions. Since an alternating voltage is applied to the electrode lines Yi (i = 1, 2, ···, m) and the electrode lines Xj (j = 1, 2, ···, n), currents flow from the electrode lines Yi (i = 1, 2, ···, m) and the electrode lines Xj (j = 1, 2, ···, n) toward the wirings and electrodes for the display function through this crossing capacitance. That is, similar to the case of the detection capacitance fluctuation type, a basic current flows through the electrode lines Yi (i = 1, 2, ···, m) and the electrode lines Xj (j = 1, 2, ···, n) even when a finger is not in contact with the glass substrate. Therefore, the contact is detected by separating and measuring the current displaced by the contact of the finger with the glass substrate from the basic current.

[0028] Also, in the contact detection type input method of the detection capacitance timely formation type described here, both the electrode lines Yi (i = 1, 2, ···, m) and the electrode lines Xj (j = 1, 2, ···, n) measure the current flowing through the wiring to which an alternating voltage is applied. This is a method called high-side current detection, which has a more complex circuit configuration compared to the low-side current detection method and involves a significant cost increase.

[0029] [Contact Detection Type Input Method of Detection Capacitance Timely Formation Type with 1:1 Connection] A conventional embodiment in which the contact detection type input function of the detection capacitance timely formation type, which enables the identification of the multi-point simultaneous contact position by connecting the detection electrode and the control circuit connection terminal one-to-one, is applied to the display screen of a fringe field switching mode liquid crystal display device will be described below with reference to the drawings. The fringe field switching mode liquid crystal display device is one of the typical high viewing angle conversion technologies widely used today in liquid crystal display devices. The technology is irrelevant to the present invention, and when applying the present invention to a liquid crystal display device, it is not limited to the fringe field mode liquid crystal display device. Therefore, a detailed description of the fringe field mode liquid crystal display device will not be given.

[0030] First, after depositing a 300-nm-thick MoW alloy on the first glass substrate by sputtering, the scanning signal line 100 is processed into a predetermined shape through a photolithography process. A plan view of the pixel at this time is shown in FIG. 22. Next, after depositing a 40-nm-thick transparent electrode layer made of indium and tin oxide (hereinafter referred to as ITO) by sputtering, the pixel electrode 101 is processed into a predetermined shape through a photolithography process. A plan view of the pixel at this time is shown in FIG. 23. Next, 300 nm of SiO2 is deposited as a gate insulating film by chemical vapor deposition, and then 10 nm of InGaZnO4 is deposited by sputtering. Subsequently, InGaZnO4 is processed into a semiconductor layer 102 of the transistor in a predetermined shape through a photolithography process. A plan view of the pixel at this time is shown in FIG. 24. Next, after depositing 200 nm of SiO2 by chemical vapor deposition, it is processed into a predetermined shape through a photolithography process to form a channel protection layer 103 of the semiconductor layer. A plan view of the pixel at this time is shown in FIG. 25. Next, through a photolithography process, SiO2 is processed into a predetermined shape to expose the MoW alloy layer at the end of the scanning signal line, which becomes the scanning signal power supply electrode, i.e., the control circuit connection terminal, and a contact hole 104 for connecting the pixel electrode 101 and the transistor is formed. A plan view of the pixel at this time is shown in FIG. 26. Next, after depositing 300 nm of AlNd alloy by sputtering, it is processed into a predetermined shape through a photolithography process to form an electrode 105 that becomes the source or drain of the transistor, an electrode 106, a display signal line 107, and a detection signal lead-out line 108 for transmitting the detection signal to the control circuit. A plan view of the pixel in which the detection signal lead-out line is arranged and the detection electrode is connected to the detection signal lead-out line 108 is shown in FIG. 27. Next, 300 nm of SiO2 is deposited as an interlayer insulating film by chemical vapor deposition, and then 3 μm of acrylic resin is coated and deposited. Subsequently, it is processed into a predetermined shape through a photolithography process to expose the respective power supply electrode portions located at the ends of the scanning signal line 100, the display signal line 107, and the detection signal lead-out line 108, i.e., the metal layers of the control circuit connection terminal portions, and a contact hole 109 for connecting the detection signal lead-out line 108 and the detection electrode is formed. A plan view of the pixel in which the detection signal lead-out line is arranged and the detection electrode is connected to the detection signal lead-out line 108 is shown in FIG. 28.Next, after forming a 40-nm ITO film by sputtering, the detection electrode 110 is processed into a predetermined shape having a slit 112 corresponding to the fringe field switching mode through a photolithography process. A plan view of a pixel in which the detection signal lead-out line at this time is arranged and the detection electrode 110 and the detection signal lead-out line 108 are connected is shown in FIG. 29. Finally, a liquid crystal layer is sandwiched together with a counter substrate on which a color filter is arranged on a second glass substrate, and a control circuit is connected to complete the product. Needless to say, the manufacturing method of this embodiment of the prior art is not limited to the method described here.

[0031] In the detection capacitance timely formation type contact detection input method of the display screen built-in type, depending on the resolution of the contact detection, the detection electrodes included in a plurality of pixels belonging to an active unit of approximately 5 mm square are electrically connected and integrated to function as one huge detection electrode. FIG. 30 is a schematic plan view showing the arrangement of pixels constituting one active unit 111. The detection electrodes 110 of each pixel are connected and integrated with the detection electrodes of adjacent pixels above, below, left, and right. Only one detection signal lead-out line 108 is arranged in the active unit 111, and the integrated detection electrode 110 is connected to the detection signal lead-out line 108 through a contact hole 109.

[0032] FIG. 31 is a conceptual diagram showing a state in which the detection electrodes electrically integrated in the active unit are connected to the control circuit connection terminals one-to-one by the detection signal lead-out lines. Since the pixels are arranged in a matrix in the liquid crystal display panel 125, the detection electrodes 120 electrically integrated and indicated by a one-dot chain line arranged in the active unit are also arranged in a matrix. Each of the electrically integrated detection electrodes 120 is connected to one end of the detection signal lead-out line 121 at an electrical connection point 122 schematically shown by a black circle, and the other end is connected to a control circuit connection terminal 123 schematically shown by a white circle. Each control circuit connection terminal 123 is connected to a control circuit 124.

[0033] When the frame frequency is 60Hz, which rewrites the screen display 60 times per second, the length of the frame period, which is the display time of one screen, is approximately 16.7ms. In the contact detection input method with an in-display screen system, about 75% of the frame period is set as the display operation period, during which the pixels operate for screen display, and the remaining about 25% of the period is set as the sensor operation period, during which the pixels operate for contact detection. In FIG. 8, particularly in both periods, the detection electrode 110 has its function greatly switched by the control circuit via the detection signal lead-out line 108.

[0034] During the display operation period, the common potential is supplied to the detection electrode 110 for the display operation. Then, a display signal potential is applied to the pixel electrode 101 from the display signal line 107 via a transistor so that the difference between the potential of the detection electrode 110 and the potential of the pixel electrode 101 becomes a predetermined voltage. By changing the alignment state of the liquid crystal molecules to a predetermined alignment state by the electric field generated at this time, the transmittance of the light incident on the liquid crystal layer is controlled to perform the display operation.

[0035] During the sensor operation period, as shown in FIG. 31, the active units arranged in a matrix are sequentially selected one row at a time in a time-division manner. That is, an alternating voltage is applied to each integrated detection electrode 120 arranged in the same selected row direction from the control circuit 124 via the control circuit connection terminal 123 and the detection signal lead-out line 121. As a result, as described with reference to FIGS. 19 to 20, a current flows through the detection signal lead-out line 121 according to the detection capacitance formed between the integrated detection electrode 120 and a finger or the like due to a contact event, and the control circuit 124 measures this current via the control circuit connection terminal 123 to detect the contact event. Also, a common potential is supplied to each integrated detection electrode 120 arranged in the non-selected row via the corresponding detection signal lead-out line 121 and control circuit connection terminal 123, and the same display operation as the display period is performed.

[0036] Figures 32 to 33 are schematic cross-sectional views of pixels arranged in the selected rows during the sensor operation period, showing the detection operation principle of this embodiment following FIGS. 19 to 20. A detection electrode 110 having a slit corresponding to the fringe field switching mode is disposed above the pixel electrode 101 with an insulating film layer 132 including a gate insulating film interposed therebetween, and a counter substrate 130 made of a liquid crystal layer 131 and glass is disposed above it. A galvanometer and an AC power source are connected in series to the detection signal lead 135, which is for conveniently showing that high-side current detection is performed by a control circuit. Here, since the potential of the pixel electrode 101 is the same as that during the display operation period and different from the potential of the detection electrode 110, an electric line of force is generated between the two electrodes, but it is omitted to avoid complication of the figure.

[0037] Figure 32 shows the case where the finger is non-contact. As already described, an AC voltage is applied to the detection signal lead 135 and the detection electrode 110, so a current flows through a capacitance formed between the detection electrode 110 and the pixel electrode 101. That is, a basic current flows through the detection signal lead 135. Figure 33 shows the case where the finger is in contact. The finger 136, that is, the human body can be regarded as having a ground potential. The electric line of force 133 emitted from the detection electrode 110 concentrates on the finger 136, and a detection capacitance 134 is formed between the detection electrode 110 and the finger 136. As a result, the current flowing through the detection signal lead 135 is displaced according to the detection capacitance 134 formed between the detection electrode 110 and the finger 136. The contact event is detected by measuring this current displacement with a control circuit.

[0038] Here, when the frame frequency is 60 Hz and the sensor operation period is 25%, the sensor operation period is approximately 4 ms, which is almost the same as the relaxation time of the display luminance change caused by the displacement of the liquid crystal molecules from the white display alignment state to the black display alignment state. It is widely known that the relaxation time of the display luminance change between intermediate gray levels is even longer. And the selection period of the active units arranged in each row direction corresponds to the sensor operation period divided by the number of active units arranged in the column direction. That is, the selection period of each active unit is sufficiently shorter than the relaxation time of the display luminance change due to the change in the voltage applied to the liquid crystal layer, and the change in the display luminance during the selection period is sufficiently small and can be ignored. That is, it can be regarded as having no influence on the image quality.

[0039] As described above, this embodiment provides a technology that enables the application of a contact detection type input method with an in - built display screen capable of identifying multiple - point simultaneous contact positions for screen sizes exceeding 7 inches, which was a conventional application limit, by using the detection capacitance timely formation type.

[0040] However, since the integrated detection electrode 120 and the control circuit connection terminal 123 in the active unit 111 must be connected one - to - one, a very large number of control circuit connection terminals 123 are required. For example, when the size of the active unit is 5 mm square, in a full - spec high - definition display with a screen size of 7 inches, the number of control circuit connection terminals 123 exceeds 500, and one dedicated driver IC is required. And the number of control circuit connection terminals 123 increases in proportion to the square of the screen size. Furthermore, for the display industry, which strongly demands reducing the number of driver ICs used even by one for cost reduction, the productization of a technology with an increasing number of control circuit connection terminals 123 is not practical.

[0041] Furthermore, although not shown in FIG. 31 to avoid complication, in reality, at least the power supply electrodes of the display signal lines 107 are arranged in the gaps in the arrangement of the control circuit connection terminals 123. That is, as the number of control circuit connection terminals 123 increases, the pitch of the control circuit connection terminals 123 and the power supply electrodes of the display signal lines 107 decreases, resulting in a decrease in the connection yield with each control circuit, or the need to increase the outer size of the liquid crystal display panel 125 to secure the arrangement locations of the control circuit connection terminals 123 and the power supply electrodes of the display signal lines 107, etc. problems become apparent. As a result, the practicality of the technology shown in this embodiment is increasingly reduced.

[0042] As a means of suppressing and reducing the increase in the number of control circuit connection terminals 123 by connecting the integrated detection electrodes 120 and the control circuit connection terminals 123 in the active unit 111 on a one-to-one basis, the introduction of a multiplexer can be considered. A multiplexer is an element circuit that outputs only the selected signal from among a plurality of input signals, and a transistor is used as the first switching element for signal selection.

[0043] [Contact Detection Input Method of Detection Capacitance Timing Formation Type with Multiplexer Introduction] By arranging a multiplexer between the region where the integrated detection electrodes 120 are arranged in a matrix and the region where the control circuit connection terminals 123 are arranged in FIG. 31, a detection capacitance timing formation type contact detection input function capable of identifying multiple simultaneous contact positions with reduced control circuit connection terminals is incorporated into the display screen of a prior art liquid crystal display device in the fringe field switching mode. The outline of this embodiment will be described below with reference to the drawings.

[0044] FIG. 34 is a schematic plan view showing the operating principle of this embodiment, and only the multiplexer 140 portion composed of transistors 141 is shown by an equivalent circuit. Here, as an example, the active units are arranged in 3 rows and 3 columns, and the multiplexer 140 is composed of the same number of transistors 141 as the active units. An integrated detection electrode 120 is arranged in each active unit. The gate electrodes, which are the control electrodes of the respective transistors 141 corresponding to the active units arranged in the row direction, are connected to the selection signal line 145, and the end of the selection signal line 145 is connected to the control circuit connection terminal 146 of the multiplexer. Selection signals are sequentially supplied to the respective selection signal lines 145 corresponding to the active units arranged in the column direction from the control circuit in a time-division manner via the control circuit connection terminal 146 of the multiplexer, and the respective transistors 141 connected to the same selection signal line 145 are synchronously switched between the ON state and the OFF state. The integrated detection electrode 120 in each active unit is connected at an electrical connection point 122 to one end of the detection signal lead 121, and the other end of the detection signal lead 121 is connected to one electrode 144 that functions as the source or drain of the corresponding transistor 141. The other electrode 143 that functions as the source or drain of each transistor is connected to the control circuit connection terminal 123 shared by the integrated detection electrodes 120 in the active units arranged in the column direction.

[0045] The integrated detection electrodes 120 in the active units arranged in the column direction that share the same control circuit connection terminal 123 are each applied with an alternating voltage from the control circuit 124 via the control circuit connection terminal 123 when the corresponding transistor 141 is in the ON state, and at the same time, detection of a contact event is performed based on measurement of the current flowing through the corresponding detection signal lead 121. As a result, the introduction of the multiplexer enables a significant reduction in the number of control circuit connection terminals 123 to the same number as the number of columns of the active units.

[0046] However, in this embodiment, each integrated detection electrode 120 in each active unit arranged in a matrix and each transistor 141 constituting the multiplexer 140 are connected one-to-one. In particular, the same number of transistors 141 are arranged in the same direction corresponding to the active units arranged in the column direction. Therefore, in order to introduce the multiplexer 140, a new area for arranging the transistors 141 in the column direction of the active units must be secured. That is, if the multiplexer 140 is arranged in the frame area, which is the area between the arrangement area of the active units and the arrangement area of the control circuit connection terminal 123, the width of the frame area will increase. In mobile terminals and the like where it is strongly required to reduce the housing size as much as possible or to increase the display screen size as much as possible for the same housing size, the increase in the frame width is an extremely serious problem.

[0047] As a means of introducing a multiplexer without increasing the frame width, as shown in FIG. 35, arranging the transistors constituting the multiplexer inside the pixel can be mentioned. In FIG. 35, the integrated detection electrode 150 and the detection signal lead-out wire 152 are electrically connected at the electrical connection point 155 via the transistor 151. The transistors 151 arranged in the active units in the row direction are sequentially selected time-divisionally by the transistor control wiring 154, and the detection current flows from the integrated detection electrode 150 toward the control circuit connection terminal 153 through the selected transistor 151 that is in the ON state. As a result, it is possible to significantly reduce the number of control circuit connection terminals 153 to the same number as the number of columns of the active units without increasing the frame area width. However, in this case, since the transistor 151, which is a pattern irrelevant to the display function, is arranged in the pixel, a decrease in display performance such as a decrease in the aperture ratio is inevitable. Therefore, in the design, it is necessary to suppress the decrease in the aperture ratio as much as possible.

[0048] In addition, the integrated detection electrode 150 and the detection signal lead-out wire 152 cross the pixel electrode 101, the scanning signal line 100, etc. for the display operation, and capacitances are formed at their respective crossing portions. Therefore, when an AC voltage is applied from the control circuit to the integrated detection electrode 150 and the detection signal lead-out wire 152 via the control circuit connection terminal 153, current flows from the integrated detection electrode 150 and the detection signal lead-out wire 152 to the capacitances formed at these crossing portions and the like, and this becomes the basic current. The capacitance of the transistor 151 formed between the electrode of the transistor 151 connected to the detection signal lead-out wire 152 and the gate electrode of the transistor 151 is electrically connected to the detection signal lead-out wire 152, and this capacitance also generates a basic current. That is, it is considered that the introduction of a multiplexer increases the basic current. As described above, when the basic current flows, the detection sensitivity of the contact event decreases. When there is variation in the basic current, the detection sensitivity of the contact event decreases more comprehensively. Alternatively, it will lead to an increase in the cost of the control circuit to suppress the decrease in sensitivity. Furthermore, when attempting to increase the detection current by increasing the size of the transistor 151 that constitutes the multiplexer to improve the detection sensitivity, it not only promotes a decrease in the aperture ratio but also leads to a further increase in the basic current due to an increase in the crossing capacitance.

[0049] As a countermeasure, it is necessary to introduce a method for reducing the basic current. As this method, there is a method of introducing a differential circuit inside the control circuit and removing the basic current by taking the difference in the currents flowing through two adjacent detection signal lead-out wires. That is, when the finger is not in contact, the basic current is removed by taking the difference in the currents flowing through two adjacent detection signal lead-out wires 121, and the detection of the finger contact event is performed by measuring the change in current when the finger comes into contact with respect to the value of this differential current. However, with this method, the variation in the basic current cannot be removed, so it is necessary that there is no variation in the basic current flowing through each detection signal lead-out wire. As described above, when there is variation in the basic current, it becomes difficult to tolerate the variation in the detection current itself. Therefore, it is desirable that the variation in the basic current be as small as possible, preferably 1% or less.

[0050] For example, if there are variations in the area of each intersection capacitance formed by the detection signal lead-out wires, variations will occur in the basic current. Although there are several factors causing variations in the intersection capacitance, all of them are smoothly distributed, so there will be no extreme difference between the capacitances formed at intersections spaced 5 mm apart. However, according to the experience of the inventor of this patent, for example, when wet etching an aluminum-based film, the widths of two wires spaced 5 mm apart frequently have variations of about ±0.1 to 0.2 μm. In recent years, the display signal line width of substrates for display screens for mobile terminals is about 3 μm. When the wiring width varies by 0.1 μm with respect to a wiring width of 3 μm, the intersection area will vary by about 3%. When the intersection capacitance of the wiring is the main factor of the basic current, the variation in the basic current will be of the same order.

[0051] Such variations in line width due to processing accuracy in the manufacturing process not only become a factor causing variations in the basic current, but at the same time, variations in the capacitance and resistance of the detection signal lead-out wires, variations in the capacitance and resistance due to size variations of the transistor 151, and variations in the detection capacitance due to variations in the detection electrodes also occur. Therefore, the detection current itself will have variations. As described above, when there are variations in the detection current itself in a state where the basic current cannot be completely subtracted, it will lead to a decrease in detection sensitivity and an increase in the cost of the control circuit for compensating this. As other important factors causing variations in the detection current itself, variations in transistor characteristics can be cited. The resistance at the interface between the semiconductor layer of the transistor and the source and drain electrodes is very sensitive to the interface state, but it is extremely difficult to precisely control the interface state in the manufacturing process. For this reason, it is difficult to make the characteristics of transistors arranged in adjacent pixels exactly the same as each other, and it can only be contained within a range of variations with a certain width, and this variation cannot be removed by a differential circuit.

[0052] The present invention has been made in view of the above points, and an object of the present invention is to provide a sensor device based on a contact detection type input method with a built-in display screen, which can be inexpensively realized for a screen size exceeding 7 inches, by reducing the base current and the variation of the base current more than before to significantly improve the sensitivity. Specifically, it is to newly derive and provide design conditions for realizing this.

Means for Solving the Problems

[0053] In order to realize a contact detection type input method with a built-in display screen applicable to a screen size exceeding 7 inches, it is first necessary to relax the constraints caused by the time constant determined by the resistance and capacitance of the current detection path derived from the screen size. Therefore, by using a detection capacitance timely formation type with a high detection sensitivity that enables non-contact detection and can obtain a large detection current, the frequency of the applied alternating voltage is reduced to relax the time constant constraint. In the detection capacitance timely formation type, in order to identify the multi-point simultaneous contact positions, the detection electrodes and the control circuit connection terminals are connected one-to-one. In order to reduce the increase in the number of control circuit connection terminals, which is a problem of this method, a multiplexer is introduced. Furthermore, as a countermeasure against the decrease in the contact event detection sensitivity due to the increase in the base current caused by the capacitance of the transistors constituting the multiplexer, a base current reduction mechanism is introduced. Since the base current is generated by applying an alternating voltage to the current detection path, the casing voltage application method proposed by the co-researchers of the present invention is used as a method to fundamentally eliminate the generation of the base current.

[0054] The casing voltage application method applies an alternating potential to the human body through the casing of a mobile terminal or the like that is touched by one hand, for example, to give an alternating potential to the other finger that touches the screen. As a result, it becomes possible to fix the potentials of the detection electrodes, detection signal lead-out lines, etc. constituting the current detection path to a reference potential, for example, the ground potential, and the generation of the base current can be fundamentally eliminated. Furthermore, since it is possible to realize a low-side current detection for measuring the current flowing through the fixed potential side path, cost reduction can be achieved by simplifying the control circuit.

[0055] Fig. 36 shows an equivalent circuit diagram used in the research leading to the present invention. The first region 200 represents a finger, and an alternating potential V is applied to the finger through the housing. The second region 201 represents a sensing electrode, and the capacitance C4 is the capacitance formed between the finger and the sensing electrode when the finger touches the glass substrate on the surface of the display screen. The third region 202 and the fourth region 203, one of which represents a sensing signal lead-out wire and the other represents a transistor constituting a multiplexer. The capacitances C1 and C2 and the resistances R1 and R2 in each region are connected to each other in a transmission line manner. The fifth region 204 represents a control circuit connection terminal, and it represents that the current I3 flowing through the resistance R2 in the fourth region 203 reaches the control circuit as a sensing current, and the resistance R2 is grounded by low-side current sensing.

[0056] In the initial stage of this research, various circuit configuration analysis models were studied for all possible parameters that could be considered as actual devices, such as the method of setting the time constant circuit model including the arrangement order of capacitances and resistances in the third region 202 and the fourth region 203. The equivalent circuit shown in Fig. 36 is the result of a series of studies and is the optimal equivalent circuit as an analysis model for this research.

[0057] Assuming that the angular frequency of the applied alternating potential in the first region 200 is ω and the imaginary unit is j, the following three equations, Equation (1) to Equation (3), hold for the currents I1, I2, and I3 in the equivalent circuit shown in Fig. 36. V = {1 / (jωC4)}I1 + {1 / (jωC1)}(I1 - I2) ······ (1) {1 / (jωC1)}(I1 - I2) = R1I2 + {1 / (jωC2)}(I2 - I3) ····· (2) {1 / (jωC2)}(I2 - I3) = R2I3 ······ (3) From Equations (1) to (3), the sensing current I3 transmitted to the control circuit is represented by the following Equation (4). I3 = α / (β + γ + 1) ······· (4) α = C4V〔{(C1 + C4)R1 + (C1 + C2 + C4)R2}ω^2 + {ω - C2(C1 + C4)R1R2ω^3}〕j β = (C2^2){(C1 + C4)^2}(R1^2)(R2^2)ω^4 γ = 〔{(C1 + C4)^2}(R1^2) + {(C1 + C2 + C4)^2}(R2^2) + 2{(C1 + C4)^2}R1R2〕ω^2 From Equation (4), the magnitude |I3| of the detected current I3 is expressed by the following Equation (5). |I3| = C4ωV / (β + γ + 1)^(1 / 2) ······ (5) Here, " ^ " represents the exponentiation symbol. For example, X^2 represents the square of X, and X^(1 / 2) represents the square root of X.

[0058] Figure 37 is a graph showing the frequency dependence of the magnitude |I3| of the detected current I3 obtained by the derived Equation (5) and by simulation using Spice for the case where C4 is 1 pF, C1 and R1 are 1 pF and 1 MΩ respectively, and C2 and R2 are 100 pF and 100 Ω respectively. The thick solid black line shows the result obtained from Equation (5), and the dashed white line shows the simulation result using Spice. The magnitude |I3| of the detected current I3 is in decibel notation. Spice is a circuit simulator with high analysis accuracy and is widely used worldwide. Since the results of both are in complete agreement, it can be said that Equation (5) has been correctly derived.

[0059] Here, consider the case of performing high-side current detection on the equivalent circuit of FIG. 36. At this time, the AC power supply in the first region 200 of FIG. 36 moves to the fifth region, and the first region representing the finger is grounded. The equivalent circuit in this case is shown in FIG. 38. Regarding the applied AC voltage frequency dependencies of the magnitude Low-|I3| of the current I3 in the low-side current detection shown in FIG. 36, the magnitude High-|I1| of the current I1 in the high-side current detection shown in FIG. 38, and the magnitude High-|I3| of the current I3, the simulation results by Spice are shown in FIG. 39. In FIG. 39, the thick black solid line indicates the magnitude Low-|I3| of the current I3 in the low-side current detection, the thin white dashed line indicates the magnitude High-|I1| of the current I1 in the high-side current detection, and the dashed-dotted line indicates the magnitude High-|I3| of the current I3 in the high-side current detection, respectively. The magnitude High-|I3| of the current I3 in the high-side current detection corresponds to the current measured by the control circuit and includes the basic current flowing out to the ground through the capacitors C1 and C2. Therefore, it is larger regardless of the frequency than the magnitude Low-|I3| of the current I3 corresponding to the measured current in the control circuit in the low-side current detection where no basic current is generated. On the other hand, the magnitude High-|I1| of the current I1 in the high-side current detection is the one excluding the basic current and corresponds to the current obtained by subtracting the basic current by introducing a basic current reduction mechanism such as a differential circuit. The current- applied AC voltage frequency characteristics of the magnitude High-|I1| of the current I1 in this high-side current detection and the magnitude Low-|I3| of the current I3 in the low-side current detection are completely identical. That is, when performing high-side current detection on a current detection circuit in which a detection signal lead-out wire and a transistor are arranged between the detection electrode and the control circuit connection terminal, when removing the basic current from the measured current by introducing a basic current reduction mechanism such as a differential circuit, the current-applied AC voltage frequency dependency of the measured current after removing the basic current coincides with the current-applied AC voltage frequency dependency of the current measured by low-side current detection for the same current detection circuit. Thereby, the derived formula (5) representing the magnitude |I3| of the detection current I3 also holds in the case of high-side current detection where the basic current is removed from the measured current by introducing a basic current reduction mechanism such as a differential circuit.

[0060] Next, in the equivalent circuit of the low-side current detection shown in FIG. 36, when the third region 202 is used as a transistor and the fourth region 203 is used as a detection signal lead-out line, that is, when the transistor is arranged in the pixel, and when the third region 202 is used as a detection signal lead-out line and the fourth region 203 is used as a transistor, that is, when the transistor is arranged in the frame region, the results obtained from Equation (5) regarding the applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 are shown in FIG. 40. In FIG. 40, the solid line indicates the case where the transistor is arranged in the pixel, and the dashed line indicates the case where the transistor is arranged in the frame region. The magnitude |I3| of the detection current I3 is expressed in amperes. Here, the detection capacitance C4 is 1 pF, the capacitance and resistance of the transistor are 1 pF and 1 MΩ respectively, and the capacitance and resistance of the detection signal lead-out line are 100 pF and 100 Ω respectively. That is, when the transistor is arranged in the pixel, C1 = 1 pF, R1 = 1 MΩ, C2 = 100 pF, R2 = 100 Ω, and when the transistor is arranged in the frame region, C1 = 100 pF, R1 = 100 Ω, C2 = 1 pF, R2 = 1 MΩ. In FIG. 40, when the transistor is arranged in the pixel, the maximum value of the magnitude |I3| of the detection current I3 is larger. If the size of the transistor, that is, the capacitance and resistance of the transistor are the same, it shows that a higher detection sensitivity to the contact event can be obtained when the transistor is arranged in the pixel than when it is arranged in the frame region. The reason for this will be explained next.

[0061] First, as shown in FIGS. 37 and 40, the graph of the applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 can be considered to be divided into a low-frequency region where the magnitude |I3| of the detection current I3 increases monotonically with the increase in frequency, a mid-frequency region where the change in the magnitude |I3| of the detection current I3 is small with the increase in frequency, and a high-frequency region where the magnitude |I3| of the detection current I3 decreases monotonically with the increase in frequency. This suggests that in Equation (5) representing the magnitude |I3| of the detection current I3, the frequency dependence of the applied AC voltage on the magnitude |I3| of the detection current I3 varies greatly depending on each frequency region.

[0062] As a result of the research, from Equation (5), in the low-frequency region of the graph of the applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3, it is given by the following Equation (6): |I3| = C4ωV ······ (6) In the mid-frequency region, it is given by the following Equation (7): |I3| = C4V / {φ^(1 / 2)} ······ (7) φ = 〔{(C1 + C4)^2}(R1^2) + {(C1 + C2 + C4)^2}(R2^2) + 2{(C1 + C4)^2}R1R2〕ω^2 In the high-frequency region, it is given by the following Equation (8): |I3| = C4V / {C2(C1 + C4)R1R2ω} ······ (8) It was found that each can be approximated. Here, " ^ " represents the exponentiation symbol. For example, X^2 represents the square of X, and X^(1 / 2) represents the square root of X.

[0063] When the third region 202 is a transistor and the fourth region is a detection signal lead wire, the relationship R1 ≫ R2 holds. At this time, it was derived from the research that the approximate formula (7) for the mid-frequency region can be simplified as the following Equation (9). |I3| = C4V / {(C1 + C4)R1} ······ (9) Similarly, when the third region 202 is a detection signal lead wire and the fourth region is a transistor, the relationship R1 ≪ R2 holds, and at this time the approximate formula (7) for the mid-frequency region can be simplified as the following Equation (10). |I3| = C4V / {(C1 + C2 + C4)R2} ······ (10) Since both C1 and R1 in Equation (9) and C2 and R2 in Equation (10) represent the capacitance and resistance of the transistor, their respective values are the same, and C1 in Equation (10) is the capacitance of the detection signal lead-out wire. Therefore, the difference between Equation (9) and Equation (10) lies in the presence or absence of the capacitance of the detection signal lead-out wire in the denominator. For Equation (10) with the capacitance of the detection signal lead-out wire, the value of the magnitude |I3| of the detection current I3 becomes smaller. Therefore, when arranging the transistor constituting the multiplexer between the detection signal lead-out wire and the control circuit connection terminal, that is, when arranging it in the border area, a larger detection current can be obtained than when arranging it between the detection electrode and the detection signal lead-out wire, that is, when arranging it in the pixel, and the detection sensitivity of the contact event becomes higher.

[0064] Generally, the resistance of a transistor can be regarded as being proportional to L / W with respect to the length L and width W of the channel that serves as the charge transport path in the semiconductor layer constituting the transistor. Also, in order to minimize the capacitance of the transistor as much as possible, the channel length L is set to the minimum processing dimension of the manufacturing process. According to research, in order for the maximum value of the detection current obtained when the transistor is arranged in the border area in FIG. 40 to be equal to the maximum value when the transistor is arranged in the pixel, considering the change in capacitance due to the change in W with the channel length L fixed, the channel width W of the transistor arranged in the border area needs to be about 100 times that when the transistor is arranged in the pixel. As a result, it was obtained. For example, when the channel width when the transistor is arranged in the pixel is 10 μm, the channel width required when the transistor is arranged in the border area is 1 mm. Therefore, when arranging the transistor constituting the multiplexer in the border area, a significant increase in the border area width is inevitable, which is not practical. This is an extremely serious problem especially in mobile terminals where the border width is desired to be made as narrow as possible. This indicates that the transistor constituting the multiplexer is preferably arranged in the pixel. Therefore, unless otherwise specified in the following description, the transistor constituting the multiplexer is assumed to be arranged in the pixel, and C1 and R1 represent the capacitance and resistance of the transistor constituting the multiplexer, and C2 and R2 represent the capacitance and resistance of the detection signal lead-out wire, respectively.

[0065] Here, taking the case where the transistors constituting the multiplexer are arranged in pixels, when C4 = 1 pF, C1 = 1 pF, R1 = 1 MΩ, C2 = 100 pF, and R2 = 100 Ω, the current- applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 obtained from Equation (5) and the approximate equations (6), (8), and (9) for each frequency region is shown in FIG. 19. In FIG. 41, the thick gray solid line represents the current- applied AC voltage frequency dependence obtained from Equation (5), the short dashed line represents the approximate equation (6) for the low- frequency region, the dotted line represents the simplified approximate equation (9) for the mid- frequency region, and the long dashed line represents the approximate equation (8) for the high- frequency region. It can be seen that each approximate equation can approximate with good accuracy. Particularly in the mid- frequency region, since the relationship R1≫R2 holds, it coincides with the simplified approximate equation (9) with very good accuracy. However, since the simplified approximate equation (9) is a constant value with respect to frequency, it can be seen that although the frequency dependence in the mid- frequency region is convex upward according to Equation (5), the change in the current value with respect to the change in frequency is extremely small. From this, it is shown that the applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 in the mid- frequency region can be regarded as substantially constant throughout the mid- frequency region and can be regarded as the maximum value without any problem. That is, the approximate equation for the magnitude |I3| of the detection current I3 for the mid- frequency region can be regarded as representing the maximum value |I3|max of the detection current without any problem.

[0066] However, the relationship R1≫R2 or the relationship R1≪R2 does not always hold. For this reason, it is generally necessary to use Equation (7) which is not simplified as the approximate equation for the magnitude |I3| of the detection current I3 for the mid- frequency region. Also, when the simplified approximate equation holds in the mid- frequency region, the simplified approximate equation and the non- simplified approximate equation coincide. Here, the dash- dotted line in FIG. 41 represents the frequency at the intersection of the approximate equations for the low- frequency region and the mid- frequency region, and the double dash- dotted line represents the frequency at the intersection of the approximate equations for the mid- frequency region and the high- frequency region. The meanings of these will be explained later.

[0067] Next, a method for setting a design point in a graph of the current-applied AC voltage frequency dependence of the magnitude |I3| of the obtained detection current I3 will be described. In actual design, it is necessary to perform an optimal design that satisfies various design conditions simultaneously. In the design of a sensor device to which the present invention is applied, specifically, in the design of a sensor device based on a contact detection type input method with a built-in display screen that can be realized at low cost for a screen size exceeding 7 inches, it is particularly required to improve the detection sensitivity of contact events as much as possible. For this purpose, first, it is necessary to set the detection current to be as large as possible, and next, it is necessary to reduce the basic current as much as possible. In addition, in order to reduce the power consumption as much as possible, it is strongly required to reduce the frequency of the AC voltage applied to the detection capacitance as much as possible. Furthermore, an improvement in aperture ratio for maintaining or improving display performance, and an improvement in yield for reducing manufacturing costs due to improved productivity are also strongly required.

[0068] In actual design, there is a trade-off relationship where if the line width of the detection signal lead wire is made thinner, the aperture ratio increases, but the yield decreases due to an increase in the disconnection defect rate. Which of the aperture ratio and the yield is prioritized is determined by the product specifications, the processing accuracy of the manufacturing process, and the operating status of the disconnection repair equipment. Also, as the basic current component included in the detection current becomes smaller due to the introduction of means for reducing the basic current, even if the detection current decreases slightly, it becomes possible to easily obtain high detection sensitivity by amplifying the displacement voltage component caused by a contact event that is important as a detection signal. Therefore, in order to perform an optimal design under such circumstances, using the expression (5) that clearly represents the current-applied AC voltage frequency dependence of the magnitude |I3| of the already derived detection current I3 and the approximate expressions (6) to (8) for each frequency region, it is necessary to search for and set a design point that can obtain a relatively large detection current while satisfying these conditions well in balance. As a result, it is quite possible that the set design point is in the low or high frequency region and the obtained detection current is significantly smaller than the maximum value of |I3|.

[0069] Therefore, in the current-applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3, a method for clarifying the boundary of each frequency region is required. However, since the expression (5) representing the current-applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 is convex upward and smoothly continuous in the entire frequency range, there are no distinct extreme values or inflection points that would serve as the boundaries of the frequency regions. Thus, as an alternative, the intersection of the approximate expressions for each frequency region will be regarded as a pseudo-inflection point. Specifically, the intersection of the approximate expression (6) for the low-frequency region and the approximate expression (7) for the mid-frequency region will be the first pseudo-inflection point, and the intersection of the approximate expression (7) for the mid-frequency region and the approximate expression (8) for the high-frequency region will be the second pseudo-inflection point. At this time, if the angular frequency corresponding to the frequency of the first pseudo-inflection point is ω1 and the angular frequency corresponding to the frequency of the second pseudo-inflection point is ω2, it has been found that ω1 and ω2 are respectively represented by the following expressions (11) and (12). ω1 = φ^(-1 / 2) ······ (11) ω2 = {φ^(1 / 2)} / {C2(C1 + C4)R1R2} ······ (12) Here, " ^ " represents the exponentiation symbol. For example, X^2 represents the square of X, and X^(1 / 2) represents the square root of X. Also, φ is represented by the following expression. φ = {(C1 + C4)^2}(R1^2) + {(C1 + C2 + C4)^2}(R2^2) + 2{(C1 + C4)^2}R1R2 In FIG. 41, the dashed-dotted line represents the value obtained by converting the angular frequency ω1 with respect to the first pseudo-bending point into a frequency, and the two-dot chain line represents the value obtained by converting the angular frequency ω2 with respect to the second pseudo-bending point into a frequency. Thus, in order to obtain high detection sensitivity, it is preferable to set a large detection current. When this is set as the top-priority design condition, the design point is set in the mid-frequency region. Therefore, if the angular frequency of the design point is ω, the setting range of ω is ω1 ≤ ω ≤ ω2. However, when the constraints due to other design conditions are strong and the design point cannot be set in the mid-frequency region, it will be set in the low-frequency region or the high-frequency region. Both regions are substantially symmetric with respect to the mid-frequency region, and the range in which the detection current can be obtained is the same in both regions. That is, the applied AC voltage frequency for obtaining the same detection current is selected from two regions, the low-frequency region and the high-frequency region. Generally, since the power consumption increases as the frequency increases, when the requirement for low power consumption is strong, it is preferable to set the design point in the low-frequency region where the frequency is low as long as the obtained detection current is the same.

[0070] Conversely, when setting the design point in the high-frequency region, in addition to requiring high power consumption to obtain the same detection current, further adverse effects occur. Originally, it is desirable that the detection capacitance formed between the finger that is not directly above the detection electrode is small, and the detection current flowing due to this detection capacitance is sufficiently small. However, when a high-frequency voltage is applied to this small detection capacitance, the detection current flowing due to this small detection capacitance increases and becomes non-negligible. As a result, the resolution of the sensor sensitivity decreases. Therefore, it is not preferable to set the design point in the high-frequency region unless there are special circumstances. From the above, it is desirable to set the angular frequency ω of the design point in the range where ω ≤ ω2.

[0071] Next, in order to increase the detected current with respect to the set angular frequency ω, it is necessary to reduce the resistance R1 of the transistors constituting the multiplexer from the approximation formula (7) for the intermediate frequency range. As already described, in order to reduce the capacitance of the transistor, since the channel length L of the transistor is generally set to the minimum processing dimension in the manufacturing process, it is necessary to widen the channel width W in order to reduce the resistance R1 of the transistor. However, widening the channel width W while keeping the channel length L fixed necessarily increases the transistor capacitance C1, resulting in a lower detected current increase effect. Furthermore, since the size of the transistor is increased, the aperture ratio is lowered, and a decrease in display performance is inevitable. Therefore, a method of reducing the resistance of the transistor can be considered by dispersing and arranging the transistors constituting the multiplexer in a plurality of pixels and connecting them in parallel to the detection signal lead-out lines. In this case, since it is possible to reduce the size of each of the transistors to be dispersed, it is a very effective means also from the viewpoint of improving the aperture ratio. Furthermore, as already described, the variation in the detected current due to the variation in the transistor characteristics caused by the non-uniformity of the interface state between the semiconductor layer and the electrode of the transistor cannot be removed even by introducing a basic current reduction mechanism such as a differential circuit. However, by dispersing and arranging the transistors and connecting them in parallel to each other, an effect of averaging the variation in the transistor characteristics can be expected. Therefore, the calculation method of the detected current when the transistors constituting the multiplexer are dispersed in a plurality of pixels will be described below.

[0072] First, consider the case where a plurality of transistors constituting the multiplexer are dispersed and arranged with respect to a single detection signal lead-out line. As a result of research, the equivalent circuit in this case can be represented by the equivalent circuit shown in FIG. 42. FIG. 42 shows an AC power supply with a potential amplitude V in FIG. 36, a detection capacitance C4 composed of a finger and a detection electrode, and a set of n transistors each having a transistor capacitance C1 and a resistance R1 connected to the detection signal lead-out line. That is, it corresponds to arranging n uniform pixels each having the same value for the detection capacitance C4, the transistor capacitance C1, and the resistance R1.

[0073] In FIG. 42, since the capacitances C1 and C4 in each pair are connected to different nodes and thus not in parallel connection, the combined capacitance cannot be simply obtained for each capacitance. However, since the potentials of the corresponding nodes in each pair in FIGS. 36 and 42 are equal, the current I1 in FIG. 42 1 , I1 2 , ···, I1 n are equal to each other and coincide with the current I1 in FIG. 36. Therefore, I1 1 = I1 2 = ··· = I1 n = I 1 can be expressed as such. Similarly, the currents I2 in FIG. 42 1 , I2 2 , ···, I2 n are also equal to each other. If this is replaced with I2 0 , then I2 1 = I2 2 = ··· = I2 n = I2 0 can be expressed as such. At this time, there is a relationship of 0 between the currents I2 and I2 I2 = nI2 0 being established. From these equations, equations (1) to (3) can be transformed as shown in the following equations (13 - 1) to (13 - 3). V = {1 / (jωC4)}I1 + {1 / (jωC1)}(I1 - I20) ······ (13 - 1) {1 / (jωC1)}(I1 - I2 0 ) = R1I2 0 + {1 / (jωC2)}(nI2 0 - I3) ······ (13 - 2) {1 / (jωC2)}(nI2 0 - I3) = R2I3 ······ (13 - 3) From equations (13 - 1) to (13 - 3), the detected current I3 is expressed by the following equation (14). I3 = α1 / (β + γ1 + 1) α1 = nC4V〔{(C1 + C4)R1+(nC1 + C2 + nC4)R2}ω^2+{ω - C2(C1 + C4)R1R2ω^3}〕j γ1={(C1 + C4)^2}(R1^2)+〔{(nC1 + C2 + nC4)^2}(R2^2)+2n{(C1 + C4)^2}R1R2〕ω^2·······(14) Here, Equation (14) is obtained by replacing C1, C4, and R1 in Equation (4) with C1 → nC1 C4 → nC4 R1 → R1 / n respectively. This replacement corresponds to replacing with the combined capacitance and combined resistance obtained as the parallel sum when C1, C4, and R1 are connected in parallel with each other. Therefore, when each set of C4, C1, and R1 connected to a single detection signal lead-out wire are equal to each other, it can be said that their respective combined capacitances and combined resistances can be conveniently treated as their respective parallel sums.

[0074] Next, consider the case where a plurality of detection signal lead-out wires are arranged in one active unit, and transistors and detection electrodes that constitute a multiplexer are arranged on each detection signal lead-out wire, and pixels of the same shape are arranged one by one. As a result of the research, the equivalent circuit in this case can be represented by the equivalent circuit shown in Fig. 43. Fig. 43 shows an AC power supply with a potential amplitude of V in Fig. 36, a detection capacitance C4 composed of fingers and detection electrodes, and one set each composed of the capacitance C1 and resistance R1 of the transistor, connected to each of the m detection signal lead-out wires. That is, it corresponds to arranging one uniform pixel having the same values for the detection capacitance C4, the capacitance C1 of the transistor, and the resistance R1 on each detection signal lead-out wire, and further, each detection signal lead-out wire is uniform with respect to each other and has the same capacitance and resistance.

[0075] In FIG. 43, the currents flowing through the m detection signal lead-out lines are combined to be I30 and transmitted to the control circuit. Similar to the case of FIG. 42, since the capacitances and resistances corresponding to each other in each detection signal lead-out line are connected to different nodes respectively and thus are not in parallel connection, it is not possible to simply obtain the combined capacitance and combined resistance of each. However, since the potentials of the corresponding nodes in each pair in FIGS. 36 and 43 are equal to each other, using the currents I1, I2, and I3 shown in FIG. 36 I1 1 =I1 2 =···=I1 m =I1 I2 1 =I2 2 =···=I2 m =I2 I3 1 =I3 2 =···=I3 m =I3 we can set. As a result, the detection current I30 is expressed by the following equation (15). I3 0 =mI3······(15) When expanding equation (15) and comparing it with equation (4), it is found that equation (15) is equivalent to replacing C1, C2, C4, R1, and R2 in equation (4) with C1→mC1 C2→mC2 C4→mC4 R1→R1 / m R2→R2 / m respectively. This replacement corresponds to replacing C1, C2, C4, and R1 and R2 with the combined capacitance and combined resistance which are the parallel sums in the case where they are connected in parallel with each other respectively. Therefore, when transistors are dispersedly arranged on a plurality of detection signal lead-out lines, when the capacitance C1 and resistance R1 of each transistor associated therewith, the detection capacitance C4, and the capacitance C2 and resistance R2 of the detection signal lead-out lines are equal to each other respectively, it can be said that the respective combined capacitances and combined resistances can be conveniently treated as the respective parallel sums. And the detection current is proportional to the number of lead-out lines to be dispersedly arranged.

[0076] Next, the method for obtaining the combined capacitance and combined resistance when the capacitance C1 of the distributed transistors, the resistances R1 and R2 of the detection signal lead-out lines, and the detection capacitance C4 are non-uniform will be described. When considering the calculation method of the detection current from a design perspective, in some cases, a calculation method that can calculate a current value that is surely lower than the actual flowing current value, even if the calculation accuracy is somewhat inferior compared to obtaining an exact detection current value, is more important. This is because the exact calculated value is the value under the condition that various conditions are fixed, and this value does not include variations due to the processing accuracy of the manufacturing process, so the variation in the current value must be separately calculated by calculation. That is, when designing based on a calculated value that is surely smaller than the actual flowing current value, the difference becomes the design margin. From this perspective, a method of uniform approximation can be considered in which the maximum value among the distributed capacitances and resistances is used as the representative value of each of C1, R1, C2, and R2, and the minimum value is used as the representative value of C4, assuming that each of C1, R1, C2, R2, and C4 has its respective representative value. However, this method may lead to an unfavorable design far from the optimal design, such as a significant decrease in the aperture ratio due to an overly large design margin being expected unnecessarily. Therefore, it is important to select representative values with an appropriate margin. Thus, a method for selecting representative values with an appropriate margin will be described.

[0077] As methods for selecting representative values for equalizing and approximating each capacitance and resistance, the following three methods can be mentioned. The first method is to use the maximum value among the individually distributed values for C1, R1, C2, and R2, and the minimum value for C4 as the representative value. Among the three methods, it is the method with the minimum detection current and the maximum design margin, and this will be called the minimum detection current approximation. In this case, if the values of a very small number of elements are significantly different from others, the margin may be overestimated, which may unnecessarily greatly reduce the aperture ratio, so it needs to be used carefully. The second method is a method of replacing a value that is extremely different from others with a value equivalent to others while ignoring its particularity when there are values that are extremely different among the individually distributed capacitances and resistances C1, R1, C2, R2, and C4, and this will be called the special element exclusion approximation. In this case, it is necessary to carefully examine the validity of excluding extremely different values. Also, it is conceivable to use the arithmetic mean value of non-excluded values as the value to be replaced. The third method is to consider using the arithmetic mean value of each as the representative value without exclusion even if there are values that are extremely different among the individually distributed capacitances and resistances C1, R1, C2, R2, and C4. This will be called the arithmetic mean approximation.

[0078] In actual design, since it is desirable to reduce the variation in the detected current, these capacitances and resistors C1, R1, C2, R2, C4 are required to be uniform, and it is difficult to deliberately make them non-uniform. As a case of making them non-uniform, for example, the pixel size may be non-uniform depending on the product specifications. That is, in recent displays, it has generally become the case that the size and aperture ratio of pixels that emit red, green, and blue light corresponding to the three primary colors of light are different for each color. For example, by reducing the size of green pixels with high visibility for humans and lowering the aperture ratio to reduce the luminance of green, and instead increasing the size of blue pixels with low visibility and increasing the aperture ratio to increase the luminance of blue, the visual luminance is improved. When trying to improve the screen luminance by increasing the aperture ratio forcibly when the pixel size is uniform, there may be drawbacks such as a yield decrease, but by changing the pixel size for each color of the pixel, it becomes possible to suppress the yield decrease, and it is possible to improve the screen luminance perceived by humans while maintaining the yield. However, since the pixel sizes are not extremely different for each color, even in such a case, as a result, each capacitance and resistors C1, R1, C2, R2, C4 also become values close to each other. For this reason, among the three uniform approximation methods, the most appropriate method is considered to be the additive average approximation.

[0079] If the variation in each capacitance and resistors C1, R1, C2, R2, C4 is large and it is not preferable to apply the additive average approximation, but the additive average approximation is applied, the value of the detected current assumed in the design and the value of the detected current actually flowing will be different. When the actually flowing detected current is larger than the design assumed value, since a higher detection sensitivity than expected in the design can be obtained, there is no particular problem. Conversely, even when the actually flowing detected current is smaller than the design assumed value, if the base current is sufficiently reduced by the introduction of the base current reduction mechanism, sufficient detection sensitivity can be obtained based on the displacement voltage component as the detection signal separated from the detected current. And in this case, although there is a possibility of causing a somewhat unnecessary decrease in the aperture ratio and a decrease in the yield, since the variation in each capacitance and resistors C1, R1, C2, R2, C4 does not become extremely large, the aperture ratio and the yield do not decrease so much as to become a problem.

[0080] Also, regarding the variation in transistor characteristics due to the interface state between the semiconductor layer and the electrode constituting the transistor, it is not uncommon for the current to vary by about 10% even when the voltage applied to the transistor is the same. This variation in current value is considered to be a major problem when the mechanism for reducing the base current is not provided. However, in the present invention, since the mechanism for reducing the base current is provided, it does not affect the detection sensitivity. In the actual manufacturing process, since the characteristics of the transistor are controlled to fall within a certain range, even if there are variations in the characteristics of individual transistors, the current that is synthesized and reaches the control circuit is considered to be averaged and substantially constant, and thus there is no problem with the detection sensitivity. Therefore, it is considered that there is no need to anticipate the variation in transistor resistance, which is a factor causing variation in transistor characteristics due to the interface state between the semiconductor layer and the electrode caused by the manufacturing process, in the design.

[0081] As described above, redefining the capacitance of each transistor constituting the multiplexer distributed in the pixels as Ctft, the resistance as Rtft, and the individual detection capacitance as Csen, when Ctft, Rtft, and Csen are each uniformly approximated, and assuming the number of distributed elements within the active unit to be analyzed as N, the combined capacitance C1, the combined resistance R1, and the combined value C4 of the detection capacitance of the transistors constituting the multiplexer distributed in the pixels are expressed as shown in the following formulas (16) to (17). C1 = NCtft ······ (16) R1 = Rtft / N ······ (17) C4 = NCsen ······ (18) Here, when the detection electrodes constituting the detection capacitance are integrated across a plurality of pixels and a plurality of transistors are connected, the value obtained by dividing the detection capacitance constituted by the integrated detection electrodes by the number of connected transistors, that is, the detection capacitance borne by one transistor, is defined as Csen.

[0082] Next, the combined resistance of the capacitance C2 of the detection signal lead will be described. In the previous descriptions, the capacitance C1 of the transistors that make up the multiplexer has been treated as a single capacitance. However, the transistor capacitance C1 is composed of the gate electrode of the transistor, the semiconductor layer disposed on the gate electrode via an insulating film, and the source electrode and drain electrode connected to the semiconductor layer. From years of research by the inventors of the present invention, since the semiconductor layer is a high resistor even when the transistor is in the ON state, the channel, which is the current path in the semiconductor layer, is represented by a resistor, with its both ends being the source electrode and the drain electrode respectively, and the gate-source capacitance composed of the respective electrodes is connected between the gate electrode and the source electrode, and an equivalent circuit in which the gate-drain capacitance composed of the respective electrodes is connected between the gate electrode and the drain electrode can represent the transistor. In this case, it is possible to set the boundary between the gate-source capacitance and the gate-drain capacitance at the center of the channel without any problem. Originally, the source and the drain represent functions rather than electrode names. The source is the source of the charge that is the source of the current flowing through the transistor, and the destination is called the drain. Therefore, due to the voltage relationship applied to the three electrodes of the transistor, the source and the drain will be interchanged at any time. Thus, the capacitance formed between the electrode that functions as the source or drain connected to the detection signal lead and the gate electrode is defined as Cgl again. In this equivalent circuit model of the transistor, the capacitance Cgl will be connected to the detection signal lead in parallel with each crossing capacitance formed in the region where the detection signal lead crosses various electrode wirings and the like. That is, the capacitance Cgl that constitutes the capacitance C1 of the transistors that make up the multiplexer will function as a part of the detection signal lead capacitance C2. As a result, it has been found that the combined value of the capacitance C2 of the detection signal lead of the analysis target active unit in the case where the transistors that make up the multiplexer are dispersedly arranged in pixels is represented by the following formula (19). C2 = Σ(i = 1…m){Nsy Cline-s + ni Cgl(Nsy - 1)} + Cline_b······(19) Here, Σ(i = 1…m) is a symbol representing the sum from i = 1 to m, m is the number of detection signal lead-out lines arranged in the active unit to be analyzed, n_i is the number of transistors connected to the i-th detection signal lead-out line in the active unit to be analyzed, Cline-s is the capacitance of the non-transistor part of the detection signal lead-out line corresponding to the length of one active unit in the detection signal lead-out line direction, Nsy is the number of active units arranged in the detection signal lead-out line direction, and Cline_b is the capacitance of the detection signal lead-out line in the frame area. Also, Cline-s, Cgl, and Cline_b are respectively values approximated by uniformization. In particular, when n_i is a constant value n, Equation (19) is represented by the following Equation (19-2). C2 = m{Nsy Cline-s + nCgl(Nsy - 1)} + Cline_b ······ (19-2)

[0083] Also, assuming that Rline-s is the resistance of the detection signal lead-out line approximated by uniformization corresponding to the length of one active unit in the detection signal lead-out line direction, and Rline_b is the resistance of the detection signal lead-out line in the frame area, and both are approximated by uniformization, the combined resistance R2 of the detection signal lead-out line of the active unit to be analyzed is represented by the following Equation (20). R2 = (Nsy Rline-s / m) + Rline_b ······ (20)

[0084] Thus, when the transistors constituting the multiplexer are dispersedly arranged in the pixels, in Equation (5) representing the magnitude |I3| of the detection current I3 and the approximate equations (6) to (8) for each frequency region, the capacitance C1 and resistance R1 of the transistor are respectively represented by Equation (16) and Equation (17), the detection capacitance C4 is represented by Equation (18), and the capacitance C2 and R2 of the detection signal lead-out line for the active unit to be analyzed are respectively replaced by Equation (19) and Equation (20).

[0085] Here, from equations (19) and (20) for obtaining the combined capacitance C2 and combined resistance R2 of the detection signal lead wire, as the position of the active unit to be analyzed is farther from the control circuit connection terminal, C2 and R2 increase. As a result, due to the increase in C2R2, the graph of the current - applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 shifts to the lower left, that is, to the low - current side and the low - frequency side. Figure 22 is a graph showing this state. The thick solid line represents the current - applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 in the active unit farthest from the control circuit connection terminal of the detection signal lead wire in the row of active units arranged in the detection signal lead wire direction. The thin solid line represents the current - applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 in the active unit closest to the control circuit connection terminal of the detection signal lead wire. The thick dashed - dotted line represents the value obtained by converting the angular frequency ω1 of the first pseudo - bending point for the active unit farthest from the control circuit connection terminal of the detection signal lead wire into frequency. The thin dashed - double - dotted line represents the value obtained by converting the angular frequency ω2 of the second pseudo - bending point for the active unit closest to the control circuit connection terminal of the detection signal lead wire into frequency, which are all results obtained from the existing analytical expressions starting from equation (5). Here, it is assumed that each of the capacitances and resistances arranged dispersedly is uniform. The capacitance Ctft of one transistor constituting the multiplexer is 50 fF, the capacitance Cgl on the side connected to the detection signal lead wire among the capacitances between the source or drain and the gate is 25 fF, the resistance Rtft of one transistor is 1 MΩ, the capacitance Cline - s of the non - transistor part of the detection signal lead wire corresponding to the length of one active unit in the detection signal lead wire direction is 1.25 pF, the resistance of the detection signal lead wire corresponding to the length of one active unit in the detection signal lead wire direction is 500 Ω, the combined value of the detection capacitances is 1 pF, the number m of detection signal lead wires arranged in the active unit is 10, the number of transistors connected to one active unit in one active unit is 10, the number of active unit arrangements in the detection signal lead wire direction is 20, and the amplitude of the applied AC potential V is 1 V. Also, both the capacitance Cline_b and the resistance Rline_b of the detection signal lead wire in the frame area are set to zero, but in fact, both are sufficiently small values compared to the value of the entire detection signal line, so it does not interfere with explaining the effect of the invention.

[0086] In FIG. 44, since the detection capacitances C4 are equal, the low-frequency regions of both active units coincide, but the middle and high-frequency regions of the active unit farthest from the control circuit connection terminal are shifted downward to the left. When the variation of the detection current is small and importance is attached to the reduction of the frequency of the applied alternating voltage, it can be said that it is desirable to set the angular frequency ω of the design point in the vicinity between the angular frequency ω1_f with respect to the first pseudo-bending point of the active unit farthest from the control circuit connection terminal and the angular frequency ω1_n with respect to the first pseudo-bending point of the active unit closest to the control circuit connection terminal. Also, when performing amplification processing on the displacement voltage component separated from the detection current, which is important as a detection signal, by an integration circuit or the like, it is desirable to set it on the high-frequency side. Therefore, the angular frequency ω2_n with respect to the second pseudo-bending point at which the detection current of the active unit closest to the control circuit connection terminal starts to decrease from near the maximum value can be regarded as the upper limit of the design point. Accordingly, it is desirable that the angular frequency ω with respect to the design point satisfies at least ω ≦ ω2_n, and the value of ω will be specifically determined according to other design conditions. However, when setting the angular frequency ω with respect to the design point to the angular frequency ω2_n with respect to the second pseudo-bending point of the active unit closest to the control circuit connection terminal, the detection current of the active unit farthest from the control circuit connection terminal will be lower than the detection current of the active unit closest to the control circuit connection terminal. However, as described above, when the basic current is reduced by the introduction of the basic current reduction mechanism, the constraints on the magnitude and variation of the detection current are relaxed. In particular, when the basic current is completely suppressed by the housing voltage application method, it does not significantly affect the detection sensitivity and the cost of the control circuit, and is acceptable as a design. Also, from this perspective, the angular frequency ω2_n can be regarded as the maximum value ω2_max among the angular frequencies ω2 with respect to the second pseudo-bending points in all the active units. That is, when the maximum of the angular frequencies ω2 with respect to the second pseudo-bending points of the plurality of arranged active units is set as ω2_max, it is desirable that the angular frequency ω with respect to the design point satisfies at least ω ≦ ω2_max.

[0087] Needless to say, it is preferable that the variation in the detection current be small even when a mechanism for reducing the basic current is provided. When substituting the equations (16) to (20) of each capacitance and resistance in the case of distributed arrangement into the approximate equation (7) for the medium frequency range and the approximate equation (8) for the high frequency range, even if the number N of distributed arrangements within each active unit is the same, the value of the detection current changes depending on the arrangement method. In particular, when the number of detection signal lead-out lines to which transistors are connected within the active unit is m, and the number of transistors connected to each detection signal lead-out line is n, it has been derived from research that the values of the approximate equations for the medium and high frequency ranges increase as m increases. This means that when the number N of distributed arrangements within the active unit is fixed, the graph of the current - applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 shifts upward to the right with respect to an increase in the number m of detection signal lead-out lines arranged within the active unit. In actual design, since the number n of transistors connected to each detection signal lead-out line is not always constant, by replacing it with the relationship between the number N of distributed arrangements within the active unit and the number m of detection signal lead-out lines to which transistors are connected within the active unit, and increasing the ratio m / N of m to N as the active unit moves away from the control circuit connection terminal, the variation in the graph of the current - frequency dependence relationship of the magnitude |I3| of the detection current I3 can be reduced. At this time, since both the number N of distributed arrangements within the active unit and the number m of detection signal lead-out lines to which transistors are connected within the active unit are integers, it goes without saying that the value of m / N does not necessarily change according to the distance from the control circuit connection terminal, and there may be cases where m / N has the same value in a plurality of active units.

[0088] Also, when emphasizing an increase in the detection current, it is desirable to set a large value for the m / N value. When representing the integer part of the square root of N as Nsqrt_int, it has been found from research that it is desirable to have an active unit with at least m ≧ Nsqrt_int.

[0089] Alternatively, it is also effective to set the synthetic detection capacitance C4 to increase as the active unit moves away from the control circuit connection terminal based on the approximate expressions (6) to (8) for each frequency range.

[0090] Next, the capacitance and resistance of the transistors constituting the multiplexer, the capacitance and resistance of the detection signal lead-out lines, and the method for calculating the detection capacitance will be described. Design means finding an optimal combination that simultaneously satisfies not only meeting the product specifications, which are the requirements from customers, but also obtaining performance that is slightly higher than that required by the product specifications, increasing the manufacturing yield in the manufacturing process as much as possible, and reducing costs as much as possible. To achieve this, a very large number of design conditions must be satisfied during the design. Therefore, the designer has to adjust a huge number of parameters in all design conditions one by one to reach the optimal design, which is the optimal solution. For example, in order to incorporate the present invention into a display device, it is necessary to satisfy new design conditions for imparting a sensor function in addition to various design conditions of the original display device. That is, the present invention provides these new design conditions. Therefore, similar to many other design conditions, it is necessary to repeatedly confirm one by one whether the design conditions provided by the present invention are satisfied during the intermediate stage of the design. Therefore, as a method for calculating the capacitance and resistance of the transistors constituting the multiplexer, which are the design parameters of the present invention, and the capacitance and resistance of the detection signal lead-out lines and the detection capacitance, although it is a simple method, a calculation method with sufficient accuracy is required in the process of reaching the optimal design. In recent years, due to the improvement in the performance of computers and simulators, software for extracting the capacitance and resistance of each part by analyzing the entire display device has been put on the market. If there is an environment where such software for extracting capacitance and resistance is used at the final stage of the design, checking the compliance of the design conditions provided by the present invention based on the capacitance and resistance extracted using this software can be expected to lead to a further optimized design. However, it takes time to extract capacitance and resistance using such a simulator. In particular, when strictly obtaining the resistance of a transistor using a simulator, it is necessary to divide the transistor into a fine mesh and perform potential analysis and carrier density analysis inside the transistor self-consistently at each time step to achieve convergence, which requires a huge amount of time. Therefore, it is not practical to do this during the design process. For this reason, it is difficult to use it during the intermediate stage of the design, and it will be implemented as a final confirmation at the end of the design.Therefore, as a method for calculating the capacitance and resistance of the transistors constituting the multiplexer, the capacitance and resistance of the detection signal lead-out line, and the detection capacitance, although it is a simple method, it is important to establish a calculation method with sufficient accuracy in the process of reaching the optimal design.

[0091] First, regarding the capacitance, since the thickness of the metal layer and the insulating film layer is very thin compared to the width of each electrode and wiring, it is acceptable to treat it as a parallel plate model. That is, it is to multiply the intersection area between conductor layers such as metal by the relative permittivity of the insulating film layer and the permittivity of vacuum, and then divide by the thickness of the insulating film layer. In particular, regarding the capacitance of the transistors constituting the multiplexer, in the present invention, since the transistors are in the ON state, a virtual electrode is considered in which the semiconductor layer and the source and drain electrodes connected thereto are regarded as having the same potential and integrated. It is acceptable to regard the area where this virtual electrode intersects with the gate electrode arranged via the insulating film as the capacitance region. Also, this transistor capacitance is divided into two at the center of the source electrode and the drain electrode, and it is acceptable to regard the capacitance located on the detection signal connection line side among the divided capacitances as functioning as the capacitance of the detection signal lead-out line. Needless to say, the accuracy can be further improved by adding the coupling capacitance formed between the calculated capacitance in the parallel plate model and the neighboring conductor layer.

[0092] Regarding the resistance of the detection signal lead-out line, it is calculated by multiplying the sheet resistance, which is a general method, by the length of the resistance calculation region and then dividing by the width of the resistance calculation region. If the line width varies depending on the location, it is acceptable to divide it into parts with different line widths, calculate the resistance for each part, and then add them together. Also, when the shape of the resistance calculation region is complex, from the perspective of ensuring the design margin, it is acceptable to use the width of the narrowest part in the direction of current flow as the representative value.

[0093] Next, a method for calculating the resistance of the transistors that make up the multiplexer will be described. A transistor can be regarded as a variable resistor whose resistance changes depending on the relative magnitudes of the potentials applied to the gate, source, and drain, which are the three electrodes that make it up. The transistor resistance fluctuates due to the AC voltage applied to the detection capacitor. FIG. 45 shows an equivalent circuit diagram for analyzing the relationship between the AC voltage applied to the detection capacitor and the transistor resistance using Spice of a circuit simulator. Although the transistor resistance Rtft is regarded as a fixed resistor, it is a model that can grasp the characteristics of the timing when the transistor capacitance fluctuates and the relative potential relationship between the electrodes when the resistance is maximized. FIG. 45 is an analysis model equivalent to FIG. 36, and the gate electrode potential Vg is connected to the transistor capacitance Ctft. Let the potential of the node where the detection capacitor Cf and the transistor capacitance Ctft are connected, that is, the node 210 corresponding to the detection electrode, be Vtft-f, and the potential of the node 211 where the capacitance Cline and the resistance Rtft of the detection signal lead are connected be Vtft-l. Since the transistor is in the ON state during the selection period when the detection current flows, Vg is set to 15V. The AC voltage Vf applied to the detection capacitor Cf is a rectangular pulse with an amplitude of 5V, a pulse width of 10 μs, and a period of 20 μs. The capacitance Cline of the detection signal lead is 100 pF, and the resistance Rline of the detection signal line is 100 Ω. Assuming the application of the case voltage addition method as the basic current reduction mechanism, the terminal side of the detection signal lead resistance Rline is grounded. FIG. 46 shows the analysis results by Spice of the potential Vtft-f of the node 210 and the potential Vtft-l of the node 211 when the detection capacitor Cf is 1 pF, the transistor capacitance Ctft is 0.02 pF, and the transistor resistance Rtft is 1 MΩ, and the applied AC potential Vf applied to the detection capacitor Cf.

[0094] In FIG. 46, the dashed line indicates the applied AC potential Vf, and the solid line indicates the Spice analysis result of the potential Vtft-f at node 210. The Spice analysis result of the potential Vtft-l at node 211 is constant at the ground potential, i.e., 0 V, and is not shown to avoid complicating the graph. The potential Vtft-f at node 210 is displaced by coupling associated with the potential change ΔVf of the applied AC potential Vf pulse, and then approaches the ground potential, i.e., 0 V, exponentially by charging and discharging during the period until the potential of the next Vf pulse changes. In particular, the change amount ΔVtft-f of the potential Vtft-f at node TFT-f at the rising edge of the Vf pulse is expressed by the following equation (21) ΔVtft-f = {Cf / (Cf + Ctft)}ΔVf ······ (21) In FIG. 24, ΔVtft-f is 4.9 V, which is consistent with the calculated value by equation (21).

[0095] In the electron transport theory of the channel, the source and drain of a transistor function such that the side where electrons move, i.e., the side with a lower potential, serves as the source, and the other side serves as the drain. Therefore, when an alternating potential is applied as in this analysis model, the source and drain reverse constantly. Let the source potential be Vs and the drain potential be Vd. Then, Vs and Vd obtained from the analysis results shown in Fig. 46 are respectively shown in Fig. 47. In Fig. 47, the dashed line represents the source potential Vs, and the solid line represents the drain potential Vd. The gate-source voltage Vgs and the source-drain voltage Vds at this time are respectively shown in Fig. 48. In Fig. 48, the thin line represents the gate-source voltage Vgs, and the thick line represents the source-drain voltage Vds. Figs. 47 and 48 show the analysis results when the resistance Rtft of the transistor is fixed. However, since neither the gate-source voltage Vgs nor the source-drain voltage Vds is a constant value as shown in Fig. 48, the actual resistance Rtft of the transistor is not constant and changes with time. That is, the analysis results of the source potential Vs, the drain potential Vd, the gate-source voltage Vgs, and the source-drain voltage Vds shown in Figs. 47 and 48 do not accurately reproduce the actual values. However, it is considered that the characteristics such as the timing of each potential change and the instantaneous potential change due to coupling can be accurately reproduced.

[0096] Here, the current Ids flowing between the source and drain of the transistor is given by the following equation (22) in the linear region (Vds < Vgs - Vth): Ids = (W / L)μCox{(Vgs - Vth)Vds - (1 / 2)(Vds^2)} ······ (22) And in the saturation region (Vds > Vgs - Vth), it is given by the following equation (23): Ids = (1 / 2)(W / L)μCox(Vgs - Vth)^2 ······ (23) and this is widely known. In Equations (22) and (23), W represents the channel width, L represents the channel length, μ represents the field-effect mobility, Cox represents the capacitance per unit area, and Vth represents the threshold voltage. When IGZO is considered as the semiconductor layer of the transistor, the threshold voltage Vth is about 0.2 V or less. If the amplitude of the AC voltage applied to the detection capacitance is about 5 V and the gate potential for turning on the transistor is about 15 V, only the current in the linear region flows between the source and drain of the transistor. Since the source-drain current Ids flows by applying the source-drain voltage Vds to the channel resistance determined by the gate-source voltage Vgs and the source-drain voltage Vds, the resistance of the channel, that is, the resistance Rtft of the transistor, can be expressed by the following Equation (24). Rtft = Vds / Ids ······ (24)

[0097] Assume again that the gate-source voltage Vgs and the source-drain voltage Vds are given as in Fig. 48. If the direction of the source-drain current Ids flowing from the transistor to the detection signal lead-out wire is defined as positive, the source-drain current Ids becomes as shown in Fig. 49 according to Equation (22) for the linear region. In Fig. 49, the thin solid line represents the gate-source voltage Vgs, the thick solid line represents the source-drain voltage Vds, and the thin dashed line represents the source-drain current Ids. The resistance Rtft of the transistor at this time becomes as shown in Fig. 50 according to Equation (24). In Fig. 50, the thin solid line represents the gate-source voltage Vgs, the thick solid line represents the source-drain voltage Vds, and the thin dashed line represents the transistor resistance Rtft. In Fig. 49, since Vgs > 15V, the transistor is always in the ON state, and the source-drain current Ids flows when the source-drain voltage Vds ≠ 0. And during the period when the gate-source voltage Vgs is large, the resistance of the channel decreases, so Ids increases. Therefore, among the currents Ids that flow twice during one cycle of the applied alternating potential Vf, the lower one, that is, the period when the gate-source voltage Vgs is flat and the source-drain voltage Vds ≠ 0, that is, the period when the detection electrode potential Vtft-f once rises and then returns to the ground potential due to the change of the applied alternating potential Vf from a low potential to a high potential, the current Ids flowing during this period is considered to limit the detection sensitivity. The resistance Rtft of the transistor during this period is larger than the resistance in other periods in Fig. 50. It was confirmed that these situations are the same even when the amplitude Vf_amp of the applied alternating potential Vf becomes extremely large, about 20V, and Ids has a mixture of the linear region and the saturation region. Therefore, since the potential Vtft-l of node 211 is the potential of the detection signal lead-out wire, if this is set as Vline, the drain potential Vd when the resistance Rtft of the transistor reaches the maximum value is given by the following Equation (25), Vd = Vline + {Cf / (Cf + Ctft)} Vf_amp ······ (25) The source potential Vs is given by the following Equation (26), Vs = Vline ······ (26) They can be represented respectively. Thus, the gate-source voltage Vgs is given by the following equation (27): Vgs = Vg - Vline ······ (27) The source-drain voltage Vds is given by the following equation (28): Vds = {Cf / (Cf + Ctft)} Vf_amp ······ (28) They can be represented respectively. Therefore, substituting Vgs and Vds given by equations (27) and (28) into equation (22) for the linear region and equation (23) for the saturation region respectively to obtain Ids, and then obtaining the resistance Rtft of the transistor from equation (24).

[0098] When using the case voltage application method as the basic current reduction mechanism, the potential Vline of the detection signal lead can be regarded as equal to the ground potential. When not using the case voltage application method, since it is equivalent to the case of swapping Vf and Vline, in equation (27), Vline can be replaced with Vf, that is, the ground potential. Based on the above, the specific invention content is shown below.

[0099] The sensor device of the present invention has at least an active unit, a detection signal lead-out wire, a first switching element, a control circuit, and a basic current reduction mechanism. The active unit includes an active circuit region arranged in a substantially matrix shape and having electrodes that function as at least one or more detection electrodes. When a detected object approaches or contacts the active unit, a capacitance is formed in a timely manner between the detection electrode and the detected object. An AC voltage is applied to the timely formed capacitance by the control circuit or the control circuit and the basic voltage reduction mechanism, and an electrical signal generated at the detection electrode is selectively transmitted to the control circuit through the detection signal lead-out wire, in which one or more are electrically connected to each other, and one or more of the first switching elements. In the sensor device that detects the approach or contact event of the detected object by signal processing in the control circuit, when the number of transistors arranged in each of the active units is N, the added average capacitance and resistance of the first switching element are Ctft and Rtft respectively, and the added average value of the timely formed capacitance is Csen, the combined capacitance C1 of the first switching element is expressed by Equation (1A) as C1 = NCtft ······ (1A) The combined resistance R1 of the first switching element is expressed by Equation (1B) as R1 = Rtft / N ······ (1B) The combined capacitance C4 of the timely formed capacitance is expressed by Equation (1C) as C4 = NCsen ······ (1C) Let the number of the detection signal lead-out lines to which the first switching element is connected in the active unit be m, the number of the first switching elements connected to the i-th detection signal lead-out line in the active unit be n_i, the additive average capacitance of the non-transistor portion of the detection signal lead-out line corresponding to the length of one active unit in the direction of the detection signal lead-out line be Cline-s, the number of the active units arranged in the direction of the detection signal lead-out line be Nsy, the additive average capacitance of the detection signal lead-out line in the frame region be Cline_b, and the additive average capacitance formed between the electrode functioning as the source or drain connected to the detection signal lead-out line and the gate electrode be Cgl. When Σ(i = 1…m) represents the symbol for the sum from i = 1 to m, the combined capacitance C2 of the detection signal lead-out line from the connection terminal of the detection signal lead-out line connected to the control circuit to the active unit is expressed by Equation (1D) as follows: C2 = Σ(i = 1…m){Nsy Cline-s + n_i Cgl(Nsy - 1)} + Cline_b ······(1D) Let the additive average resistance of the detection signal lead-out line corresponding to the length of one active unit in the direction of the detection signal lead-out line be Rline-s, and the additive average resistance of the detection signal lead-out line in the frame region be Rline_b. Then, the combined resistance R2 of the detection signal lead-out line from the connection terminal of the detection signal lead-out line connected to the control circuit to the active unit is expressed by Equation (1E) as follows: R2 = (Nsy Rline-s / m) + Rline_b ······(1E) Let ω2 for the active unit be expressed by Equation (1F) assuming that "^" represents the exponentiation symbol: ω2 = {φ^(1 / 2)} / {C2(C1 + C4)R1R2} ······(1F) φ = {(C1 + C4)^2}(R1^2) + {(C1 + C2 + C4)^2}(R2^2) + 2{(C1 + C4)^2}R1R2 When the maximum value among ω2 for each of the active units arranged in the approximate matrix shape is ω2_max, the angular frequency ω of the alternating voltage applied to the capacitances formed in a timely manner is ω ≦ ω2_max shall be satisfied.

[0100] In the active unit arranged in the approximate matrix shape, when the number of transistors arranged in one active unit is N, the number of the detection signal lead-out lines to which the transistors are connected is m, and the integer part of the square root of N is represented as Nsqrt_int, there is an active unit in which at least m≧Nsqrt_int.

[0101] As the position of the active unit is farther from the control circuit connection terminal, the value of the ratio m / N of m to N increases.

[0102] As the position of the active unit is farther from the control circuit connection terminal, C4 increases.

[0103] The basic current reduction mechanism has a function of applying an alternating current potential to the housing of the sensor device and a function of applying a reference potential to the control circuit connection terminal.

[0104] When the amplitude of the alternating current voltage applied to the timely formation capacitance due to the contact event is Vf_amp, the electrode potential of the timely formation capacitance on the side to which the alternating current potential is applied is Vf, the gate electrode potential of the transistor in the selection period is Vg, and the potential of the detection signal lead-out line is Vline, Vds is expressed by Equation (2A), Vds={Csen / (Csen+Ctft)}Vf_amp······(2A) When Vline is the reference potential, Vgs is expressed by Equation (2B), Vgs=Vg-Vline······(2B) When Vline is not the reference potential, Vgs is expressed by Equation (2C), Vgs=Vg-Vf······(2C) When the threshold voltage of the transistor is Vth, the field effect mobility is μ, the capacitance per unit area is Cox, the channel length is L, the channel width is W, and "^" is the exponentiation symbol, when Vds≦Vgs-Vth, Ids is expressed by Equation (2D) Ids = (W / L)μCox{(Vgs - Vth)Vds - (1 / 2)(Vds^2)} ······ (2D) When Vds > Vgs - Vth, when Ids is expressed by Equation (2E), Ids = (1 / 2)(W / L)μCox(Vgs - Vth)^2 ······ (2E) The weighted average resistance Rtft of the transistor is Rtft = Vds / Ids is defined as such.

Advantages of the Invention

[0105] According to the present invention, a sensor device is provided with a contact detection input method of a display screen built-in type capable of identifying a multi-point simultaneous contact position applicable to a screen size exceeding 7 inches, which significantly reduces the number of control circuit connection terminals without increasing the frame width or significantly reducing the aperture ratio.

Brief Description of the Drawings

[0106]

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Embodiments for Carrying Out the Invention

[0107] [Embodiment 1] As an embodiment of the sensor device of the present invention, a contact detection type input function of a detection capacitance timely formation type that enables identification of a multi-point simultaneous contact position by connecting detection electrodes and control circuit connection terminals one-to-one is applied to the display screen of a liquid crystal display device in a fringe field switching mode. The case where transistors constituting a multiplexer are dispersedly arranged in pixels to reduce the number of control circuit connection terminals will be described below with reference to the drawings. Specifically, it is applied to a 15.6-inch diagonal liquid crystal display device in which the pixel size is 90 μm × 270 μm and an active unit composed of (3 × 16) × 16 such pixels is arranged in 80 × 45.

[0108] The sensor device of this embodiment can be fabricated by the same manufacturing process as the conventional example described with reference to FIG. 8. First, after depositing a 200-nm-thick MoW alloy film on the first glass substrate by sputtering, a scanning signal line 300 that transmits a selection signal to the display transistor 314 and the gate electrodes of transistors 340 that form a multiplexer and a control wiring 330 for switching the ON and OFF states of the transistor 340 are processed into a predetermined shape through a photolithography process. A plan view of the pixel at this time is shown in FIG. 51. Next, after depositing a 40-nm-thick transparent electrode layer made of indium tin oxide (hereinafter referred to as ITO) by sputtering, the pixel electrode 301 is processed into a predetermined shape through a photolithography process. A plan view of the pixel at this time is shown in FIG. 52. Next, SiO2 is deposited as a gate insulating film by chemical vapor deposition to a thickness of 300 nm, and then InGaZnO4 is deposited by sputtering to a thickness of 10 nm. Subsequently, through a photolithography process, the InGaZnO4 is processed into semiconductor layers 302 and 332 of the transistors 314 and 340, respectively, with a predetermined shape. A plan view of the pixel at this time is shown in FIG. 53. Next, after depositing SiO2 by chemical vapor deposition to a thickness of 200 nm, the channel protection layers 303 and 333 of the semiconductor layer are processed into a predetermined shape through a photolithography process. A plan view of the pixel at this time is shown in FIG. 54. Next, through a photolithography process, SiO2 is processed into a predetermined shape to expose the MoW alloy layers at the ends of the scanning signal line 300 and the control wiring 330, and a power supply electrode, that is, a control circuit connection terminal, and a contact hole 304 for connecting the pixel electrode 301, the display transistor 314 are formed. A plan view of the pixel at this time is shown in FIG. 55. Next, after depositing a 300-nm-thick AlNd alloy film by sputtering, it is processed into a predetermined shape through a photolithography process to form electrodes 305 and 306 that serve as the source or drain of the transistor, a display signal line 307, and a detection signal lead-out line 308 for transmitting the detection signal to the control circuit. A plan view of the pixel in which the detection signal lead-out line 308 and the transistor 340 that forms a multiplexer are arranged at this time is shown in FIG. 56.Next, SiO2 was deposited as an interlayer insulating film by chemical vapor deposition to a thickness of 300 nm. Subsequently, an acrylic resin was coated and formed to a thickness of 3 μm, and then processed into a predetermined shape by a photolithography process to expose the respective power supply electrode portions located at the ends of the scanning signal line 300, the control wiring 330, the display signal line 307, and the detection signal lead-out line 308, that is, the metal layer of the control circuit connection terminal. Also, a contact hole 339 for connecting the transistor constituting the multiplexer and the detection electrode is formed. A plan view of the pixel in which the detection signal lead-out line 308 and the transistor 340 constituting the multiplexer are arranged at this time is shown in FIG. 57. Next, after depositing 40 nm of ITO by sputtering, the detection electrode 310 is processed into a predetermined shape having a slit 312 corresponding to the fringe field switching mode by a photolithography process. A plan view of the pixel in which the detection signal lead-out line 308 and the transistor 340 constituting the multiplexer are arranged at this time is shown in FIG. 58. Here, the relative dielectric constant of SiO2 is 5.0, the relative dielectric constant of the acrylic resin insulating film is 3.5, the resistivity of AlNd is 4.5 μΩ·cm, the thickness of each glass substrate is 0.5 mm, and the relative dielectric constant is 5.0. At this time, the capacitance per unit area Cox of the gate insulating film is 1.5 F / m^2. Next, a liquid crystal layer is sandwiched between the first glass substrate and a counter substrate having a color filter disposed on the second glass substrate. Subsequently, a control circuit is connected to each control circuit connection terminal, and together with a backlight disposed on the back, it is housed in a housing to be completed. Also, it goes without saying that the manufacturing method of the sensor device in this embodiment is not limited to what has been described here. For example, regarding the acrylic resin described with reference to FIG. 57, it may be processed so as to be disposed only in the vicinity of the intersection portion of the detection electrode 310, the display signal line 307, and the detection signal lead-out line 308.

[0109] FIG. 59 is a schematic diagram showing the state of pixels arranged in one active unit in the sensor device of the present embodiment. In FIG. 59, in the active unit 350, three pixels corresponding to RGB arranged in an RGB vertical stripe are grouped together to form an integrated detection electrode 351, and 16 transistors 340 that form a multiplexer and 16 detection signal lead-out lines 308 are arranged for each group of G pixels. The integrated detection electrode 351 is connected to the detection signal lead-out line 308 via a contact hole 339 and a transistor 340 that forms a multiplexer. As a result, each integrated detection electrode 351 arranged in the active unit 350 functions as if it is a single larger integrated detection electrode. When the transistor 340 that forms the multiplexer is in the ON state, the integrated detection electrode 351 and the detection signal lead-out line are electrically connected. Also, in FIG. 59, since the pixel sizes are uniformly set, the G pixels with high luminance visibility have a reduced aperture ratio within an allowable range due to the arrangement of the detection signal lines 308.

[0110] FIG. 60 is a conceptual diagram showing the schematic structure of the multiplexer in the present embodiment. In FIG. 60, the area surrounded by the broken line represents the active unit 350, the solid line represents the control wiring 330 that transmits the selection signal for switching the ON state and OFF state of the transistors 340 constituting the multiplexer, and the dashed-dotted line represents the detection signal lead-out line 308. The 16 control wirings 330 arranged in each active unit 350 are shared with the active units arranged in the horizontal direction and are connected to each other near the connection terminal 361 for connecting to the Y-control circuit 360 that controls the control wiring 330 and are electrically integrated. Similarly, the 16 detection signal lead-out lines 308 arranged in each active unit 350 are shared with the active units arranged in the vertical direction and are connected to each other near the connection terminal 363 for connecting to the X-control circuit 362 that controls the detection signal lead-out line 308 and are electrically integrated. Thereby, by sequentially applying the selection signal to each bundle of the control wiring 330 in a time-division manner during the sensor operation period, all the transistors 340 arranged in each active unit 350 arranged in the row direction are synchronously turned on, and the detection electrodes 351 of each active unit are electrically connected to the detection signal lead-out line 308 arranged in each corresponding active unit 350 via the transistors 340, and the detection signals emitted from the detection electrodes of each active unit are transmitted to the X-control circuit 362 through the corresponding connection terminals 363, and signal processing for detecting the contact event is performed. Also, although not shown in FIG. 60 to avoid complication, actually, at least the connection terminals of the scanning signal lines for transmitting the selection signal to the display function transistors are arranged in the gaps in the arrangement of the Y-control circuit connection terminals 361, and at least the connection terminals of the display signal lines are arranged in the gaps in the arrangement of the X-control circuit connection terminals 363.

[0111] Here, the capacitance of one transistor 340 constituting the multiplexer is set to 0.072 pF. The capacitance Cline_s and resistance Rline_s of the detection signal lead wire with respect to the length of one active unit, that is, 0.27 mm × 16 = 4.32 mm, are set to 0.52 pF and 120 Ω, respectively. The capacitance Cf formed between the detection electrode in which three RGB pixels are integrated and the finger via the second glass substrate is set to 0.003 pF. The amplitude Vf_amp of the alternating voltage applied to the detection capacitance Cf by the housing voltage addition method is a rectangular continuous pulse of 1 V. The reference potential applied to the detection signal lead wire is 0 V, and the potential applied to the gate electrode, that is, the control wiring 330, is set to 15 V to turn on the transistor 340 constituting the multiplexer. At this time, from Equation (27), the gate-source voltage Vgs of the transistor 340 constituting the multiplexer at the rising edge of the applied alternating voltage pulse that determines the detection current is 15 V, and from Equation (28), the source-drain voltage Vds is 0.039 V. Also, assuming the threshold voltage Vth of the transistor 340 constituting the multiplexer is 0.2 V, since Vds < Vgs - Vth, when the field-effect mobility is 10 cm^2 / Vs, from Equation (22) for the source-drain current in the linear region and Equation (24) for the relationship between the source-drain current and the transistor resistance, the resistance of one transistor 340 constituting the multiplexer is 0.46 MΩ. Further, in this embodiment, the number m of detection signal lead wires to which the transistor 340 constituting the multiplexer is connected and the number n of connections of the transistor 340 constituting the multiplexer connected to one detection signal lead wire in each active unit are both 16. The distributed arrangement number of the transistors 340 constituting the multiplexer is 256, and the number Nsy_max of active units arranged in the detection signal line direction is 45.As a result, the combined capacitance C1 and combined resistance R1 of the transistor 340 constituting the multiplexer in the unit to be analyzed are obtained from Equation (16) and Equation (17) respectively, the combined detection capacitance C4 is obtained from Equation (18), and the combined capacitance C2 and combined resistance R2 of the detection signal lead-out line with respect to the active unit to be analyzed are obtained from Equation (19-2) and Equation (20) respectively. By substituting these into Equation (5), the current - applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 can be obtained, and by substituting these into Equation (12), the angular frequency ω2 with respect to the second pseudo - inflection point in the current - applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 can be obtained. However, although both the capacitance Cline_b and resistance Rline_b of the detection signal lead - out line in the frame area are set to zero, since both are actually sufficiently small values with respect to the value of the entire detection signal line, it does not interfere with explaining the effect of the invention.

[0112] Figure 61 is a graph showing the current - applied AC voltage frequency dependence of the magnitude |I3| of the detection current I3 calculated in this embodiment. The thin line shows the result for the active unit closest to the control - circuit connection terminal 363 of the detection signal lead - out line 308, and the thick line shows the result for the active unit farthest from the control - circuit connection terminal 363 of the detection signal lead - out line 308. From Figure 61, when the case - voltage application method is applied, the constraint conditions for the variation in the magnitude |I3| of the detection current I3 are greatly relaxed, and when a design that emphasizes setting a high frequency for the applied AC voltage is performed, the design point may be set at the angular frequency ω2 for the active unit closest to the control - circuit connection terminal 363 of the detection signal lead - out line 308 where the angular frequency ω2 with respect to the second pseudo - inflection point is maximized. The frequency corresponding to ω2 in this case is approximately 2.5 GHz. Conversely, when a design that emphasizes minimizing the variation in the magnitude |I3| of the detection current I3 without applying the case - voltage application method is performed, it can be said that it is preferable to set it near the angular frequency ω1 with respect to the first pseudo - inflection point for the active unit farthest from the control - circuit connection terminal 363 of the detection signal lead - out line 308. The frequency corresponding to ω1 in this case is approximately 550 kHz.

[0113] FIG. 62 shows a graph in which two results of parameter changes are added to reduce the variation in the current-applied AC voltage frequency dependence with respect to the magnitude |I3| of the detection current I3 caused by the distance from the control circuit connection terminal 363 of the detection signal lead wire 308 shown in FIG. 32. The thick dashed line shows the result when the dispersion arrangement number N is reduced to 64 and m = 16, n = 4 for the active unit closest to the control circuit connection terminal 363 of the detection signal lead wire 308. The thick dashed-dotted line shows the result when the dispersion arrangement number N is reduced to 32 and m = 8, n = 4 for the active unit closest to the control circuit connection terminal 363 of the detection signal lead wire 308. The result of m = 16, n = 4 is shifted downward with respect to m = n = 16, and the result of m = 8, n = 4 is further shifted downward. As a result of the study, it was found that these shifts are due to a decrease in the dispersion arrangement number N. That is, the mid-frequency region of the current-applied AC voltage frequency dependence with respect to the magnitude |I3| of the detection current I3 for the active unit closest to the control circuit connection terminal 363 of the detection signal lead wire 308 is flat in a relatively wide range, which is due to the fact that the CR time constant determined by the combined capacitance C1 and combined resistance R1 of the transistors 340 constituting the multiplexer limits the detection current in the mid-frequency region. Within the range where the CR time constant limits the speed, even if m and n are changed while fixing the dispersion arrangement number N = mn to reduce the combined capacitance C2 and combined resistance R1 of the detection signal lead wire, the detection current does not change.

[0114] FIG. 63 is a graph showing the result of the current-applied AC voltage frequency dependence with respect to the magnitude |I3| of the detection current I3 when the size of the active unit is reduced to 8×8 sets with three RGB pixels in a set, with the dispersion arrangement number N fixed at 64, for the active unit farthest from the control circuit connection terminal 363 of the detection signal lead wire 308, with the combined capacitance C2 and combined resistance R2 of the detection signal lead wire being strongly rate-limiting. In FIG. 34, the dashed line shows the case of m = 16, n = 4, the solid line shows the case of m = 8, n = 8, and the dashed-dotted line shows the case of m = 4, n = 16. From FIG. 63, it can be seen that if the dispersion arrangement number N is the same, the larger the number m of the detection signal lead wires to which the transistors constituting the multiplexer are connected, the larger the magnitude |I3| of the detection current I3.

[0115] [Embodiment 2] Next, the sensor device according to Embodiment 2 of the present invention will be described with reference to FIGS. 64 to 65. The sensor device according to Embodiment 2 is different from the sensor device according to Embodiment 1 in that a virtual synthetic active unit composed of a plurality of active units is provided. As shown in FIG. 64, one virtual synthetic active unit V5612 includes a plurality of active units 350-51, 350-52, 350-61, and 350-62 adjacent to each other. In the present embodiment, one virtual synthetic active unit V5612 is composed of 2 (X direction) × 2 (Y direction) active units adjacent to each other, and 3 (X direction) × 2 (Y direction) virtual synthetic active units V5612 are arranged. However, the number of active units constituting the virtual synthetic active unit V5612 and the number of virtual synthetic active units V5612 arranged are not limited to this. For example, one virtual synthetic active unit may be composed of 3 (X direction) × 2 (Y direction), 2 (X direction) × 3 (Y direction), or 3 (X direction) × 3 (Y direction) active units. In addition, except that the X-addition control circuit 462 and the Y-addition control circuit 460 of Embodiment 2 are replaced by the X-control circuit 362 and the Y-control circuit 360 of Embodiment 1 shown in FIG. 60, which have different configurations from them, Embodiment 2 is the same as Embodiment 1. Therefore, the same reference numerals are given to the same parts as those in Embodiment 1, and only the configurations different from those in Embodiment 1 will be described.

[0116] As shown in FIG. 64, the two transistor control wirings 330-1 and 330-2 of the virtual synthetic active unit V5612 are connected to the Y-addition control circuit 460, and the two detection signal lead-out lines 308-5 and 308-6 are connected to the X-addition control circuit 462.

[0117] The Y-addition control circuit 460 is different from the Y-control circuit 360 of Embodiment 1 in that when detecting the region A1 touched by the finger, it is configured to collectively select and activate the two transistor control wirings 330-1 and 330-2, that is, apply a potential that turns on all the connected transistors simultaneously.

[0118] As shown in FIG. 65, the X-addition control circuit 462 includes an X-control circuit 362 and a detection signal addition circuit 450. The detection signal addition circuit 450 includes input terminals X1 to X6 and output terminals XA1 to X6. The input terminals X1 to X6 are respectively connected to each of the six detection signal lead-out lines 308-1 to 308-6, and the output terminals XA1 to X6 are respectively connected to the connection terminals 363 of the X-control circuit 362.

[0119] The detection signal addition circuit 450 includes transistors MX11, MX22, MX33, MX44, MX55, and MX66 as second switching elements respectively connected between the input terminals X1 to X6 and the output terminals XA1 to X6, which constitute a plurality of transistors 440 forming a multiplexer. The detection signal addition circuit 450 also includes transistors MX12, MX23, MX34, MX45, and MX56 as third switching elements connected between adjacent input terminals X1 to X6. When the virtual synthesis active unit V5612 is configured, the transistors MX55 and MX56 are controlled to be in the ON state and the transistor MX66 is controlled to be in the OFF state, so that the detection signals S5 and S6 of the two detection signal lead-out lines 308-5 and 308-6 are added at the output terminal XA5.

[0120] Next, the operation of the virtual synthesis active unit V5612 will be described with reference to FIG. 66. FIG. 66 schematically shows the timing chart of the signals of each part in FIGS. 64 to 65 when detecting the area A1 where the finger touches.

[0121] First, when connection terminals Y1 and Y2, which are connected to transistor control wirings 330-1 and 330-2 connected to active units 350-51, 350-52, 350-61, and 350-62 constituting virtual synthesis active unit V5612 including region A1, are collectively selected and activated at time t1 by Y-addition control circuit 460, transistors M51, M52, M61, and M62 of the four active units 350-51, 350-52, 350-61, and 350-62 connected to the transistor control wirings 330-1 and 330-2 turn on.

[0122] The detection signal is a current resulting from a change in the potential of the detection electrode in response to a change in the potential of a finger to which an alternating potential is applied by the housing voltage application method. Detection signals S51 and S52 respectively represent detection signals flowing through transistors M51 and M52, and an alternating current flows when each transistor is in the on state. Then, detection signals S51 and S52 are respectively read from transistors M51 and M52 to detection signal lead 308-5, added together, and the combined detection signal S5 (S51 + S52) propagates to input terminal X5 of detection signal addition circuit 450.

[0123] Similarly, detection signals S61 and S62 respectively represent detection signals flowing through transistors M61 and M62, and an alternating current flows when each transistor is in the on state. Then, detection signals S61 and S62 are respectively read from transistors M61 and M62 to detection signal lead 308-6, added together, and the combined detection signal S6 (S61 + S62) propagates to input terminal X6 of detection signal addition circuit 450.

[0124] Since the combined detection signals S5 and S6 are controlled such that transistor MX66 is in the off state and transistors MX55 and MX56 are in the on state in detection signal addition circuit 450, S5 and S6 are added together at output terminal XA5 to generate combined detection signal SA5 (S51 + S52 + S61 + S62).

[0125] Finally, the composite detection signal SA5 is input to the connection terminal 363 of the X-control circuit 362. That is, the composite detection signal SA5, which is the combination of the detection signals of the four active units 350-51, 350-52, 350-61, and 350-62, is input to the X-control circuit 362. Thereby, the virtual composite active unit V5612 can generate the composite detection signal SA5 in which the detection signals of the individual active units are added. The composite detection signal SA5 input to the X-control circuit 362 is subjected to detection signal processing by the method described with reference to FIGS. 6 to 12, for example. Therefore, in the sensor device of Embodiment 2 in which the virtual composite active unit V5612 is configured, even when the detection signals of the individual active units are small, the area A1 where the finger touches can be detected. Also, in Embodiment 2, the time to activate the connection terminals Y1 and Y2 connected to the transistor control wirings 330-1 and 330-2 is made to coincide with the timing to turn on transistors such as MX55 and MX56 in the detection signal addition circuit 450. This is to avoid complication of the explanation, and actually, there may be cases where the timings are not made to coincide by appropriately shifting the timings in consideration of the delay due to the CR time constant of each wiring. However, whether or not these timings coincide does not affect the explanation of the effects of the present invention.

[0126] [Modification Example of Embodiment 2] In Embodiment 2, the case where one virtual composite active unit V5612 is composed of four active units connected to two adjacent transistor control wirings 330-1 and 330-2 and two adjacent detection signal lead-out lines 308-5 and 308-6 has been described. However, as described above, the number of the active units is not limited to four. That is, the number of transistor control wirings and detection signal lead-out lines connected to the plurality of active units constituting the virtual composite active unit is not limited to two, and they do not have to be equal to each other.

[0127] For example, when one virtual composite active unit is composed of three adjacent active units and is connected to three detection signal lead-out lines 308-4, 308-5, and 308-6, the detection signal addition circuit 450 may be controlled such that transistors MX55 and MX66 are in the OFF state and MX44, MX45, and MX56 are in the ON state, as shown in FIG. 67. In this case, the synthesized detection signal SA4 obtained by adding the detection signals read out on the three detection signal lead-out lines 308-4, 308-5, and 308-6 is input from the output terminal XA4 to the X-control circuit 362. As a result, even when the detection signals of the individual active units are small, the region A1 where the finger has touched can be detected.

[0128] Thus, according to Embodiment 2 of the present invention and its modified examples, within the range satisfying the above ω≦ω2_max, a virtual composite active unit of an arbitrary rectangular size can be formed from a plurality of active units, and a synthesized detection signal obtained by adding the detection signals of the individual active units can be generated, so that the detection sensitivity can be improved.

[0129] [Embodiment 3] Next, the sensor device according to Embodiment 3 of the present invention will be described with reference to FIGS. 68 to 69. In the sensor device of Embodiment 3, the configuration of the X-addition control circuit is particularly different from that of the sensor device of Embodiment 2. Embodiment 3 is the same as Embodiment 2 except that the X-addition control circuit 562 of Embodiment 3 is replaced by the X-addition control circuit 462 of Embodiment 2 shown in FIG. 65, which has a different configuration. Therefore, the same reference numerals are given to the same parts as those in Embodiment 2, and only the configuration different from that of Embodiment 2 will be described.

[0130] As shown in FIG. 68, the X-addition control circuit 562 includes an X-control circuit 362A and a detection signal addition circuit 550. The detection signal addition circuit 550 includes input terminals X1 to 6 and output terminals XB1 to 6. The input terminals X1 to 6 are connected to the respective six detection signal lead-out lines 308-1 to 6, and the output terminals XB1 to 6 are connected to the respective connection terminals 363 of the X-control circuit 362A.

[0131] The detection signal addition circuit 550 includes, as a plurality of transistors 540 constituting a multiplexer, each of the transistors MX11, MX22, MX33, MX44, MX55, and MX66 as a second switching element connected between the input terminals X1 to 6 and the addition terminals XA1 to 6, and each of the transistors MX12, MX23, MX34, MX45, and MX56 as a third switching element connected between adjacent input terminals X1 to 6.

[0132] Each of the addition terminals XA1 to 6 of the detection signal addition circuit 550 includes capacitors CB1 to 6 as charge storage means connected at one end via each of M1C1 to 6 of the plurality of transistors 540 constituting the multiplexer. Also, each of the addition terminals XA1 to 6 is connected to each of the output terminals XB1 to 6 via each of M2C1 to 6 of the plurality of transistors 540 constituting the multiplexer. Incidentally, the other ends of the capacitors CB1 to 6 are grounded.

[0133] Next, the operation of the detection signal addition circuit 550 according to Embodiment 3 of the present invention will be described.

[0134] First, as shown in FIG. 68, the detection signal addition circuit 550 is set to an addition mode in which the transistors MX55, MX56, and M1C5 are controlled to be in the ON state and the transistor MX66 is controlled to be in the OFF state. Similar to FIG. 66, the virtual synthesis active unit V5612 has the terminals Y1 and Y2 to which the transistor control wirings 330-1 and 330-2 are connected activated at time t1 by the Y-addition control circuit 460. Then, since the transistors MX55 and MX56 of the detection signal addition circuit 550 are controlled to be in the ON state and the transistor MX66 is controlled to be in the OFF state, the detection signals S5 and S6 read out on the two detection signal lead-out lines 308-5 and 308-6 are added at the addition terminal XA5 to generate a synthesized detection signal SA5. Then, the synthesized detection signal SA5 is stored as detection charge in the capacitor CB5 via the transistor M1C5.

[0135] Next, after a predetermined time has elapsed, the detection signal addition circuit 550 is set to a detection mode that controls the transistors M1C5 and M2C5 to be in the ON state and MX55 to be in the OFF state, as shown in FIG. 69. Then, the detection charge of the combined detection signal SA5 stored in the capacitor CB5 is input to the X-control circuit 362A via the transistor M2C5 and the output terminal XB5. As a result, the virtual combined active unit V5612 can generate the combined detection signal SA5 obtained by adding the detection charges of the individual active units, so that even when the detection signals of the individual active units are small, the area A1 where the finger has made contact can be detected. More specifically, the transistor M1C5 and the capacitor CB5 form a kind of integration circuit. That is, due to the application of the housing voltage, the finger makes an AC potential fluctuation, and the detection signals S5 and S6 resulting from this are AC currents whose phases are synchronized with the potential fluctuation of the finger. The combined detection signal SA5 is also an AC current with a synchronized phase. By turning on the transistor M1C5 in synchronization with the positive polarity period of the combined detection signal SA5, the detection charge is additively accumulated in the capacitor CB5, and as the amount of detection charge accumulated increases, the electrode potential of the capacitor CB5 connected to the transistor M1C5 increases additively. By comparing the values of the charge amount accumulated in the capacitor CB5 via the transistors M1C5 and M2C5, or the electrode potential of the capacitor CB5 connected to the transistor M1C5, between when the finger is in contact and when it is not in contact using the X-control circuit 362A, a contact event can be detected.

[0136] Thus, according to the third embodiment of the present invention, within the range satisfying the above ω ≦ ω2_max, the virtual combined active unit V5612 is formed from a plurality of active units, and the combined detection signal SA5 obtained by adding the detection charges of the individual active units can be generated, so that the detection sensitivity can be improved.

[0137] Also, the detection signals S5 and S6 of the two detection signal lead-out lines 308-5 and 308-6 are once stored in the capacitor CB5 via the transistor M1C5, and the total amount of detected charge integrated within a predetermined time is temporarily input to the X-control circuit 362A via the transistor M2C5. Therefore, the transistor control wirings 330-1 and 330-2 that control the active units 350-51, 350-52, 350-61, and 350-62 constituting the virtual synthesis active unit V5612 do not need to be simultaneously activated and only need to be activated within a predetermined time. That is, if the timing of the Y-addition control circuit 460 that activates the transistor control wirings 330-1 and 330-2 of the virtual synthesis active unit V5612 is designed in a line sequential manner similar to the display operation period, it is not necessary to include in the Y-addition control circuit 460 a function of switching the control signal output timing of the transistor control wirings 330-1 and 330-2 between the display operation and the detection function operation. Therefore, cost reduction can be achieved by simplifying the circuit configuration. In this case, since the period during which the transistors M51, M52, M61, and M62 are in the ON state is halved, the amount of charge stored in the capacitor CB5 is also halved. However, in addition to the reduction of the basic current by the chassis voltage addition method, within the range satisfying the above ω≦ω2_max, the detection sensitivity can be appropriately maintained by increasing the number of transistors arranged in the active units 350-51, 350-52, 350-61, and 350-62.

[0138] Also, in Embodiments 2 to 3, each of the plurality of transistors 440 and 540 that constitute the multiplexer of the detection signal addition circuits 450 and 550 can be appropriately controlled to be in an ON state or an OFF state according to, for example, the rectangular size of the virtual composite active unit V5612 composed of a plurality of active units. The control electrodes of the transistors 440 and 540 are connected to the output terminals of the register circuit and are configured to be set by appropriately rewriting the stored register values. Thereby, by changing the number of active units constituting the virtual composite active unit by a touch panel operation or the like, the detection sensitivity and the resolution, which is the minimum area that can be detected, can be appropriately changed. Further, the control electrode may be set by a programmable (rewritable) means such as a fuse, SRAM (static random access memory), DRAM (dynamic random access memory), flash memory, etc. instead of the register circuit. Thereby, since the number of detection signal lead-out lines 308-1 to 6 added in the detection signal addition circuits 450 and 550 can be easily changed, the detection sensitivity can be easily changed.

[0139] Also, in Embodiments 2 to 3, the detection signal addition circuits 450 and 550 have been described as being arranged inside the X-addition control circuits 462 and 562. However, the present invention is not limited to this, and for example, the same effects can be obtained even if they are arranged in a circuit region inside the touch panel outside the X-addition control circuits 462 and 562.

[0140] So far, the present invention has been described with reference to the specific embodiments shown in the drawings. However, the present invention is not limited to the embodiments shown in the drawings, and it goes without saying that any known configuration can be adopted as long as the effects of the present invention can be achieved.

[0141] Also, so far, the present invention has been described in the case where a transistor as a first switching element constituting a multiplexer is arranged in a pixel. However, it goes without saying that the present invention is also applicable when it is arranged in a frame area.

Explanation of Reference Numerals

[0142] 100 Scanning signal line 101 Pixel electrode 102 Semiconductor layer 103 Channel protection layer 104 Pixel electrode - transistor connection contact hole 105 Source or drain electrode 106 Source or drain electrode 107 Display signal line 108 Detection signal lead-out wire 109 Detection signal lead-out wire - detection electrode connection contact hole 110 Detection electrode 111 Active unit 112 Detection electrode slit 120, 120a~120c Detection electrodes 121, 121a~121c Detection signal lead-out wires 122 Electrical connection point 123, 123a~123c Control circuit connection terminals 124 Control circuit 125 Liquid crystal display panel 130 Opposite substrate (Fig. 11) 131 Liquid crystal layer 132 Insulating film layer 133 Electric lines of force 134 Detection capacitance 135 Detection signal lead-out wire 136 Finger 140 Multiplexer 141 Transistor 142 Gate electrode 143 Source or drain electrode 144 Source or drain electrode 145 Selection signal line 146 Multiplexer control circuit connection terminal 150 Detection electrode 151 Transistor 152 Detection signal lead-out wire 153 Control circuit connection terminal 154 Transistor control wiring 155 Electrical connection point 200, 200a First region 201 Second region 202 Third region 203 Fourth region 204, 204a Fifth region 210 Detection electrode potential node 211 Detection signal lead-out line potential node 300 Scanning signal line 301 Pixel electrode 302 Semiconductor layer 303 Channel protection layer 304 Pixel electrode - transistor connection contact hole 305 Source or drain electrode 306 Source or drain electrode 307 Display signal line 308 Detection signal lead-out line 310 Detection electrode 312 Detection electrode slit 314 Display function main transistor 330 Transistor control wiring 332 Semiconductor layer 333 Channel protection layer 335 Source or drain electrode 336 Source or drain electrode 339 Detection signal lead-out line - transistor connection contact hole 340 Transistor constituting multiplexer (first switching element) 350 Active unit 351 Three - pixel integrated detection electrode 360 Y - control circuit 361 Y - control circuit connection terminal 362, 362A X - control circuit 363 X - control circuit connection terminal 440, 540 Transistor constituting multiplexer (second and third switching elements) 450, 550 Detection signal addition circuit 460 Y - addition control circuit 462 and 562 X-Addition Control Circuit Additive average capacitance of Ctft transistor Additive average resistance of Rtft transistor Additive average value of Csen timely formed capacitance Additive average capacitance of non-transistor part of detection signal lead corresponding to length of one active unit Additive average capacitance of detection signal lead in border area Nsy Number of active units arranged in detection signal lead direction Additive average capacitance formed between electrode functioning as source or drain connected to detection signal lead and gate electrode Synthetic capacitance of transistor Synthetic resistance of transistor Synthetic capacitance of detection signal lead Synthetic resistance of detection signal lead Synthetic detection capacitance Angular frequency for first pseudo-bending point Angular frequency for second pseudo-bending point ω2_max The maximum value of ω2 for each active unit N Number of transistors arranged in active unit m Number of detection signal leads to which transistors in active unit are connected n Number of transistors connected to detection signal lead in active unit n_i Number of transistors connected to the i-th detection signal lead in active unit Nsqrt_int Integer part of square root of N Vf Electrode potential to which applied AC voltage is applied Vf_amp Amplitude of applied AC voltage Vline Potential of detection signal lead μ Field effect mobility Vth Threshold voltage L Channel length W Channel width Cox Unit area capacitance Vgs Gate-source voltage Vds Drain-source voltage Vg Gate electrode potential V5612 Virtual synthetic active unit CB1~6 Capacitor

Claims

1. An active unit having an active circuit region arranged at least substantially in a matrix shape and including at least one or more detection electrodes; a first switching element connected to the detection electrode; one or more detection signal lead-out lines connected in parallel to each other and connected to the detection electrode via the first switching element; and a control circuit connected to the detection signal lead-out line. When a detected object approaches or contacts the active unit, a capacitance is timely formed between the detection electrode and the detected object, and an AC voltage is applied to the timely formed capacitance by the control circuit, so that an electrical signal generated at the detection electrode is selectively transmitted to the control circuit via the detection signal lead-out line and the first switching element. In a sensor device that detects an approach or contact event of the detected object by signal processing in the control circuit, where the number of arrangements of the first switching element in the active unit is N, the capacitance and resistance of the α (α = 1... N) -th first switching element are Ctf t_α and Rtf t_α respectively, and the timely formed capacitance is Csen_α, the combined capacitance C1 of the first switching element is expressed by Equation (1A) as C1 = Σ(α = 1... N) Ctf t_α......(1A) the combined resistance R1 of the first switching element is expressed by Equation (1B) as R1 = 1 / {Σ(α = 1... N) 1 / Rtf t_α}......(1B) the combined capacitance C4 of the timely formed capacitance is expressed by Equation (1C) as C4 = Σ(α = 1... N) Csen_α......(1C) Let the number of the detection signal lead-out lines to which the first switching element is connected in the active unit be m, the number of the first switching elements connected to the i-th (i = 1... m) detection signal lead-out line in the active unit be n_i, the number of the active units arranged in the direction of the detection signal lead-out line be Nsy, the capacitance of the non-switching element part of the detection signal lead-out line corresponding to the length of one active unit from the connection terminal of the detection signal lead-out line connected to the control circuit in the j-th (j = 1... Nsy) detection signal lead-out line direction be Cline-s_ij, the capacitance formed between the electrode functioning as the source or drain of the k-th (k = 1... n_i) first switching element connected to the detection signal lead-out line in the active unit and the gate electrode be Cgl_ijk, the capacitance of the detection signal lead-out line in the frame region which is the region outside the active unit be Clin_b, when Σ(i = 1... m) is a symbol representing the sum from i = 1 to m, etc., the combined capacitance C2 of the detection signal lead-out line from the connection terminal of the detection signal lead-out line to the active unit is expressed by Equation (1D) as C2 = Σ(i = 1... m) {Σ(j = 1... Nsy) Cline-s_ij + Σ(j = 1... Nsy - 1) {Σ(k = 1... n_i) Cgl_ijk}} + Clin_b......(1D) Let the resistance of the detection signal lead-out line corresponding to the length of one active unit of the j-th (j = 1... Nsy) active unit connected to the i-th (i = 1... m) detection signal lead-out line be Rline-s_ij, and the resistance of the detection signal lead-out line in the frame region be Rline_b. When this is the case, the combined resistance R2 of the detection signal lead-out line from the connection terminal of the detection signal lead-out line connected to the control circuit to the active unit is expressed by Equation (1E) as R2 = 1 / {Σ(i = 1... m) {1 / {Σ(j = 1... Nsy) Rline-s_ij}} + Rline_b......(1E) Let ω2 for each active unit be expressed by Equation (1F) with "^" representing the exponentiation symbol as ω2 = {φ^(1 / 2)} / {C2(C1 + C4)R1R2}......(1F) φ = {(C1 + C4)^2}(R1^2) + {(C1 + C2 + C4)^2}(R2^2) + 2{(C1 + C4)^2}R1R2 When the maximum value of ω2 among the ω2 values for each of the active units is defined as ω2_max, the angular frequency ω of the AC voltage applied to the timely formed capacitance satisfies ω ≦ ω2_max A sensor device characterized in that A plurality of virtual synthetic active units arranged in a substantially matrix form, each of which is constituted by a plurality of adjacent active units; A sensor device comprising detection signal addition means for adding electrical signals generated at respective detection electrodes of the active units constituting the virtual synthetic active unit. **Claim 2** The detection signal addition means is connected to the detection signal lead-out lines connected to the respective detection electrodes of the active units constituting the virtual synthetic active unit, and a second switching element capable of controlling selective conduction of current between the plurality of detection signal lead-out lines, and a third switching element capable of controlling selective conduction of current between the detection signal lead-out line and the control circuit. The sensor device according to claim 1, characterized in that it comprises. **Claim 3** The control electrode of the first switching element is connected to a transistor control wiring for controlling selective conduction of current between the detection electrode and the detection signal lead-out line. The plurality of transistor control wirings corresponding to the virtual synthetic active units are configured to be collectively selected when detecting an approach or contact event of the object to be detected. The sensor device according to claim 2, characterized in that. **Claim 4** The second switching element, the third switching element, and the collectively selected transistor control wiring are programmably connected to a register. The sensor device according to claim 3, characterized in that. **Claim 5** The detection signal addition means is connected to the detection signal lead-out line, and includes charge storage means capable of storing charges of electrical signals generated at the plurality of detection electrodes of the active unit when the transistor control wiring is selected, and transmitting the stored charges of the charge storage means to the control circuit at a predetermined timing. The sensor device according to claim 1, characterized in that. **Claim 6** In the active units arranged in the substantially matrix form, when the integer part of the square root of N is represented as Nsqrt_int, m satisfies at least m ≧ Nsqrt_int. The sensor device according to claim 1, characterized in that. **Claim 7** The sensor device according to claim 1, wherein the ratio m / N of m to N increases as the position of the active unit moves away from the connection terminal of the detection signal lead wire connected to the control circuit.

8. The sensor device according to claim 1, wherein C4 increases as the position of the active unit moves away from the connection terminal of the detection signal lead wire connected to the control circuit.

9. The sensor device further comprises a basic current reduction mechanism, The sensor device according to claim 1, wherein the basic current reduction mechanism is configured to apply the AC voltage to the housing of the sensor device and apply a reference voltage to the connection terminal of the detection signal lead wire connected to the control circuit.

10. When the amplitude of the AC voltage applied to the timely formation capacitance due to the contact event is Vf_amp, the electrode potential of the timely formation capacitance on the side to which the AC voltage is applied is Vf, the gate electrode potential of the α (α = 1... N) -th first switching element in the selection period is Vg, and the potential of the detection signal lead wire is Vline, Vds is expressed by Equation (2A), Vds = {Csen_α / (Csen_α + Ctft_α)} Vf_amp...... (2A) When Vline is the reference potential, Vgs is expressed by Equation (2B), Vgs = Vg - Vline...... (2B) When Vline is not the reference potential, Vgs is expressed by Equation (2C), Vgs = Vg - Vf...... (2C) When the threshold voltage of the transistor is Vth, the field - effect mobility is μ, the capacitance per unit area is Cox, the channel length is L, the channel width is W, and "^" is the exponentiation symbol, when Vds ≤ Vgs - Vth, Ids is expressed by Equation (2D) Ids = (W / L) μCox{(Vgs - Vth) Vds - (1 / 2) (Vds^2)}...... (2D) When Vds > Vgs - Vth, when Ids is expressed by Equation (2E), Ids = (1 / 2) (W / L) μCox (Vgs - Vth)^2...... (2E) The resistance Rtft_α of the first switching element is Rtft_α = Vds / Ids The sensor device according to claim 1, characterized in that.

11. The sensor device according to claim 1, wherein at least one of the detection electrodes has two or more first switching elements connected thereto.

12. The sensor device according to claim 1, wherein in the active unit, at least two of the detection signal lead-out wires are short-circuited.

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