On-board charging device

Abstract

This device, which consists of a non-isolated circuit, suppresses leakage current and other issues in an on-board charging device that charges the on-board DC power supply using power from an external AC power supply, compared to conventional designs. [Solution] An on-board charging device 10 for a vehicle equipped with an inverter 5 that converts power between AC flowing through the multi-phase coils 7 of a rotating electric machine 70 and DC from a DC power supply 3, charges the DC power supply 3 of the vehicle with power supplied from an external AC power supply 4. The device comprises an AC / DC converter 1 connected to the external AC power supply 4 via a pair of input boost inductors Le and equipped with a T-type power factor correction circuit, and an active decoupling circuit 2 configured using the coils 7 and the inverter 5. The T-type power factor correction circuit includes a split decoupling capacitor Cdc in which two capacitors are connected in series between the positive and negative terminals of the DC, with the positive terminal connected to the positive terminal of the inverter 5 and the negative terminal connected to the negative terminal of the inverter 5.

Description

[Technical Field]

[0001] This invention relates to an in-vehicle charging device. [Background technology]

[0002] In recent years, electric vehicles and plug-in hybrid vehicles have become widespread and are equipped with an onboard charger that charges the DC power supply that powers the traction motor (the source of power for the wheels) using an external AC power supply while the DC power supply is mounted on the vehicle. Traditionally, these onboard chargers have sometimes used circuits with isolation transformers. However, in order to shorten the charging time of the large-capacity onboard battery, increasing the voltage of the external power supply increases the voltage rating and current capacity of the isolation transformer, and the size of the isolation transformer also increases, leading to increased costs. In addition, charging devices that use isolation transformers generally have an active bridge circuit using switching elements, and the total number of switching elements used in the onboard charger tends to be large.

[0003] Japanese Patent Publication No. 2024-115454 discloses an on-board charging device that charges a DC power supply while mounted on a vehicle without using an isolation transformer (in the background art, the reference numerals in parentheses indicate the references to the document). As shown in Figure 3 of the said document, this on-board charging device (10) comprises an AC-DC converter (1) connected to an external AC power supply (4) and a DC-DC converter (2) connected to a DC power supply (3) to be charged. The DC-DC converter (2) is configured with a bidirectional chopper comprising a leg with switching elements connected in series, an inductor with one end connected to the midpoint of the leg, and an output capacitor connected between the other end of the inductor and the negative DC terminal. The leg of this bidirectional chopper is configured to also function as a multi-phase inverter (5) that drives a traction motor (70), and the inductor of the bidirectional chopper is configured to also function as a multi-phase coil (7) of the traction motor (70). The output capacitor (C2) of the bidirectional chopper is located between the neutral point (7N), where multiple phase coils (7) are connected to each other, and the DC negative electrode. [Prior art documents] [Patent Documents]

[0004] [Patent Document 1] Japanese Patent Publication No. 2024-115454 [Overview of the project] [Problems that the invention aims to solve]

[0005] The on-board charging device disclosed in the above-mentioned published patent is a non-isolated circuit configured without using an isolation transformer, thus eliminating the need for an isolation transformer itself and making it easier to keep the circuit size of the surrounding circuits small. Furthermore, by sharing the traction motor drive system (inverter, coil), the circuit size and additional costs when mounting the on-board charging device on a vehicle can be kept small. On the other hand, considering that charging a DC power supply is performed by a person, similar to refueling, it is preferable to suppress leakage current from the on-board charging device and the voltage drop that occurs when such leakage current flows through objects in contact with the vehicle (hereinafter referred to as "leakage current, etc." as appropriate). While suppressing leakage current is easier with an isolated circuit, it is preferable to further reduce leakage current in an on-board charging device using a non-isolated circuit that allows for smaller circuit size.

[0006] In light of the above, it is desirable to suppress leakage current and other issues in an on-board charging device that is composed of a non-isolated circuit and charges the on-board DC power supply with power from an external AC power supply, compared to conventional designs. [Means for solving the problem]

[0007] In view of the above, the on-board charging device comprises a multi-phase coil connected to each other at a neutral point, a rotating electric machine that serves as a driving force source for the wheels, an inverter that converts power between DC and multi-phase AC, and a DC power supply connected to the inverter, and charges the DC power supply of a vehicle with power supplied from an external AC power supply, wherein the on-board charging device comprises an AC-DC converter that converts power between AC power on the external AC power supply side and a first DC power, a storage capacitor, the inverter that functions as a charge / discharge control circuit for the storage capacitor, and the multi-phase coil that functions as a storage inductor connecting the inverter and the storage capacitor. The AC-DC converter comprises an active decoupling circuit that generates a second DC power for charging the DC power supply by reducing the pulsation component from the first DC power, the AC-DC converter being connected to the external AC power supply via a pair of input boost inductors, the AC-DC converter comprising a T-type power factor correction circuit, the T-type power factor correction circuit comprising a split decoupling capacitor in which two capacitors are connected in series between the positive and negative DC electrodes, the positive terminal of the split decoupling capacitor being connected to the positive terminal of the inverter, and the negative terminal of the split decoupling capacitor being connected to the negative terminal of the inverter.

[0008] Relatively low-frequency leakage current to ground is caused by the second harmonic of the frequency of the external AC power supply. This leakage current can be reduced by splitting the input boost inductor at the input section of the AC / DC converter into a pair of input boost inductors, which cancels out the current phases flowing through them. Relatively high-frequency leakage current to ground is caused by the switching of the switching elements in the on-board charging device and is generated by voltage fluctuations at the midpoint of the AC / DC converter. One possible method to suppress this leakage current is to place a floating filter between the node between the load connected to the DC power supply and the DC power supply, and between the node between the AC / DC converter and the external AC power supply. However, with this configuration, by providing a T-type power factor correction circuit and a divided decoupling capacitor, the voltage at the midpoint can be stabilized and high-frequency leakage current to ground can be reduced. In other words, high-frequency leakage current to ground can be suppressed with a small-scale configuration without using a floating filter. Thus, with this configuration, leakage currents and the like in an on-board charging device that charges the on-board DC power supply with power from an external AC power supply can be suppressed more than in conventional systems.

[0009] Further features and advantages of the on-board charging device will become clear from the following description of exemplary and non-limiting embodiments, which will be illustrated with reference to the drawings. [Brief explanation of the drawing]

[0010] [Figure 1] Schematic circuit block diagram of a drive control system for a rotating electric machine. [Figure 2] Schematic block diagram of an in-vehicle charging system [Figure 3] Block diagram showing a circuit model of a charging system with an additional filter to suppress leakage current. [Figure 4] Circuit block diagram showing the configuration principle of the in-vehicle charging device of the first embodiment. [Figure 5] Circuit block diagram showing the configuration of the in-vehicle charging device of the first embodiment. [Figure 6] Waveform diagram showing the control signal of the switch section of the first embodiment. [Figure 7] Logic circuit diagram for generating the control signal of the switch section of the first embodiment [Figure 8] Diagram showing the current flow in the AC-DC converter of the first embodiment (positive half-cycle of grid voltage and grid current, charging the input boost inductor) [Figure 9] Diagram showing the current flow in the AC-DC converter of the first embodiment (positive half-cycle of grid voltage and grid current, charging the split decoupling capacitor) [Figure 10] Diagram showing the current flow in the AC-DC converter of the first embodiment (negative half-cycle of grid voltage and grid current, charging the input boost inductor) [Figure 11] Diagram showing the current flow in the AC-DC converter of the first embodiment (negative half-cycle of grid voltage and grid current, charging the split decoupling capacitor) [Figure 12] Diagram showing the current flow in the active decoupling circuit of the first embodiment (charging the storage inductor from the split decoupling capacitor) [Figure 13] Diagram showing the current flow in the active decoupling circuit of the first embodiment (charging the storage capacitor from the storage inductor) [Figure 14] Diagram showing the current flow in the active decoupling circuit of the first embodiment (charging the storage inductor from the storage capacitor) [Figure 15] Diagram showing the current flow in the active decoupling circuit of the first embodiment (charging the split decoupling capacitor from the storage inductor) [Figure 16] Circuit block diagram showing the configuration principle of the in-vehicle charging device of the second embodiment [Figure 17] Circuit block diagram showing the configuration of the in-vehicle charging device of the second embodiment [Figure 18] Waveform diagram showing the control signal of the switch section of the second embodiment [Figure 19] Logic circuit diagram for generating the control signal of the switch section of the second embodiment [Figure 20] Diagram showing the current flow in the AC-DC converter of the second embodiment (positive half-cycle of grid voltage and grid current, charging the input boost inductor) [Figure 21] Diagram showing the current flow in the AC-DC converter of the second embodiment (positive half-cycle of grid voltage and grid current, charging the split decoupling capacitor) [Figure 22] Diagram showing the current flow in the AC-DC converter of the second embodiment (negative half-cycle of grid voltage and grid current, charging the input boost inductor) [Figure 23] Diagram showing the current flow in the AC-DC converter of the second embodiment (negative half-cycle of grid voltage and grid current, charging the split decoupling capacitor) [Figure 24] Circuit block diagram showing the configuration principle of the in-vehicle charging device of the third embodiment [Figure 25] Circuit block diagram showing the configuration of the in-vehicle charging device of the third embodiment [Figure 26] Waveform diagram showing the control signal of the switch section of the third embodiment [Figure 27] Logic circuit diagram for generating the control signal of the switch section of the third embodiment [Figure 28] Diagram showing the current flow in the AC-DC converter of the third embodiment (positive half-cycle of grid voltage and grid current, charging the input boost inductor) [Figure 29] Diagram showing the current flow in the AC-DC converter of the third embodiment (positive half-cycle of grid voltage and grid current, charging the first split decoupling capacitor) [Figure 30] Diagram showing the current flow in the AC-DC converter of the third embodiment (negative half-cycle of grid voltage and grid current, charging the input boost inductor) [Figure 31] Diagram showing the current flow in the AC-DC converter of the third embodiment (negative half-cycle of grid voltage and grid current, charging the second split decoupling capacitor) [Figure 32]Figure 28 shows the current flow including the DC link balancer circuit (charging the input boost inductor during a positive half-cycle of grid voltage and grid current). [Figure 33] Figure 29 shows the current flow including the DC link balancer circuit (charging the first split decoupling capacitor during a positive half-cycle of grid voltage and grid current). [Figure 34] Figure 30 shows the current flow including the DC link balancer circuit (grid voltage and grid current are negative for half a cycle, charging the input boost inductor). [Figure 35] Figure 31 shows the current flow including the DC link balancer circuit (grid voltage and grid current are negative for half a cycle, charging the second split decoupling capacitor). [Figure 36] Circuit block diagram showing an example of a conventional in-vehicle charging device. [Figure 37] Circuit block diagram showing another example of a conventional in-vehicle charging device. [Figure 38] Circuit block diagram showing an example of an in-vehicle charging device equipped with an active decoupling circuit. [Figure 39] A functional block diagram obtained by modifying the circuit block diagram shown in Figure 38. [Figure 40] Diagram showing current flow in an active decoupling circuit (during storage capacitor charging) [Figure 41] Diagram showing current flow in an active decoupling circuit (during storage capacitor charging) [Figure 42] Diagram showing current flow in an active decoupling circuit (during storage capacitor discharge) [Figure 43] Diagram showing current flow in an active decoupling circuit (during storage capacitor discharge) [Modes for carrying out the invention]

[0011] The embodiments of the on-board charging device will be described below with reference to the drawings. The circuit block diagram in Figure 1 schematically shows the drive control system of the rotating electric machine 70, and the block diagram in Figure 2 shows an example of an on-board charging system equipped with the on-board charging device 10. The on-board charging device 10 is a device that charges the DC power supply 3 provided in the vehicle 9, as shown in Figure 1, while it is mounted on the vehicle. As shown in Figure 1, the vehicle 9 in this embodiment is equipped with a rotating electric machine 70, which is a traction motor that serves as a driving force source for the wheels of a hybrid vehicle or electric vehicle, an inverter 5 that converts power between DC and multi-phase AC (in this case, 3-phase), and a DC power supply 3 connected to the inverter 5. It should be noted that this does not preclude the vehicle 9 from also being equipped with other driving force sources such as an internal combustion engine (not shown) in addition to the rotating electric machine 70. Furthermore, the rotating electric machine 70 can function as both an electric motor (traction motor) and a generator.

[0012] As shown in Figure 2, the onboard charging device 10 is a device that charges a DC power supply 3 with power supplied from an external AC power supply 4 (grid power supply, single-phase in this case), and is a device known as an onboard charger. As will be described in detail later, in the preferred embodiments illustrated in this specification (the first to third embodiments), the onboard charging device 10 is configured by sharing the inverter 5 and the coil 7 of the rotating electric machine 70.

[0013] As shown in Figure 1, the vehicle 9 is equipped with a control device 8 that controls a rotating electric machine 70. The control device 8 drives the rotating electric machine 70 by performing current feedback control. The rotating electric machine 70 to be driven is an Interior Permanent Magnet Synchronous Motor (IPMSM) having a stator with multiple phases (N phases, where N is an arbitrary natural number; here, a 3-phase configuration with N=3 is exemplified) of coils 7 (stator coils) and a rotor with field magnets (here, permanent magnets). Such a configuration is well known, and the illustration of the stator, rotor, permanent magnets, etc. is omitted. Also, here, a Y-type (star-type) configuration in which the 3-phase coils 7 are short-circuited at the neutral point 7N is exemplified. However, the rotating electric machine 70 may also have, for example, two sets of 3-phase coils 7 and be driven by a 6-phase AC.

[0014] As shown in Figure 1, the drive control system for the rotating electric machine 70 includes an inverter 5. The inverter 5 is connected to the AC rotating electric machine 70 and the DC power supply 3 to convert power between multi-phase AC and DC. The inverter 5 has U-phase legs, V-phase legs, and W-phase legs as multi-phase legs. Each leg is formed by connecting an upper switching element 5U located on the positive terminal side and a lower switching element 5L located on the negative terminal side in series. The midpoint of each leg, i.e., the connection point between the upper switching element 5U and the lower switching element 5L, is connected to the coil 7 of each phase. Specifically, the midpoint of the U-phase leg is connected to the U-phase coil 7u, the midpoint of the V-phase leg is connected to the V-phase coil 7v, and the midpoint of the W-phase leg is connected to the W-phase coil 7w. The DC side of the inverter 5 is provided with a smoothing capacitor (DC link capacitor 6) that smooths the voltage between the positive and negative terminals (DC link voltage).

[0015] The DC power supply 3 is composed of, for example, a rechargeable secondary battery such as a lithium-ion battery or an electric double-layer capacitor. In this embodiment, when the rotating electric machine 70 is a traction motor, the DC power supply 3 is a high-voltage, high-capacity DC power supply, and its rated power supply voltage is, for example, 200 to 800 [V]. In most cases, the DC power supply 3 is configured to be able to disconnect and reconnect its electrical connection to the inverter 5 and the rotating electric machine 70 by operating the vehicle's main switch (ignition switch), etc. For example, the DC power supply 3 is connected to the inverter 5 and the rotating electric machine 70 via a main contactor 11 composed of relays, etc. The DC power supply 3 is also equipped with a current sensor (battery current sensor 31) that detects the input and output current to the DC power supply 3 (battery current Ibat: see Figure 39) and a voltage sensor (battery voltage sensor 32) that detects the terminal voltage of the DC power supply 3 (battery voltage Vbat: see Figure 39, etc.).

[0016] Although not shown in the diagram, if the DC power source 3 is, for example, a lithium-ion battery, a Battery Management System (BMS) is often provided. Secondary batteries such as lithium-ion batteries are composed of multiple cells (battery cells). A Battery Management System is a battery management and control system that performs the following functions: (1) preventing overcharging and over-discharging of cells, (2) preventing overcurrent from flowing through cells, (3) managing the temperature of cells, (4) calculating the State of Charge (SOC), and (5) equalizing the cell voltage. The battery current sensor 31 and battery voltage sensor 32 mentioned above may be configured as part of the Battery Management System.

[0017] As shown in Figure 1, the inverter 5 is configured with multiple switching elements. It is preferable to use power semiconductor elements capable of high-frequency operation, such as IGBTs (Insulated Gate Bipolar Transistors), power MOSFETs (Metal Oxide Semiconductor Field Effect Transistors), SiC-MOSFETs (Silicon Carbide - Metal Oxide Semiconductor FETs), SiC-SITs (SiC - Static Induction Transistors), and GaN-MOSFETs (Gallium Nitride - MOSFETs), as the switching elements. In the various circuits illustrated in this specification, including the circuit of the on-board charging device 10, unless otherwise specified, the examples illustrate configurations in which FETs are used as switching elements. Furthermore, in this embodiment, a freewheeling diode is connected in parallel to each switching element, with the forward direction being from the negative electrode to the positive electrode (from the lower stage to the upper stage). Naturally, when using a body diode or when using reverse conduction in a wide-bandgap semiconductor transistor composed of enhancement-mode gallium nitride (E-GaN), it is not necessary to have a freewheeling diode of this form.

[0018] As shown in Figure 1, the inverter 5 is controlled by the control device 8. The control device 8 is constructed with logic circuits such as a microcomputer as its core components. For example, the control device 8 drives the rotating electric machine 70 via the inverter 5 by performing current feedback control using the vector control method, based on the target torque (torque command) of the rotating electric machine 70 provided as a request signal from other control devices such as the vehicle control device 90, which is one of the higher-level control devices. In the vector control method, the currents flowing through the coils 7 of each phase (U-phase current, V-phase current, W-phase current in this embodiment) are transformed into vector components of the d-axis, which is the direction of the magnetic field generated by the permanent magnets arranged in the rotor, and the q-axis, which is perpendicular to the d-axis (a direction advanced by π / 2 in electrical angle with respect to the direction of the magnetic field), and feedback control is performed. The coordinate system to which the coordinate transformation is performed is called the dq-axis orthogonal coordinate system. The operating voltage of the logic circuit elements such as the microcomputer is approximately 3.3 to 5 volts. In this embodiment, for the sake of simplicity, the diagram is omitted, but the control signals generated by the logic circuit elements are transmitted to the inverter 5 via the drive circuit. Alternatively, one could consider that the control device 8 includes a drive circuit.

[0019] The actual current flowing through the coils 7 of each phase of the rotating electric machine 70 is detected by a current sensor (motor current sensor 81), and the control device 8 acquires the detection results. In addition, the magnetic pole position (electric angle) and the rotational speed (angular velocity) of the rotor of the rotating electric machine 70 at each point in time are detected by a rotation sensor 82, such as a resolver or an inductive position sensor, and the control device 8 acquires the detection results. The control device 8 performs current feedback control using the detection results of the motor current sensor 81 and the rotation sensor 82. The control device 8 is configured to have various functional parts for current feedback control, and each functional part is realized through the cooperation of hardware such as a microcomputer and software (program).

[0020] As described above, the rotating electric machine 70, which is connected to the DC power supply 3 via the inverter 5, can also function as a generator and charge the DC power supply 3. For example, in a hybrid vehicle, it is possible to supply mechanical energy to the rotating electric machine 70 using power from an internal combustion engine or the like, causing the rotating electric machine 70 to generate electricity. However, opportunities for power generation are infrequent, and the DC power supply 3 may not be sufficiently charged. Also, in an electric vehicle that is only equipped with a rotating electric machine 70 as a driving force source, power generation is limited to mechanical energy from the wheels during coasting, etc., and the DC power supply 3 is often not sufficiently charged. Furthermore, even in a hybrid vehicle, it may be more energy efficient to supply power from an external source than to have the rotating electric machine 70 generate power. For this reason, it is preferable that the DC power supply 3 be configured to be rechargeable by an external power source while it is mounted on the vehicle 9.

[0021] As shown in Figure 2, the on-board charging device 10 of this embodiment charges a DC power supply 3 with power supplied from an external AC power supply 4. For example, the on-board charging device 10 is configured to include a front-end circuit 1 on the side of the external AC power supply 4 and a back-end circuit 2 on the side of the DC power supply 3. The on-board charging device 10 includes a grid contactor 13 that selectively connects the front-end circuit 1 and the external AC power supply 4. The grid contactor 13 is also configured, for example, by a relay. When the grid contactor 13 is closed, the grid contactor 13 becomes conductive, and the external AC power supply 4 and the on-board charging device 10 (front-end circuit 1) are electrically connected.

[0022] The following describes various forms of the on-board charging device 10, including conventional circuit configurations. As will be described later, all forms of the on-board charging device 10 are equipped with switching elements, and these switching elements are also controlled by the control device 8, which drives and controls the rotating electric machine 70 via the inverter 5. Furthermore, as will be described later, in addition to the main contactor 11 and grid contactor 13 mentioned above, the on-board charging device 10 also includes a function switching contactor 12, which will be described later. These contactors are also controlled by the control device 8, or by the control device 8 and the vehicle control device 90. The control device 8 is used for both driving control of the rotating electric machine 70 and charging control of the DC power supply 3.

[0023] In many cases, an EMI filter, such as the one labeled "EF1" in Figure 3, is placed between the external AC power supply 4 and the on-board charging device 10 to reduce EMI (Electromagnetic Interference) noise. However, for the sake of simplicity in illustration and explanation, the EMI filter is not shown in the illustration, including in the conventional circuit configuration. Also, the external AC power supply 4 is often equipped with a residual current protection device such as an RCD, but for the same reason, it is not shown in the illustration.

[0024] An in-vehicle charging device 10 using an isolation transformer is known. However, if the voltage of the external AC power supply 4 is increased in order to shorten the charging time of the DC power supply 3 with a large power capacity, the voltage rating and current capacity of the isolation transformer also increase, and the size of the isolation transformer also increases, leading to an increase in cost. In addition, charging devices using isolation transformers generally often include an active bridge circuit using switching elements, and the number of switching elements used in the entire in-vehicle charging device 10 tends to be large.

[0025] Therefore, an on-board charging device 10 has been proposed that charges the DC power supply 3 while mounted on the vehicle without using an isolation transformer. Figure 36 shows a conventional configuration example of such an on-board charging device 10. This on-board charging device 10 includes an AC-DC converter which is a front-end circuit 1 and a DC-DC converter which is a back-end circuit 2.

[0026] The AC-DC converter in front-end circuit 1 is configured with an input boost inductor Le and a full-bridge circuit using switching elements. In AC-DC conversion, a phase difference occurs between the voltage phase and the current phase due to inductive and capacitive impedances, and this phase difference reduces the power factor. The AC-DC converter is configured with switching elements, and by controlling the switching elements, for example, the current phase can be adjusted, the phase difference can be compensated for, and the power factor can be improved. In other words, the AC-DC converter is configured to function as a power factor correction (PFC) circuit that improves the power factor of the DC power converted from the AC power supplied from the external AC power source 4.

[0027] An AC-DC converter equipped with a full-bridge circuit can function as a bidirectional converter capable of AC-DC conversion and DC-AC conversion. For example, an on-board charging device 10 can be used as a device having two functions: one for charging a DC power source 3 with power supplied from an external AC power source 4, and the other for supplying AC power to devices outside the vehicle using the power stored in the DC power source 3. In recent years, it has been proposed to use the DC power source 3 of electric vehicles and hybrid vehicles as an emergency power source in the event of a disaster. By equipping the AC-DC converter with a full-bridge circuit, the DC power source 3 can be used as such an emergency power source.

[0028] As shown in Figure 36, the AC-DC converter can be configured by sharing the drive system circuit of the rotating electric machine 70. Specifically, one leg of the full-bridge circuit is configured using the inverter 5, and the input boost inductor Le can be configured using the multi-phase (in this case, 3-phase) coils 7 of the rotating electric machine 70. The 3-phase legs constituting the inverter 5 are connected in parallel, forming an apparent single leg, and together with the single-phase leg, they form a full-bridge circuit. The multi-phase coils 7 are short-circuited at the neutral point 7N and can be considered as a single inductor. The input boost inductor Le is configured using the multi-phase coils 7. As will be described later, the input boost inductor Le is composed of coil 7 and an additional inductor La. The neutral point 7N of coil 7 is connected to one terminal of the external AC power supply 4 via the additional inductor La, and the midpoint of the single-phase leg in the full-bridge circuit is connected to the other terminal of the external AC power supply 4.

[0029] The backend circuit 2, a DC-DC converter, is formed independently without sharing the drive system circuit of the rotating electric machine 70. The DC-DC converter is configured as a DC-DC converter (chopper) comprising a leg with switching elements connected in series to be switched and controlled complementaryly, an output inductor Lo with one end connected to the midpoint of the leg, and an output capacitor Co connected between the other end of the output inductor Lo and the negative terminal of the DC. The backend circuit 2 is a bidirectional converter (bidirectional chopper) that functions as a buck converter (buck chopper) in the direction from the frontend circuit 1 to the DC power supply 3, and as a boost converter (boost chopper) in the direction from the DC power supply 3 to the frontend circuit 1.

[0030] The on-board charging device 10 includes a main contactor 11 and a grid contactor 13, as well as a function switching contactor 12. The function switching contactor 12 is also configured, for example, by a relay. The main contactor 11 includes a contact (first main contactor 11a) that is closed when power is supplied from the DC power supply 3 to the rotating electric machine 70 via the inverter 5, and a contact (second main contactor 11b) that is closed when power is supplied to the DC power supply 3 via the on-board charging device 10. The function switching contactor 12 selectively switches between a first function that connects the inverter 5 and the DC power supply 3 to form a circuit that drives the rotating electric machine 70 as shown in Figure 1, and a second function that connects the AC / DC converter (inverter 5) of the front-end circuit 1 and the DC / DC converter of the back-end circuit 2 to form the on-board charging device 10. The function switching contactor 12 includes a contact (first function contactor 12a) that is closed when the first function is selected, and a contact (second function contactor 12b) that is closed when the second function is selected.

[0031] In this configuration, as described above, the coil 7 of the rotating electric machine 70 is used as the input boost inductor Le in the front-end circuit 1. However, the rotating electric machine, which is the driving force source of the vehicle, is designed to have a high power density and therefore has a small inductance. For this reason, in this configuration, when AC to DC conversion occurs in the front-end circuit 1, the peak value of the harmonic current of the grid current of the external AC power supply 4 tends to be large. In other words, the total harmonic distortion (THD) of the input grid current tends to be high.

[0032] While increasing the control frequency of the AC / DC converter can reduce total harmonic distortion, this also necessitates shortening the control period (control frequency) of the control device 8. Switching at high frequencies increases losses in the switching elements of the DC / DC converter, leading to a decrease in the efficiency of the on-board charging device 10. Furthermore, it becomes necessary to use a microcomputer or similar device capable of high-speed operation as the control device 8, leading to an increase in the cost of the on-board charging device 10. For this reason, to compensate for the insufficient inductance of coil 7, the necessary inductance is secured as the input boost inductor Le by using coil 7 and an additional inductor La. In other words, while coil 7 is used as the input boost inductor Le, the need for an additional inductor La tends to limit the cost reduction effect.

[0033] Figure 37 shows an example of a conventional on-board charging device 10 that improves upon this point. In this on-board charging device 10, the AC-DC converter of the front-end circuit 1, among the AC-DC converter and DC-DC converter, is configured without using the drive system circuit of the rotating electric machine 70 (the coil 7 of the rotating electric machine 70 and the inverter 5 that drives the rotating electric machine 70). The input boost inductor Le is configured with an inductor of appropriate constants.

[0034] The DC-DC converter in the backend circuit 2, which does not require a higher inductance than the input boost inductor Le, is configured using the same drive system circuit as the rotating electric machine 70. Specifically, in the DC-DC converter, a leg in which switching elements are connected in series to be switched complementaryly is configured using an inverter 5. As described above, the three-phase leg constituting the inverter 5 appears to constitute a single leg. The output inductor Lo is configured using the multi-phase (three-phase in this case) coils 7 of the rotating electric machine 70. The neutral point 7N of the coil 7 and the output capacitor Co are connected via a function switching contactor 12.

[0035] This on-board charging device 10 also includes a main contactor 11, a function switching contactor 12, and a grid contactor 13. The main contactor 11, like the on-board charging device 10 described above with reference to Figure 36, has two contacts: a first main contactor 11a and a second main contactor 11b. The function is also the same, so a detailed explanation is omitted. The function switching contactor 12, like the on-board charging device 10 described above with reference to Figure 36, is a contactor that selectively switches between a first function and a second function, but is composed of one contact. When the first function is selected, the function switching contactor 12 is opened, and when the second function is selected, the function switching contactor 12 is closed.

[0036] As shown in Figures 36 and 37, a decoupling capacitor is connected between the front-end circuit 1 and the back-end circuit 2, that is, between the AC-DC converter and the DC-DC converter. This decoupling capacitor is the same DC link capacitor 6 used in the rotating electric machine drive system shown in Figure 1. The pulsating component of the DC power generated by the AC-DC converter (the AC component Prpl, described later: see Figure 39, etc.) is reduced by utilizing the smoothing effect of the decoupling capacitor. Since the voltage used to charge the DC power supply 3 is higher than the voltage output from the DC power supply 3, the DC link capacitor 6 requires a higher capacity (voltage rating, etc.) than when acting as a smoothing capacitor in the drive system of the rotating electric machine 70. As a result, the capacitance of the DC link capacitor 6 needs to be larger than when it is used only in the drive control system of the rotating electric machine, which may increase the size of the DC link capacitor 6 and thus the cost.

[0037] Furthermore, in Figures 36 and 37, the main contactor 11 has two contacts: a first main contactor 11a and a second main contactor 11b. In particular, a large current flows through the second main contactor 11b, which is closed during charging. For this reason, the component size of the main contactor 11 tends to be relatively large, which increases the overall size of the on-board charging device 10 and easily increases the system cost.

[0038] Figure 38 shows an example of an on-board charging device 10 in view of this point. Compared with the on-board charging device 10 illustrated in Figure 37, this on-board charging device 10 has the same configuration of the front-end circuit 1 (AC-DC converter), but the back-end circuit 2 is configured as an active decoupling circuit instead of a bidirectional chopper. Furthermore, the on-board charging device 10 shown in Figure 38 does not have a decoupling capacitor (DC link capacitor 6) between the front-end circuit 1 and the back-end circuit 2 (or at the final stage of the front-end circuit 1).

[0039] Specifically, the on-board charging device 10 includes an AC-DC converter (front-end circuit 1) connected to an external AC power source 4 that converts AC power from the external AC power source 4 into first DC power, and an active decoupling circuit (back-end circuit 2) that generates second DC power from the first DC power to charge the DC power source 3 by reducing the pulsation component. The active decoupling circuit comprises a storage capacitor Cs, a leg (inverter 5) which functions as a charge / discharge control circuit for the storage capacitor Cs by connecting a complementary upper switching element 5U and a lower switching element 5L in series, and a multi-phase coil 7 which functions as a storage inductor Ls connecting the leg (inverter 5) and the storage capacitor Cs. The storage capacitor Cs is connected between the neutral point 7N of the coil 7 and the negative electrode of the DC.

[0040] In the on-board charging device 10 illustrated in Figure 37, the backend circuit 2 is simply a DC-DC converter, so the frontend circuit 1 and the backend circuit 2 can operate independently. However, in the on-board charging device 10 illustrated in Figure 38, the ripple generated in the frontend circuit 1 is dynamically reduced by the backend circuit 2. Therefore, the backend circuit 2 needs to operate in accordance with the ripple. In other words, the frontend circuit 1 (AC-DC converter) and the backend circuit 2 (active decoupling circuit) operate in coordination. To put it another way, the switching elements of the active decoupling circuit are controlled to operate in conjunction with the AC-DC converter.

[0041] This on-board charging device 10 also includes a main contactor 11, a function switching contactor 12, and a grid contactor 13. Unlike the on-board charging device 10 described above with reference to Figures 36 and 37, the main contactor 11 is configured to have only one contact. Therefore, it is easier to miniaturize the main contactor 11 compared to those devices. The main contactor 11 is closed in both cases: when power is supplied from the DC power supply 3 to the rotating electric machine 70 via the inverter 5, and when power is supplied to the DC power supply 3 via the on-board charging device 10. The function switching contactor 12 is a contactor that selectively switches between a first function and a second function, similar to the on-board charging device 10 described above with reference to Figure 37, and is configured by a single contactor. When the first function is selected, the function switching contactor 12 is open, and when the second function is selected, the function switching contactor 12 is closed. When the function switching contactor 12 is closed, the neutral point 7N of the coil 7 and the DC negative electrode are connected via the storage capacitor Cs.

[0042] When the inverter 5 functions as an on-board charging device 10, the main contactor 11 is controlled to close and the function switching contactor 12 is also controlled to close. That is, with both the main contactor 11 and the function switching contactor 12 closed, power is converted between the external AC power supply 4 and the DC power supply 3 via an active decoupling circuit including the AC / DC converter and the inverter 5. When the inverter 5 is used to drive the rotating electric machine 70, the main contactor 11 is controlled to close and the function switching contactor 12 and the grid contactor 13 are controlled to open. That is, with both the main contactor 11 closed and the function switching contactor 12 open, power is converted between the DC power supply 3 and the rotating electric machine 70 via the inverter 5.

[0043] In this way, the function switching contactor 12 makes it possible to realize a circuit that can switch between the function of driving and controlling the rotating electric machine 70 and the function of charging the DC power supply 3 with an external AC power supply 4 simply by changing the connection configuration of the coil 7. In other words, the on-board charging device 10 can be configured with a simple configuration.

[0044] The circuit block diagram in Figure 39 is a functional breakdown of the circuit block diagram in Figure 38 to facilitate understanding of the on-board charging device 10, which includes an AC / DC converter and an active decoupling circuit. The AC / DC converter boosts the voltage while adjusting the power factor of the grid current Igrid supplied from the external AC power supply 4 to a value close to "1". However, the first DC power output from the AC / DC converter has a pulsating component superimposed on it, which is due to the AC frequency of the external AC power supply 4. Specifically, the pulsation of the second harmonic component of the AC frequency of the external AC power supply 4 is superimposed. The active decoupling circuit reduces this pulsation of the second harmonic component and supplies the second DC power, which has had its pulsation suppressed, preferably the second DC power from which the pulsation has been removed, to the DC power supply 3 to charge the DC power supply 3.

[0045] Here, let the AC voltage supplied from the external AC power supply 4 be "grid voltage Vgrid" and the AC current be "grid current Igrid". If the amplitudes of each (the value from the amplitude center to the wave height, which is a positive value; the same applies to "_peak" below) are "Vgrid_peak" and "Igrid_peak", then the grid voltage Vgrid and grid current Igrid are expressed by the following equations (1) and (2).

[0046]

number

number

[0047] Assuming that the input power Pin input to the AC-DC converter from the external AC power source 4 is converted to a first DC power with a power factor of "1", the relationship between the input power Pin and the first DC power is given by equations (3) to (5) below. Equation (3) shows that the first DC power contains a DC component Pdc and an AC component Prpl. Equation (4) shows the DC component Pdc of the first DC power, and equation (5) shows the AC component Prpl, which is the pulsating component of the first DC power.

[0048]

number

number

number

[0049] The active decoupling circuit functions by flowing a current corresponding to the AC component Prpl shown in equation (5) into the active decoupling circuit so that the output power Pout to the DC power supply 3 consists only of the DC component Pdc shown in equation (4) (equation (6)). Therefore, the active decoupling circuit operates to match the capacitor power Pcs, which is the power in the storage capacitor Cs, with the AC component Prpl of the first DC power, as shown in equation (7) below.

[0050]

number

number

[0051] Since the relationship between the capacitor voltage Vcs and the capacitor current Ics is given by equation (8) below, the capacitor power Pcs can be controlled by appropriately controlling the capacitor current Ics.

[0052]

number

[0053] Figures 40 to 43 show the current flow in the active decoupling circuit. Figures 40 and 41 show the current flow when the storage capacitor Cs is being charged, with the capacitor current Ics flowing through the storage capacitor Cs being a positive value (with the direction of the arrow shown in Figure 39 being positive). Figures 42 and 43 show the current flow when the storage capacitor Cs is being discharged, with the capacitor current Ics flowing through the storage capacitor Cs being a negative value. In this in-vehicle charging device 10, the decoupling effect by the active decoupling circuit is achieved by using a discontinuous current (discontinuous current) due to pulse width modulation, rather than a so-called continuous current (continuous current).

[0054] As described above, the inverter 5 is configured with multiple legs (three in this case) in parallel, corresponding to the number of phases of the multi-phase (three in this case) AC, where an upper switching element 5U and a lower switching element 5L are connected in series between the positive and negative DC electrodes. The inverter 5 is configured with three legs. When the inverter 5 functions as part of the on-board charging device 10, multiple upper switching elements 5U and multiple lower switching elements 5L are switched simultaneously. In other words, the inverter 5, which has multiple phase legs corresponding to multi-phase AC, can be used as an inverter (switching circuit) with single-phase legs. Because the current flows distributed across multiple legs, wear and tear on the inverter 5 can be easily suppressed even when the inverter 5 is used as part of the on-board charging device 10.

[0055] Furthermore, since approximately the same magnitude of current flows through each of the multiple-phase coils 7, excluding individual differences and errors, almost no torque is generated in the rotor even when the coils 7 are also used in the on-board charging device 10. Since charging with the DC power supply 3 is performed when the vehicle is stopped, it is undesirable for torque to be generated in the rotor. For example, it is conceivable to interrupt the power transmission from the rotor to the wheels using a clutch, or to restrict the rotation of the output components connected to the wheels or the rotor using a so-called parking brake. However, in these cases, it can lead to increased complexity of the vehicle's drive system mechanism or create mechanical loads on the gears of the vehicle's drive system. However, when the coils 7 are used as part of the on-board charging device 10, the inverter 5 is controlled so that no torque is generated in the rotor, so almost no mechanical load is created on the gears of the vehicle's drive system.

[0056] The inverter 5, which functions as a charge / discharge control circuit for the storage capacitor Cs, controls the capacitor current command I *The inverter 5 is controlled by a modulated pulse generated by the control device 8 based on CS. The inverter 5 is controlled by the modulated pulse to one of three states: a first state in which all upper switching elements 5U are ON and all lower switching elements 5L are OFF; a second state in which all upper switching elements 5U are OFF and all lower switching elements 5L are ON; and a third state in which all upper switching elements 5U and all lower switching elements 5L are OFF. In the third state, all switching elements 5S of the inverter 5 are OFF.

[0057] Figure 40 shows the current flow when the capacitor current Ics flowing through the storage capacitor Cs increases in a positive value, with the direction of the arrow shown in Figure 39 being positive, thereby charging the storage capacitor Cs. The switching element of the inverter 5 is controlled to be in the first state. The AC component Prpl of the first DC power is consumed to charge the storage capacitor Cs, and approximately only the DC component Pdc of the first DC power is supplied to the DC power supply 3 as the second DC power.

[0058] Figure 41 shows the current flow when the storage capacitor Cs is being charged while the capacitor current Ics decreases to a positive value. The inverter switching elements are controlled to be in the third state. The capacitor current Ics decreases by recirculating between the storage capacitor Cs and the freewheeling diode connected in parallel with the lower switching element 5L, or by recirculating through the lower switching element 5L (when using the reverse conduction of the body diode or E-GaN transistor).

[0059] In other words, during the phase in which the storage capacitor Cs is charged, the lower switching element 5L is controlled to be constantly off, and the upper switching element 5U is switched by pulse width modulation control.

[0060] Figure 43 shows the current flow when the capacitor current Ics is negative and the absolute value of the capacitor current Ics is decreasing while the storage capacitor Cs is being discharged. The inverter's switching elements are controlled to be in the third state. The current flowing due to the charge discharged from the storage capacitor Cs flows to the DC power supply 3 through the freewheeling diode connected in parallel to the upper switching element 5U, or through the upper switching element 5U (when utilizing the reverse conduction of the body diode or E-GaN transistor). That is, the AC component Prpl of the first DC power is reduced by the charge discharged from the storage capacitor Cs, and approximately only the DC component Pdc of the first DC power is supplied to the DC power supply 3 as the second DC power.

[0061] Figure 42 shows the current flow when the capacitor current Ics is negative and the absolute value of the capacitor current Ics is increasing, discharging the storage capacitor Cs. The inverter switching elements are controlled to enter the second state. The absolute value of the capacitor current Ics increases while recirculating between the lower switching element 5L and the storage capacitor Cs.

[0062] In other words, during the discharge phase of the storage capacitor Cs, the upper switching element 5U is controlled to be constantly off, contrary to the charging phase, while the lower switching element 5L is switched by pulse width modulation control.

[0063] Furthermore, Capacitor current command I * The phase of cs coincides with the phase of the grid voltage Vgrid and the phase of the grid current Igrid. The grid voltage Vgrid and grid current Igrid are expressed by equations (1) and (2) above, and the AC component Prpl of the input power Pin is expressed by equation (5) above, so the capacitor current command I * cs is the capacitor current command I * The amplitude of cs is I *cs_peak is expressed by the following equation (9). Note that the phase of the capacitor voltage Vcs lags the phase of the capacitor current Ics by π / 2. The capacitor voltage Vcs is expressed by the following equation (10), where Vcs_peak is the amplitude of the capacitor voltage Vcs.

[0064]

number

number

[0065] As explained above, the inverter 5 controls the switching of multiple upper-stage switching elements 5U and multiple lower-stage switching elements 5L so that the phase of the current flowing through the storage capacitor Cs matches the phase of the AC current in the AC-DC converter. The active decoupling circuit attenuates the harmonic components of the AC in the AC-DC converter by passing them through the storage capacitor Cs. The attenuation effect can be enhanced by matching the phase of the current flowing through the storage capacitor Cs with the phase of the AC current in the AC-DC converter.

[0066] Referring to Figures 38 to 43, the above-described on-board charging device 10 is a non-isolated circuit constructed without the use of an isolation transformer, making it easier to keep the circuit size of the isolation transformer itself and its surrounding circuits small. Furthermore, by sharing the drive system (inverter 5, coil 7) of the rotating electric machine 70, the circuit size and additional costs when mounting the on-board charging device 10 on the vehicle 9 can also be kept small. In addition, the number of contacts of the contactor is also fewer than that of the on-board charging device 10 exemplified in Figure 37, making it easier to miniaturize the contactor. Moreover, since there is no capacitor placed between the front-end circuit 1 and the back-end circuit 2, it is not necessary to set the voltage rating and capacity of the DC link capacitor 6 according to the specifications of the on-board charging device 10, which is less likely to lead to an increase in size and cost.

[0067] On the other hand, considering that the DC power supply 3 is charged by a person, it is preferable that the leakage current from the on-board charging device 10 and the voltage drop that occurs when leakage current flows through the contacts with the vehicle 9 (hereinafter referred to as "leakage current, etc." as appropriate) be suppressed to the smallest possible extent. Suppression of leakage current is easier to achieve with a circuit using an isolation transformer, but the on-board charging device 10 of this embodiment, as illustrated below, has a non-isolated circuit configuration while still having a structure that sufficiently suppresses the generation of leakage current and the voltage drop caused by the flow of leakage current.

[0068] The following describes three circuit configurations of the first, second, and third embodiments. Before describing each embodiment, a circuit model for suppressing leakage current will be described with reference to Figure 3. The inventors verified the leakage current using a model in which a three-phase AC power supply 41 is connected to the on-board charging device 10 via a grid simulator 42 and a Line Impedance Stabilization Network (LISN) 43, as shown in Figure 3. The symbol "UG" represents the utility ground, the symbol "FG" represents the frame ground, the symbol "SG" represents the chassis ground of the vehicle 9, and the symbol "GLC" represents the leakage current to ground. The symbol "30" is a resistive load that also includes the DC power supply 3 to be charged. The symbol "HIM" is a human impedance model that assumes an object in contact with the vehicle 9, such as a human body. The symbol "Vbody" represents the voltage drop across an object in contact with the vehicle 9.

[0069] On the input side of the on-board charging device 10, i.e., between the external AC power supply 4 and the on-board charging device 10, the input common-mode EMI filter EF1 (AC-side common-mode EMI filter) consists of a first common-mode inductive filter Lcm_ac1, a second common-mode inductive filter Lcm_ac2, a first capacitive Y filter Cy_ac1, and a second capacitive Y filter Cy_ac2. On the output side of the on-board charging device 10, i.e., between the on-board charging device 10 and the resistive load 30 (DC power supply 3), the output common-mode EMI filter EF2 (DC-side common-mode EMI filter) consists of a third common-mode inductive filter Lcm_dc and a third capacitive Y filter Cy_dc. The ground of the Y filter is the chassis ground SG of the vehicle 9.

[0070] Floating filters FF are provided between the input common-mode EMI filter EF1 and the on-board charging device 10, and between the on-board charging device 10 and the output common-mode EMI filter EF2. A floating common-mode inductive filter Lf (AC-side common-mode inductive filter) and a first floating capacitive filter Cfac (AC-side floating capacitive filter) are positioned between the input common-mode EMI filter EF1 and the on-board charging device 10, and a second floating capacitive filter Cfdc (DC-side floating capacitive filter) is positioned between the on-board charging device 10 and the output common-mode EMI filter EF2. The negative terminal of the first floating capacitive filter Cfac and the negative terminal of the second floating capacitive filter Cfdc are connected so that their reference potentials are the same. The reference potentials of the first floating capacitive filter Cfac and the second floating capacitive filter Cfdc are floating grounds independent of various grounds (frame ground FG, chassis ground SG, etc.).

[0071] Leakage current to ground consists of relatively low-frequency leakage current and relatively high-frequency leakage current. Low-frequency leakage current to ground is caused by the second harmonic of the frequency of the external AC power supply 4. High-frequency leakage current is caused by the switching of the switching elements in the onboard charging device 10 and is generated by voltage fluctuations at the midpoint of the AC / DC converter in the front-end circuit 1. The "midpoint" corresponds to the midpoint of the split decoupling capacitor Cdc, which will be described later.

[0072] Experiments and simulations conducted by the inventors using this model revealed the following: (1) Low-frequency leakage current can be reduced by evenly distributing the input boost inductor Le at the input to the front-end circuit 1 across two AC lines from the single-phase external AC power supply 4. (2) High-frequency leakage current can be reduced by setting circuit constants corresponding to the switching frequency in the passive elements constituting the floating filter FF so that high-frequency leakage current passes through the floating filter FF.

[0073] Based on these findings, the inventors devised a new circuit configuration to reduce leakage current in the on-board charging device 10. In principle, this new circuit configuration is applicable to any of the circuit configurations shown in Figure 37, Figures 38 and 39. However, considering that (1) the circuit configurations shown in Figures 38 and 39 have been confirmed to have less leakage current to ground compared to the circuit configuration shown in Figure 37, and (2) they are advantageous in terms of the number of contacts of the contactor and the capacitance of the DC link capacitor 6 as described above, the following description will explain an example of how the configuration is applied to the on-board charging device 10 (configuration shown in Figures 38 and 39) which is equipped with an active decoupling circuit as the backend circuit 2.

[0074] Below, three embodiments—the first, second, and third embodiments—are given as examples of preferred circuit configurations. The first embodiment will be described with reference to Figures 4 to 15, the second embodiment with reference to Figures 16 to 23, and the third embodiment with reference to Figures 24 to 35. When describing matters common to the first, second, and third embodiments, they may be collectively referred to as "this embodiment." The on-board charging device 10 of this embodiment includes a front-end circuit 1 that functions as an AC-DC converter and a back-end circuit 2 that functions as an active decoupling circuit. The configuration and basic operation of the active decoupling circuit are as described above with reference to Figures 38 to 43, so when illustrating each embodiment, the explanation is omitted unless specifically mentioned.

[0075] The detailed configuration will be described for each embodiment, but the in-vehicle charging device 10 of this embodiment includes a front-end circuit 1 (AC-DC converter) that converts power between the AC power on the external AC power source 4 and the first DC power, and a back-end circuit 2 (active decoupling circuit) that generates a second DC power for charging the DC power source 3 by reducing the pulsation component from the first DC power. The active decoupling circuit is configured as described above with reference to Figures 38 to 43, and includes a storage capacitor Cs, a leg (inverter 5) that functions as a charge / discharge control circuit for the storage capacitor Cs, in which a complementary upper switching element 5U and a lower switching element 5L are connected in series, and a multi-phase coil 7 that functions as a storage inductor Ls connecting the leg (inverter 5) and the storage capacitor Cs (see also Figures 5, 17, and 25).

[0076] For example, as shown in Figures 4 and 5, the AC-DC converter is connected to an external AC power supply 4 via a pair of input boost inductors Le (see also Figures 16, 17, 24, and 25). The AC-DC converter also includes a T-type power factor correction circuit, which includes a T-type connection section 1T (the specific configuration will be described in each embodiment). As shown in the schematic diagram of Figure 4, the T-type power factor correction circuit includes a divided decoupling capacitor Cdc, in which two capacitors (first decoupling capacitor Cdc1, second decoupling capacitor Cdc2) are connected in series between the positive and negative terminals of the DC circuit. The first decoupling capacitor Cdc1 and the second decoupling capacitor Cdc2 have the same electrical specifications (capacitance, etc.). As shown in Figures 5, 17, and 25, the positive terminal of the divided decoupling capacitor Cdc is connected to the positive terminal of the inverter 5, and the negative terminal of the divided decoupling capacitor Cdc is connected to the negative terminal of the inverter 5.

[0077] In the front-end circuit 1, the DC voltage between the positive and negative poles is divided by the first decoupling capacitor Cdc1 and the second decoupling capacitor Cdc2. Therefore, high voltage ratings are not required for individual capacitors. Furthermore, the DC link capacitor 6 in the drive control system of the rotating electric machine 70 shown in Figure 1 can be shared with the divided decoupling capacitor Cdc.

[0078] As described above, relatively low-frequency leakage current to ground is reduced by splitting the input boost inductor Le at the input section of the AC / DC converter into a pair of input boost inductors Le (first input boost inductor Le1, second input boost inductor Le2), causing the current phases flowing through them to cancel each other out. Relatively high-frequency leakage current to ground can be reduced by a floating filter FF as described above. In this embodiment, instead of a floating filter FF, a T-type power factor correction circuit and a divided decoupling capacitor Cdc are used to reduce high-frequency leakage current to ground.

[0079] For example, as shown in Figures 4 and 5, the T-type power factor correction circuit comprises a leg with two switch sections (S5, S6) connected in series, and a midpoint switch section MS connected between the midpoint of the leg and the midpoint of the split decoupling capacitor Cdc. The T-type connection section 1T is formed by the leg and an arm including the midpoint switch section MS. As described above, high-frequency leakage current is caused by the switching of the switching elements in the on-board charging device 10. Therefore, high-frequency leakage current can be reduced by maintaining the voltage at the midpoint of the DC voltage (DC link voltage) of the AC-DC converter in the front-end circuit 1, i.e., the voltage at the midpoint of the split decoupling capacitor Cdc, at half the DC link voltage. The midpoint switch section MS makes it easier to appropriately stabilize the voltage at the midpoint of the split decoupling capacitor Cdc, which is the voltage at the midpoint of the AC-DC converter.

[0080] Thus, the in-vehicle charging device 10 of this embodiment can form a non-isolated charging circuit with a small circuit configuration by (1) reducing low-frequency leakage current to ground by equally dividing the input boost inductor Le and making it function as a common-mode inductive filter, and (2) reducing high-frequency leakage current to ground by replacing the floating filter FF with a T-type power factor correction circuit equipped with a divided decoupling capacitor Cdc.

[0081] Furthermore, in common to the first, second, and third embodiments, the on-board charging device 10 of this embodiment can be used as a device having two functions: a function for charging the DC power supply 3 with power supplied from an external AC power supply 4 (AC-DC conversion), and a function for supplying AC power to devices outside the vehicle using the power stored in the DC power supply 3 (DC-AC conversion). In recent years, it has been proposed to use the DC power supply 3 of electric vehicles and hybrid vehicles as an emergency power supply in the event of a disaster. By having an AC-DC converter equipped with a full-bridge circuit, the DC power supply 3 can be used as such an emergency power supply.

[0082] Here, the on-board charging device 10 of the first embodiment will be described with reference to Figures 4 to 15. The circuit block diagram in Figure 4 shows the configuration principle of the on-board charging device 10, and the circuit block diagram in Figure 5 shows the configuration of the on-board charging device 10 that shares the drive control system of the rotating electric machine. Figures 4 and 5 differ in whether or not the backend circuit 2 shares the drive control system of the rotating electric machine, but are otherwise the same. Therefore, in describing the operation, the configuration shown in the principle diagram of Figure 4 will be used for explanation.

[0083] As shown in Figures 4 and 5, the on-board charging device 10 of the first embodiment has a T-type power factor correction circuit with three legs: a first leg, a second leg, and a third leg. The first leg is configured with one bidirectional switch section BS in which two switch sections (fifth switch S5, sixth switch S6) are connected in series. The first leg is positioned between a pair of input terminals on the AC side. The second leg is configured with two switch sections (first switch S1, second switch S2) connected in series between the positive and negative DC electrodes. The third leg is configured with two switch sections (third switch S3, fourth switch S4) connected in series between the positive and negative DC electrodes. The second leg and the third leg form a full bridge circuit. The pair of input boost inductors Le comprises a first input boost inductor Le1 connected between the first end of the external AC power supply 4 and the first end of the first leg, and a second input boost inductor Le2 connected between the second end of the external AC power supply 4 and the second end of the first leg. The seventh switch S7, which serves as the midpoint switch MS, is connected between the midpoint of the first leg and the midpoint of the split decoupling capacitor Cdc. Additionally, the first end of the first leg is connected to the midpoint of the second leg, and the second end of the first leg is connected to the midpoint of the third leg.

[0084] The first switch S1, second switch S2, third switch S3, and fourth switch S4 form an active full-wave rectifier circuit using a full-bridge circuit. The first leg, which includes the fifth switch S5 and sixth switch S6, and the seventh switch S7, which is a midpoint switch MS, and the arm connecting the midpoint of the first leg to the midpoint of the split decoupling capacitor Cdc form a T-connection section 1T.

[0085] The front-end circuit 1 of the in-vehicle charging device 10 of the first embodiment is configured with seven switch sections. Of these, the T-type power factor correction circuit core, the T-type connection section 1T, which is composed of the fifth switch S5, the sixth switch S6, and the seventh switch S7, can be formed with only two switching elements by configuring the bidirectional switch section BS of the fifth switch S5 and the sixth switch S6 with one bidirectional switching element, and the seventh switch S7 with one monodirectional switching element. Here, a bidirectional switching element is a bidirectional switching element made of one semiconductor element, which can allow current to flow in both directions when both switch sections are ON, and can block current from flowing in either direction when both switch sections are OFF. In other words, with this configuration, the front-end circuit 1 can be configured with a very small number of switching elements, making it easy to reduce costs. Furthermore, the first switch S1, the second switch S2, the third switch S3, and the fourth switch S4 are each composed of monodirectional switching elements.

[0086] The fifth switch S5 and the sixth switch S6, which constitute the bidirectional switch section BS, may be connected in opposite directions to the configurations shown in Figures 4 and 5. In other words, in the configurations shown in Figures 4 and 5, the fifth switch S5 and the sixth switch S6 are arranged so that the cathodes of their diodes face each other, but they may also be arranged so that the anodes of their diodes face each other. Specifically, Figures 4 and 5 illustrate a drain-common connection configuration in which the drains of the fifth switch S5 and the sixth switch S6 face each other, but they may also be connected in a source-common configuration where the sources face each other.

[0087] Furthermore, the seventh switch S7, which constitutes the midpoint switch section MS connected between the midpoint of the leg formed by the series connection of the fifth switch S5 and the sixth switch S6, and the midpoint of the split decoupling capacitor Cdc, may also be connected in the opposite direction to the configuration shown in Figures 4 and 5. In other words, in the configuration shown in Figures 4 and 5, the seventh switch S7 is positioned so that the midpoint of the leg formed by the series connection of the fifth switch S5 and the sixth switch S6 faces the cathode of the diode of the seventh switch S7, but it may also be positioned so that the midpoint faces the anode of the diode. For example, Figures 4 and 5 illustrate a drain-common connection configuration in which the drains of the fifth switch S5, the sixth switch S6, and the seventh switch S7 are connected facing each other, but it may also be a source-common connection configuration in which the sources of the fifth switch S5, the sixth switch S6, and the seventh switch S7 are connected facing each other.

[0088] The in-vehicle charging device 10 of the first embodiment has the following excellent features: (1) a floating filter FF is not required; (2) the T-connection section 1T can be formed with a very small number (2) switching elements; (3) it is equipped with a power factor correction circuit, enabling AC-DC conversion with high efficiency; and (4) the input boost inductor can be set to an appropriate value, making it easy to keep the total harmonic distortion of the grid current low.

[0089] FIG. 6 shows the control signals of each switch section. The upper row shows the carrier CA for generating the control signal, the voltage command corresponding to the voltage “V AN ” at node “A” shown in FIGS. 4 and 5, and the voltage command corresponding to the voltage “V BN ” at node “B”. The control device 8 compares the voltage command with the carrier CA and generates a pulse that becomes active (ON) when the voltage command is greater than the carrier CA. FIG. 7 shows a logic circuit diagram for generating the control signals of each switch section. The control device 8 generates a signal “SG1” based on the voltage command corresponding to the voltage “V AN ” and generates a signal “SG2” based on the voltage command corresponding to the voltage “V BN ”.

[0090] The signal “SG1” directly becomes the control signals of the first switch S1 and the fourth switch S4, and the signal “SG2” directly becomes the control signals of the second switch S2 and the third switch S3. That is, in the full-bridge circuit constituting the full-wave rectifier circuit, the switch sections arranged at the diagonal positions are switched simultaneously.

[0091] The control signal of the fifth switch S5 is a signal obtained by inverting the logic of the signal “SG2”, and the control signal of the sixth switch S6 is a signal obtained by inverting the logic of the signal “SG1”. Also, the control signal of the seventh switch S7 is the logical product of the control signal of the fifth switch S5 and the control signal of the sixth switch S6. In other words, the control signal of the seventh switch S7 is a control signal that becomes effective when both the first switch S1 and the fourth switch S4 are in the off state and when both the second switch S2 and the third switch S3 are in the off state. From the waveform diagram in FIG. 6, when the first switch S1 and the fourth switch S4 are controlled to be in the off state, the second switch S2 and the third switch S3 are also controlled to be in the off state. Also, when the second switch S2 and the third switch S3 are controlled to be in the off state, the first switch S1 and the fourth switch S4 are also controlled to be in the off state. Therefore, the seventh switch S7 is controlled to be in the on state when all the switch sections constituting the full-bridge circuit are in the off state.

[0092] Figures 8 to 11 show the current flow in front-end circuit 1 (AC-DC converter with T-type power factor correction circuit). The circled switch sections are the switches that are controlled to the ON state. Figures 8 and 9 show the current during a positive half-cycle of grid voltage and grid current. Figure 8 shows the case when energy is charged to the input boost inductor Le(Le1,Le2), and Figure 9 shows the case when energy is charged to the split decoupling capacitor Cdc.

[0093] As shown in Figure 8, when the fifth switch S5 and sixth switch S6 of the first leg are controlled to the ON state, and the seventh switch S7 is also controlled to the ON state, a closed circuit is formed from the external AC power supply 4 to the external AC power supply 4, passing through the first input boost inductor Le1, the fifth switch S5, the sixth switch S6, and the second input boost inductor Le2. As a result, energy is transferred from the external AC power supply 4 to the input boost inductor Le. At this time, the first switch S1, the second switch S2, the third switch S3, and the fourth switch S4, which constitute the full bridge circuit, are controlled to the OFF state. Also, with the seventh switch S7 controlled to the ON state, the midpoint of the first leg and the midpoint of the split decoupling capacitor Cdc are connected.

[0094] As shown in Figure 9, when the first switch S1 and the fourth switch S4 are controlled to be ON at the same time, and the other switches are controlled to be OFF, a closed circuit is formed from the external AC power supply 4 through the first input boost inductor Le1, the first switch S1, the split decoupling capacitor Cdc, the fourth switch S4, and the second input boost inductor Le2 to the external AC power supply 4. As a result, energy is transferred from the input boost inductor Le (first input boost inductor Le1) to the split decoupling capacitor Cdc.

[0095] Figures 10 and 11 show the grid voltage and grid current during a negative half-cycle. Figure 10 also shows the case where energy is charged to the input boost inductor Le(Le1,Le2), and Figure 11 shows the case where energy is charged to the split decoupling capacitor Cdc.

[0096] As shown in Figure 10, when the fifth switch S5 and sixth switch S6 of the first leg are controlled to the ON state, and the seventh switch S7 is also controlled to the ON state, a closed circuit is formed from the external AC power supply 4 through the second input boost inductor Le2, the sixth switch S6, the fifth switch S5, and the first input boost inductor Le1 to the external AC power supply 4. As a result, energy is transferred from the external AC power supply 4 to the input boost inductor Le. All other switches are controlled to the OFF state. Also, with the seventh switch S7 controlled to the ON state, the midpoint of the first leg and the midpoint of the split decoupling capacitor Cdc are connected.

[0097] As shown in Figure 11, when the second switch S2 and the third switch S3 are controlled to be ON at the same time, and the other switches are controlled to be OFF, a closed circuit is formed from the external AC power supply 4 through the second input boost inductor Le2, the third switch S3, the split decoupling capacitor Cdc, the second switch S2, and the first input boost inductor Le1 to the external AC power supply 4. As a result, energy is transferred from the input boost inductor Le (second input boost inductor Le2) to the split decoupling capacitor Cdc.

[0098] Figures 12 to 15 show the current flow in backend circuit 2 (active decoupling circuit). Figures 12 and 13 show the current during a positive half-cycle of grid voltage and grid current.

[0099] Figure 12 shows the current flow when energy is transferred from the split decoupling capacitor Cdc to the storage inductor Ls, thereby charging the storage inductor Ls. The first backend switch S21 (corresponding to the upper switching element 5U of the inverter 5) is controlled to be ON, and the second backend switch S22 (corresponding to the lower switching element 5L of the inverter 5) is controlled to be OFF. A closed circuit is formed from the split decoupling capacitor Cdc, through the first backend switch S21, the storage inductor Ls, and the storage capacitor Cs, back to the split decoupling capacitor Cdc. The ripple energy of the second harmonic component of the grid current is bypassed to the active power decoupling circuit while the DC power supply 3 is supplied with a charging current (battery current Ibat).

[0100] Figure 13 shows the current flow when energy is transferred from the storage inductor Ls to the storage capacitor Cs, and the storage capacitor Cs is charged. Both the backend first switch S21 and the backend second switch S22 are controlled to the off state. As a result, a closed circuit is formed from the storage inductor Ls, through the storage capacitor Cs and the freewheeling diode of the switching element constituting the backend second switch S22, back to the storage inductor Ls.

[0101] Figure 14 shows the current flow when energy is transferred from the storage capacitor Cs to the storage inductor Ls, thereby charging the storage inductor Ls. The backend second switch S22 is controlled to the ON state, and the backend first switch S21 is controlled to the OFF state. This forms a closed circuit from the storage capacitor Cs, through the storage inductor Ls and the backend second switch S22, back to the storage capacitor Cs.

[0102] Figure 15 shows the current flow when energy is transferred from the storage inductor Ls to the split decoupling capacitor Cdc, and the split decoupling capacitor Cdc is charged. Both the backend first switch S21 and the backend second switch S22 are controlled to the off state. As a result, a closed circuit is formed from the storage capacitor Cs, through the storage inductor Ls, the freewheeling diode which is a switching element constituting the backend first switch S21, and the split decoupling capacitor Cdc, to the storage capacitor Cs.

[0103] Next, the on-board charging device 10 of the second embodiment will be described with reference to Figures 16 to 23. The circuit block diagram in Figure 16 shows the configuration principle of the on-board charging device 10, and the circuit block diagram in Figure 17 shows the configuration of the on-board charging device 10 that shares a rotating electric drive system. Figures 16 and 17 differ in whether or not the backend circuit 2 shares a rotating electric drive system; otherwise, they are the same. Therefore, in describing the operation, the configuration shown in the principle diagram of Figure 16 will be used for explanation.

[0104] As shown in Figures 16 and 17, the in-vehicle charging device 10 of the second embodiment includes a full-bridge circuit in which the T-type power factor correction circuit comprises a first leg in which two switch units (first phase first switch Su1, first phase fourth switch Su4) are connected in series between the positive and negative DC electrodes, and a second leg in which two switch units (second phase first switch Sv1, second phase fourth switch Sv4) are connected in series between the positive and negative DC electrodes. The pair of input boost inductors Le comprises a first input boost inductor Le1 connected between the first end of the external AC power supply 4 and the midpoint of the first leg, and a second input boost inductor Le2 connected between the second end of the external AC power supply 4 and the midpoint of the second leg.

[0105] The midpoint switch section MS comprises a first bidirectional switch section BS1, in which two switch sections (first phase third switch Su3, first phase second switch Su2) are connected in series to form one bidirectional switch section BS, and a second bidirectional switch section BS2, in which two switch sections (second phase third switch Sv3, second phase second switch Sv2) are connected in series to form one bidirectional switch section BS. The first bidirectional switch section BS1 is connected between the midpoint of the first leg and the midpoint of the split decoupling capacitor Cdc, and the second bidirectional switch section BS2 is connected between the midpoint of the second leg and the midpoint of the split decoupling capacitor Cdc.

[0106] The pair of switch sections (the pair of Su3 and Su2, and the pair of Sv3 and Sv2) constituting the first bidirectional switch section BS1 and the second bidirectional switch section BS2 may be connected in the opposite direction to the configuration shown in Figures 16 and 17. That is, in the configuration shown in Figures 16 and 17, the first phase third switch Su3 and the first phase second switch Su2 are arranged so that the anodes of their diodes face each other, and the second phase third switch Sv3 and the second phase second switch Sv2 are arranged so that the anodes of their diodes face each other, but they may also be arranged so that the cathodes of their diodes face each other. In other words, Figures 16 and 17 illustrate a source common connection configuration in which the first phase third switch Su3 and the first phase second switch Su2 are connected with their sources facing each other, but they may also be connected in a drain common configuration with their drains facing each other. Similarly, Figures 16 and 17 illustrate a source common connection configuration in which the second-phase third switch Sv3 and the second-phase second switch Sv2 are connected with their sources facing each other. However, a drain common connection configuration in which the drains are connected with each other is also possible.

[0107] The T-type power factor correction circuit comprises two T-type connections 1T. The first T-type connection 1T consists of a first leg in which two switch sections (first phase first switch Su1, first phase fourth switch Su4) are connected in series, and an arm that connects the midpoint of the first leg to the midpoint of the split decoupling capacitor Cdc via a first midpoint switch section MS1 ​​(first phase third switch Su3, first phase second switch Su2). The second T-type connection 1T consists of a second leg in which two switch sections (second phase first switch Sv1, second phase fourth switch Sv4) are connected in series, and an arm that connects the midpoint of the second leg to the midpoint of the split decoupling capacitor Cdc via a second midpoint switch section MS2 (second phase third switch Sv3, second phase second switch Sv2). In other words, in the second embodiment, the T-type power factor correction circuit can be said to include two sets of T-type connections 1T connected in parallel to the midpoint of the split decoupling capacitor.

[0108] The front-end circuit 1 of the in-vehicle charging device 10 in the second embodiment is configured with eight switch sections. Of these, the first bidirectional switch section BS1, which consists of a first-phase third switch Su3 and a first-phase second switch Su2, can be configured with one bidirectional switching element, and the second bidirectional switch section BS2, which consists of a second-phase third switch Sv3 and a second-phase second switch Sv2, can be configured with one bidirectional switching element. In other words, in the second embodiment, the core of the T-type power factor correction circuit can be configured using two bidirectional switching elements. Even including the full-bridge circuit, the front-end circuit 1 can be formed by providing six switching elements: four monodirectional switching elements and two bidirectional switching elements. In other words, in the second embodiment as well, the front-end circuit 1 can be configured with a very small number of switching elements, making it easy to reduce costs.

[0109] Similar to the first embodiment, the in-vehicle charging device 10 of the second embodiment also has the following excellent features: (1) a floating filter FF is not required; (2) the core of the T-type power factor correction circuit can be formed with a very small number (2) switching elements; (3) because it is equipped with a power factor correction circuit, AC-DC conversion is possible with high efficiency; and (4) the input boost inductor can be set to an appropriate value, making it easy to keep the total harmonic distortion of the grid current low.

[0110] Figure 18 shows the control signals for each switch section. The upper row shows the control signals for the first phase switch section (Su1, Su2, Su3, Su4), and the lower row shows the control signals for the second phase switch section (Sv1, Sv2, Sv3, Sv4). The top row of both the upper and lower sections shows the carrier CA and voltage command for generating the control signals, similar to Figure 6. The phase of the voltage command differs by 180 degrees (=π) between the first and second phases (the waveform is inverted in the amplitude direction). The pulse generation method is clear from the explanation in Figure 6, so a detailed explanation is omitted.

[0111] Figure 19 shows the logic circuit diagram for generating control signals for each switch section. The control device 8 generates signals "SG1" and "SG2". Signal "SG1" directly becomes the control signal for the first phase first switch Su1 and the second phase fourth switch Sv4, and signal "SG2" directly becomes the control signal for the first phase fourth switch Su4 and the second phase first switch Sv1. In other words, in the full-bridge circuit that constitutes the full-wave rectifier circuit, the switch sections arranged in a cross-connection position are switched simultaneously.

[0112] The control signals for the first phase third switch Su3 and the second phase second switch Sv2 are signals with the logic of signal "SG1" inverted, and the control signals for the first phase second switch Su2 and the second phase third switch Sv3 are signals with the logic of signal "SG2" inverted. As shown in the waveform diagram of Figure 18, the first phase third switch Su3, second phase second switch Sv2, first phase second switch Su2, and second phase third switch Sv3, which constitute the midpoint switch section MS, are controlled to turn ON when the first phase first switch Su1, first phase fourth switch Su4, second phase first switch Sv1, and second phase fourth switch Sv4, which constitute the full bridge circuit, are all OFF.

[0113] Figures 20 to 23 show the current flow in front-end circuit 1 (AC-DC converter with T-type power factor correction circuit). Figures 20 and 21 show the current during a positive half-cycle of grid voltage and grid current. Figure 20 also shows the case when energy is charged to the input boost inductor Le(Le1,Le2), and Figure 21 shows the case when energy is charged to the split decoupling capacitor Cdc.

[0114] As shown in Figure 20, when the first phase first switch Su1, the first phase fourth switch Su4, the second phase first switch Sv1, and the second phase fourth switch Sv4 constituting the full bridge circuit are all controlled to the OFF state, and the first phase third switch Su3, the first phase second switch Su2, the second phase second switch Sv2, and the second phase third switch Sv3 constituting the midpoint switch section MS are all controlled to the ON state, a closed circuit is formed from the external AC power supply 4 to the external AC power supply 4, passing through the first input boost inductor Le1, the first phase third switch Su3, the first phase second switch Su2, the second phase second switch Sv2, the second phase third switch Sv3, and the second input boost inductor Le2. As a result, energy is transferred from the external AC power supply 4 to the input boost inductor Le. The midpoint of the split decoupling capacitor Cdc is connected to the midpoint of the first bidirectional switch section BS1 and the second bidirectional switch section BS2 in the closed circuit.

[0115] As shown in Figure 21, when the first phase first switch Su1 and the second phase fourth switch Sv4 are controlled to be ON simultaneously, and the other switches are controlled to be OFF, a closed circuit is formed from the external AC power supply 4, passing through the first input boost inductor Le1, the first phase first switch Su1, the split decoupling capacitor Cdc, the second phase fourth switch Sv4, and the second input boost inductor Le2 to the external AC power supply 4. As a result, energy is transferred from the input boost inductor Le (first input boost inductor Le1) to the split decoupling capacitor Cdc.

[0116] Figures 22 and 23 show the grid voltage and grid current during a negative half-cycle. Figure 22 also shows the case where energy is charged to the input boost inductor Le(Le1,Le2), and Figure 23 shows the case where energy is charged to the split decoupling capacitor Cdc.

[0117] As shown in Figure 22, when the first phase first switch Su1, the first phase fourth switch Su4, the second phase first switch Sv1, and the second phase fourth switch Sv4 constituting the full bridge circuit are all controlled to the OFF state, and the first phase third switch Su3, the first phase second switch Su2, the second phase second switch Sv2, and the second phase third switch Sv3 constituting the midpoint switch section MS are all controlled to the ON state, a closed circuit is formed from the external AC power supply 4 to the external AC power supply 4, passing through the second input boost inductor Le2, the second phase third switch Sv3, the second phase second switch Sv2, the first phase second switch Su2, the first phase third switch Su3, and the first input boost inductor Le1. As a result, energy is transferred from the external AC power supply 4 to the input boost inductor Le. The midpoint of the split decoupling capacitor Cdc is connected to the midpoint of the first bidirectional switch section BS1 and the second bidirectional switch section BS2 in the closed circuit.

[0118] As shown in Figure 23, when the second-phase first switch Sv1 and the first-phase fourth switch Su4 are controlled to be ON simultaneously, and the other switches are controlled to be OFF, a closed circuit is formed from the external AC power supply 4 through the second input boost inductor Le2, the second-phase first switch Sv1, the split decoupling capacitor Cdc, the first-phase fourth switch Su4, and the first input boost inductor Le1 to the external AC power supply 4. As a result, energy is transferred from the input boost inductor Le (second input boost inductor Le2) to the split decoupling capacitor Cdc.

[0119] The operation of the backend circuit 2 (active decoupling circuit) is the same as described above for the first embodiment, so the explanation will be omitted.

[0120] Next, the on-board charging device 10 of the third embodiment will be described with reference to Figures 24 to 35. The circuit block diagram in Figure 24 shows the configuration principle of the on-board charging device 10, and the circuit block diagram in Figure 25 shows the configuration of the on-board charging device 10 that shares the drive control system of the rotating electric machine. Figures 24 and 25 differ in whether or not the backend circuit 2 shares the drive control system of the rotating electric machine, but are otherwise the same. Therefore, in describing the operation, the configuration shown in the principle diagram of Figure 24 will be used for explanation.

[0121] As shown in Figures 24 and 25, the in-vehicle charging device 10 of the third embodiment further includes a DC link voltage balancer circuit VB in the AC-DC converter of the front-end circuit 1. The T-type power factor correction circuit has a first leg in which a switch section (first switch S1, fourth switch S4) is connected in series between the positive and negative DC electrodes. The pair of input boost inductors Le includes a first input boost inductor Le1 connected between the first end of the external AC power supply 4 and the midpoint of the first leg, and a second input boost inductor Le2 connected between the second end of the external AC power supply 4 and the midpoint of the split decoupling capacitor Cdc. The midpoint switch section MS has a bidirectional switch section BS in which two switch sections (second switch S2, third switch S3) are connected in series. The bidirectional switch section BS is connected between the midpoint of the first leg and the midpoint of the split decoupling capacitor Cdc. The DC link voltage balancer circuit VB includes a second leg in which two switch sections (fifth switch S5, sixth switch S6) are connected in series between the positive and negative DC terminals, and a balancer inductor Lb connected between the midpoint of the split decoupling capacitor Cdc and the midpoint of the second leg.

[0122] A T-shaped connection section 1T is formed by a first leg in which a first switch S1 and a second switch S2 are connected in series, and an arm that connects the midpoint of the first leg and the midpoint of the split decoupling capacitor Cdc via a midpoint switch section MS.

[0123] The T-type power factor correction circuit of the on-board charging device 10 in the third embodiment is configured with four switch sections. Including the DC link voltage balancer circuit VB, the front-end circuit 1 is configured with six switch sections. Of these, the bidirectional switch section BS, which consists of the second switch S2 and the third switch S3, can be configured with one bidirectional switching element. That is, in the third embodiment, the core of the T-type power factor correction circuit is configured with one bidirectional switching element, and the T-type power factor correction circuit can be configured with three switching elements: one bidirectional switching element and two monodirectional switching elements. Even including the DC link voltage balancer circuit VB, the front-end circuit 1 can be formed by providing five switching elements: one bidirectional switching element and four monodirectional switching elements. That is, in the third embodiment as well, the front-end circuit 1 can be configured with a very small number of switching elements, making it easy to reduce costs.

[0124] Similar to the first and second embodiments, the in-vehicle charging device 10 of the third embodiment also has the following excellent features: (1) a floating filter FF is not required; (2) although an additional circuit (DC link voltage balancer circuit VB) is required, the core of the T-type power factor correction circuit can be formed with a very small number (1) switching element; (3) because it is equipped with a power factor correction circuit, AC-DC conversion is possible with high efficiency; and (4) the input boost inductor can be set to an appropriate value, making it easy to keep the total harmonic distortion of the grid current low.

[0125] Figure 26 shows the control signals for the first switch S1, the second switch S2, the third switch S3, and the fourth switch S4. The top row shows the carrier CA and voltage command for generating the control signals, similar to Figure 6. The method for generating pulses is clear from the explanation in Figure 6, so a detailed explanation is omitted.

[0126] Figure 27 shows a logic circuit diagram for generating control signals. The control device 8 generates signals "SG1" and "SG2". Signal "SG1" directly becomes the control signal for the first switch S1, and signal "SG2" directly becomes the control signal for the second switch S2. The control signal for the third switch S3 is the inverted logic of signal "SG1", and the control signal for the fourth switch S4 is the inverted logic of signal "SG2".

[0127] Figures 28 to 31 show the current flow in front-end circuit 1 (AC-DC converter with T-type power factor correction circuit). Figures 28 and 29 show the current during a positive half-cycle of grid voltage and grid current. Figure 28 shows the case when energy is charged to the input boost inductor Le(Le1,Le2), and Figure 29 shows the case when energy is charged to the first decoupling capacitor Cdc1.

[0128] As shown in Figure 28, when the first switch S1 and the fourth switch S4 constituting the half-bridge circuit are both controlled to the OFF state, and the third switch S3 and the second switch S2 constituting the midpoint switch section MS are both controlled to the ON state, a closed circuit is formed from the external AC power supply 4 to the external AC power supply 4, passing through the first input boost inductor Le1, the third switch S3, the second switch S2, and the second input boost inductor Le2. As a result, energy is transferred from the external AC power supply 4 to the input boost inductor Le. Since the third switch S3 and the second switch S2 constituting the midpoint switch section MS are both ON, the midpoint of the first leg and the midpoint of the split decoupling capacitor Cdc are connected.

[0129] As shown in Figure 29, when the first switch S1 is controlled to the ON state and the other switches are controlled to the OFF state, a closed circuit is formed from the external AC power supply 4 through the first input boost inductor Le1, the first switch S1, the first decoupling capacitor Cdc1, and the second input boost inductor Le2 to the external AC power supply 4. As a result, energy is transferred from the input boost inductor Le (first input boost inductor Le1) to the first decoupling capacitor Cdc1.

[0130] Figures 30 and 31 show the grid voltage and grid current during a negative half-cycle. Figure 30 also shows the case when energy is charged to the input boost inductor Le(Le1,Le2), and Figure 31 shows the case when energy is charged to the second decoupling capacitor Cdc2.

[0131] As shown in Figure 30, when the first switch S1 and the fourth switch S4 constituting the half-bridge circuit are both controlled to the OFF state, and the third switch S3 and the second switch S2 constituting the midpoint switch section MS are both controlled to the ON state, a closed circuit is formed from the external AC power supply 4 to the external AC power supply 4, passing through the second input boost inductor Le2, the second switch S2, the third switch S3, and the first input boost inductor Le1. As a result, energy is transferred from the external AC power supply 4 to the input boost inductor Le. Since the third switch S3 and the second switch S2 constituting the midpoint switch section MS are both ON, the midpoint of the first leg and the midpoint of the split decoupling capacitor Cdc are connected.

[0132] As shown in Figure 31, when the fourth switch S4 is controlled to the ON state and the other switches are controlled to the OFF state, a closed circuit is formed from the external AC power supply 4 through the second input boost inductor Le2, the second decoupling capacitor Cdc2, the fourth switch S4, and the first input boost inductor Le1 to the external AC power supply 4. As a result, energy is transferred from the input boost inductor Le (second input boost inductor Le2) to the second decoupling capacitor Cdc2.

[0133] As described above with reference to Figures 28 to 31, in the front-end circuit 1 of the third embodiment, of the divided decoupling capacitors Cdc, the first decoupling capacitor Cdc1 on the DC positive side is charged only during half-cycles (half-periods in electrical angle) when the grid current is positive, and the second decoupling capacitor Cdc2 on the DC negative side is charged only during half-cycles when the grid current is negative. For example, when the DC power supply 3 is charged with a constant battery current Ibat, fluctuations are observed in the terminal voltages of the two capacitors. The capacitance and voltage rating of the capacitors must be determined taking this fluctuation into consideration. Providing a margin in capacitance and voltage rating may lead to increased cost and size. To suppress fluctuations in terminal voltage, a DC link voltage balancer circuit VB is added to the front-end circuit 1. As described above, in order to suppress high-frequency leakage current to ground, it is important to maintain the voltage at the midpoint of the divided decoupling capacitor Cdc at half the voltage between the positive and negative terminals (DC link voltage). In the third embodiment, a DC link voltage balancer circuit VB is added to maintain the voltage at the midpoint of the split decoupling capacitor Cdc at "DC link voltage / 2".

[0134] As described above, the DC link voltage balancer circuit VB includes a second leg in which a switch section (fifth switch S5, sixth switch S6) is connected in series between the positive and negative DC terminals, and a balancer inductor Lb connected between the midpoint of the split decoupling capacitor Cdc and the midpoint of the second leg. The operation of the DC link voltage balancer circuit VB will be described below with reference to Figures 32 to 35.

[0135] Figures 32 to 35 show the current flow in the front-end circuit 1, including the DC link voltage balancer circuit VB. Figure 32 shows the current flow corresponding to Figure 28, Figure 33 shows the current flow corresponding to Figure 29, Figure 34 shows the current flow corresponding to Figure 30, and Figure 35 shows the current flow corresponding to Figure 31.

[0136] As described above with reference to Figure 28, when the first switch S1 and the fourth switch S4 are both controlled to the OFF state, and the third switch S3 and the second switch S2 are both controlled to the ON state, the input boost inductor Le is charged. As shown in Figure 32, at this time, the fifth switch S5 located on the positive side of the second leg is controlled to the ON state, and the sixth switch S6 located on the negative side is controlled to the OFF state. As a result, a closed loop is formed from the first decoupling capacitor Cdc1 through the fifth switch S5 and balancer inductor Lb to the first decoupling capacitor Cdc1, and the balancer inductor Lb is charged.

[0137] As described above with reference to Figure 29, when the first switch S1 is controlled to the ON state and the third switch S3 and fourth switch S4 are controlled to the OFF state, the first decoupling capacitor Cdc1 is charged. As shown in Figure 33, at this time, the sixth switch S6 located on the negative side of the second leg is controlled to the ON state and the fifth switch S5 located on the positive side is controlled to the OFF state. As a result, a closed loop is formed from the balancer inductor Lb through the second decoupling capacitor Cdc2 and the sixth switch S6 to the balancer inductor Lb, and energy is transferred from the balancer inductor Lb to the second decoupling capacitor Cdc2.

[0138] During the charging phase of the first decoupling capacitor Cdc1, the terminal voltage of the first decoupling capacitor Cdc1 tends to be higher than the terminal voltage of the second decoupling capacitor Cdc2, and the difference in terminal voltages between the two capacitors tends to become large. During the charging phase of the first decoupling capacitor Cdc1, energy is supplied to the second decoupling capacitor Cdc2 from the balancer inductor Lb, which can suppress the large difference in terminal voltages between the two capacitors. In other words, the voltage imbalance between the terminals of the two decoupling capacitors can be suppressed.

[0139] As described above with reference to Figure 30, when the first switch S1 and the fourth switch S4 are both controlled to the OFF state, and the third switch S3 and the second switch S2 are both controlled to the ON state, the input boost inductor Le is charged. As shown in Figure 34, at this time, the sixth switch S6 located on the negative side of the second leg is controlled to the ON state, and the fifth switch S5 located on the positive side is controlled to the OFF state. As a result, a closed loop is formed from the second decoupling capacitor Cdc2 through the balancer inductor Lb and the sixth switch S6 to the second decoupling capacitor Cdc2, and the balancer inductor Lb is charged.

[0140] As described above with reference to Figure 31, when the fourth switch S4 is controlled to the ON state and the third switch S3 and fourth switches S4 are controlled to the OFF state, the second decoupling capacitor Cdc2 is charged. As shown in Figure 35, at this time, the fifth switch S5 located on the positive side of the second leg is controlled to the ON state and the sixth switch S6 located on the negative side is controlled to the OFF state. As a result, a closed loop is formed from the balancer inductor Lb through the fifth switch S5 and the first decoupling capacitor Cdc1 to the balancer inductor Lb, and energy is transferred from the balancer inductor Lb to the first decoupling capacitor Cdc1.

[0141] During the charging phase of the second decoupling capacitor Cdc2, the terminal voltage of the second decoupling capacitor Cdc2 tends to be higher than the terminal voltage of the first decoupling capacitor Cdc1, and the difference in terminal voltages between the two capacitors tends to become large. During the charging phase of the second decoupling capacitor Cdc2, energy is supplied to the first decoupling capacitor Cdc1 from the balancer inductor Lb, which can suppress the large difference in terminal voltages between the two capacitors. In other words, the voltage imbalance between the terminals of the two decoupling capacitors can be suppressed.

[0142] The operation of the backend circuit 2 (active decoupling circuit) is the same as described above for the first embodiment, so the explanation will be omitted.

[0143] The following is a brief summary of the in-vehicle charging device (10) described above.

[0144] In one embodiment, the on-board charging device (10) is an on-board charging device (10) that charges the DC power supply (3) of a vehicle (9) equipped with a rotating electric machine (70) that serves as a driving force source for the wheels, an inverter (5) that converts power between DC and multiphase AC, and a DC power supply (3) connected to the inverter (5), with power supplied from an external AC power supply (4), wherein the on-board charging device (10) is an AC-DC converter (1) that converts power between AC power on the external AC power supply (4) side and first DC power, a storage capacitor (Cs), the inverter (5) that functions as a charge / discharge control circuit for the storage capacitor (Cs), and a multiphase coil (Ls) that functions as a storage inductor (Ls) connecting the inverter (5) and the storage capacitor (Cs). The AC / DC converter (1) is configured to include a coil (7) and an active decoupling circuit (2) that generates a second DC power to charge the DC power supply (3) by reducing the pulsation component from the first DC power, and the AC / DC converter (1) is connected to the external AC power supply (4) via a pair of input boost inductors (Le(Le1,Le2)), and the AC / DC converter (1) is equipped with a T-type power factor correction circuit, and the T-type power factor correction circuit is equipped with a split decoupling capacitor (Cdc) in which two capacitors (Cdc1,Cdc2) are connected in series between the positive and negative terminals of the DC, the positive terminal of the split decoupling capacitor (Cdc) is connected to the positive terminal of the inverter (5), and the negative terminal of the split decoupling capacitor (Cdc) is connected to the negative terminal of the inverter (5).

[0145] Relatively low-frequency leakage current to ground is caused by the second harmonic of the frequency of the external AC power supply (4). This leakage current is reduced by splitting the input boost inductor (Le) at the input to the AC / DC converter (1) into a pair of input boost inductors (Le(Le1,Le2)), causing the current phases flowing through them to cancel each other out. Relatively high-frequency leakage current to ground is caused by the switching of the switching elements in the onboard charging device (10) and is generated by voltage fluctuations at the midpoint of the AC / DC converter (1). As a method to suppress this leakage current, it is conceivable to place a floating filter (FF) between the node between the load connected to the DC power supply (3) and the DC power supply (3), and between the node between the AC / DC converter (1) and the external AC power supply (4). However, with this configuration, by providing a T-type power factor correction circuit and a split decoupling capacitor (Cdc), the voltage at the midpoint can be stabilized and the high-frequency leakage current to ground can be reduced. In other words, high-frequency leakage current to ground can be suppressed with a small-scale configuration without using a floating filter. Thus, with this configuration, leakage current and the like in an on-board charging device (10) that charges an on-board DC power supply (3) with power from an external AC power supply (4) can be suppressed more than in conventional systems.

[0146] Furthermore, the on-board charging device (10) preferably includes a T-type power factor correction circuit comprising a leg with two switch units connected in series, and a midpoint switch unit (MS) connected between the midpoint of the leg and the midpoint of the split decoupling capacitor (Cdc).

[0147] With this configuration, the midpoint switch (MS) makes it easier to properly stabilize the midpoint voltage of the split decoupling capacitor (Cdc), which is the midpoint voltage of the AC-DC converter (1).

[0148] Furthermore, the on-board charging device (10) includes a full-bridge circuit in which the T-type power factor correction circuit comprises a first leg having a bidirectional switch section (BS) in which two switch sections (S5, S6) are connected in series, and a second leg and a third leg, each having two switch sections (S1-S2, S3-S4) connected in series between the positive and negative DC electrodes, and the pair of input boost inductors (Le) are connected between the first end of the external AC power supply (4) and the first end of the first leg. The device comprises a first input boost inductor (Le1) connected in between, and a second input boost inductor (Le2) connected between the second end of the external AC power supply (4) and the second end of the first leg, wherein the midpoint switch section (MS,S7) is connected between the midpoint of the first leg and the midpoint of the divided decoupling capacitor, the first end of the first leg is connected to the midpoint of the second leg, and the second end of the first leg is connected to the midpoint of the third leg.

[0149] With this configuration, a T-type circuit (1T) can be constructed with a small configuration consisting of two switch sections: one bidirectional switch section (BS) and one midpoint switch section (MS, S7). Even including the four switch sections (S1, S2, S3, S4) in a full-bridge circuit, a T-type power factor correction circuit can be constructed with a small configuration of six switches (BS, S1, S2, S3, S4, S7).

[0150] Furthermore, the on-board charging device (10) includes a full-bridge circuit in which the T-type power factor correction circuit has a first leg and a second leg, each having two switch units (Su1-Su2, Sv3-Sv4) connected in series between the positive and negative DC electrodes, and the pair of input boost inductors (Le) includes a first input boost inductor (Le1) connected between the first end of the external AC power supply (4) and the midpoint of the first leg, and a second input boost inductor (Le2) connected between the second end of the external AC power supply (4) and the midpoint of the second leg, and the midpoint switch unit (MS) has two switch units (Su3, Su2) connected in series. The device comprises a first bidirectional switch section (BS1) that forms one bidirectional switch section (BS), and a second bidirectional switch section (BS2) in which two switch sections (Sv3, Sv2) are connected in series to form one bidirectional switch section (BS), wherein the first bidirectional switch section (BS1) is connected between the midpoint of the first leg and the midpoint of the split decoupling capacitor (Cdc), and the second bidirectional switch section (BS2) is connected between the midpoint of the second leg and the midpoint of the split decoupling capacitor (Cdc).

[0151] With this configuration, a T-type power factor correction circuit can be constructed with a small configuration consisting of six switches (Su1, BS1, Su4, Sv1, BS2, Sv4) in total, comprising two bidirectional switch sections (BS1, BS2) and four switch sections (Su1, Su4, Sv1, Sv4) in a full-bridge circuit.

[0152] Furthermore, the on-board charging device (10) further comprises the AC / DC converter (1) with a DC link voltage balancer circuit (VB), the T-type power factor correction circuit with a first leg in which a switch section (S1, S4) is connected in series between the positive and negative DC electrodes, and the pair of input boost inductors (Le) with a first input boost inductor (Le1) connected between the first end of the external AC power supply (4) and the midpoint of the first leg, and a second input boost inductor (Le2) connected between the second end of the external AC power supply (4) and the midpoint of the split decoupling capacitor (Cdc), and The midpoint switch section (MS) preferably comprises a bidirectional switch section (BS) in which two switch sections (S2, S3) are connected in series, and the bidirectional switch section (BS) is connected between the midpoint of the first leg and the midpoint of the split decoupling capacitor (Cdc). The DC link voltage balancer circuit (VB) preferably comprises a second leg in which switch sections (S5, S6) are connected in series between the positive and negative DC electrodes, and a balancer inductor (Lb) connected between the midpoint of the split decoupling capacitor (Cdc) and the midpoint of the second leg.

[0153] With this configuration, a DC link voltage balancer circuit (VB) equipped with two switch sections (S5, S6) suppresses fluctuations in the midpoint voltage of the divided DC link capacitor (Cdc), and a T-type power factor correction circuit can be constructed in a small configuration of five switches (S1, BS, S4, S5, S6) consisting of one bidirectional switch section (BS) and two switch sections (S1, S4).

[0154] Furthermore, the on-board charging device (10) is preferably configured such that the inverter (5) has multiple legs in parallel, corresponding to the number of phases of the multi-phase AC, with an upper switching element (5U) and a lower switching element (5L) connected in series between the DC positive electrode and the negative electrode, and when the inverter (5) functions as the active decoupling circuit (2), multiple upper switching elements (5U) and multiple lower switching elements (5L) are simultaneously switched and controlled, and the storage capacitor (Cs) is preferably connected between the neutral point (7N) of the coil (7) and the DC negative electrode.

[0155] With this configuration, the ripple remaining in the first DC power converted by the AC-DC converter (1) can be appropriately reduced by the active decoupling circuit (2). Furthermore, by using the coil (7) and inverter (5) in the active decoupling circuit (2), the increase in mounting space required to install the on-board charging device (10) on the vehicle (9) can be suppressed, and the increase in component costs can also be suppressed. [Explanation of Symbols]

[0156] 1: Front-end circuit (AC / DC converter), 1T: T-connection section, 2: Back-end circuit (active decoupling circuit), 3: DC power supply, 4: External AC power supply, 5: Inverter, 7: Coil, 7N: Neutral point, 9: Vehicle, 10: On-board charging device, 70: Rotating electric machine, BS: Bidirectional switch section, BS1: First bidirectional switch section, BS2: Second bidirectional switch section, Cdc: Split decoupling capacitor, Cs: Storage capacitor, Lb: Balancer inductor, Le: Input boost inductor, Le1: First input boost inductor, Le2: Second input boost inductor, Ls: Storage inductor, MS: Midpoint switch section, VB: DC link voltage balancer circuit, Vdc: DC link voltage

Claims

1. An on-board charging device for a vehicle comprising a rotating electric machine that serves as a driving force source for the wheels and has multiple phase coils connected to each other at a neutral point, an inverter that converts power between DC and multiple phase AC, and a DC power supply connected to the inverter, the DC power supply of the vehicle is charged by power supplied from an external AC power supply, An AC-DC converter that converts power between the AC power from the external AC power source and the first DC power, The system comprises a storage capacitor, an inverter that functions as a charge / discharge control circuit for the storage capacitor, and a multi-phase coil that functions as a storage inductor connecting the inverter and the storage capacitor, and an active decoupling circuit that generates a second DC power for charging the DC power supply by reducing the pulsation component from the first DC power, The AC-DC converter is connected to the external AC power supply via a pair of input boost inductors. The AC / DC converter is equipped with a T-type power factor correction circuit. The T-type power factor correction circuit includes a split decoupling capacitor in which two capacitors are connected in series between the positive and negative DC electrodes. An in-vehicle charging device in which the positive terminal of the divided decoupling capacitor is connected to the positive terminal of the inverter, and the negative terminal of the divided decoupling capacitor is connected to the negative terminal of the inverter.

2. The T-type power factor correction circuit has the following features: A leg with two switch sections connected in series, The in-vehicle charging device according to claim 1, further comprising a midpoint switch section connected between the midpoint of the leg and the midpoint of the split decoupling capacitor.

3. The T-type power factor correction circuit has the following features: A first leg has a bidirectional switch section in which two switch sections are connected in series, The full bridge circuit includes a second leg and a third leg, each having two switch sections connected in series between the positive and negative DC electrodes. The pair of input boost inductors A first input boost inductor is connected between the first end of the external AC power supply and the first end of the first leg, A second input boost inductor is connected between the second end of the external AC power supply and the second end of the first leg, The midpoint switch section is connected between the midpoint of the first leg and the midpoint of the split decoupling capacitor. The first end of the first leg and the midpoint of the second leg are connected. The in-vehicle charging device according to claim 2, wherein the second end of the first leg and the midpoint of the third leg are connected.

4. The T-type power factor correction circuit has the following features: It features a full-bridge circuit with a first leg and a second leg, each having two switch sections connected in series between the positive and negative DC electrodes. The pair of input boost inductors A first input boost inductor is connected between the first end of the external AC power supply and the midpoint of the first leg, A second input boost inductor is connected between the second end of the external AC power supply and the midpoint of the second leg, The aforementioned midpoint switch section A first bidirectional switch section in which two switch sections are connected in series to form one bidirectional switch section, It comprises a second bidirectional switch section in which two switch sections are connected in series to form one bidirectional switch section, The first bidirectional switch section is connected between the midpoint of the first leg and the midpoint of the split decoupling capacitor. The in-vehicle charging device according to claim 2, wherein the second bidirectional switch section is connected between the midpoint of the second leg and the midpoint of the split decoupling capacitor.

5. The AC-DC converter further includes a DC link voltage balancer circuit, The T-type power factor correction circuit includes a first leg in which a switch unit is connected in series between the positive and negative DC electrodes. The pair of input boost inductors A first input boost inductor is connected between the first end of the external AC power supply and the midpoint of the first leg, The system comprises a second input boost inductor connected between the second end of the external AC power supply and the midpoint of the split decoupling capacitor, The aforementioned midpoint switch section comprises a bidirectional switch section in which two switch sections are connected in series. The bidirectional switch section is connected between the midpoint of the first leg and the midpoint of the split decoupling capacitor. The DC link voltage balancer circuit is, A second leg has a switch connected in series between the positive and negative terminals of the DC current, The in-vehicle charging device according to claim 2, further comprising a balancer inductor connected between the midpoint of the split decoupling capacitor and the midpoint of the second leg.

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