Method for controlling a DC-DC power converter

Abstract

To detect a current and a voltage in a secondary side circuit of a DC-DC power conversion device with a smaller number of sensors.SOLUTION: A primary side circuit (1) includes a circuit containing switching elements R+, R-, S+, and S-. A secondary side circuit (2) includes: a diode rectification circuit containing four diodes U+, U-, V+, and V- to which resonance capacitors are connected respectively in parallel; and a smooth capacitor C2 connected to the rectification circuit in parallel. In the secondary side circuit (2), a leakage inductance L of a transformer Tr and a resonance circuit of a resonance capacitor are formed. A current i2 and a voltage v2 of the secondary side of the transformer Tr are detected from the leakage inductance L of the transformer Tr, a capacity Cr of the resonance capacitor, and a primary side voltage v1 of the transformer Tr.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present invention relates to a method for controlling a DC-DC power converter. [Background technology]

[0002] For example, a DC-DC power converter that transmits power from the primary side to the secondary side of a transformer is used as a non-contact charger that charges a secondary battery such as a lithium-ion battery in a non-contact manner. As such a DC-DC power converter, a dual active bridge (DAB) type DC-DC power converter is known that transmits power from the primary side to the secondary side of a transformer by using the phase difference between the current waveform on the primary side and the current waveform on the secondary side of the transformer.

[0003] In a DC-DC power converter of the DAB system disclosed in Patent Document 1 below, the switching phase difference between semiconductor switching elements in a primary side circuit and a secondary side circuit is controlled to enable soft switching. [Prior art documents] [Patent documents]

[0004] [Patent Document 1] JP 2016-152687 A Summary of the Invention [Problem to be solved by the invention]

[0005] When charging a secondary battery using the above-mentioned DC-DC power converter, CCCV charging (Constant Current Constant Voltage) is generally used to prevent overcharging. In this case, it is necessary to detect the real-time current and voltage in the secondary circuit. In the power converter of the above-mentioned Patent Document 1, the current and voltage of the secondary circuit can be detected by a current sensor and a voltage sensor provided in the secondary circuit.

[0006] On the other hand, with the recent trend towards electrifying vehicles, cheaper charging equipment is required, so when charging is performed using a DC-DC power converter, it is preferable to have as few sensors as possible for controlling the current in the secondary circuit.

[0007] Therefore, an object of the present invention is to detect the current and voltage in the secondary circuit of a DC-DC power converter using fewer sensors. [Means for solving the problem]

[0008] In Japanese Patent Application No. 2019-211006 filed by the present applicant, a DC-DC power converter is proposed in which a primary side circuit and a secondary side circuit are connected via a transformer. The primary side circuit has a circuit including a switching element. The secondary side circuit has a diode rectifier circuit including four diodes each having a resonant capacitor connected in parallel, and a smoothing capacitor connected in parallel to the rectifier circuit. In the secondary side circuit, a resonant circuit is formed between the leakage inductance of the transformer and the resonant capacitor.

[0009] In a DC-DC power converter having the above circuit configuration, the current and voltage on the secondary side can be controlled only by the inverter operation (operation of the switching element) of the primary side circuit, so the switching element of the secondary side circuit can be omitted. In this circuit configuration, the voltage on the secondary side of the transformer can be detected from the leakage inductance of the transformer and the capacitance of the resonant capacitor (capacitor) of the secondary side circuit. The current on the secondary side of the transformer can be detected from the voltage on the secondary side of the transformer thus obtained and the voltage on the primary side of the transformer detected by a sensor provided in the primary side circuit.

[0010] The present invention, which has been made based on the above findings, is characterized in that in the control method for a DC-DC power converter having the above circuit configuration, the current and voltage of the secondary circuit are detected from the leakage inductance of the transformer, the capacitance of the resonant capacitor of the secondary circuit, and the voltage on the primary side of the transformer. This makes it possible to detect the current and voltage of the secondary circuit using only the voltage sensor provided in the primary circuit, without providing a current sensor or voltage sensor in the secondary circuit.

[0011] The above DC-DC power converter may have a configuration including, for example, an H-bridge circuit having four of the switching elements, and a capacitor connected in parallel with the H-bridge circuit. Effect of the Invention

[0012] As described above, according to the present invention, the current and voltage on the secondary side of the transformer of a DC-DC power converter can be detected with fewer sensors. [Brief description of the drawings]

[0013] [Figure 1] 1 is a circuit diagram of a power converter according to a first embodiment of the present invention. [Diagram 2] 4A to 4C are diagrams showing the conductive state of each switch and the voltage and current waveforms of a high-frequency transformer. [Diagram 3] 4A to 4C are diagrams illustrating the commutation operations of the secondary-side diode rectifier circuit. [Figure 4] FIG. 11 is a diagram showing the characteristics of the output power Pout versus the ratio fs / fo of the frequency of the high-frequency transformer to the resonant frequency. [Diagram 5] 13 is a diagram showing the operation mode of the circuit when commutating from switches R-, S+ to switches R+, S- in the primary side circuit, and the primary voltage waveform v1 of the high-frequency transformer. [Figure 6] FIG. 4 is a circuit diagram of a power converter according to a second embodiment of the present invention. [Figure 7] FIG. 11 is an operational waveform diagram illustrating a method for controlling output power according to mode switching timing (mode 2-2). [Figure 8] FIG. 11 is an operational waveform diagram for explaining a method of controlling output power according to mode switching timing (mode 2-3). [Figure 9] 11 is a graph showing output power characteristics with respect to a period Td during which the primary voltage v1 is 0. FIG. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0014] Hereinafter, embodiments of the present invention will be described with reference to the drawings. However, none of the following embodiments are intended to provide a restrictive interpretation in determining the gist of the present invention. Also, the same or similar components are given the same reference symbols, and descriptions thereof may be omitted. The primary side soft switching circuit may be, for example, an H-bridge circuit or a half-bridge circuit, but is not limited thereto, and any circuit may be used.

[0015] The basic feature of the DC-DC power converter of this embodiment is that it employs a circuit configuration in which a primary circuit that generates a square wave or the like using a circuit having switching elements, and a secondary circuit that is made up of only passive elements and is made up of a combination of a rectifier circuit and an LC resonant circuit, are electromagnetically coupled by a transformer. This has the advantage that, despite the simple circuit configuration, the supplied power can be adjusted by the switching frequency of the primary circuit, and because the secondary circuit is made up of only passive elements, the primary circuit side and the secondary circuit side can be separated by the transformer core.

[0016] In the circuit configuration described above, the voltage on the secondary side of the transformer can be detected from the leakage inductance of the transformer and the capacitance of the resonant capacitor of the secondary side circuit. The current on the secondary side of the transformer can be detected from the voltage on the secondary side of the transformer thus detected and the voltage on the primary side of the transformer detected by the voltage sensor. This makes it possible to detect the voltage and current on the secondary side without providing a current sensor or a voltage sensor in the secondary side circuit, thereby reducing the number of sensors provided in the circuit compared to the conventional case. A specific circuit diagram will be described below with reference to the drawings.

[0017] (First embodiment) Fig. 1 shows a circuit diagram of a power converter (10) of the first embodiment. This circuit is a unidirectional isolated DC-DC power converter, in which a primary circuit (1) is provided with an H-bridge circuit, and a secondary circuit (2) is composed of a diode rectifier circuit, which are coupled by a high-frequency transformer Tr. Note that the primary circuit and the secondary circuit may be simply referred to as the "primary side" and the "secondary side", respectively. The high-frequency transformer Tr may be simply referred to as the "transformer".

[0018] The primary side H-bridge circuit consists of four switching elements R+, R-, S+, and S-. Diodes are connected in anti-parallel to the switching elements. The parasitic capacitance (stray capacitance) of the switching elements is C s The primary side H-bridge circuit is the input DC voltage V in A high frequency square wave AC voltage V 1 Convert to.

[0019] The leakage inductance of the high-frequency transformer Tr is represented as L, and the leakage inductance of the entire high-frequency transformer Tr is expressed as a converted value on the secondary side. When the leakage inductance is small, a reactor is connected in series to the transformer, and the leakage inductance of the high-frequency transformer itself and the inserted reactor are included in the leakage inductance L (inductance L). When the number of turns of the primary and secondary windings of the transformer are n1 and n2, respectively, the primary voltage v 1 The quadratic conversion value of is v 1 '(=v 1 / a).

[0020] The secondary diode rectifier circuit consists of four diodes U+, U-, V+, and V-, each with a resonant capacitor connected in parallel, and a smoothing capacitor C 2 Here, the capacitance of the resonant capacitor C r is a relatively large value (e.g., several tens of nF to 1 μF, specifically, 100 nF to 1 μF, typically 500 nF to 1 μF) compared to the parasitic capacitance of the diode (e.g., several nF to 10 nF). The secondary side diode rectifier circuit converts the high frequency square wave voltage into the output DC voltage Vout Convert to.

[0021] In addition, by using a diode (a diode designed to have a large capacitance) with a sufficiently large parasitic capacitance (for example, several tens of nF to 1 μF), the resonant capacitor (C r ) can be built into the diode. In this case, there is no need to provide a resonance capacitor outside the diode, and the diode can be made compact.

[0022] The secondary circuit shown in this embodiment is characterized by an inductance L and a capacitor (C r ) and other configurations can be changed as appropriate depending on the application. For example, in Fig. 1, a DC power supply is connected to the secondary output, but a DC load may also be connected. For example, if it is used as a charging circuit for a secondary battery, it is represented as a DC power supply as shown in Fig. 1, but if it is used as a DC-DC converter that converts power from railway overhead lines to the railway, it is represented as a DC load.

[0023] FIG. 2 shows the conduction state of each switch in the isolated DC-DC power conversion circuit of FIG. 1 and the voltage and current waveforms of the high-frequency transformer. The horizontal axis in FIG. 2 is time. In FIG. 2, for the sake of simplicity, the turn ratio of the high-frequency transformer is set to a=1, and further, when the input and output DC voltages are equal (V in =V out In the primary side H-bridge circuit, the switching phase of the R and S phases is shifted by 180 degrees, and the duty of each switch is 50%, so that the input DC voltage V in is the amplitude, and the frequency f s (=1 / 2T s ;Half-period of high-frequency waveform T s ) square wave AC voltage to the primary voltage v 1 The primary voltage v in Figure 2 1 The waveform of the primary voltage v 1 The operation modes according to the change of the secondary voltage v are shown as <Mode 1-1> to <Mode 1-4>. 2 is the primary voltage v 1If the excitation current of the high frequency transformer is assumed to be sufficiently small compared to the primary and secondary currents and is ignored, then the primary current i of the high frequency transformer 1 and the secondary current i 2 The primary current i in Figure 2 is equal to 1 , secondary current i 2 The details of the waveform will be explained later. The primary current i 1 , secondary current i 2 Along with the waveforms, each section from <Mode 2-1> to <Mode 2-4> is shown as the operating mode associated with the commutation of the secondary side diode rectifier circuit.

[0024] Figure 3 shows the commutation operation of the secondary diode rectifier circuit. It shows the circuit operation in each mode when the diodes U- and V+ are commutated to the diodes U+ and V-, respectively, in the secondary diode rectifier circuit when the primary voltage is switched from negative to positive by the switching of the H-bridge circuit. In <Mode 2-1> before the commutation in Figure 3, the primary switches R- and S+ are conductive, and the primary voltage v 1 ' is the input DC voltage -V in (=-V out ) and the secondary current i 2 Diodes U- and V+ conduct and a negative current -I n is flowing. Secondary voltage v 2 is the output DC voltage -V out Since the primary and secondary voltages are equal, the constant value -I n Secondary current i 2 The voltage across the parallel capacitor of diodes U- and V+ is zero, and the voltage across the parallel capacitor of diodes U+ and V- is the output DC voltage V out The battery is charged at time t=t in Figure 2. 2 At this point, the switch in the H-bridge circuit is switched from R-, S+ to R+, S-, and the primary voltage v 1 '-V in From V in When the current flow changes to <Mode 2-2>, the circuit transitions to <Mode 2-2>. Even when the circuit transitions to <Mode 2-2> in Fig. 3, diodes U- and V+ continue to conduct due to the continuity of the secondary current caused by inductance L. The voltage equation for the secondary circuit in <Mode 2-2> is given by the following equation.

[0025]

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[0026] Here, the primary voltage v in <Mode 2-2> 1 '=V in =V out , secondary current initial value i 2 (t 2 )=-I n Substituting this into equation (1), the secondary current i 2 (t) is obtained by the following equation.

[0027]

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[0028] As shown in Figure 2, in <Mode 2-2>, the secondary current i 2 (t) increases toward zero at a constant rate. 2 (t) is time t=t 3 When it becomes zero, <Mode 2-2> ends. 2 =t 3 -t 2 is given by (4).

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[0029] time t=t 3 and the secondary current i 2 (t 3 When the time t=t, all the diodes in the secondary circuit become non-conductive, and the system moves to <Mode 2-3> in Fig. 3. <Mode 2-3> starts at time t=t 3 The initial voltages of the parallel capacitors of the diodes U- and V+ are both zero, and the initial voltages of the parallel capacitors of the diodes U+ and V- are both V outIn mode 2-3, the inductor L and four capacitors (C r ) resonant circuit is formed. Due to the symmetry of the circuit, the current i 2 / 2 flows through each capacitor. The voltage equation in <Mode 2-3> is given by the following equation.

[0030]

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[0031] Primary voltage v 1 '=V in =V out , secondary current initial value i 2 (t 3 ) = 0 into equation (5) and solve it, we get the secondary current i 2 (t) is obtained by the following equation.

[0032]

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[0033] Secondary current i 2 (t) is the resonant angular frequency ωo(=2πf o =1 / √(LC r ) flows the secondary voltage v 2 is the secondary current i in Eq. (6). 2 Using (t), it is obtained by the following formula:

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[0034] The secondary current i in Eqs. (6) and (7) 2 , secondary voltage v 2 The secondary current i has a sinusoidal waveform as shown in Figure 2. 2 , secondary voltage v 2 are I n , V out At time t=t, the voltages of the parallel capacitors of the diodes U+ and V- also become zero. 4In Fig. 2, at time t=t 4 The phase at is π / 2, and the secondary current i 2 The amplitude of I n and the period T of <Mode 2-3> 3 =t 4 -t 3 is obtained from equations (6) and (7) as follows:

[0035]

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[0036] (3) Secondary current i of <Mode 2-2> 2 , the period T of <Mode 2-2> in equation (4) 2 can be rewritten as follows by substituting equation (8).

[0037]

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[0038] time t=t 4 When the voltages of the parallel capacitors of diodes U+ and V- become zero, diodes U+ and V- become conductive, and the mode shifts to <Mode 2-4> in Figure 3. The voltage equation of the secondary circuit in <Mode 2-4> is given by the following equation.

[0039]

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[0040] Here, the primary voltage v in <Mode 2-4> 1 '=V in =V out , secondary current initial value i 2 (t 4 )=I nSubstituting this into equation (12), the secondary current i 2 (t) is obtained by the following equation.

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[0041] As shown in Fig. 2, in <Mode 2-4>, the secondary current i 2 (t) becomes a constant value. When the H-bridge circuit switches and the primary voltage changes from positive to negative, <Mode 2-4> ends.

[0042] Due to the operation of the secondary side circuit, each diode switches when the parallel capacitor voltage is zero. In other words, there is no recovery loss in the diode, so there is no power loss and the efficiency is extremely high. Output power P out is the output current i out Using this, the half-period T of the high-frequency transformer s The average power of is given by the following equation:

[0043]

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[0044] Output power P out The control of the frequency f of the high frequency transformer s The output power P in Eq. (14) can be adjusted by out The frequency of the high-frequency transformer f s (=1 / 2T s ) and resonant frequency f o (=1 / (2π√(LC r Rewriting using ), we get:

[0045]

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[0046] Figure 4 shows the ratio of the frequency of the high-frequency transformer to the resonant frequency, f s / f oOutput power P out The resonance frequency f o is determined by the circuit parameters and is a constant value, and the transformer frequency f s By increasing the output power P out In Fig. 4, the ratio of the frequency of the high-frequency transformer to the resonant frequency (f s / f o ) = 1 / 4 of the rated output P out The output power P in Eq. (15) is normalized to 1. out The maximum frequency (f s / f o ) max is the secondary current i in Fig. 2 2 In the waveform (t), the current value I n When the period of is zero, it is obtained by the following equation.

[0047]

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[0048] The maximum frequency of the high-frequency transformer is (f s / f o ) max = 1.22, and in this case, the output power can be reduced to 0.23 of the rated output in Figure 4. s , (f s / f o ) max If the frequency is set higher than the value determined by, then equation (15) does not hold, but the output power P out It is possible to reduce the

[0049] Next, the soft switching commutation of the H-bridge circuit will be explained.

[0050] In Figure 1, the parallel capacitor C of the switching element of the primary H-bridge circuit s However, in order to perform soft switching of the switch, a separate capacitor may be connected externally in parallel. In the following, the parallel capacitance of the switching element is defined as Cs This is explained as follows.

[0051] Figure 5 shows the circuit operation mode when commutating from switches R-, S+ to switches R+, S- in the primary side H-bridge circuit and the primary voltage waveform v of the high frequency transformer. 1 Shows.

[0052] In <Mode 1-1> before commutation, both switches R- and S+ are conductive, and the primary voltage v 1 is the negative input DC voltage -V in The primary current i 1 As a negative constant current -I n flows through the switches R- and S+. The voltages of the parallel capacitors of the switches R- and S+ are both zero, and the voltages of the parallel capacitors of the switches R+ and S- are both equal to the input DC voltage V in When the switches R- and S+ are turned off, the parallel capacitor voltage is zero, so the switches R- and S+ are zero voltage switched (ZVS).

[0053] By turning off the switches R- and S+, the circuit transitions to <Mode 1-2> in Fig. 5. In <Mode 1-2>, the primary current i 1 is a negative current -I n and four parallel capacitors C s The voltage of the parallel capacitor of the switches R- and S+ increases from zero, and the voltage of the parallel capacitor of the switches R+ and S- decreases. Due to the change in these capacitor voltages, the primary voltage v 1 is the negative input DC voltage -V in to positive input DC voltage V in The primary voltage v 1 is the positive input DC voltage V inWhen this occurs, the parallel capacitor voltages of the switches R+ and S- both become zero, and the parallel diodes of the switches R+ and S- become conductive. The parallel diodes of the switches R+ and S- become conductive, causing a transition to <Mode 1-3>. During the period of <Mode 1-3>, a gate signal is applied to make the switches R+ and S- conductive. Since the parallel diodes are in a conductive state, even if a gate signal is applied to the switches, they are reverse-biased and do not become conductive. As already mentioned, the primary current i 1 increases at a constant rate toward zero. Then, the primary current i 1 When the primary current i goes from negative to positive, it transitions to <Mode 1-4>. 1 When the sign of changes from negative to positive and the switches R+ and S- are conductive, the parallel capacitor voltage remains zero, and ZVS can be achieved. Soft switching can be performed similarly for other switching, reducing switching losses.

[0054] As described above, according to the first embodiment, it is possible to obtain the effect of efficiently reducing switching loss with a simple circuit configuration.

[0055] As a further effect, the control of the output power can be easily performed, and the controllability of the low output side is particularly improved. As described above, the output power can be adjusted by the switching frequency according to equation (15), but P out When the value of is 0.23 or less, the controllability becomes poor and the output fluctuates greatly with a slight frequency fluctuation, but according to the first embodiment, it becomes possible to control the power on the output side without relying on the formula (15). As a result, it becomes possible to control the supply power to be extremely small, for example, after the charge level of the output power supply target, such as a storage battery, exceeds a certain value. A specific method for this will be described in detail in the third embodiment.

[0056] Second embodiment: Circuit configuration using a primary side half-bridge circuit FIG. 6 shows a circuit in which the primary side H-bridge circuit in FIG. 1 in the first embodiment is replaced with a half-bridge circuit (1'). The input DC voltage is 2V inThen, two capacitors C1 are connected in series to provide an input DC voltage of 2V. in The DC neutral point is set at V. in One of the two input terminals of the high frequency transformer is connected to the R phase output terminal, and the other is connected to the DC neutral point. The half bridge consists of two switching elements R+ and R-. The switching elements have built-in anti-parallel diodes, and the parasitic capacitance of the switching elements is s It states that:

[0057] If we ignore the S-phase switching signal, the operating waveforms of the isolated DC-DC power conversion circuit using the H-bridge circuit shown in Figure 2 can be directly applied to the operating waveforms of the circuit using the half-bridge circuit in Figure 6. In other words, by turning on the switching element R+, the primary voltage v of the high-frequency transformer 1 is connected to the upper capacitor C1 of the input DC voltage, and the capacitor voltage V in By turning on the switching element R-, the primary voltage v of the high frequency transformer 1 is connected to the lower capacitor C1 of the input DC voltage and the negative capacitor voltage -V in become.

[0058] Therefore, just like when using an H-bridge circuit, the primary voltage v 1 As the amplitude V in In the circuit using the half-bridge circuit, the high-frequency transformer and secondary circuit are the same as the circuit using the H-bridge circuit in Figure 1, and the secondary voltage v in Figure 2 is obtained. 2 , primary current i 1 , secondary current i 2 The following waveforms are obtained. In the secondary circuit, highly efficient operation is possible without generating recovery loss in the diode.

[0059] As described in the first embodiment, the output power P out The control of the frequency of the high-frequency transformer f s (=1 / 2T s) can be adjusted. In addition, soft switching of the primary side half bridge circuit can be achieved.

[0060] Let us explain the commutation from switch R- to switch R+ in the half-bridge circuit of Figure 6. When switch R- is in a conducting state, the primary voltage v 1 is the negative input DC voltage -V in And the primary current i 1 is a negative current -I n flows through switch R-. Also, the voltage of the parallel capacitor of switch R- is zero. When switch R- is turned off, the voltage of the parallel capacitor of switch R- is zero, and ZVS is realized. The primary current i 1 Then, a negative current -I n When this current flows, the voltage across the parallel capacitor of switch R+ becomes V in When the primary voltage v decreases to zero, the parallel diode of switch R+ becomes conductive. 1 is the positive input DC voltage V in and the primary current i 1 Also negative current -I n The primary current i 1 If a conductive signal is given to switch R+ while is negative, the primary current i 1 Even when changes from negative to positive and the current flows from the diode to the switch R+, the voltage of the parallel capacitor is zero and ZVS is realized. Therefore, soft switching can be realized in all commutations, and switching losses can be reduced.

[0061] The secondary circuit shown in this embodiment is the same as that in the first embodiment, and has an inductance L and a capacitance C r It is characterized by using resonance with the secondary side. Therefore, other configurations can be changed appropriately depending on the application. For example, in FIG. 6, a DC power supply is connected to the secondary side output, but a DC load may also be connected. For example, if it is used as a charging circuit for a secondary battery, it is represented as a DC power supply as shown in FIG. 6, but if it is used as a DC-DC converter that converts power from railway overhead lines to the railway, it is represented as a DC load.

[0062] As described above, a DC-DC power converter can be configured using a half-bridge circuit as the primary side circuit. According to the second embodiment, the configuration of the primary side circuit is simpler than that of the first embodiment, and a smaller or less expensive DC-DC power converter can be obtained. Note that the control of the output power can be adjusted by the switching frequency according to equation (15).

[0063] (Third embodiment) d A method for controlling power transmission by controlling As described in the first and second embodiments, in either circuit configuration, the power of the secondary circuit can be controlled by changing the switching frequency of the primary circuit according to equation (15). However, according to the circuit configuration described in the first embodiment, the primary voltage v 1 The period T when d Since it is possible to control the secondary power without using frequency control, it is possible to control the secondary power.

[0064] In this embodiment, in the circuit described in the first embodiment, the primary voltage v 1 The period T when d A power control method implemented by controlling the above will be described.

[0065] (1) Power reduction control method in <Mode 2-2> The frequency f of the high-frequency transformer in the unidirectional isolated DC-DC power conversion circuit in Figure 1 s When the output power P out This section explains how to control the primary voltage v by the switching pattern of the primary H-bridge circuit. The operating waveform in Figure 2 is at maximum output power, and the primary voltage v 1 As the amplitude V in The output power P out The basic idea of ​​power reduction is to adjust the switching timing of the S phase switch for the period T d delays the primary voltage v 1 In the square wave waveform, the voltage is zero during the period Td Set the primary voltage v 1 In the operation waveform of Figure 7, the primary voltage v 1 The period T during which d A new mode has been added to the period T d The waveforms other than the above are the same as those at maximum output power in Figure 2. In the operating waveforms in Figure 7, the primary voltage v 1 The period T during which d <Mode 2-21>, Primary voltage V 1 V in The period when the primary voltage v 1 '=0, secondary current initial value i 2 (t 2 )=-I n , (8) respectively, and the secondary current i 2 (t) is obtained by the following equation.

[0066]

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[0067] That is, the gradient of the secondary current in <Mode 2-21> of Eq. (17) is di 2 Since / dt is half the slope of <mode 2-2> in equation (2), the maximum period T d is the period T of <Mode 2-2> 2 =√(LC r ) is twice as long as the period T 21 =T d The range of is given by the following formula:

[0068]

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[0069] Here, time t=t in equation (18)2 +T d Substituting the above, the secondary current initial value i 2 (t 2 +T d ) is obtained by the following equation:

[0070]

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[0071] In addition, the primary voltage v 1 '=V in =V out , the secondary current initial value i 2 (t 2 +T d ), (8), and the secondary current i 2 (t) is obtained by the following equation.

[0072]

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[0073] End time t of <Mode 2-22> 3 Then, the secondary current i in Eq. (21) 2 (t 3 )=0, so the end time t 3 and period T of <Mode 2-22> 22 is given by the following equation:

[0074]

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[0075] Based on the derived secondary current waveforms for all modes, the output power P out Calculating the primary voltage v 1 The period T when d The output power P outcan be controlled.

[0076]

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[0077] (2) Power reduction control method in <Mode 2-3> Primary voltage v 1 The period T when d is 2π√(LC r ) or more, the primary voltage v 1 becomes zero, so the secondary current i 2 The resonant waveform of the primary voltage v 1 The period T when d is 2π√(LC r ) or more, the secondary voltage v 2 , primary current i 1 , secondary current i 2 The waveforms of each of the above are shown in Fig. 8. 3 and 4 Between the primary voltage v 1 The period T during which 31 <Mode 2-31>, Primary voltage V 1 V in Period T 32 In addition, the secondary current i 2 Current value -I m The absolute value of I m is I in Eq. (8). n (>I m ) is smaller than

[0078] <Mode 2-1> in Figure 8 becomes the circuit connection of <Mode 2-1> before commutation in Figure 3, and the primary side switches R- and S+ are conductive, and the primary voltage v 1 ' is the input DC voltage -V in is applied, and the secondary current i 2 Diodes U- and V+ conduct and a negative current -I m At time t=t in Figure 8, 2At this point, the switch in the H-bridge circuit is switched from R- to R+, causing the primary voltage v 1 '-V in When the voltage changes from 0 to 0, the mode changes to <Mode 2-21>. The primary voltage v 1 '=V in =V out , secondary current initial value i 2 (t 2 )=-I m Substituting this into equation (1), the secondary current i 2 (t) is obtained by the following equation.

[0079]

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[0080] As shown in Fig. 8, in <Mode 2-21>, the secondary current i 2 (t) increases toward zero at a constant rate. 2 (t) is time t=t 3 When it reaches zero, <Mode 2-21> ends. 21 =t 3 -t 2 is given by equation (27).

[0081]

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[0082] time t=t 3 and the secondary current i 2 (t 3 When the time t becomes 0, all the diodes in the secondary circuit become non-conductive, and the system moves to <Mode 2-31> in Fig. 3. <Mode 2-31> starts at time t=t 3 The initial voltages of the parallel capacitors of the diodes U- and V+ are both zero, and the initial voltages of the parallel capacitors of the diodes U+ and V- are both V outIn mode 2-3, the inductor L and four capacitors (C r ) resonant circuit is formed. Due to the symmetry of the circuit, the current i 2 / 2 flows through each capacitor, and the voltage equation (5) holds. 1 '=0, secondary current initial value i 2 (t 3 ) = 0 into equation (5) and solve it, we get the secondary current i 2 (t) is obtained by the following equation.

[0083]

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[0084] Secondary voltage v 2 is the secondary current i 2 Using (t), it is obtained by the following formula:

[0085]

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[0086] The secondary current i in Eqs. (28) and (29) 2 , secondary voltage v 2 As shown in Figure 8, the waveform of the sinusoidal wave is 3 +T 31 The secondary voltage v 2 , secondary voltage v 2 are obtained by the following equations, respectively:

[0087]

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[0088] time t=t 3 +T 31 When the switch of the H-bridge circuit is switched from S+ to S-, the primary voltage v 1 =V inincreases stepwise, resulting in a new resonant operation. The voltage equation for <Mode 2-32> is given by the following equation.

[0089]

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[0090] Primary voltage v 1 '=V in =V out , secondary current initial value i 2 (t 3 +T 31 ) into equation (32) and solve it, we get the secondary current i 2 (t) is obtained by the following equation.

[0091]

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[0092] The term in the first line of equation (33) is time t=t 3 +T 31 With the primary voltage v 1 '=V out The term in the second line is the resonant current due to the change to 3 +T 31 The third line is an equation expressing these two terms as one resonant current. The secondary voltage v of <Mode 2-32> 2 is the secondary current i in Eq. (33). 2 Using (t), it is obtained by the following formula:

[0093]

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[0094] time t=t 4 and the secondary voltage v 2 (t 4 )=V out At time t=t, the voltage across the parallel capacitors of diodes U+ and V- becomes zero, diodes U+ and V- become conductive, and <Mode 2-32> ends. 4Then, the second term of equation (34) becomes zero, and the period T 32 Using the period t 4 -t 3 =T 31 +T 32 Therefore, the period T can be expressed as follows: 32 is required.

[0095]

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[0096] time t=t 4 Using equation (35), for the period T 31 is given as a function of

[0097]

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[0098] (36) 4 Substituting this into equation (33), the secondary current i 2 (t 4 )=I m is obtained by the following equation.

[0099]

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[0100] Period T 31 If the length increases, the secondary current value I m becomes smaller, and the transmission power can be reduced. 31 =π√(LC r ) when the secondary current value I m = 0, so in the period T 31 is zero to π√(LC r ) range. <Mode 2-4> period t>t 4 Then, the voltage equation (12) is established, and the primary voltage v 1 '=V in =V out From di 2 / dt=0, and the secondary current i 2 (t 4 )=I m In equation (27), I in equation (37) flows. m Substituting, period T 21 is obtained by the following equation.

[0101]

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[0102] Period T during which the primary voltage is zero d is the period T 31 Using the above, it is obtained as follows:

[0103]

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[0104] Period T 31 The maximum value of π√(LC r ) The maximum period during which the primary voltage is zero when d max is obtained by the following formula:

[0105]

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[0106] The secondary voltage v in Figure 8 2 and the secondary current i 2 From period T s Output power P out is expressed by the following equation.

[0107]

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[0108] In equation (41), I in equation (37) m , T in Eq. (35) 32 , T in Eq. (38) 21 Substituting the above, the output power P out is expressed by the following formula: is expressed by the following formula:

[0109]

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[0110] FIG. 9 shows the period T d The output power P of Eq. (42) out Figure 9 shows the ratio of the frequency of the high-frequency transformer to the resonant frequency, f s / f o =π√(LC r ) / T s = 1 / 4, T s =4π√(LC r ) The zero voltage period of the primary voltage T d The longer the secondary current I m decreases, and the output power P out Therefore, the zero voltage period T of the primary voltage is also reduced. d Therefore, the output power P out can be controlled.

[0111] In the above explanation, the turn ratio of the high frequency transformer is assumed to be a = 1, and the input and output DC voltages are assumed to be equal. in =V out The case where the input / output DC voltage is V in / a=V out Similar waveforms are obtained when the input / output DC voltage is V in / a≠V out In the case of (a), when the secondary current waveform is a constant value in the above description, an error may occur such as an inclination of the secondary current waveform due to the difference between the primary and secondary voltages, but the basic function described can be obtained in the same way.

[0112] (Fourth embodiment) - Output power control and separation of primary and secondary circuits - In the circuit configurations of the present invention shown in Figures 1 and 6, the secondary side circuits are both composed of passive elements and do not need to be controlled. Therefore, the DC voltage V in and the primary current i 1 Detects the frequency f of the primary circuit. sor the period of zero voltage T d The output power P out In addition, if the primary winding iron core and secondary winding iron core of the high frequency transformer can be separated, it becomes possible to physically separate the primary circuit and the secondary circuit. The primary circuit and the secondary circuit can be combined and used only when transmitting power.

[0113] For example, by placing the primary circuit, which supplies power, on the ground side and the secondary circuit on the vehicle side, and bringing the primary and secondary cores of a high-frequency transformer close together only during power transmission (charging the vehicle), power transmission (charging) is possible. The transformer's cores are physically separated (independent) except when power is being transmitted, but during power transmission (charging), the primary and secondary cores are coupled by the electromagnetic force acting between the cores, making power transmission possible. In this way, it can also be used for non-radiative magnetic field coupling type contactless power transmission.

[0114] (Secondary current and voltage detection) In conventional DC-DC power converters, a current sensor and a voltage sensor are provided in the secondary circuit to detect the current and voltage on the secondary side. In contrast, in the above power converter (10), the secondary circuit (2) is a passive circuit that can be adjusted by the switching frequency of the primary circuit (1), so the leakage inductance L of the transformer Tr and the capacitance C of the resonant capacitor of the secondary circuit (2) are r and the voltage V of the primary circuit (1) 1 From this, the current i in the secondary circuit (2) 2 and voltage v 2 can be detected.

[0115] Specifically, in the power converter (10), the secondary current value i of the transformer Tr is 2 is expressed by the following formula. L is the voltage difference between both ends of the transformer Tr (the primary voltage v 1 and the secondary voltage v 2 ) and C is a constant.

[0116]

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[0117] The primary circuit (1) of the power converter (10) has a primary voltage v 1 A voltage sensor (3) is provided to detect the leakage inductance L of the transformer Tr and the capacitance C of the resonant capacitor of the secondary circuit (2) (see FIG. 1 and FIG. 6). r From this, the voltage v on the secondary side of the transformer Tr 2 The voltage v on the secondary side of the transformer Tr calculated in this way can be calculated. 2 and the voltage v on the primary side of the transformer Tr measured by the voltage sensor (3). 1 Therefore, the voltage difference between the primary and secondary sides of the transformer Tr is V L Then, in the above formula, the leakage inductance L of the transformer Tr and the voltage difference V L By substituting, the current i on the secondary side of the transformer Tr 2 can be calculated.

[0118] For example, in the third embodiment described above, the output current value I m {=i 2 (t 4 )} is expressed by the above formula (37). In the formula, the leakage inductance L and the capacitance of the resonance capacitor Cr are known, and the output DC voltage V out (=V in / a) is the primary voltage V detected by the voltage sensor (3). in and the turns ratio a. Therefore, T in the formula 31 If is known, the output current value I m From FIG. 8, T 31 =T d -T 21 It is. 21 is the output DC voltage V out (=V in / a), leakage inductance L, and previous output current value I m (In this embodiment, the output current value I mis calculated intermittently at short intervals, and the previous output current value I m " is the previously calculated output current value I m This can be calculated by substituting .

[0119] In this case, the output current value I m The procedure for calculating is as follows. (1) Start from time 0. (2) Current from the previous time 21 Request. (3) From the voltage waveform, T d Measure (calculate the time it takes for the voltage to rise) and calculate T 31 Request. (4)(37) T 31 , V out , L, C r Substituting the above, the output current value I m Request. By repeating the above calculations during power conversion (charging), the output current value can be calculated from the voltage value detected by the voltage sensor (3). [Industrial Applicability]

[0120] The power converter (10) according to the present invention is used, for example, in a charger for secondary batteries used in various applications. For example, the power converter (10) according to the present invention is used when charging a secondary battery mounted on an automated guided vehicle in a non-contact manner. In this case, a primary side circuit (1) is provided in a charging station, and a secondary side circuit (2) is provided in the automated guided vehicle. A voltage sensor (3) provided in the primary side circuit (1) of the charging station detects a primary voltage v 1 By simply measuring the voltage v of the secondary circuit (2) of the automatic guided vehicle, 2 and current i 2 Since the secondary side voltage v detected in this way can be used, there is no need to provide a current sensor and a voltage sensor in the secondary side circuit (2). 2 and current i 2 Based on this, the input voltage V in By controlling this, it is possible to charge the battery just enough.

[0121] In addition to the above, the power converter according to the present invention can be widely used in all product fields, such as railways and other industrial equipment, and has a wide range of applicable applications and extremely high industrial applicability. [Explanation of symbols]

[0122] 10 Power Converter 1 Primary side circuit (H-bridge circuit) 1' Primary circuit (half-bridge circuit) 2 Secondary circuit 3 Voltage Sensor C1 Capacitor C2 (smoothing) capacitor Cr Resonant capacitor capacity Cs parasitic capacitance U+, U-, V+, V- Diodes R+, R-, S+, S- Switching elements Tr Transformer L Inductance

Claims

1. A control method for a DC-DC power converter in which a primary side circuit and a secondary side circuit are connected via a transformer, comprising the steps of: The primary side circuit has a circuit including a switching element and a sensor that measures a voltage on the primary side of the transformer, the secondary side circuit has a diode rectifier circuit including four diodes each having a resonant capacitor connected in parallel, and a smoothing capacitor connected in parallel to the rectifier circuit; In the secondary side circuit, a resonant circuit is formed between the leakage inductance of the transformer and the resonant capacitor, Calculating a voltage on a secondary side of the transformer from a leakage inductance of the transformer and a capacitance of the resonant capacitor; obtaining a voltage difference between the primary side voltage and the secondary side voltage of the transformer from the calculated secondary side voltage of the transformer and the primary side voltage of the transformer detected by the sensor; A control method for a DC-DC power converter, comprising the steps of: calculating a current on a secondary side of the transformer based on a leakage inductance of the transformer and the voltage difference.

2. 2. The method for controlling a DC-DC power converter according to claim 1, wherein the primary side circuit comprises an H-bridge circuit having four of the switching elements, and a capacitor connected in parallel with the H-bridge circuit.

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