AC motor control device, AC motor drive device, and electric equipment using the same

Abstract

To provide an AC motor control device and an AC motor driving device in which a motor current fundamental wave component can be reproduced with high precision, and a motor machine using the AC motor control device and the AC motor driving device.SOLUTION: An AC motor control device (3) controls a power conversion device (2) outputting AC power to an AC motor (1). The AC motor control device includes: a PWM signal generation unit (8) that generates a control signal which is formed of a PWM pulse signal and is for controlling the power conversion device, in accordance with an AC voltage command generated as a result of vector control; and a current correction unit (17) that corrects a current value of a motor current reproduced on the basis of a DC busbar current flowing through a DC side of the power conversion device so that the current value is brought close to a fundamental wave of the motor current by multiplying the current value by a correction coefficient. The vector control is performed on the basis of the current value corrected by the current correction unit.SELECTED DRAWING: Figure 1

Description

Technical Field

[0001] The present invention relates to an AC motor control device for controlling an AC motor, an AC motor drive device including this AC motor control device for driving an AC motor, and an electric appliance using this AC motor drive device.

Background Art

[0002] In various fields such as general industry, home appliances, automobiles, and railways, the high-speed rotation of AC motors is progressing for the purpose of miniaturization and high output.

[0003] For the high-speed rotation control of an AC motor, synchronous PWM control that synchronizes the phase relationship between a carrier wave and a voltage command and changes the PWM carrier frequency and the number of pulses for each electrical angular frequency (see, for example, Patent Document 1) is used. According to synchronous PWM control, the generation of harmonics is suppressed.

[0004] With the miniaturization of AC motors, for further miniaturization of devices to which AC motors are applied, a sensorless technology for estimating the magnetic pole position of an AC motor without using a sensor is applied. As the sensorless technology, a conventional technique for estimating the magnetic pole position based on the calculated value of the axis error (Δθc) between the real axis (d-q axis) and the control axis (dc-qc axis) of an AC motor is known (see, for example, Patent Document 2).

[0005] Also, for the detection of motor current, a technique for reproducing three-phase motor current based on the shunt current flowing through the DC bus of an inverter without using a phase current sensor (see, for example, Patent Document 3 (FIGS. 3 and 4)) is used.

Prior Art Documents

Patent Documents

[0006]

Patent Document 1

Patent Document 2

Patent Document 3

[0007] In synchronous PWM control, current ripple increases in the motor current at high speeds of AC motors. In particular, as AC motors become smaller and their inductance decreases, current ripple becomes more pronounced. As a result, the error between the motor current reproduced based on the shunt current and the fundamental wave component increases.

[0008] Such current detection errors reduce the accuracy of current control in AC motors. Furthermore, in the position sensorless control based on the aforementioned axis error, the axis error is calculated using the dq axis current value (see "(3)" in Patent Document 2), which reduces the accuracy of magnetic pole position estimation.

[0009] Therefore, the present invention provides an AC motor control device, an AC motor drive device, and an electric motor device using the same, which can reproduce the fundamental wave component of the motor current with high accuracy. [Means for solving the problem]

[0010] To solve the above problems, the AC motor control device according to the present invention controls a power converter that outputs AC power to an AC motor, and comprises a PWM signal generation unit that generates a control signal for controlling the power converter, consisting of a PWM pulse signal, in response to an AC voltage command generated by vector control, and a current correction unit that corrects the current value to approximate the fundamental wave of the motor current by multiplying the current value of the motor current reproduced based on the DC bus current flowing on the DC side of the power converter by a correction coefficient, and the vector control is performed based on the current value corrected by the current correction unit. Furthermore, the system has one of the following means: The first means is that the current correction unit sets a correction coefficient based on the current value and the average value of the current values. The second means is that the PWM signal generation unit generates a control signal by synchronous PWM control, the current correction unit sets a correction coefficient for a predetermined time, and after the predetermined time has elapsed, the current correction unit corrects the current value by multiplying the current value by the set correction coefficient.

[0011] To solve the above problems, the AC motor drive device according to the present invention comprises a power converter that outputs AC power to an AC motor, and a control device that controls the power converter, wherein the control device is the AC motor control device according to the present invention.

[0012] To solve the above problems, the electric device according to the present invention comprises a mechanical load driven by an AC motor, a power converter that outputs AC power to the AC motor, and a control device that controls the power converter, wherein the control device is the AC motor control device according to the present invention. [Effects of the Invention]

[0013] According to the present invention, the fundamental wave component of the motor current can be reproduced with high precision.

[0014] Other issues, configurations, and effects not mentioned above will be clarified by the following description of the embodiments. [Brief explanation of the drawing]

[0015] [Figure 1] This is a functional block diagram showing the configuration of the AC motor drive device, which is Example 1. [Figure 2] This is a coordinate plane diagram showing the relationship between coordinate axes and axis errors in a rotating coordinate system. [Figure 3] This graph shows an example of the relationship between the rotation frequency command value fr* and the carrier frequency fs in synchronous PWM control. [Figure 4] This waveform diagram shows an example of shunt current Is and motor currents Iu, Iv, and Iw. [Figure 5] This is a circuit diagram of the main circuit of a power converter, showing an example of the flow state of the shunt current Is. [Figure 6] This waveform diagram shows an example of the motor current reproduced by the current processing unit 9 (Figure 1). [Figure 7] This is a waveform diagram showing the reproduced current Iu(n). [Figure 8]It is a functional block diagram showing the configuration of the current average value calculation unit 16 in the first embodiment. [Figure 9] It is a waveform diagram showing the average reproduction current Iu’(n) output by the current average value calculation unit 16. [Figure 10] It is a functional block diagram showing the configuration of the current correction unit 17 in the first embodiment. [Figure 11] It is a functional block diagram showing the configuration of the current correction unit 17 according to the first modification example. [Figure 12] It is a functional block diagram showing the configuration of the current correction unit 17 according to the second modification example. [Figure 13] It is a functional block diagram showing the configuration of the AC motor drive device according to the second embodiment.

Mode for Carrying Out the Invention

[0016] Hereinafter, embodiments of the present invention will be described with reference to the drawings according to the following first and second embodiments.

[0017] In each figure, components with the same reference numerals indicate the same components or components having similar functions.

Examples

[0018] FIG. 1 is a functional block diagram showing the configuration of the AC motor drive device according to the first embodiment of the present invention.

[0019] The AC motor drive device according to this embodiment includes a power conversion device 2 that includes a three-phase inverter circuit and applies a three-phase AC voltage to an AC motor 1, a DC power supply 18 that supplies DC power to the power conversion device 2, a shunt resistor 19 that detects a DC bus current flowing between the DC power supply 18 and the power conversion device 2 on the DC side of the power conversion device 2, and a control device 3 that controls the speed of the AC motor 1 by controlling the switching of the power conversion device 2 based on the detected value of the DC bus current.

[0020] In this embodiment, a permanent magnet synchronous motor is used as the AC motor 1. The AC motor 1 drives a mechanical load 21 such as a blower or a compressor.

[0021] The control device 3 generates control signals Sup, Sun, Svp, Svn, Swp, and Swn for controlling the switching of the power converter 2 by vector control based on the detected value of the DC bus current detected by the shunt resistor 19. The control signals Sup, Sun, Svp, Svn, Swp, and Swn control the on / off switching of the semiconductor switching elements (IGBTs in Figure 5) in the U-phase upper arm, U-phase lower arm, V-phase upper arm, V-phase upper arm, W-phase upper arm, and W-phase lower arm of the three-phase inverter circuit (see Figure 5), respectively. As a result, the power converter 2 converts the DC power input from the DC power supply 18 into three-phase AC power and outputs a three-phase AC voltage to the AC motor 1.

[0022] In vector control, a coordinate transformation is performed between the rotating coordinate system and the fixed coordinate system according to the rotational position of the AC motor 1. In this embodiment, the rotational position of the AC motor 1 is estimated by position sensorless control as described later, without using a rotational position sensor (e.g., an encoder) or a magnetic pole position sensor (e.g., a Hall element).

[0023] Furthermore, in this embodiment, the motor current used in vector control and position sensorless control is detected based on the DC bus current using a so-called single-shunt resistor method, without using a phase current sensor (e.g., a CT). As will be described later, in this embodiment, the three-phase motor currents reproduced from the DC bus current are corrected to extract the fundamental wave components of the motor currents for each phase. These fundamental wave components are used in vector control and position sensorless control. This improves the accuracy and stability of the control of the AC motor 1.

[0024] In this embodiment, the control device 3 includes a computer system such as a microcomputer, and the computer system executes a predetermined program to perform vector control, position sensorless control, and motor current detection based on DC bus current.

[0025] The following describes the operation related to vector control in the control device 3, using Figure 1 as an example.

[0026] The three-phase (U, V, and W phase) motor currents Iu'', Iv'', Iw'' (output of the current correction unit 17), which are reproduced and corrected from the DC bus current, are converted by the coordinate transformation unit 10 into d-axis current Idc and q-axis current Iqc in a rotating coordinate system with the dc-qc axes set as coordinate axes within the control device 3.

[0027] The current command generation unit 5 generates a q-axis current command Iq* and a d-axis current command Id* from the q-axis current Iqc without using a speed controller (ASR) or PI controller. In this embodiment, the current command generation unit 5 generates Iq* from Iqc via a first-order lag element and changes Id* to a predetermined value according to the magnitude of Iqc.

[0028] The rotation speed command generation unit 4 generates a rotation speed command ωr* for the AC motor 1. For example, the rotation speed command generation unit 4 may output a pre-set rotation speed value as ωr*, or it may output a rotation speed value selected from a plurality of pre-set rotation speed values ​​as ωr*. The rotation speed command generation unit 4 may also generate ωr* based on a pre-set rotation speed pattern.

[0029] The vector control unit 6 generates the d-axis voltage command Vdc* and the q-axis voltage command Vqc* using a predetermined voltage equation based on the rotational speed command ωr*, the q-axis current command Iq*, and the d-axis current command Id*. The voltage equation is expressed using the motor constants of the AC motor 1 (d-axis inductance, q-axis inductance, winding resistance, and induced voltage constant), but a detailed explanation is omitted as this is publicly known.

[0030] The coordinate transformation unit 7 converts the d-axis voltage command Vdc* and q-axis voltage command Vqc* in a rotating coordinate system with the dc-qc axes as coordinate axes into three-phase (U, V, and W phase) voltage commands Vu*, Vv*, and Vw* in a fixed axis.

[0031] Here, we will describe the position sensorless control in this embodiment.

[0032] In this embodiment, the control device 3 performs position sensorless control using the axis error estimation unit 11 and the PLL control unit 12 shown in Figure 1.

[0033] Figure 2 is a coordinate plane diagram showing the relationship between coordinate axes and axis error (Δθ) in a rotating coordinate system.

[0034] As shown in Figure 2, the axis error Δθ is defined as the phase difference between the dc-qc axis, which is the control coordinate axis, and the dq axis, which is the actual coordinate axis set on the rotor of the AC motor 1.

[0035] In this embodiment, the direction of the main magnetic flux of the permanent magnets in the rotor of the AC motor 1 is set to the d-axis.

[0036] In Figure 2, "ω1" represents the rotational speed of the rotor of the AC motor 1 (an estimated value in this embodiment).

[0037] The axis error estimation unit 11 (Figure 1) estimates the axis error Δθ by calculating the right-hand side of equation (1) based on the rotation speed ω1 (estimated value), d-axis voltage command Vdc* and q-axis voltage command Vqc*, d-axis current Idc and q-axis current Iqc.

[0038] Δθ = tan -1 {(Vdc*-R·Idc+ω1·Lq·Iqc) / (Vqc*-R·Iqc-ω1·Lq·Idc)}…(1) In equation (1), R and Lq are the winding resistance and q-axis inductance of the AC motor 1, respectively.

[0039] It is well known that in a permanent magnet motor such as AC motor 1, the shaft error Δθ is expressed by equation (1). Therefore, a further detailed explanation of equation (1) is omitted.

[0040] The PLL control unit 12 (Figure 1) uses PLL (Phase Locked Loop) control to estimate a rotational speed ω1 that makes the axis error Δθ zero, based on the axis error estimation unit 11.

[0041] The integration unit 13 (Figure 1) calculates the rotational position θdc of the rotor of the AC motor 1 by integrating the rotational speed ω1 calculated by the PLL control unit 12. This rotational position θdc is used for coordinate transformation in the coordinate transformation units 7 and 10.

[0042] As described above, the control device 3 converts the d-axis voltage command Vdc* and q-axis voltage command Vqc* generated by vector control into three-phase AC voltage commands Vu*, Vv*, and Vw* using the coordinate transformation unit 7. Furthermore, as described below, the control device 3 generates control signals Sup, Sun, Svp, Svn, Swp, and Swn according to Vu*, Vv*, and Vw*.

[0043] The PWM signal generation unit 8 (Figure 1) generates control signals Sup, Sun, Svp, Svn, Swp, Swn, which consist of PWM pulse signals, based on three-phase (U, V, and W phase) voltage commands Vu*, Vv*, Vw*. In this embodiment, the PWM signal generation unit 8 generates Sup, Sun, Svp, Svn, Swp, Swn, which consist of PWM pulse signals, by comparing the carrier wave Sc (for example, a triangular wave) generated by the carrier generation unit 15 with Vu*, Vv*, Vw*.

[0044] In this embodiment, in the high-speed range of the AC motor 1, the PWM signal generation unit 8 generates control signals Sup, Sun, Svp, Svn, Swp, and Swn by synchronous PWM control.

[0045] In synchronous PWM control, the period of the carrier wave is an integer multiple of the period of the sinusoidal voltage command, and the phases of the carrier wave and the voltage command are synchronized. In synchronous PWM control, when the output frequency of the power converter 2 is changed, the carrier frequency is also changed according to the output frequency. At this time, the number of pulses in the PWM pulse signal per half-period of the voltage command remains constant. Furthermore, in synchronous PWM control, the number of pulses may be switched depending on the range of the output frequency.

[0046] Figure 3 is a graph showing an example of the relationship between the rotation frequency command value fr* (=ωr* / 2π) and the carrier frequency fs in synchronous PWM control that switches the number of pulses.

[0047] In this example, in the low-speed range, asynchronous PWM control generates the PWM pulse signal. In the medium-to-high-speed range, it switches to synchronous PWM control, and the number of pulses decreases as the frequency range increases. fs is set to 3n times fr* (n: a natural number). This suppresses the generation of harmonics.

[0048] As shown in Figure 1, in this embodiment, the synchronous PWM control unit 14 calculates the carrier period Ts in synchronous PWM control based on the d-axis voltage command Vdc* and q-axis voltage command Vqc*, the rotation speed command ωr*, and the rotation position θdc, and commands the carrier generation unit 15 to generate a carrier wave Sc such that the carrier period is Ts and the phase of the carrier wave and the voltage command are synchronized.

[0049] The carrier generation unit 15 generates an interrupt signal Trg at the timing when the carrier wave Sc reaches its maximum and minimum values. The control device 3 executes control processing related to vector control, position sensorless control, synchronous PWM control, and motor current detection in response to Trg.

[0050] Next, the detection of motor current using a single shunt resistor method in this embodiment will be explained using Figures 1 and 4-10.

[0051] As shown in Figure 1, the control device 3 includes a current detection unit 20, a current processing unit 9, a current average value calculation unit 16, and a current correction unit 17 in order to detect the motor current of the AC motor 1.

[0052] The current detection unit 20 calculates the DC bus current in the power converter 2 based on the voltage drop across the shunt resistor 19 and outputs the calculated value as the shunt current Is.

[0053] The current processing unit 9 uses the three-phase AC voltage commands Vu*, Vv*, Vw* and the control signals Sup, Sun, Svp, Svn, Swp, Swn to reproduce the three-phase motor currents Iu, Iv, Iw from the shunt current Is output by the current detection unit 20.

[0054] The current average calculation unit 16 calculates the average values ​​Iu', Iv', and Iw' of the three-phase motor current reproduced at a certain point in time and the three-phase motor current reproduced at a previous point in time. In other words, the current average calculation unit 16 calculates the moving average value of the three-phase motor current at two consecutive points in time.

[0055] The current correction unit 17 compares the moving average values ​​Iu', Iv', Iw' of the three-phase motor currents with the three-phase motor currents Iu, Iv, Iw, and calculates correction coefficients to correct Iu, Iv, Iw. Furthermore, the current correction unit 17 corrects Iu, Iv, Iw using the calculated correction coefficients and outputs the corrected three-phase motor currents Iu'', Iv'', Iw''.

[0056] Through the means described above, current ripple is removed from the three-phase motor currents Iu, Iv, Iw (reproduced values), and a three-phase motor current Iu'', Iv'', Iw'', close to the fundamental wave component is obtained. In this embodiment, such three-phase motor currents Iu'', Iv'', Iw'', are converted into d-axis current Idc and q-axis current Iqc by the coordinate transformation unit 10 (Figure 1), and these Idc and Iqc are used for vector control and position sensorless control. This improves the accuracy and stability of vector control and position sensorless control.

[0057] Figure 4 is a waveform diagram showing an example of shunt current Is and motor currents Iu, Iv, Iw. In Figure 4, the waveforms of the three-phase AC voltage commands Vu*, Vv*, Vw* and carrier wave Sc, as well as the control signals Sup, Svp, Swp, are also shown.

[0058] The detection timing shown in Figure 4 indicates the timing at which the current detection unit 20 (Figure 1) detects the shunt current Is flowing through the shunt resistor 19. This detection timing corresponds, for example, to the timing at which the A / D conversion function of the microcomputer constituting the control device 3 is activated. Furthermore, this detection timing is before and after the timing at which the control signal of the intermediate phase among the three-phase AC voltage commands Vu*, Vv*, and Vw* changes.

[0059] The shunt current Is flowing through the shunt resistor 19 at the timing before and after the timing when the control signal of the intermediate phase changes will be explained using Figure 5.

[0060] Figure 5 is a circuit diagram of the main circuit of power converter 2 (Figure 1), showing an example of the flow state of the shunt current Is.

[0061] The intermediate phase is the V phase. That is, in Figure 4, the case is "Vu*>Vv*>Vw*".

[0062] The semiconductor switching elements (IGBTs in this embodiment) that constitute the main three-phase inverter circuit are switched ON and OFF by the control signals Sup, Sun, Svp, Svn, Swp, and Swn. In each phase, the semiconductor switching elements of the upper arm and the lower arm are switched ON and OFF in a complementary manner.

[0063] In Mode 1, the semiconductor switching elements in the U-phase upper arm, V-phase upper arm, and W-phase upper arm are in the ON state, while the other semiconductor switching elements are in the OFF state. In Figure 4, this corresponds to the case where Sup, Svp, and Swp are all "1", i.e., ON control signals. In this Mode 1, no shunt current Is flows through the shunt resistor 19 (Is=0).

[0064] In Mode 2, the semiconductor switching elements in the U-phase upper arm, V-phase upper arm, and W-phase lower arm are in the ON state, while the other semiconductor switching elements are in the OFF state. In Figure 4, this corresponds to the case where Sup and Svp are "1", i.e., ON control signals, and Swp is "0", i.e., OFF control signal. In this Mode 2, a shunt current Is equal to the W-phase motor current flows through the shunt resistor 19 (Is = -Iw: the direction toward the AC motor 1 is considered the positive direction).

[0065] In Mode 3, the semiconductor switching elements in the U-phase upper arm, V-phase lower arm, and W-phase lower arm are in the ON state, while the other semiconductor switching elements are in the OFF state. In Figure 4, this corresponds to the case where Sup is "1", i.e., an ON control signal, and Svp and Swp are "0", i.e., OFF control signals. In this Mode 3, a shunt current Is equal to the U-phase motor current flows through the shunt resistor 19 (Is = Iu).

[0066] In Mode 4, the semiconductor switching elements in the U-phase lower arm, V-phase lower arm, and W-phase lower arm are in the ON state, while the other semiconductor switching elements are in the OFF state. In Figure 4, this corresponds to the case where Sup, Svp, and Swp are all "0", i.e., OFF control signals. In this Mode 1, no shunt current Is flows through the shunt resistor 19 (Is=0).

[0067] In modes 2 and 3 described above, Is corresponds to the timing before and after the timing when the control signal Svp changes from 1 (ON) to 0 (OFF) during the period when the V phase is the intermediate phase, as shown in Figure 4. Therefore, by detecting Is at such timings, -Iw and Iu can be detected. In this case, Iv can be calculated from the three-phase current relationship "Iu + Iv + Iw = 0". In other words, Iu, Iv, and Iw can be reconstructed from Is.

[0068] As described above, the current detection unit 20 (Figure 1) detects the shunt current Is at the timing before and after the timing when the control signal of the intermediate phase among the three-phase AC voltage commands Vu*, Vv*, and Vw* changes. At this time, in order to ensure the simultaneity of detecting the two-phase motor currents detected based on Is, the detection timing of the shunt current Is is set to be near the peak or bottom of Is, as shown in Figure 4. Therefore, as shown in Figure 4, the detected motor currents (Iu, Iv, Iw) are values ​​near the peak or bottom of the current ripple in the motor current.

[0069] Therefore, the motor currents Iu, Iv, and Iw reproduced from the shunt current Is by the current processing unit 9 (Figure 1) contain detection errors with respect to the fundamental wave component of the motor current. In particular, low-inductance motors, which are advantageous for high speeds, have large current ripples, and thus the detection errors become larger.

[0070] Figure 6 is a waveform diagram showing an example of the motor current reproduced from the shunt current Is by the current processing unit 9 (Figure 1).

[0071] In Figure 6, Iu(n) is the U-phase motor current reproduced from the shunt current Is (hereinafter referred to as the "reproduced current"). In Figure 4, the actual U-phase motor current (hereinafter referred to as the "actual current") Iu and the fundamental wave component of Iu (hereinafter referred to as the "fundamental wave component") Iuo are shown together. In Figure 6, the waveforms for one period are shown for Iu(n), Iu, and Iuo, respectively.

[0072] In Figure 6, the reproduced current Iu(n) has a stepped waveform. This is because the control device 3 performs control processing in discrete time using a microcomputer. The plotted points in the waveform of Iu(n) indicate the time when the current processing unit 9 reproduced the motor current from the shunt current Is detected by the current detection unit 20.

[0073] As each plotted point indicates, the reproduced current Iu(n) represents the actual current Iu including current ripple, but it has a detection error with respect to the fundamental wave component Iuo. Therefore, using the d-axis current Idc and q-axis current Iqc converted from the reproduced current Iu(n) may reduce the accuracy and stability of vector control and sensorless control.

[0074] Therefore, in this embodiment, the control device 3 (Figure 1) corrects the reproduced current In(n) using the current average value calculation unit 16 and the current correction unit 17 in order to obtain a current value close to the fundamental wave component.

[0075] The correction of the reproducible current Iu(n) in this embodiment will be described below. The control device 3 corrects the reproducible currents (Iu(n), Iv(n), Iw(n)) for each of the U, V, and W phases using similar means, but the correction of Iu(n) will be described below.

[0076] As shown in Figure 6 above, the waveform of the fundamental wave component Iuo passes near the center of the current ripple. Therefore, in this embodiment, as a correction means, first, the average value of the reproduced current Iu(n) is calculated by the current average value calculation unit 16 (Figure 1), as described below.

[0077] Figure 7 is a waveform diagram showing the reproduced current Iu(n). Note that Iu(n) shown in Figure 7 is the same as Iu(n) shown in Figure 6.

[0078] When the reproduced current Iu(n) shown in Figure 7 is obtained, the control device 3 performs synchronous PWM control with 3 pulses as shown in Figure 3. As a result, Iu(n) changes in a stepwise manner over one cycle, as shown in the figure, and sequentially shows six current values ​​Iu(1), Iu(2), Iu(3), Iu(4), Iu(5), and Iu(6).

[0079] Figure 8 is a functional block diagram showing the configuration of the current average value calculation unit 16 in this embodiment 1.

[0080] The current averaging calculation unit 16 adds the input reproduced current Iu(n) and the output of the delay unit 162, i.e., Iu(n-1) which was input one time point before Iu(n) was input, using the adder 161. Furthermore, the current averaging calculation unit 16 divides the added value from the adder 161 by 2 and outputs the divided value as the averaged reproduced current Iu'(n) (hereinafter referred to as the "average reproduced current"). In other words, the current averaging calculation unit 16 calculates the average value of the input reproduced current Iu(n) and Iu(n-1) which was input one time point before Iu(n) was input (=(Iu(n)+Iu(n-1)) / 2 (arithmetic mean)) and outputs the calculated average value as the average reproduced current Iu'(n).

[0081] In this embodiment, the current values ​​of the reproduced current for one cycle, i.e., the current values ​​Iu(1) to Iu(6) shown in Figure 7, are sequentially input to the current average value calculation unit 16, and the average reproduced current Iu'(n) for one cycle is calculated.

[0082] Figure 9 is a waveform diagram showing the average reproduced current Iu'(n) output by the current average value calculation unit 16.

[0083] In Figure 9, as in Figure 6, the waveforms for one period are shown for both the actual current Iu and the fundamental wave component Iuo.

[0084] As shown in Figure 9, the average reproduced current Iu'(n) has a smaller error with respect to the fundamental wave component Iuo than the reproduced current Iu(n) (Figure 6).

[0085] As shown in Figure 6 above, the waveform of the fundamental wave component Iuo passes near the center of the current ripple. Therefore, by calculating the average value, or median, of the current values ​​(Iu(n) and Iu(n-1)) at two consecutive time points of the reproduced current Iu(n) which includes the current ripple, an average reproduced current Iu'(n) that is closer to the fundamental wave component than Iu(n) can be obtained.

[0086] In this embodiment, the average reproduced current Iu'(n) is further corrected by the current correction unit 17 (Figure 1).

[0087] Figure 10 is a functional block diagram showing the configuration of the current correction unit 17 in this embodiment 1.

[0088] The current correction unit 17 receives the reproduced current Iu(n) and the average reproduced current Iu'(n) as inputs, and the divider 171 calculates the correction coefficient K(n) (=Iu'(n) / Iu(n)).

[0089] The divider 171 calculates a correction coefficient K(n) for a fixed time Tk that is pre-set in the time setter 173, and holds the calculated K(n). In this embodiment, Tk is set to a time corresponding to one cycle of the reproduced current Iu(n).

[0090] Note that Tk can be a time period equivalent to multiple cycles, but it only needs to be a time period equivalent to at least one cycle. This is because the current ripple associated with synchronous PWM control periodically repeats the same waveform.

[0091] The current correction unit 17 uses the switch 172 to select either the calculated correction coefficient K(n) or "1" set by the constant setter 174 as a coefficient to be multiplied by the reproduced current Iu(n).

[0092] The switch 172 has a node T to which K(n) is input and a node F to which the constant "1" is input.

[0093] When the current correction unit 17 begins the correction process, the switch 172 connects its own output to node F. At this time, the switch 172 selects the constant "1" and outputs it.

[0094] When a certain time Tk set by the time setter 173 has elapsed since the current correction unit 17 started executing the correction process, the switch 172 switches the node to which its output is connected from F to T. At this time, the switch 172 sequentially selects and outputs the correction coefficient K(n) for one cycle of the reproduced current Iu(n) held by the divider 171.

[0095] The current correction unit 17 multiplies the reproduced current Iu(n) by the coefficient selected by the switch 172, i.e., either the correction coefficient K(n) or "1", and outputs the multiplied value as the corrected reproduced current Iu''. That is, from the start of the correction process until the calculation of the correction coefficient K(n) is completed, the current correction unit 17 outputs Iu(n) as Iu'', and once the calculation of K(n) is completed, it outputs K(n)·Iu(n) as Iu''(n), which is closer to the fundamental wave component Iuo than Iu(n).

[0096] Similarly, the corrected and reproduced currents Iv''(n) and Iw''(n) for the V-phase and W-phase are calculated using the same method.

[0097] By using a corrected reproducible current (Iu'', Iv''(n), Iw''(n)) with reduced error from the fundamental wave component in vector control and sensorless control, the accuracy and stability of these controls are improved.

[0098] Here, as a comparative example, we will describe a case where the control device 3 directly uses the average reproducible current for vector control or sensorless control at the instant it is calculated.

[0099] As mentioned above, the average reproduced current is calculated using the current value of the reproduced current at a given point in time and the current value of the reproduced current at the point just before that point. Therefore, the d-axis current and q-axis current converted from phase-lag currents are used for control. This makes it difficult to ensure the accuracy and stability of the control.

[0100] In contrast, in the above embodiment, the correction coefficient K(n) is calculated for a certain period of time, and after that period has elapsed, the reproduced current is corrected using the calculated correction coefficient K(n). This improves the accuracy and stability of the control.

[0101] Furthermore, when using the corrected reproducible current, the rotational position θdc (rotational phase) used in the coordinate transformation units 7 and 10 (Figure 1) may be set to the phase near the center of the current ripple. This improves the accuracy and stability of vector control and sensorless control. For example, the control device 3 selects such a phase based on table data showing the relationship between the three-phase AC voltage command and the phase.

[0102] As described above, according to this embodiment 1, a correction coefficient is calculated based on the average reproduced current, which is the average value of the current value at a certain point in time and the current value at the point one point before that point in the reproduced current, and the reproduced current. The corrected reproduced current is calculated by multiplying the reproduced current by the calculated correction coefficient. In this corrected reproduced current, the error with the fundamental wave component is reduced compared to the reproduced current. As a result, the accuracy and stability of the control are improved by controlling the AC motor based on the corrected reproduced current.

[0103] Therefore, stable high-speed driving becomes possible even with low-inductance motors that have large current ripple.

[0104] Furthermore, this embodiment 1 is not limited to the case where the number of pulses in synchronous PWM control is 3, but can also be applied to other numbers of pulses.

[0105] Next, the first and second modifications of Example 1 will be described with reference to Figures 11 and 12.

[0106] In Example 1, during the operation of the AC motor 1, a correction coefficient is calculated for a predetermined time (at least one cycle of the reproduced current), and after the predetermined time has elapsed, the reproduced current is corrected using the calculated correction coefficient. In contrast, in the first and second modifications, a table of correction coefficients is provided in advance, and the reproduced current is corrected based on this table data.

[0107] Figure 11 is a functional block diagram showing the configuration of the current correction unit 17 according to the first modified example.

[0108] The current correction unit 17 in the first modified example has table data 176 that shows a reference value for the correction coefficient. In this modified example, the table data 176 shows the correspondence between the current value of the three-phase motor current detected simultaneously using a phase current sensor, for example, a CT (Current Transformer), and the reference value of the correction coefficient calculated based on this current value in the same manner as in Example 1.

[0109] The current correction unit 17 adjusts K(n) by referring to the table data 176 and comparing the correction coefficient K(n), which is calculated in the same manner as in Example 1, with a reference value of the correction coefficient corresponding to the current value of the reproduced current (Iu(n) in Figure 11). The current correction unit 17 multiplies the reproduced current by the adjusted correction coefficient K'(n) and outputs the multiplied value as the corrected reproduced current (Iu''(n) in Figure 11).

[0110] Similar to Example 1, the current correction unit 17 sets K'(n) for a certain period of time Tk (for example, at least one cycle of the reproduced current) after starting the correction process, and holds the set K'(n) in, for example, the table data 176. After the certain period of time Tk has elapsed, the current correction unit 17 switches the coefficient multiplied by the reproduced current from 1 to K'(n), and corrects the reproduced current using the held K'(n) sequentially.

[0111] Figure 12 is a functional block diagram showing the configuration of the current correction unit 17 according to a second modified example.

[0112] The current correction unit 17 according to the second modification has a table data 177 that shows the relationship between the voltage value of the three-phase AC voltage command, the current value of the fundamental wave component, and a correction coefficient calculated in the same manner as in Example 1.

[0113] The current correction unit 17 receives a three-phase AC voltage command (Vu* in Figure 12) and an average reproduced current (Iu'(n) in Figure 12), and refers to the table data 177 to select a correction coefficient K(n) corresponding to the voltage value of the input three-phase AC voltage command and the current value of the input average reproduced current. The current correction unit 17 multiplies the reproduced current (Iu(n) in Figure 12) by K(n) and outputs the multiplied value as the corrected reproduced current (Iu''(n) in Figure 12).

[0114] Similar to Example 1, the current correction unit 17 sets K(n) for a certain period Tk (for example, at least one cycle of the reproduced current) after starting the correction process, and holds the set K(n) in, for example, the table data 177. After the certain period Tk has elapsed, the current correction unit 17 switches the coefficient multiplied by the reproduced current from 1 to K(n), and corrects the reproduced current using the held K(n) sequentially. [Examples]

[0115] Figure 13 is a functional block diagram showing the configuration of an AC motor drive device, which is Embodiment 2 of the present invention.

[0116] The differences from Example 1 will be explained below.

[0117] As shown in Figure 13, the AC motor 1 drives the blower 22 in the vacuum cleaner as a mechanical load.

[0118] Furthermore, the AC motor drive device according to this second embodiment is equipped with an operation switch 23, as shown in Figure 13. The operation switch 23 selects one of the following operating modes for the vacuum cleaner: "strong," "weak," or "off."

[0119] The rotation speed command generation unit 4 generates a rotation speed command ωr* according to the operating mode selected by the operation switch 23. For example, for the operating modes "strong", "weak", and "off", it generates rotation speed commands ωr1*, ωr2* (<ωr*1) and zero, respectively.

[0120] According to this embodiment 2, a small AC motor can be controlled stably and with high precision, enabling high-speed operation. This allows the small AC motor to be applied to a vacuum cleaner, thereby enabling the vacuum cleaner to be miniaturized.

[0121] Furthermore, the AC motor drive devices according to Example 1 and Example 2 can be applied not only to vacuum cleaners, but also to other electric equipment in which a mechanical load is driven by an AC motor, such as electric compressors, electric vehicles, and electric railway vehicles.

[0122] It should be noted that the present invention is not limited to the embodiments described above, and various modifications are included. For example, the embodiments described above are explained in detail to make the present invention easier to understand, and are not necessarily limited to those having all the configurations described. In addition, it is possible to add, delete, or replace some of the configurations in each embodiment with other configurations. [Explanation of Symbols]

[0123] 1...AC motor 2…Power converter 3…Control device 4…Rotation speed command generation unit 5...Current command generation section 6…Vector control unit 7... Coordinate transformation section 8...PWM signal generation section 9...Current Processing Unit 10... Coordinate transformation section 11...Axis error estimator 12…PLL Control Unit 13...Integrator part 14... Synchronous PWM control unit 15…Carrier generation unit 16...Current average value calculation unit 17...Current correction unit 18…DC power supply 19... Shunt resistor 20...Current detection unit 21... Load 22... Blower 23... Operation switch 161... Adder 162... Delay device 171...Divider 172... Switch 173... Time setting device 174... Constant setting device 175... Multiplier 176,177… Table data

Claims

1. An AC motor control device for controlling a power converter that outputs AC power to an AC motor, A PWM signal generation unit generates a control signal for controlling the power converter, consisting of a PWM pulse signal, in response to an AC voltage command generated by vector control. A current correction unit that corrects the current value of the motor current reproduced based on the DC bus current flowing on the DC side of the power converter by multiplying the current value by a correction coefficient to bring it closer to the fundamental wave of the motor current, Equipped with, The vector control is performed based on the current value corrected by the current correction unit. The current correction unit is characterized in that it sets the correction coefficient based on the current value and the average value of the current values.

2. In the AC motor control device according to claim 1, The current correction unit is characterized by calculating the correction coefficient by dividing the current value by the average value.

3. In the AC motor control device according to claim 1, An AC motor control device characterized in that the rotational position used for coordinate transformation in the vector control is estimated by sensorless control.

4. In the AC motor control device according to claim 1, The AC motor control device is characterized in that the PWM signal generation unit generates the control signal by synchronous PWM control.

5. An AC motor control device for controlling a power converter that outputs AC power to an AC motor, A PWM signal generation unit generates a control signal for controlling the power converter, consisting of a PWM pulse signal, in response to an AC voltage command generated by vector control. A current correction unit that corrects the current value of the motor current reproduced based on the DC bus current flowing on the DC side of the power converter by multiplying the current value by a correction coefficient to bring it closer to the fundamental wave of the motor current, Equipped with, The vector control is performed based on the current value corrected by the current correction unit. The PWM signal generation unit generates the control signal by synchronous PWM control, The current correction unit sets the correction coefficient for a predetermined period of time. The current correction unit is characterized in that, after a predetermined time has elapsed, it corrects the current value by multiplying it by the set correction coefficient.

6. In the AC motor control device according to claim 5, The AC motor control device is characterized in that the predetermined time is at least one cycle of the motor current.

7. In the AC motor control device according to claim 1, The current correction unit is characterized by setting the correction coefficient based on table data.

8. In the AC motor control device according to claim 7, The AC motor control device is characterized in that the table data shows the correspondence between the motor current detected using a phase current sensor and the reference value of the correction coefficient.

9. In the AC motor control device according to claim 7, The AC motor control device is characterized in that the table data shows the relationship between the AC voltage command, the fundamental wave, and the correction coefficient.

10. A power converter that outputs AC power to an AC motor, A control device for controlling the power converter, In an AC motor drive system equipped with, The control device is an AC motor control device according to claim 1 or claim 5, characterized in that the control device is an AC motor control device according to claim 1 or claim 5.

11. AC motor and, A mechanical load driven by the aforementioned AC motor, A power converter that outputs AC power to an AC motor, A control device for controlling the power converter, In electric equipment equipped with, The electric motor device is characterized in that the control device is an AC motor control device according to claim 1 or claim 5.

12. In the electric device according to claim 11, The aforementioned mechanical load is a blower in an electric vacuum cleaner, and the electric device is characterized in that.

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