Current-resonant DC / DC converter

Abstract

To provide a current resonance type DC / DC converter capable of corresponding to a wider range of output voltages without adding an element and a circuit and without complicating a software configuration for control.SOLUTION: A current resonance type DC / DC converter 1 includes a main circuit part 10 including a primary side switching circuit 11, a primary side resonance circuit 12 and a secondary side rectifier circuit 13, and a control part 20. The control part 20 performs frequency modulation control, output suppression conversion control decreasing output, and output rise conversion control rising output. The output suppression conversion control is performed based on a first control amount calculated using a drive frequency. The output rise conversion control is performed based on a second control amount calculated using the drive frequency.SELECTED DRAWING: Figure 1

Description

Technical Field

[0001] The present invention relates to a current resonance type DC / DC converter.

Background Art

[0002] In recent years, LLC type and CLLC type (or also called CLLLC type) current resonance type DC / DC converters have attracted attention. LLC type and CLLC type current resonance type DC / DC converters have advantages such as being small-sized, highly efficient, having a small number of components, and being able to reduce costs, and thus are used in various applications. For example, they are used as the power supply for a rapid charger that charges the battery of an electric vehicle, and as the power supply for a bidirectional V2H (Vehicle to Home) system that charges and discharges the battery of an electric vehicle.

[0003] In LLC type and CLLC type current resonance type DC / DC converters, output control is performed by frequency modulation control, but there is a problem that the output voltage range that can be output is narrow. In frequency modulation control, when reducing the output voltage, it is necessary to increase the drive frequency to cope, but when the drive frequency is increased, the loss increases, and there is also a problem that the output cannot be reduced to zero even if the drive frequency is increased. Furthermore, in frequency modulation control, when the input voltage is low, sufficient boosting cannot be performed, and there is a problem that a desired output voltage cannot be obtained.

[0004] For example, the CHAdeMO standard, which is the charging standard for electric vehicles, requires corresponding to a wide range of output voltages from 150 [V] to 450 [V]. Therefore, in a conventional current resonance type DC / DC converter used in a rapid charger, in addition to frequency modulation control, burst control (intermittent control) is performed when reducing the output, but there is a problem that the output current ripple becomes large when performing intermittent control.

[0005] Patent Document 1 proposes a method for controlling a device by switching between frequency modulation control and phase shift control to accommodate a wide range of output voltages. This method involves increasing the drive frequency to the maximum using frequency modulation control to reduce the output, then starting phase shift control to further reduce the output. However, this method requires separate frequency modulation control and phase shift control, and since the control amount of frequency modulation control and the control amount of phase shift control are unrelated, the control becomes complex, leading to the problem of a complex software configuration.

[0006] Patent Document 2, like Patent Document 1, performs control by switching between frequency modulation control and phase shift control, but proposes a method to suppress output changes associated with switching by performing phase shift control while performing frequency modulation control within a predetermined switching range. However, since this method also switches between two control modes, it has the same problems as the method in Patent Document 1, as well as the difficulty of controlling two control variables simultaneously.

[0007] Furthermore, in V2H systems requiring bidirectional operation, the desired output voltage cannot be obtained if the battery voltage is low during discharge operation. Therefore, a bidirectional buck-boost converter and a bidirectional CLLC current-resonant converter are combined to perform buck-boost operation. This results in challenges such as reduced efficiency, increased number of components, and increased power supply size.

[0008] Patent Document 3 proposes a current-resonant DC / DC converter that includes a boost (voltage boosting) circuit in addition to the LLC resonant circuit. However, this configuration requires separate frequency modulation control and boost control, resulting in a complex software configuration due to the need to switch between two control modes. Furthermore, the number of components increases due to the addition of a switching element for the boost.

[0009] Patent Document 4 proposes a method in which the drive circuit is a current resonant circuit with a full bridge configuration, in which the drive circuit is controlled so that the turn-off of the switching element on the upper arm of the first leg and the turn-off of the switching element on the lower arm of the second leg have a phase difference, and the amount of phase shift is increased as the drive frequency increases. In this method, when the input / output voltage ratio is large and the drive frequency is below the resonant frequency, the amount of phase shift is fixed to a minimum value to perform a boost operation. When the input / output voltage ratio becomes small, the amount of phase shift is increased as the drive frequency increases from the resonant frequency to the maximum drive frequency, and when the drive frequency reaches the maximum drive frequency, the drive frequency is fixed and only the amount of phase shift is increased to perform a step-down operation.

[0010] However, the method described in Patent Document 4 has the problem that, because phase shift control is performed under conditions where the drive frequency is above the resonant frequency, a large resonant current is interrupted, and soft switching may not be achieved, resulting in reduced efficiency. In addition, when the input / output voltage ratio is large, the drive frequency is set lower than the resonant frequency to perform the boost operation, but depending on the input / output conditions, the boost operation may not be sufficiently performed, and there is a problem that it cannot handle a wide range of output voltages. [Prior art documents] [Patent Documents]

[0011] [Patent Document 1] Patent No. 6898511 [Patent Document 2] International Publication No. 2022 / 153723 [Patent Document 3] U.S. Patent Application Publication No. 2015 / 0162840 [Patent Document 4] Japanese Patent Publication No. 2017-99182 [Overview of the project] [Problems that the invention aims to solve]

[0012] The present invention has been made in view of the above circumstances, and its objective is to provide a current-resonant DC / DC converter that can handle a wide range of output voltages without adding elements or circuits, and without complicating the control software configuration. [Means for solving the problem]

[0013] To solve the above problems, the current resonant DC / DC converter according to the present invention is Transformer circuit and, A primary side switching circuit provided on the primary side of the transformer circuit, comprising at least one primary side leg, and comprising a set of primary side switching elements in which the primary side legs are connected in series, A primary side resonant circuit, which includes a resonant coil and a resonant capacitor, is provided between the primary side leg and the transformer circuit, A secondary rectifier circuit provided on the secondary side of the transformer circuit, comprising at least one secondary side leg, the secondary side leg comprising a set of rectifier means connected in series, the rectifier means comprising a diode or a secondary side switching element, A main circuit section comprising, A control unit that controls the primary switching element and / or the secondary switching element, A current resonant DC / DC converter including, The control unit, Frequency modulation control is used to control the output of the main circuit section using the drive frequency as a control variable when the drive frequency of the primary switching element is between a first frequency and a second frequency smaller than the first frequency. At least one of the following controls: an output suppression conversion control that reduces the output using a first control amount calculated using the drive frequency when the drive frequency exceeds the first frequency, and an output boost conversion control that increases the output using a second control amount calculated using the drive frequency when the drive frequency falls below the second frequency; It is characterized by performing the following:

[0014] In this configuration, the frequency modulation control, the output suppression conversion control based on the first control quantity, and the output boost conversion control based on the second control quantity are all frequency controls that include the drive frequency in the control quantity. That is, in this configuration, there is no need to switch to another control mode for control, and there is no need to take measures against changes in output associated with the switching of the control mode, and control can be performed based only on frequency control. Therefore, according to this configuration, it is possible to cope with a wide range of output voltages without adding elements and circuits and without complicating the software configuration for control.

[0015] In the current resonance type DC / DC converter, <000,0078>The output suppression conversion control is Frequency PWM conversion control for controlling the pulse width of the PWM signal for the primary side switching element or the secondary side switching element based on the first control quantity, Frequency intermittent conversion control for controlling the standby period during which the on / off operation of the primary side switching element is not performed based on the first control quantity, It can be configured to include frequency phase shift conversion control for controlling the phase shift amount between the primary side legs based on the first control quantity.

[0016] In the current resonance type DC / DC converter, for example, The output boost conversion control is frequency boost conversion control for short-circuiting the secondary side rectifier circuit by controlling the on-period of the secondary side switching element of the secondary side rectifier circuit during the period when a resonance current flows through the primary side resonance circuit based on the second control quantity.

[0017] In the current resonance type DC / DC converter, The first control quantity is calculated by an operation of a value including the drive frequency and a predetermined first gain, The control unit during the output suppression conversion control can be configured to vary the value of the first frequency and / or the first gain according to the input / output conditions regarding the input voltage, the input / output voltage ratio, or the output power of the main circuit unit.

[0018] Also, in the current resonance type DC / DC converter, the second control amount is calculated by an operation of a value including the driving frequency and a predetermined second gain, the control unit during the output boost conversion control can be configured to vary the value of the second frequency and / or the second gain according to input / output conditions regarding the input voltage, the input / output voltage ratio, or the output power of the main circuit unit.

[0019] In the current resonance type DC / DC converter, the control unit during the output suppression conversion control performs control such that the first control amount does not exceed a predetermined first maximum control amount, the control unit during the output boost conversion control can be configured to perform control such that the second control amount does not exceed a predetermined second maximum control amount.

[0020] In the current resonance type DC / DC converter, the main circuit unit includes a resonance coil and a resonance capacitor, the primary side switching circuit includes two primary side legs each consisting of a first leg and a second leg connected in parallel, the secondary side rectifying circuit includes two secondary side legs each consisting of a third leg and a fourth leg connected in parallel, and the rectifying means of each leg consists of the secondary side switching element, or only the rectifying means constituting the upper and lower arms of the fourth leg consists of the secondary side switching element, or only the rectifying means constituting the lower arm of the third leg and the lower arm of the fourth leg consists of the secondary side switching element, and an anti-parallel diode can be connected in parallel to the current path of the secondary side switching element.

[0021] In the current resonance type DC / DC converter, the control unit performs frequency phase shift conversion control as the output suppression conversion control, The control unit during the frequency phase shift conversion control can be configured to determine the drive frequency by comparing the output value of the output with a target value, and to set the phase shift amount as the difference between the drive frequency and the first frequency multiplied by a predetermined first gain, thereby creating a phase difference between the first leg and the second leg equal to the phase shift amount.

[0022] In the current resonant DC / DC converter, The control unit performs frequency boost conversion control as the output boost conversion control, During the frequency boost conversion control, the control unit determines the drive frequency by comparing the output value of the output with a target value, and sets the boost amount to a value obtained by multiplying the difference between the second frequency and the drive frequency by a predetermined second gain, thereby giving the secondary switching element an on period equal to the boost amount.

[0023] In the current resonant DC / DC converter, The main circuit section includes a resonant coil and a resonant capacitor, The primary-side switching circuit includes two primary-side legs, a first leg and a second leg, which are connected in parallel. The secondary rectifier circuit includes two secondary legs, a third leg and a fourth leg, connected in parallel, and the rectifier means of each leg consists of the secondary switching element. The control unit performs frequency PWM conversion control as the output suppression conversion control. The control unit during the frequency PWM conversion control can determine the drive frequency by comparing the output value of the output with a target value, calculate the phase difference which is the first control quantity using the drive frequency, and configure the control unit to shift the on-timing of the secondary switching element by the phase difference relative to the on-timing of the primary switching element. [Effects of the Invention]

[0024] According to the present invention, a current-resonant DC / DC converter capable of handling a wide range of output voltages can be provided without adding elements or circuits, and without complicating the control software configuration. [Brief explanation of the drawing]

[0025] [Figure 1] This is a diagram showing a current-resonant DC / DC converter according to the first embodiment. [Figure 2] This figure shows the relationship between the drive frequency, phase shift amount, and boost amount during discharge operation in a current-resonant DC / DC converter according to the first embodiment. [Figure 3] This is a control flow diagram of a current-resonant DC / DC converter according to the first embodiment. [Figure 4] This is a control block diagram of a current-resonant DC / DC converter according to the first embodiment. [Figure 5] This is a waveform diagram of the switching element drive during frequency modulation control of a current-resonant DC / DC converter according to the first embodiment. [Figure 6] This diagram shows the current flow of a current-resonant DC / DC converter according to the first embodiment, where (A) is the flow during mode 1 in Figure 5, and (B) is the flow during mode 3 in Figure 5. [Figure 7] This is a waveform diagram of the switching element drive during frequency phase shift conversion control of a current resonant DC / DC converter according to the first embodiment. [Figure 8] This diagram shows the current flow of a current-resonant DC / DC converter according to the first embodiment, where (A) is the flow during mode 2 of Figure 7 and (B) is the flow during mode 4 of Figure 7. [Figure 9] This is a waveform diagram of the switching element drive during frequency boost conversion control of the current resonant DC / DC converter according to the first embodiment. [Figure 10]This diagram shows the current flow of a current-resonant DC / DC converter according to the first embodiment, where (A) is the flow during mode 1B in Figure 9, and (B) is the flow during mode 3B in Figure 9. [Figure 11] This figure shows a current-resonant DC / DC converter according to the first modified example. [Figure 12] This figure shows a current-resonant DC / DC converter according to the second modified example. [Figure 13] This is a waveform diagram of the switching element drive during frequency boost conversion control of a current resonant DC / DC converter according to the second modified example. [Figure 14] This diagram shows the current flow in a current-resonant DC / DC converter according to the second modified example, where (A) is the flow during the mode 1B period in Figure 13, and (B) is the flow during the mode 3B period in Figure 13. [Figure 15] This figure shows a current-resonant DC / DC converter according to the third modified example. [Figure 16] This is a waveform diagram of the switching element drive during frequency PWM conversion control of a current resonant DC / DC converter according to the third modified example. [Figure 17] This diagram shows the current flow of a current-resonant DC / DC converter according to the third modified example, where (A) is the flow during mode 1' in Figure 16, and (B) is the flow during mode 2' in Figure 16. [Figure 18] This diagram shows the current flow in a current-resonant DC / DC converter according to the third modified example, where (A) is the flow during mode 3' of Figure 16, and (B) is the flow during mode 4' of Figure 16. [Modes for carrying out the invention]

[0026] Hereinafter, embodiments of the current-resonant DC / DC converter according to the present invention will be described with reference to the attached drawings.

[0027] [First Embodiment] Figure 1 shows a bidirectional CLLC current-resonant DC / DC converter 1 according to a first embodiment of the present invention. The current-resonant DC / DC converter 1 consists of a main circuit section 10 and a control section 20. The main circuit section 10 includes a transformer circuit Tr, a primary-side switching circuit 11, a primary-side resonant circuit 12, a secondary-side rectifier circuit 13, a secondary-side resonant circuit 14, and terminals T1 to T4.

[0028] The transformer circuit Tr consists of one or more high-frequency isolation transformers. The high-frequency isolation transformer comprises a primary coil and a secondary coil. The primary coil is connected to the primary switching circuit 11 via a primary resonant circuit 12, and the secondary coil is connected to the secondary rectifier circuit 13 via a secondary resonant circuit 14.

[0029] The primary-side switching circuit 11 is a full-bridge circuit consisting of switching elements Q1 to Q4 (corresponding to the "primary-side switching elements" of the present invention). That is, the primary-side switching circuit 11 includes a first leg and a second leg (corresponding to the "primary-side leg" of the present invention) connected in parallel, with a pair of switching elements Q1 and Q2 connected in series in the first leg forming an upper and lower arm, and a pair of switching elements Q3 and Q4 connected in series in the second leg forming an upper and lower arm. Note that although switching elements Q1 to Q4 are each composed of one switching element, they may be composed of two or more switching elements connected in parallel for current distribution.

[0030] For the switching elements Q1 to Q4, power semiconductors such as IGBTs (insulated-gate bipolar transistors), SiC (silicon carbide), and MOSFETs (metal-oxide-semiconductor field-effect transistors) using GaN (gallium nitride) can be used. Furthermore, for an LLC-type drive circuit, it is desirable that the current paths of the switching elements Q1 to Q4 have regenerative reverse-circuit diodes (hereinafter referred to as diodes) D1 to D4 and soft-switching partial resonant capacitors (hereinafter referred to as capacitors) C1 to C4 connected in parallel. Diodes D1 to D4 may be built-in diodes of the switching elements Q1 to Q4 or external diodes. Capacitors C1 to C4 may be parasitic capacitances of the switching element Q1, external capacitors, or a combination thereof.

[0031] The primary resonant circuit 12 includes a resonant coil Lr1 and a resonant capacitor Cr1. The resonant coil Lr1 and the resonant capacitor Cr1, together with the excitation coil and primary coil of the transformer circuit Tr, constitute an LLC-type resonant circuit. Note that the excitation coil of the transformer circuit Tr is omitted from the illustration as it is included in the transformer circuit Tr. The resonant coil Lr1 may consist only of the leakage flux of the transformer circuit Tr, or it may consist of the leakage flux and an individual coil.

[0032] The secondary rectifier circuit 13 is a full-bridge circuit consisting of switching elements Q5 to Q8 (corresponding to the "secondary switching elements" of the present invention). That is, the secondary rectifier circuit 13 includes a third leg and a fourth leg (corresponding to the "secondary legs" of the present invention) connected in parallel, with a pair of switching elements Q5 and Q6 connected in series in the third leg forming the upper and lower arms, and a pair of switching elements Q7 and Q8 connected in series in the fourth leg forming the upper and lower arms. Note that although switching elements Q5 to Q8 are each composed of one switching element, they may be composed of two or more switching elements connected in parallel for current distribution.

[0033] For example, the same power semiconductors as those used for switching elements Q1 to Q4 can be used for switching elements Q5 to Q8. In addition, reverse-circuit diodes (hereinafter referred to as diodes) D5 to D8 and partial resonant capacitors (hereinafter referred to as capacitors) C5 to C8 are connected in parallel to the current path of switching elements Q5 to Q8. Note that diodes D5 to D8 may be external diodes or built-in diodes of switching elements Q5 to Q8, and capacitors C5 to C8 may be parasitic capacitors of switching elements Q5 to Q8, and may not be present when the rectifier circuit is operating. Thus, the secondary rectifier circuit 13 has the same configuration as the primary switching circuit 11, and the switching elements Q5 to Q8, diodes D5 to D8, and capacitors C5 to C8 have the same configuration as the primary switching elements Q1 to Q4, diodes D1 to D4, and capacitors C1 to C4.

[0034] The secondary resonant circuit 14 includes a resonant coil Lr2 and a resonant capacitor Cr2. The resonant coil Lr2 and the resonant capacitor Cr2, together with the excitation coil and secondary coil of the transformer circuit Tr, constitute an LLC-type resonant circuit. In other words, the secondary resonant circuit 14 has the same configuration as the primary resonant circuit 12, and the main circuit section 10 is a symmetrical circuit with the transformer circuit Tr in between.

[0035] The control unit 20 includes a control circuit for controlling the switching elements Q1 to Q4 of the primary-side switching circuit 11 and the switching elements Q5 to Q8 of the secondary-side rectifier circuit 13. The control circuit of the control unit 20 consists of a control processor, memory, and peripheral circuits, and can be, for example, a microcontroller or a DSP. The control unit 20 also includes a detection circuit consisting of sensors and an AD converter for detecting voltages and currents necessary for control, but the description and illustration of the detection circuit are omitted.

[0036] The control unit 20 monitors the input (input current, input voltage, or input power) of the main circuit unit 10 and controls the switching (on / off) of the switching elements Q1 to Q8 so that the output (output current, output voltage, or output power) of the main circuit unit 10 becomes a desired value.

[0037] Specifically, the control unit 20 performs frequency modulation control, frequency phase shift conversion control (corresponding to the "output suppression conversion control" of the present invention), and frequency boost conversion control (corresponding to the "output boost conversion control" of the present invention). All of these controls are frequency controls. Frequency phase shift conversion control differs from conventional phase shift control in that the control variable is the phase shift amount calculated using the drive frequency (switching frequency). Frequency boost conversion control differs from conventional boost control in that the control variable is the boost amount calculated using the drive frequency (switching frequency).

[0038] Hereinafter, it is assumed that a rechargeable battery (for example, an electric vehicle battery) is connected to terminals T1 and T2, and the DC terminals of a DC / AC inverter are connected to terminals T3 and T4. Furthermore, the main circuit section 10 performs a discharge operation that generates a bus voltage (output voltage) V2 with the DC / AC inverter based on the battery voltage (input voltage) V1.

[0039] During discharge operation, the control unit 20 monitors the input voltage V1 and controls the drive-side switching elements Q1-Q4 and the rectifier-side switching elements Q5-Q8 so that the output voltage V2 becomes a desired value. For example, during frequency modulation control, the control unit 20 turns the switching elements Q1-Q4 on and off at an appropriate drive frequency and performs synchronous rectification control or diode rectification control on the switching elements Q5-Q8. In the following explanation, for simplicity, it will be assumed that the switching elements Q5-Q8 are turned off and diode rectification is performed (diode rectification control is performed).

[0040] Fig. 2 shows the relationship between the driving frequency f of the switching elements Q1 to Q4 during the discharge operation, the phase shift amount θ of the frequency phase shift conversion control, and the boost amount ψ of the frequency boost conversion control. The phase shift amount θ corresponds to the "first control amount" of the present invention, and the boost amount ψ corresponds to the "second control amount" of the present invention. Here, the phase shift amount θ is the phase difference between the first leg (switching elements Q1, Q2) and the second leg (switching elements Q3, Q4) of the primary side switching circuit 11, and the boost amount ψ is the value represented in [°] of the period during which any one of the switching elements Q5 to Q8 of the secondary side rectifying circuit 13 is turned on and short-circuited during the period when the resonance current flows through the primary side resonance circuit 12.

[0041] When the driving frequency f is included in the range from a predetermined second frequency fψs (90 [kHz] in Fig. 2) to a first frequency fθs (140 [kHz] in Fig. 2), the control unit 20 performs frequency modulation control for controlling the output of the main circuit unit 10 using the driving frequency f as a control amount. When increasing the output during frequency modulation control, the control unit 20 decreases the driving frequency f, and when decreasing the output, the control unit 20 increases the driving frequency f. In Fig. 2, the high resonance frequency fr determined by the resonance coil Lr1 and the resonance capacitor Cr1 is 125 kHz, and the low resonance frequency f0 including the excitation coil of the transformer circuit Tr is 30 kHz. That is, it is set such that f0 < fψs < fr < fθs in operation.

[0042] As the output decreases, the driving frequency f increases. When the driving frequency f exceeds 140 [kHz], which is the first frequency fθs, the control unit 20 starts the frequency phase shift conversion control. The control unit 20 sets the phase shift amount θ, which is the control amount of the frequency phase shift conversion control, to a value obtained by multiplying the difference between the driving frequency f at that time and the first frequency fθs by a first gain Kθ. The first gain Kθ is a predetermined frequency phase shift conversion gain. When increasing the output during the frequency phase shift conversion control, the control unit 20 decreases the phase shift amount θ, and when decreasing the output, the control unit 20 increases the phase shift amount θ.

[0043] As the output further decreases, the phase shift amount θ increases, and if the drive frequency f exceeds the maximum phase shift drive frequency fθh of 225 [kHz], the control unit 20 fixes the phase shift amount θ to 170°, which is the maximum phase shift amount θh (corresponding to the "first maximum control amount" of the present invention). The phase shift amount θ [°] is the angle with respect to one period T (T=360°) of the drive frequency f, and is a value in the range of 0° to a maximum of 180°.

[0044] On the other hand, as the output increases, the drive frequency f decreases, and when the drive frequency f falls below the second frequency fψs, which is 90 kHz, the control unit 20 starts frequency boost conversion control. The control unit 20 sets the boost amount ψ, which is the control amount for frequency boost conversion control, to a value obtained by multiplying the difference between the second frequency fψs and the drive frequency f at that time by the second gain Kψ. The second gain Kψ is a predetermined frequency boost conversion gain, and when the input voltage V1, which is the battery voltage V1x, is V11 (in Figure 2, V11 = 300 V), it becomes Kψ(V11), and when the battery voltage V1x is V12 (in Figure 2, V12 = 150 V), it becomes Kψ(V12) (however, Kψ(V12) > Kψ(V11)). During frequency boost conversion control, the control unit 20 increases the boost amount ψ when increasing the output, and decreases the boost amount ψ when decreasing the output.

[0045] As the output increases further, the boost amount ψ increases, and when the drive frequency f falls below the maximum boost drive frequency fψh of 67 [kHz], the control unit 20 fixes the boost amount ψ to the maximum boost amount ψh (corresponding to the "second maximum control amount" of the present invention). In Figure 2, the maximum boost amount ψh(V11) when the battery voltage V1x is V11 is 15°, and the maximum boost amount ψh(V12) when the battery voltage V1x is V12 is 45°.

[0046] In frequency boost conversion control, switching elements Q5 and Q6 are turned on to perform boost drive. However, when boost drive is performed, the resonant current is short-circuited by the secondary rectifier circuit 13, and a large amount of energy is accumulated in the resonant coils Lr1 and Lr2 in a short time. Therefore, the output changes significantly in response to changes in the boost pulse width (boost amount ψ) used to turn on the switching elements Q5 and Q6. As a result, control may become unstable depending on the battery voltage V1x. In this embodiment, however, the second gain Kψ and the maximum boost amount ψh are varied according to the battery voltage V1x, so the slope of the boost amount ψ can be adjusted in conjunction with the battery voltage V1x, making it possible to take a large boost ratio according to the input and output conditions while avoiding control instability.

[0047] For example, the control unit 20 may pre-store predetermined input / output conditions (input voltage V1, input / output voltage ratio V1 / V2, or output power) and a second gain Kψ and maximum boost amount ψh associated with those input / output conditions in memory, acquire input / output information with a detection circuit, read the second gain Kψ and maximum boost amount ψh from memory based on the detected input / output information, and start frequency boost conversion control. In this way, by varying the second gain Kψ and maximum boost amount ψh according to the input / output conditions, an optimal boost rate can be obtained even when the input voltage V1 is low, and the range of the outputtable output voltage V2 can be appropriately expanded. In addition, the second frequency fψs may be varied according to the input / output conditions, similar to the second gain Kψ and maximum boost amount ψh.

[0048] Furthermore, the first gain Kθ, the maximum phase shift amount θh, and the first frequency fθs may be varied according to the input / output conditions. For example, the control unit 20 may pre-store predetermined input / output conditions and the first gain Kθ, maximum phase shift amount θh, and first frequency fθs associated with those input / output conditions in memory, acquire input / output information with the detection circuit, read the first gain Kθ, maximum phase shift amount θh, and first frequency fθs from memory based on the detected input / output information, and start frequency phase shift conversion control.

[0049] Figure 3 shows a control flow diagram of the transition control between frequency modulation control, frequency phase shift conversion control, and frequency boost conversion control by the control circuit of the control unit 20. Before performing the transition control, the control unit 20 performs frequency modulation control, turning switching elements Q1 to Q4 on and off at an appropriate drive frequency f, and turning off switching elements Q5 to Q8 to cause the secondary rectifier circuit 13 to perform diode rectification.

[0050] When the control unit 20 starts transition control (S1), it acquires input / output information using a detection circuit (S2). The input / output information includes at least one output value of the input voltage V1, input current, output voltage V2, and output current.

[0051] The control unit 20, having acquired input / output information, reads control parameters from memory (S3). The control parameters include the first frequency fθs, the first gain Kθ, the maximum phase shift amount θh, the second frequency fψs, the second gain Kψ, and the maximum boost amount ψh. The second gain Kψ and the maximum boost amount ψh will have different values ​​depending on the input voltage V1, as shown in Figure 2. Note that the fixed parameters among the control parameters only need to be read once at the start of the transition control.

[0052] Next, the control unit 20 compares the output value (output current value, output voltage value, or output power value) included in the input / output information with the target value (target output current value, target output voltage value, or target output power value) and determines the drive frequency f of the switching elements Q1 to Q4 so that the output value approaches the target value (S4). For example, if the input / output information includes the output value of output voltage V2 and the target value is the target output voltage value, the control unit 20 compares the output value of output voltage V2 with the target output voltage value and determines the drive frequency f so that the output value of output voltage V2 approaches the target output voltage value.

[0053] If the output value is smaller than the target value, the control unit 20 decreases the drive frequency f to increase the output. On the other hand, if the output value is larger than the target value, the control unit 20 increases the drive frequency f to decrease the output. The target value may be, for example, a value pre-set in the current resonant DC / DC converter 1, a value determined from input / output conditions, or a value input from an external device.

[0054] The control unit 20 determines whether the drive frequency f determined in step S4 is greater than the first frequency fθs read out in step S3 (S5). If the drive frequency f is not greater than the first frequency fθs (NO in S5), the control unit 20 determines whether the drive frequency f determined in step S4 is less than the second frequency fψs read out in step S3 (S6).

[0055] If the drive frequency f is not smaller than the second frequency fψs (NO in S6), the control unit 20 performs frequency modulation control. During frequency modulation control, the control unit 20 sets the phase difference between the switching elements Q1 and Q2 of the first leg and the switching elements Q3 and Q4 of the second leg to zero, and turns the switching elements Q1 to Q4 on and off at the drive frequency f determined in step S4. At the same time, it turns off the switching elements Q5 and Q6 of the third leg and the switching elements Q7 and Q8 of the fourth leg, causing the secondary rectifier circuit 13 to perform diode rectification (S7).

[0056] In step S5, if the drive frequency f is greater than the first frequency fθs (YES in S5), the control unit 20 performs frequency phase shift conversion control. During frequency phase shift conversion control, the control unit 20 calculates the phase shift amount θ by multiplying the difference between the drive frequency f determined in step S4 and the first frequency fθs by the first gain Kθ read out in step S3 (S8).

[0057] The control unit 20, having calculated the phase shift amount θ, determines whether the phase shift amount θ is greater than the maximum phase shift amount θh read out in step S3 (S9). If the phase shift amount θ is greater than the maximum phase shift amount θh (YES in S9), the control unit 20 fixes the phase shift amount θ to the maximum phase shift amount θh by changing the phase shift amount θ calculated in step S8 to the maximum phase shift amount θh (S10), and proceeds to the next step S11. If the phase shift amount θ is less than or equal to the maximum phase shift amount θh (NO in S9), the control unit 20 proceeds to step S11 with the phase shift amount θ remaining as the phase shift amount θ calculated in step S8.

[0058] In step S11, the control unit 20 drives the primary switching circuit 11 with a phase shift. Specifically, the control unit 20 shifts the phase of the second leg relative to the phase of the first leg by a phase shift amount θ, and turns the switching elements Q1 to Q4 on and off at the drive frequency f determined in step S4. At the same time, it turns off the switching elements Q5 and Q6 of the third leg and the switching elements Q7 and Q8 of the fourth leg, causing the secondary rectifier circuit 13 to perform diode rectification (S11).

[0059] In step S6, if the drive frequency f is smaller than the second frequency fψs (YES in S6), the control unit 20 performs frequency boost conversion control. During frequency boost conversion control, the control unit 20 calculates the boost amount ψ by multiplying the difference between the second frequency fψs and the drive frequency f determined in step S4 by the second gain Kψ read out in step S3 (S12).

[0060] The control unit 20, having calculated the boost amount ψ, determines whether the boost amount ψ is greater than the maximum boost amount ψh read out in step S3 (S13). If the boost amount ψ is greater than the maximum boost amount ψh (YES in S13), the control unit 20 fixes the boost amount ψ to the maximum boost amount ψh by changing the boost amount ψ calculated in step S12 to the maximum boost amount ψh (S14), and proceeds to the next step S15. If the boost amount ψ is less than or equal to the maximum boost amount ψh (NO in S13), the control unit 20 proceeds to step S15 with the boost amount ψ remaining the boost amount ψ calculated in step S12.

[0061] In step S15, the control unit 20 boost-drives the secondary rectifier circuit 13. Specifically, the control unit 20 sets the phase difference between the first leg and the second leg to zero and turns the switching elements Q1 to Q4 on and off at the drive frequency f determined in step S4. At the same time, it turns on the switching elements Q5 and Q6 of the third leg by a boost amount ψ in synchronization with the first leg, and turns off the switching elements Q7 and Q8 of the fourth leg to perform diode rectification (S15).

[0062] After steps S7, S11, and S15, the control unit 20 determines whether to continue control (S16). The control unit 20 makes the determination in step S16 based, for example, on whether it has received a control termination command from an external device. If the control unit 20 has not received a control termination command, it determines to continue the transition control (YES in S16), returns to step S2, and repeats the process from step 2 onward. If the control unit 20 has received a control termination command, it determines not to continue the transition control (NO in S16), and terminates the transition control (S17).

[0063] Figure 4 shows the control block diagram of the frequency phase shift conversion control and frequency boost conversion control by the control unit 20. Note that the phase shift amount θ[°] and boost amount ψ[°] are angles when the period T of the drive frequency f[Hz] is 360°.

[0064] The control unit 20 comprises a frequency phase shift conversion control block 30 and a frequency boost conversion control block 40. The frequency phase shift conversion control block 30 includes a first calculation unit 31, a first multiplication unit 32, and a first clamping unit 33, while the frequency boost conversion control block 40 includes a second calculation unit 41, a second multiplication unit 42, and a second clamping unit 43.

[0065] When the output is reduced, the control unit 20 increases the drive frequency f of the switching elements Q1 to Q4. If the drive frequency f is greater than the first frequency fθs, the first calculation unit 31 outputs the difference between the drive frequency f and the first frequency fθs to the first multiplication unit 32. The first multiplication unit 32 multiplies the above difference by the first gain Kθ to calculate the phase shift amount θ and outputs the phase shift amount θ to the first clamping unit 33. The first clamping unit 33 sets the upper limit of the phase shift amount θ as the maximum phase shift amount θh and outputs the phase shift amount θ[°]=(f-fθs)×Kθ≦θh.

[0066] The control unit 20 drives the primary switching circuit 11 with a phase shift based on the phase shift amount θ[°]=(f-fθs)×Kθ≦θh. Specifically, the control unit 20 switches the switching elements Q3 and Q4 of the second leg on and off by shifting the phase of the on / off state of the switching elements Q1 and Q2 of the first leg by the above phase shift amount θ (phase shift amount θ with respect to period T).

[0067] In the frequency-phase-shift conversion control block 30, the drive frequency f increases when the output is reduced, so the difference (f-fθs) output from the first calculation unit 31 increases. Until the drive frequency f reaches the maximum phase-shift drive frequency fθh (see Figure 2), the phase-shift amount θ increases uniformly with increasing drive frequency f, and when the drive frequency f exceeds the maximum phase-shift drive frequency fθh, the phase-shift amount θ is fixed at the maximum phase-shift amount θh. The control unit 20 is designed to prevent the drive frequency f from increasing beyond the maximum phase-shift drive frequency fθh.

[0068] When the control unit 20 increases the output, it reduces the drive frequency f of the switching elements Q1 to Q4. If the drive frequency f is smaller than the second frequency fψs, the second calculation unit 41 outputs the difference between the second frequency fψs and the drive frequency f to the second multiplication unit 42. The second multiplication unit 42 calculates the boost amount ψ by multiplying the above difference by the second gain Kψ(V1x) and outputs the boost amount ψ to the second clamp unit 43. The second clamp unit 43 sets the upper limit of the boost amount ψ as the maximum boost amount ψh(V1x) and outputs the boost amount ψ[°]=(fψs-f)×Kψ(V1x)≦ψh(V1x). The second gain Kψ(V1x) and the maximum boost amount ψh(V1x) are quantities that vary with the input voltage V1, which is the battery voltage V1x (see, for example, V11 and V12 in Figure 2).

[0069] The control unit 20 boosts the secondary rectifier circuit 13 based on the boost amount ψ[°]=(fψs-f)×Kψ(V1x)≦ψh(V1x). Specifically, the control unit 20 turns on the switching elements Q5 and Q6 of the third leg in synchronization with the switching elements Q1 and Q2 of the first leg by a boost amount ψ (a boost amount ψ relative to the period T), and turns off the switching elements Q7 and Q8 of the fourth leg to perform diode rectification.

[0070] In the frequency boost conversion control block 40, the drive frequency f decreases when the output is increased, so the difference (fψs-f) output from the second calculation unit 41 increases. Until the drive frequency f decreases and reaches the maximum boost drive frequency fψh (see Figure 2), the boost amount ψ increases uniformly with the decrease in drive frequency f, and when the drive frequency f falls below the maximum boost drive frequency fψh, the boost amount ψ is fixed at the maximum boost amount ψh (V1x). Note that the maximum boost drive frequency fψh is a frequency higher than the resonant frequency f0, and the turns ratio (boost ratio) of the transformer circuit Tr is designed from the input / output conditions so that the drive frequency f does not fall below the maximum boost drive frequency fψh. The control unit 20 controls the drive frequency f so that it does not fall below the maximum boost drive frequency fψh.

[0071] Figure 5 shows the drive waveform diagrams for switching elements Q1 to Q8 during frequency modulation control. Switching element Q1 turns on when the drive waveform is high and turns off when the drive waveform is low. Similarly, switching elements Q2 to Q8 turn on when the drive waveform is high and turn off when the drive waveform is low.

[0072] As shown in Figure 5, the drive waveforms of switching elements Q1 and Q2 in the first leg and the drive waveforms of switching elements Q3 and Q4 in the second leg are in phase (zero phase difference). The drive waveforms of switching elements Q5 and Q6 in the third leg and the drive waveforms of switching elements Q7 and Q8 in the fourth leg are always low.

[0073] The control unit 20 sets the phase difference between the switching elements Q1 and Q2 of the first leg and the switching elements Q3 and Q4 of the second leg to zero, driving the switching elements Q1 to Q4 with a period To and an on-duty cycle D (for example, D = 50%), and turns off the switching elements Q5 and Q6 of the third leg and the switching elements Q7 and Q8 of the fourth leg, causing the secondary rectifier circuit 13 to perform diode rectification.

[0074] Furthermore, the control unit 20 alternately turns on and off the switching elements Q1 and Q2 constituting the first leg with a phase difference of 180°, and alternately turns on and off the switching elements Q3 and Q4 constituting the second leg with a phase difference of 180°. Note that there is a predetermined dead time between the on / off timing of switching element Q1 and switching element Q2, and there is also a predetermined dead time between the on / off timing of switching element Q3 and switching element Q4, but for the sake of simplicity, the illustration of the dead time is omitted. The same applies to the following figures.

[0075] Figure 6(A) shows the current flow in the main circuit section 10 during mode 1 in Figure 5, and Figure 6(B) shows the current flow in the main circuit section 10 during mode 3 in Figure 5. Note that the currents shown in Figure 6 are the resonant current and the load current, and the excitation current is not shown. Also, the currents during the on / off switching of switching elements Q1 to Q4 (transient currents) are not shown.

[0076] During mode 1 in Figure 5, switching elements Q1 and Q4 are turned on, while switching elements Q2 and Q3 are turned off. As shown in Figure 6(A), a battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q1 and Q4. As a result, on the rectifier side, a load current flows between terminals T3 and T4 via diodes D5 and D8.

[0077] During mode 3 in Figure 5, switching elements Q1 and Q4 are off, and switching elements Q2 and Q3 are on. As shown in Figure 6(B), a battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q2 and Q3, in the opposite direction to mode 1. As a result, on the rectifier side, a load current flows between terminals T3 and T4 via diodes D6 and D7.

[0078] Figure 7 shows the drive waveforms of switching elements Q1 to Q8 during frequency phase shift conversion control. Figure 8(A) shows the current flow in the main circuit 10 during mode 2 of Figure 7, and Figure 8(B) shows the current flow in the main circuit 10 during mode 4 of Figure 7. Note that the currents shown in Figure 8 are the resonant current and load current, while the excitation current and transient current are not shown.

[0079] As shown in Figure 7, the drive waveforms of switching elements Q3 and Q4 in the second leg are phase-shifted by a phase shift amount θ compared to the drive waveforms of switching elements Q1 and Q2 in the first leg. The drive frequency f of switching elements Q1 to Q4 has increased by the amount obtained by dividing the phase shift amount θ by the first gain Kθ, and the period Tθ is shorter than the period To in Figure 5. In addition, the drive waveforms of switching elements Q5 and Q6 in the third leg and the drive waveforms of switching elements Q7 and Q8 in the fourth leg are always Low and OFF.

[0080] The state of switching elements Q1 to Q4 during the Mode 1 and Mode 3 periods in Figure 7 is the same as in the Mode 1 and Mode 3 cases in Figure 5, and the current flow in the main circuit section 10 during those periods is the same as in Figures 6 (A) and (B). However, since each period of Mode 1 and Mode 3 in Figure 7 is shortened by at least the phase shift amount θ, the resonant current and load current are reduced compared to the Mode 1 and Mode 3 cases in Figure 5.

[0081] During the Mode 2 period in Figure 7, switching elements Q1 and Q3 are ON, and switching elements Q2 and Q4 are OFF. As shown in Figure 8(A), the resonant current on the drive side continues to flow through switching elements Q1 and Q3, but gradually decreases to zero. Similarly, the load current on the rectifier side continues to flow through diodes D5 and D8, but gradually decreases to zero.

[0082] During mode 4 in Figure 7, switching elements Q2 and Q4 are turned on, while switching elements Q1 and Q3 are turned off. As shown in Figure 8(B), the resonant current on the drive side continues to flow through switching elements Q2 and Q4, but gradually decreases to zero. Similarly, the load current on the rectifier side continues to flow through diodes D6 and D7, but gradually decreases to zero.

[0083] Figure 9 shows the drive waveforms of switching elements Q1 to Q8 during frequency boost conversion control. Figure 10(A) shows the current flow in the main circuit 10 during mode 1B in Figure 9, and Figure 10(B) shows the current flow in the main circuit 10 during mode 3B in Figure 9. Note that the currents shown in Figure 10 are the resonant current and load current, while the excitation current and transient current are not shown.

[0084] As shown in Figure 9, the drive waveforms of the switching elements Q1 and Q2 in the first leg and the drive waveforms of the switching elements Q3 and Q4 in the second leg are in phase (zero phase difference). The drive frequency f of switching elements Q1 to Q4 is reduced by the amount obtained by dividing the boost amount ψ by the second gain Kψ, and the period Tψ is longer than the period To in Figure 5.

[0085] The drive waveform of switching element Q6 in the third leg becomes High in synchronization with the drive waveform of switching element Q1 in the first leg (or the drive waveform of switching element Q4 in the second leg), and becomes Low after the boost amount ψ period (mode 1B period) has elapsed. The drive waveform of switching element Q5 in the third leg becomes High in synchronization with the drive waveform of switching element Q2 in the first leg (or the drive waveform of switching element Q3 in the second leg), and becomes Low after the boost amount ψ period (mode 3B period) has elapsed. The drive waveforms of switching elements Q7 and Q8 in the fourth leg are always Low.

[0086] The states of switching elements Q1 to Q8 during the modes 1 and 3 periods in Figure 9 are the same as those in the modes 1 and 3 periods in Figure 5, and the current flow in the main circuit section 10 during those periods is the same as in Figures 6 (A) and (B).

[0087] During mode 1B in Figure 9, switching elements Q1, Q4, and Q6 are turned on, while switching elements Q2, Q3, Q5, Q7, and Q8 are turned off. As shown in Figure 10(A), the battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q1 and Q4. The rectifier side is short-circuited, and a current path is formed through switching element Q6 and diode D8, allowing current to flow continuously. As a result, a large resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1, and a large amount of energy is accumulated. This accumulated energy flows as a load current between terminals T3 and T4 in the next mode 1. This causes the main circuit section 10 to perform a voltage boost operation.

[0088] During mode 3B in Figure 9, switching elements Q2, Q3, and Q5 are turned on, while switching elements Q1, Q4, Q6-Q8 are turned off. As shown in Figure 10(B), the battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q2 and Q3, in the opposite direction to mode 1B. The rectifier side is short-circuited, and a current path is formed through switching element Q5 and diode D7, allowing current to flow continuously. As a result, a large resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1, and a large amount of energy is accumulated. This accumulated energy flows as a load current between terminals T3 and T4 in the next mode 3. This causes the main circuit section 10 to perform a boost operation.

[0089] As described above, when the drive frequency f exceeds the first frequency fθs in an attempt to reduce the output of the current resonant DC / DC converter 1, it performs frequency phase shift conversion control with a phase shift amount θ equivalent to (f-fθs)×Kθ. During modes 2 and 4 of the frequency phase shift conversion control, the battery voltage V1 is not applied to the drive side, and the main circuit section 10 performs output suppression operation (step-down operation).

[0090] Furthermore, when the current-resonant DC / DC converter 1 attempts to increase its output and the drive frequency f falls below the second frequency fψs, it performs frequency boost conversion control with a boost amount ψ equivalent to (fψs-f)×Kψ. During the modes 1B and 3B periods of frequency boost conversion control, energy is accumulated due to the resonant current and converted into load current during the following modes 1 and 3 periods. In other words, the main circuit section 10 performs an output boost operation (voltage boost operation).

[0091] Since both the phase shift amount θ in frequency phase shift conversion control and the boost amount ψ in frequency boost conversion control are control variables correlated with the drive frequency f, the current resonant DC / DC converter 1 does not require switching to another control mode, and no countermeasures are needed for output changes associated with switching control modes, allowing control based solely on frequency control. Therefore, the current resonant DC / DC converter 1 makes it possible to handle a wide range of output voltages without adding elements or circuits, and without complicating the control software configuration.

[0092] Furthermore, in the current-resonant DC / DC converter 1, the control parameters (first frequency fθs, first gain Kθ, maximum phase shift amount θh, second frequency fψs, second gain Kψ, maximum boost amount ψh) can be changed depending on the input / output conditions (e.g., input voltage V1, input / output voltage ratio V1 / V2, or output power). This allows switching elements Q1 to Q4 to be switched on and off in a region where soft switching is possible during frequency-phase-shift conversion control, thereby improving the power conversion efficiency of the main circuit section 10. In addition, during frequency-boost conversion control, the slope of the boost amount ψ can be adjusted in conjunction with the battery voltage V1x, thus avoiding unstable control.

[0093] Up to this point, we have described the case where the main circuit unit 10 performs a discharge operation of the battery connected between terminals T1 and T2. However, when the main circuit unit 10 performs a battery charge operation, the secondary rectifier circuit 13 and secondary resonant circuit 14 become the drive-side circuits, and the primary switching circuit 11 and primary resonant circuit 12 become the rectifier-side circuits. During the charge operation, V1 becomes the output voltage and V2 becomes the input voltage. During the charge operation, the control unit 20 controls the drive-side secondary rectifier circuit 13 in the same way as the primary switching circuit 11 during the discharge operation, and during the charge operation, it controls the rectifier-side primary switching circuit 11 in the same way as the secondary rectifier circuit 13 during the discharge operation.

[0094] [First variation] Figure 11 shows a current-resonant DC / DC converter 1A according to a first modification of the first embodiment. The current-resonant DC / DC converter 1A consists of a main circuit section 10A and a control section 20A. The main circuit section 10A includes a transformer circuit Tr, a primary-side switching circuit 11, a primary-side resonant circuit 12, a secondary-side rectifier circuit 13A, and terminals T1 to T4.

[0095] The current-resonant DC / DC converter 1A is configured to perform unidirectional power transmission from terminals T1 and T2 (V1 side) to terminals T3 and T4 (V2 side). Compared to the first embodiment, the current-resonant DC / DC converter 1A differs in that it does not have a secondary resonant circuit 14, has a secondary rectifier circuit 13A instead of a secondary rectifier circuit 13, and has a control unit 20A instead of a control unit 20, but all other aspects are the same.

[0096] The secondary rectifier circuit 13A has the same configuration as the secondary rectifier circuit 13 of the first embodiment, except that the fourth leg consists only of diodes D7 and D8 connected in series.

[0097] The control unit 20A controls the switching elements Q1 to Q6 so that the main circuit unit 10A performs unidirectional power transmission from terminals T1 and T2 (V1 side) to terminals T3 and T4 (V2 side). In the first embodiment, the control during discharge operation was described with switching elements Q7 and Q8 turned off, so the control of the control unit 20A is the same as the control during discharge operation in the first embodiment. That is, the control unit 20A includes the configuration of the control unit 20 for performing control during discharge operation, and performs frequency modulation control, frequency phase shift conversion control, and frequency boost conversion control in the same way as the control unit 20.

[0098] [Second variation] Figure 12 shows a current-resonant DC / DC converter 1B according to a second modification of the first embodiment. The current-resonant DC / DC converter 1B consists of a main circuit section 10B and a control section 20B. The main circuit section 10B includes a transformer circuit Tr, a primary-side switching circuit 11, a primary-side resonant circuit 12, a secondary-side rectifier circuit 13B, and terminals T1 to T4.

[0099] The current-resonant DC / DC converter 1B is configured to perform unidirectional power transmission from terminals T1 and T2 (V1 side) to terminals T3 and T4 (V2 side). Compared to the first embodiment, the current-resonant DC / DC converter 1B differs in that it does not have a secondary resonant circuit 14, has a secondary rectifier circuit 13B instead of a secondary rectifier circuit 13, and has a control unit 20B instead of a control unit 20, but all other aspects are the same.

[0100] The secondary rectifier circuit 13B has the same configuration as the secondary rectifier circuit 13 of the first embodiment, except that the upper arm of the third leg consists only of diode D5 and the upper arm of the fourth leg consists only of diode D7.

[0101] The control unit 20B controls the switching elements Q1-Q4, Q6, and Q8 so that the main circuit unit 10B performs unidirectional power transmission from terminals T1 and T2 (V1 side) to terminals T3 and T4 (V2 side). The control unit 20B includes the configuration of the control unit 20 for controlling the discharge operation, and performs frequency modulation control, frequency phase shift conversion control, and frequency boost conversion control, similar to the control unit 20. However, it differs from the first embodiment in that it turns on the switching elements Q6 and Q8 during frequency boost conversion control.

[0102] Figure 13 shows the drive waveforms of switching elements Q1-Q4, Q6, and Q8 during frequency boost conversion control. Figure 14(A) shows the current flow in the main circuit section 10B during mode 1B in Figure 13, and Figure 14(B) shows the current flow in the main circuit section 10B during mode 3B in Figure 13. Note that the currents shown in Figure 14 are the resonant current and load current, while the excitation current and transient current are not shown.

[0103] During mode 1B in Figure 13, switching elements Q1, Q4, and Q6 are turned on, while switching elements Q2, Q3, and Q8 are turned off. As shown in Figure 14(A), the battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q1 and Q4. The rectifier side is short-circuited, and a current path is formed through switching element Q6 and diode D8, allowing current to flow continuously. As a result, a large resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1, and a large amount of energy is accumulated. This accumulated energy flows as a load current between terminals T3 and T4 in the next mode 1. This causes the main circuit section 10B to perform a voltage boost operation.

[0104] During mode 3B in Figure 13, switching elements Q2, Q3, and Q8 are turned on, while switching elements Q1, Q4, and Q6 are turned off. As shown in Figure 14(B), the battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q2 and Q3, in the opposite direction to mode 1B. The rectifier side is short-circuited, and a current path is formed through switching element Q8 and diode D6, allowing current to flow continuously. As a result, a large resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1, and a large amount of energy is accumulated. This accumulated energy flows as a load current between terminals T3 and T4 in the next mode 3. This causes the main circuit section 10B to perform a voltage boost operation.

[0105] [Third variation] Figure 15 shows a current-resonant DC / DC converter 1C as a third modified example of the first embodiment. The current-resonant DC / DC converter 1C consists of a main circuit section 10C and a control section 20C. The main circuit section 10C includes a transformer circuit Tr, a primary-side switching circuit 11, a primary-side resonant circuit 12, a secondary-side rectifier circuit 13C, and terminals T1 to T4.

[0106] The current-resonant DC / DC converter 1C is configured to perform unidirectional power transmission from terminals T1 and T2 (V1 side) to terminals T3 and T4 (V2 side). Compared to the first embodiment, the current-resonant DC / DC converter 1C differs in that it does not have a secondary resonant circuit 14, has a secondary rectifier circuit 13C instead of a secondary rectifier circuit 13, and has a control unit 20C instead of a control unit 20, but is otherwise common to both.

[0107] The secondary rectifier circuit 13C has the same configuration as the secondary rectifier circuit 13 of the first embodiment, except that diodes D5 to D8 are not connected to the current paths of each switching element Q5 to Q8.

[0108] The control unit 20C has the same configuration as the control unit 20 of the first embodiment, except that it performs frequency PWM conversion control instead of frequency phase shift conversion control. The control unit 20C performs frequency modulation control, frequency PWM conversion control, and frequency boost conversion control.

[0109] Figure 16 shows the drive waveforms of switching elements Q1 to Q8 during frequency PWM conversion control. Figure 17(A) shows the current flow in the main circuit section 10C during mode 1' of Figure 16, and Figure 17(B) shows the current flow in the main circuit section 10C during mode 2' of Figure 16. Furthermore, Figure 18(A) shows the current flow in the main circuit section 10C during mode 3' of Figure 16, and Figure 18(B) shows the current flow in the main circuit section 10C during mode 4' of Figure 16.

[0110] As shown in Figure 16, the drive waveforms of the switching elements Q1 and Q2 in the first leg and the drive waveforms of the switching elements Q3 and Q4 in the second leg are in phase (zero phase difference), and are common to the drive waveform during frequency modulation control in the first embodiment shown in Figure 5, except for the period To. The drive waveforms of the switching elements Q5 and Q6 in the third leg and the drive waveforms of the switching elements Q7 and Q8 in the fourth leg are in phase (zero phase difference), but the timing of going from Low to High (turn-on timing) is delayed by a phase difference θ' relative to the drive waveforms of switching elements Q1 to Q4.

[0111] During frequency PWM conversion control, the control unit 20C controls the pulse width of the drive waveform (PWM signal) of switching elements Q5 to Q8 by calculating the phase difference θ' using the drive frequency f, similar to the phase shift amount θ in the first embodiment. The drive frequency f is determined by the same process as in step S4 in Figure 3. The phase difference θ' is, for example, the value obtained by multiplying the difference between the drive frequency f and the first frequency (the frequency at which frequency PWM conversion control is started) by a predetermined first gain (however, the value is different from the first gain Kθ). Since the phase difference θ' increases uniformly with increasing drive frequency f, the pulse width decreases uniformly with increasing drive frequency f. Therefore, the period Tθ' of switching elements Q1 to Q4 decreases compared to the period To in Figure 5. Note that if the drive frequency f exceeds a predetermined maximum PWM drive frequency, the phase difference θ' may be fixed to a predetermined maximum phase difference.

[0112] During mode 1', as shown in Figure 17(A), a battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q1 and Q4. As a result, on the rectifier side, a load current flows between terminals T3 and T4 via switching elements Q5 and Q8. During the next mode 2' period, a voltage opposite to that in mode 1' is applied to the resonant coil Lr1 in the primary switching circuit 11, so the excitation current rapidly decays and a current opposite to that in mode 1' flows. However, in the secondary rectifier circuit 13C, switching elements Q5 to Q8 are turned off, so no current flows to the load side, as shown in Figure 17(B).

[0113] During mode 3', as shown in Figure 18(A), a battery voltage V1 is applied between terminals T1 and T2, and a resonant current flows through the resonant coil Lr1 and resonant capacitor Cr1 via switching elements Q2 and Q3, in the opposite direction to mode 1'. As a result, on the rectifier side, a load current flows between terminals T3 and T4 via switching elements Q6 and Q7. During the next mode 4' period, a voltage opposite to that in mode 3' is applied to the resonant coil Lr1 in the primary switching circuit 11, so the excitation current rapidly decays, and a current opposite to that in mode 3' flows. However, in the secondary rectifier circuit 13C, switching elements Q5 to Q8 are turned off, so as shown in Figure 18(B), no current flows to the load side.

[0114] Thus, in an attempt to reduce the output, the current resonant DC / DC converter 1C performs frequency PWM conversion control, which controls the pulse width of switching elements Q5 to Q8 by a phase difference θ' calculated using the drive frequency f, when the drive frequency f exceeds a preset first frequency. During modes 2' and 4' of the frequency PWM conversion control, the load current is limited on the rectifier side, and output suppression (step-down operation) is performed in the main circuit section 10C.

[0115] Since the phase difference θ' in frequency PWM conversion control is a control variable correlated with the drive frequency f, the current resonant DC / DC converter 1C does not require switching to a different control mode, nor does it require countermeasures for output changes associated with switching control modes, allowing control based solely on frequency control. Therefore, the current resonant DC / DC converter 1C makes it possible to handle a wide range of output voltages without adding elements or circuits, and without complicating the control software configuration.

[0116] In the current-resonant DC / DC converter 1C, when frequency boost conversion control is performed, switching elements Q7 and Q8 are off. In mode 1B of the first embodiment, switching element Q6 is turned on, and the secondary rectifier circuit 13 is short-circuited by the conduction of diode D8. In mode 3B of the first embodiment, switching element Q5 is turned on, and the secondary rectifier circuit 13 is short-circuited by the conduction of diode D7 to perform the boost operation. However, in the third modified example 3, since diodes D7 and D8 are not provided in the secondary rectifier circuit 13C, for example, switching element Q8 may be turned on in synchronization with switching element Q1 during the period corresponding to mode 1B of the first embodiment, and switching element Q7 may be turned on in synchronization with switching element Q3 during the period corresponding to mode 3B of the first embodiment to cause a similar current flow and perform the boost operation.

[0117] [Other variations] The embodiments and modified versions of the current-resonant DC / DC converter according to the present invention have been described above, but the present invention is not limited to the embodiments and modified versions described above.

[0118] For example, the current-resonant DC / DC converter 1 according to the first embodiment performs both frequency-phase-shift conversion control and frequency-boost conversion control, but it may also perform only one of the two. The same applies to the current-resonant DC / DC converters 1A and 1B according to the first and second modified examples. Furthermore, the current-resonant DC / DC converter 1C according to the third modified example performs both frequency-PWM conversion control and frequency-boost conversion control, but it may also perform only one of the two.

[0119] For output suppression conversion control related to output suppression operation (step-down operation), if a control variable calculated using the drive frequency f is used, control other than frequency phase shift conversion control or frequency PWM conversion control may be performed. For example, frequency intermittent conversion control that causes the switching element to perform intermittent operation (burst operation) may be performed. In the case of frequency intermittent conversion control, it is preferable to use the standby period during which the switching element does not perform on / off operation as the control variable, and to calculate this standby period using the drive frequency f and a predetermined gain.

[0120] Similarly, for output boost conversion control related to output boost operation (voltage boost operation), if the control variable is calculated using the drive frequency f, control other than frequency boost conversion control may be performed.

[0121] Regarding frequency phase shift conversion control, in the above embodiment, the drive circuit (primary side switching circuit 11 during discharge operation) was driven with a phase shift. However, this method is not limited to any other method as long as the phase shift control is performed using a control amount calculated using the drive frequency f. Similarly, regarding frequency boost conversion control, in the above embodiment, the secondary side rectifier circuit was short-circuited to perform a boost operation. However, this method is not limited to any other method as long as the frequency boost conversion control is performed using a control amount calculated using the drive frequency f. For example, a separate boost circuit may be added to the primary or secondary side circuit, and the short-circuit operation may be performed using the added boost circuit. Furthermore, regarding frequency PWM conversion control, in the third modified example, PWM control was performed using the secondary side rectifier circuit 13C. However, the primary side switching circuit 11 may be PWM controlled to control the resonant current period.

[0122] Regarding the configuration of the main circuit of the present invention, the above embodiments and modifications show examples of LLC or CLLC systems with a full bridge configuration of two parallel legs, but the invention is not limited to these. Any LLC or CLLC system with a single half-bridge configuration may be used, as long as it performs output suppression (downward voltage operation) or output boost (upward voltage operation) based on a control amount calculated using the drive frequency f. A 3-phase LLC or 3-phase CLLC system with three parallel legs may also be used, as may a multi-parallel multi-phase LLC or multi-phase CLLC system with four or more parallel legs may also be used.

[0123] In the case of a half-bridge CLLC system, for example, frequency PWM conversion control can be performed on the drive-side switching element using the pulse width of the PWM signal's on-period (the period during which resonant current flows) as the control variable, and frequency boost conversion control can be performed on the rectifier-side switching element using the on-period of the drive signal (the short-circuit time within the period during which load current flows), i.e., the boost amount, as the control variable. Each control variable is calculated using the drive frequency f.

[0124] In the case of a 3-phase CLLC system, for example, frequency PWM conversion control can be performed on the drive-side switching element using the pulse width of the PWM signal's on-period (the period during which resonant currents of each phase with a 120° phase shift flow) as the control variable, or frequency phase shift conversion control can be performed using the phase difference (phase shift amount) between the legs of the other two phases relative to the leg of one phase as the control variable. Similarly, frequency boost conversion control can be performed on the rectifier-side switching element using the on-period of the drive signal (short-circuit time during the period in which load currents of each phase flow), i.e., the boost amount, as the control variable. Each control variable is calculated using the drive frequency f.

[0125] In the case of a multiphase CLLC system, frequency PWM conversion control can be performed on the drive-side switching element using the pulse width of the PWM signal's on-period (the period during which the resonant current of each phase flows) as the control variable, and frequency boost conversion control can be performed on the rectifier-side switching element using the boost amount, i.e., the short-circuit time within the on-period (the period during which the load current of each phase flows) of the drive signal as the control variable. Each control variable is calculated using the drive frequency f. [Explanation of Symbols]

[0126] 1. 1A~1C Current Resonant DC / DC Converter 10, 10A~10C main circuit section 11 Primary switching circuit 12 Primary side resonant circuit 13, 13A~13C secondary rectifier circuit 14 Secondary side resonant circuit 20, 20A~20C Control Unit 30 Frequency Phase Shift Transformation Control Block 31 1st calculation section 32 First Multiplication Section 33. First clamp section 40 Frequency Boost Conversion Control Block 41 2nd calculation section 42 Second Multiplication Section 43 Second clamp section

Claims

1. Transformer circuit and, A primary side switching circuit provided on the primary side of the transformer circuit, comprising at least one primary side leg, and comprising a set of primary side switching elements in which the primary side legs are connected in series, A primary side resonant circuit, which includes a resonant coil and a resonant capacitor, is provided between the primary side leg and the transformer circuit, A secondary rectifier circuit provided on the secondary side of the transformer circuit, comprising at least one secondary side leg, the secondary side legs being connected in series with a set of rectifier means, the rectifier means comprising a diode or a secondary side switching element, A main circuit section comprising, A control unit that controls the primary switching element and / or the secondary switching element, A current resonant DC / DC converter including, The control unit, Frequency modulation control is used to control the output of the main circuit section using the drive frequency as a control variable when the drive frequency of the primary switching element is between a first frequency and a second frequency smaller than the first frequency. At least one of the following controls: an output suppression conversion control that reduces the output using a first control amount calculated using the drive frequency when the drive frequency exceeds the first frequency, and an output increase conversion control that increases the output using a second control amount calculated using the drive frequency when the drive frequency falls below the second frequency; Execute A current-resonant DC / DC converter characterized by the following features.

2. The output suppression conversion control described above is: Frequency PWM conversion control that controls the pulse width of the PWM signal to the primary-side switching element or the secondary-side switching element based on the first control variable, Frequency intermittent conversion control that controls the standby period during which the primary switching element does not perform on / off operation based on the first control variable, or This includes frequency phase shift conversion control that controls the amount of phase shift between the primary side legs based on the first controlled variable. The current resonant DC / DC converter according to feature 1.

3. The output boost conversion control is a frequency boost conversion control that short-circuits the secondary rectifier circuit by controlling the on-period of the secondary switching element of the secondary rectifier circuit during the period in which a resonant current flows through the primary resonant circuit, based on the second control amount. The current resonant DC / DC converter according to feature 1.

4. The first control variable is calculated by a calculation between a value including the drive frequency and a predetermined first gain. During the output suppression conversion control, the control unit varies the value of the first frequency and / or the first gain according to the input / output conditions relating to the input voltage, input / output voltage ratio, or output power of the main circuit. The current resonant DC / DC converter according to feature 1.

5. The second control variable is calculated by a calculation between a value including the drive frequency and a predetermined second gain. During the output boost conversion control, the control unit varies the value of the second frequency and / or the second gain according to the input / output conditions relating to the input voltage, input / output voltage ratio, or output power of the main circuit. The current resonant DC / DC converter according to feature 1.

6. During the output suppression conversion control, the control unit performs control such that the first control amount does not exceed a predetermined first maximum control amount. During the output increase conversion control, the control unit performs control such that the second control amount does not exceed a predetermined second maximum control amount. The current resonant DC / DC converter according to feature 1.

7. The main circuit section includes a resonant coil and a resonant capacitor, The primary-side switching circuit includes two primary-side legs, a first leg and a second leg, which are connected in parallel. The secondary rectifier circuit includes two secondary legs, a third leg and a fourth leg, connected in parallel, wherein the rectifier means of each leg consists of a secondary switching element, or only the rectifier means constituting the upper and lower arms of the fourth leg consists of a secondary switching element, or only the rectifier means constituting the lower arm of the third leg and the lower arm of the fourth leg consists of a secondary switching element, and a reverse-circuit diode is connected in parallel to the current path of the secondary switching element. The current resonant DC / DC converter according to feature 1.

8. The control unit performs frequency phase shift conversion control as the output suppression conversion control. During the frequency-phase-shift conversion control, the control unit determines the drive frequency by comparing the output value with the target value, and sets the phase shift amount as the difference between the drive frequency and the first frequency multiplied by a predetermined first gain, thereby creating a phase difference between the first leg and the second leg equal to the phase shift amount. The current resonant DC / DC converter according to feature 7.

9. The control unit performs frequency boost conversion control as the output boost conversion control, During the frequency boost conversion control, the control unit determines the drive frequency by comparing the output value of the output with a target value, and sets the boost amount to a value obtained by multiplying the difference between the second frequency and the drive frequency by a predetermined second gain, thereby giving the secondary switching element an on period equal to the boost amount. The current resonant DC / DC converter according to feature 7.

10. The main circuit section includes a resonant coil and a resonant capacitor, The primary-side switching circuit includes two primary-side legs, a first leg and a second leg, which are connected in parallel. The secondary rectifier circuit includes two secondary legs consisting of a third leg and a fourth leg connected in parallel, and the rectifier means of each leg consists of the secondary switching element. The control unit performs frequency PWM conversion control as the output suppression conversion control. During the frequency PWM conversion control, the control unit determines the drive frequency by comparing the output value with the target value, calculates the phase difference which is the first control quantity using the drive frequency, and shifts the on-timing of the secondary switching element relative to the on-timing of the primary switching element by the phase difference. The current resonant DC / DC converter according to feature 1.

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