Control device for a 3-phase AC motor

Abstract

To provide a control device for a three-phase AC motor comprising a q-axis current controller capable of ensuring high command follow-up performance equal to PI control while maintaining high disturbance suppression performance and stability that I-P control has.SOLUTION: A control device 1 for controlling a current of a three-phase AC motor through d-q axis control comprises a q-axis current controller 2 which defines an acceleration torque command value τf calculated on the basis of a position command value X as one of input and outputs a controller output voltage Δeq to be added to a q-axis voltage command value vq*. The q-axis current controller 2 calculates an acceleration q-axis current command value iqf* which is a current command value corresponding to the acceleration torque command value τf on the basis of the acceleration torque command value τf and adds a value, which is obtained by multiplying the acceleration q-axis current command value iqf* by an IP coefficient KIP indicating a switching ratio of PI control and I-P control and a q-axis proportional gain Gqp, to the controller output voltage Δeq.SELECTED DRAWING: Figure 2

Description

[Technical Field]

[0001] This specification discloses a control device for controlling a three-phase AC motor (e.g., a synchronous motor or an induction motor) to control at least one of the speed, position, and transmitted torque of an axis (e.g., a feed axis or spindle) of an NC machine tool or other control object in general industrial machinery. [Background technology]

[0002] Generally, control devices used for axis control in NC machine tools employ a three-phase AC motor (hereinafter referred to as "AC motor" or "motor") as the actuator, and achieve sufficient control stability and high position command tracking performance by configuring multiple control loops for position, speed, and current in response to position command values ​​output from a higher-level device.

[0003] Figure 9 is a block diagram showing an example of a conventional control device 200 using a three-phase AC motor as the drive motor. The conventional control device 200 will be described below. A position command value X is output to the conventional control device 200 from a higher-level device (not shown). The rotation angle θm of the motor 300, output from the position detector 301 connected to the motor 300, is a position detection value indicating the position of the load 302 (such as a drive table) connected to and driven by the motor. This is subtracted from the position command value X by the subtractor 50, and its output is the position deviation DIF.

[0004] The position deviation DIF is amplified by the position loop gain Kp times in the position deviation amplifier 52. Meanwhile, the position command value X is differentiated with respect to time in the differentiator 51 (where s corresponds to the Laplace transform operator), and the sub-velocity command value ω f The output is as follows: The adder 53 outputs the sub-speed command value ω f The output of the position deviation amplifier 52 is added to obtain the final speed command value ω m * The following will be output.

[0005] Differentiator 54 calculates the velocity detection value ω by differentiating the position detection value θm with respect to time. mis output. In the subtracter 55, the speed command value ω m * is subtracted from the speed detection value ω m . The speed deviation Δω m which is the output of the subtracter 55 is PI (Proportional Integral) amplified by the speed controller 57. On the other hand, the sub speed command value ω f is differentiated with respect to time by the differentiator 56, and further multiplied by J to obtain the motor acceleration torque command value τ f . Here, J is the total moment of inertia, that is, the motor inertia moment + the load inertia moment. The adder 58 adds the output Δτ c of the speed controller 57 (hereinafter referred to as "controller output torque Δτ c ") and the acceleration torque command value τ f to obtain the final motor torque command value τ c * .

[0006] In the case of a permanent magnet synchronous motor (generally classified into a surface magnet synchronous motor (SPMSM) and an interior magnet synchronous motor (IPMSM)) or a reluctance synchronous motor (SynRM), the d-q axis control arithmetic unit 61 calculates and outputs the q-axis current command value i c * and the d-axis current command value i m from the N-τ (speed-torque) characteristic of the motor and the speed detection value ω q * with respect to the motor torque command value τ d * .

[0007] On the other hand, in the case of an induction motor (IM), the d-q axis control arithmetic unit 61 calculates the d-axis 2nd order magnetic flux (φdr / M) from the d-axis current detection value i m and the field weakening characteristic of the induction motor, and calculates and outputs the d-axis current command value i d * from the d-axis current detection value i d . Then, the q-axis current command value i c * is calculated and output from the torque command value τ q * and the d-axis 2nd order magnetic flux (φdr / M). Further, the q-axis current command value i d is calculated from the d-axis current detection value iq * From this, the slip angular velocity ωs is calculated, and the electrical angular velocity ω described later is calculated. re By adding these together (not shown in the diagram), the current angular velocity ω and its time integral, the current phase angle θ, are calculated and output.

[0008] Furthermore, the d-axis voltage feedforward V is output from the dq-axis control calculation unit 61. dff And, q-axis voltage feedforward V qff This consists of decoupling voltage compensation values ​​between the d-axis and q-axis (not shown) and is used to improve the current control response.

[0009] The position detection value θm is multiplied by the motor counter pole number p in the multiplier 59 to obtain the electrical angle θre. The differentiator 60 differentiates the electrical angle θre with respect to time to output the electrical angular velocity ωre. (Normally, in a synchronous motor, the electrical angular velocity ωre = current angular velocity ω.) The motor's U-phase current i u And, V-phase current i v This is detected by the U-phase current detection circuit 70 and the V-phase current detection circuit 71.

[0010] Furthermore, W phase current i w is, i w =-(i u +i v It can be calculated as follows: 3-phase current is i u +i v +i w Since it becomes =0, typically two of the three phases are detected, and the remaining one phase is calculated. The 3-phase to dq converter 66 has a U-phase current i u and V-phase current i v And from the electrical angle θre (current phase angle θ in the case of IM), the d-axis current detection value i is obtained by coordinate transformation. d And, the q-axis current detection value i q The result is calculated and output.

[0011] The subtractor 62 controls the d-axis current command value i d * From the d-axis current detection value i d Subtracting this, the d-axis current error Δi d The d-axis current controller 63 calculates the d-axis current error Δi.d An error amplifier that amplifies Δe using PI (proportional-integral) amplification, d This is the output voltage of the d-axis current controller 63.

[0012] The subtractor 64 controls the q-axis current command value i q * From the q-axis current detection value i q Subtracting this, the q-axis current error Δi q The q-axis current controller 65 calculates the q-axis current error Δi. q An error amplifier that amplifies Δe q This is the output voltage of the q-axis current controller 65 (hereinafter referred to as "controller output voltage Δe"). q It is called "..."

[0013] The operation of the q-axis current controller 65 as an error amplifier is generally expressed by equations (1) and (2).

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[0014] Figure 10 shows another configuration example of the q-axis current controller 65. q-axis current error Δi q (1-K) is at amplifier 80. IP After being amplified by a factor of ) then proportional gain G is achieved by proportional amplifier 81. qp It is amplified by two. On the other hand, the q-axis current error Δi q The integral gain G qi The current is integrally amplified by the integral amplifier 82 and added to the output of the proportional amplifier 81 by the adder 83. q-axis current detection value i q At amplifier 84, K IP After doubling the amplification, proportional gain G is achieved with proportional amplifier 85. qp It is amplified by a factor of two. The subtractor 86 subtracts the output of the proportional amplifier 85 from the output of the adder 83, and the controller output voltage Δe q Outputs.

[0015] The operation of the q-axis current controller 65 shown in Figure 10 can be expressed by equation (3). [Numerical] Here, the IP coefficient K IP is a variable that takes a value in the range of 0 ≤ K IP ≤ 1 according to the control state. When K IP = 0, it is PI control, and when K IP = 1, it is I-P control.

[0016] The adder 67 adds the output voltage Δe of the d-axis current controller 63 d and the d-axis voltage feedforward V dff to output the d-axis voltage command value v d * Similarly, the adder 68 adds the controller output voltage Δe of the q-axis current controller 65 q and the q-axis voltage feedforward V qff to output the q-axis voltage command value v q *

[0017] The d-q → 3-phase converter 69 converts the d-axis voltage command value v d * and the q-axis voltage command value v q * and the electrical angle θre (in the case of IM, the current phase angle θ) into the voltage command values of each phase of U, V, and W by coordinate transformation, and amplifies the power with a PWM inverter (not shown) to output the phase voltages (v u , v v , v w ) of the motor drive. Each output phase voltage is applied to each phase of the motor to generate each phase current.

[0018] The above is an example of the conventional control device 200, and its configuration has been described. However, in order to control the position detection value θm of the load 302, which is the control target, with high precision according to the position command value X, it is necessary to set the control performance of the q-axis current controller for torque control, which is the smallest loop, especially high in the current control loop.

[0019] Hereinafter, the control performance of the q-axis current control loop will be described using FIG. 10. Generally, PI control (K​IP When (at = 0), it has high command following performance and excellent responsiveness, so it is suitable for servo applications. However, when the control gains (G qp and G qi ) are increased, control oscillation is likely to occur and stability decreases.

[0020] [[ID=�]]On the other hand, in I-P control (K IP = 1), it has high stability and does not easily generate control oscillation even when the control gain is increased in order to suppress voltage disturbance. However, its command following performance is low and it is suitable for regulator applications. That is, in the control device, sufficient command following performance cannot be achieved.

[0021] Also, in regulator applications, the q-axis current error Δi q is small, and in servo applications, Δi q tends to be large. Therefore, according to the magnitude of the q-axis current error Δi q |Δi q |, when |Δi q | → small (0), K IP → 1, and when |Δi q | → large, K IP → 0, the IP coefficient K IP may be variably controlled. However, in recent control devices, not only command following performance but also high disturbance suppression performance and stability are increasingly required even during acceleration and deceleration when |Δi q | becomes large.

Summary of the Invention

Problems to be Solved by the Invention

[0022] In a control device for torque, speed, and position of a feed axis or a main axis that uses d-q axis control for current control of a three-phase AC motor, to provide a control device for a three-phase AC motor equipped with a q-axis current controller that can ensure high command following performance equivalent to that of PI control while maintaining the high disturbance suppression performance and stability of I-P control.

Means for Solving the Problems

[0023] Acceleration torque command value τ for a 3-phase AC motor f Therefore, the acceleration q-axis current command value i qf * Calculate and, as proportional compensation, K IP ·i qf * ·G qp The output voltage Δe of the q-axis current controller q Add it to the total. [Effects of the Invention]

[0024] In the control device for a three-phase AC motor according to the present invention, the IP coefficient K IP Under the condition i = 1, the acceleration q-axis current command value i qf * The difference between the q-axis current detection value iq and (i qf * -i q ) with a q-axis proportional gain G qp By amplifying it and using it as the proportional component of the q-axis current controller, the high disturbance suppression performance and stability of IP control can be maintained, and the q-axis current detection value i q The acceleration q-axis current command value i qf * This can provide high command tracking performance.

[0025] Furthermore, the q-axis current command value i q * The velocity detection value ω included m The mechanical vibration component mixed in is G qp Therefore, it is not amplified. As a result, the response gain of the q-axis current controller with respect to the mechanical resonant frequency can be significantly reduced compared to PI control, and consequently, the vibration suppression filters such as notch filters inserted into the speed control loop can be reduced. [Brief explanation of the drawing]

[0026] [Figure 1] This is a block diagram showing an example configuration of a control device using a three-phase AC motor as the drive motor according to the present invention. [Figure 2] This is a block diagram showing an example configuration of the q-axis current controller of the present invention. [Figure 3]This is a block diagram showing an example configuration of the coefficient / gain setting device of the present invention. [Figure 4] This is a block diagram model for determining the frequency characteristics of the q-axis current controller of the present invention. [Figure 5] This graph shows an example of the frequency characteristics of the q-axis current controller of the present invention at KIP=0. [Figure 6] This graph shows an example of the frequency characteristics of the q-axis current controller of the present invention at KIP=1. [Figure 7] This is a block diagram showing a second configuration example of a control device using a three-phase AC motor as the drive motor according to the present invention. [Figure 8] This is a block diagram showing an example configuration of the speed controller of the present invention. [Figure 9] This is a block diagram showing an example configuration of a control device that uses a conventional three-phase AC motor as the drive motor. [Figure 10] This is a block diagram showing an example configuration of a conventional q-axis current controller. [Modes for carrying out the invention]

[0027] The best mode for carrying out the present invention will be described below using examples (hereinafter referred to as embodiments). Figure 1 is a block diagram showing an example of the configuration outline of a control device 1 for a three-phase AC motor according to the present invention. The control device 1 is, for example, a computer having a processor and memory. The parts of the control device 1, excluding the q-axis current controller 2, are the same as those in the conventional examples described above, and are therefore given the same numbers and their descriptions are omitted.

[0028] The q-axis current controller 2 controls the q-axis current command value i, which is the output of the dq-axis control calculation unit 61. q * And the acceleration torque command value τ f And, the q-axis current detection value i q With and as inputs, the controller output voltage Δe q The output is shown. Figure 2 is a block diagram showing an example configuration of the q-axis current controller 2 of the present invention. Hereafter, the operation of the q-axis current controller 2 in the present invention will be explained using Figure 2.

[0029] The q-axis current controller 2 has a different τ compared to the conventional q-axis current controller 65. f →i qf * It comprises a converter 3, an amplifier 4, and an adder 5. f →i qf * The converter 3 receives the acceleration torque command value τ f The input is the acceleration q-axis current command value i qf * It outputs the acceleration torque command value τ. f This is the value obtained by multiplying the position command value X by the total moment of inertia J, which is the second time derivative of the position command value X. Therefore, the acceleration torque command value τ f This value is independent of the position detection value θm.

[0030] Amplifier 4 controls the acceleration q-axis current command value i qf * to K IP It is amplified by two. Adder 5 controls the q-axis current error Δi in amplifier 80. q (1-K IP The output amplified by ) times is added to the output value of amplifier 4. The output of adder 5 is then used to obtain the proportional gain G of proportional amplifier 81. qp It is amplified by two. The output of the proportional amplifier 81 is added to or subtracted from other values ​​in the adder 83 and subtractor 86, and the controller output Δe q Therefore, in the q-axis current controller 2 of Figure 2, the acceleration q-axis current command value i qf * In addition, the IP coefficient K IP And the q-axis proportional gain G qp The value obtained by multiplying by the controller output voltage Δe q It can be said that this is being added to the total.

[0031] Here, τ f →i qf *The converter 3 uses different conversion formulas depending on the type of motor. For example, it is calculated using formula (4) for induction motors (IM) and formula (5) for surface magnet synchronous motors (SPMSM). Although similar conversion formulas exist for embedded magnet synchronous motors (IPMSM) and reluctance synchronous motors (SynRM) based on their torque generation principles, a detailed description is omitted. i qf * =τ f / {p·M0(φdr / M)} ; where p is the number of pole pairs of motors and M0 is the mutual inductance ·····(4) i qf * =τ f / (p·Φf) ; Permanent magnet magnetic flux φf ·····(5)

[0032] The input-output relationship of the q-axis current controller 2 can be expressed as equation (6).

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[0033] Equation (6) can be simplified to equation (7).

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[0034] q-axis current command value i q * This includes the velocity detection value ω of the velocity loop. m It contains resonant mechanical vibration components that have been introduced from there. q * The acceleration q-axis current command value i qf * Then, it is divided into the q-axis current command value portion that includes the mechanical vibration component (hereinafter referred to as "vibration component-containing q-axis current command value ivb") and defined by equation (8).

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[0035] Equation (7) can be expressed as equation (9) using the definition in equation (8).

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[0036] A major feature of the q-axis current controller 2 of the present invention is the (1-K) of the q-axis current command value ivb containing the oscillation component. IP ) times, proportional amplification (G qp This refers to being subject to )

[0037] From equation (9), the output of the q-axis current controller 2 of the present invention can be summarized as follows.

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[0038] Next, the configuration and operation of the coefficient / gain setter 10 shown in Figure 2 will be described. Figure 3 is a block diagram showing an example configuration of the coefficient / gain setter 10 of the present invention. In this example, the coefficient / gain setter 10 has a q-axis current error Δi q This will be the input. K IP0 The setting device 11 has a q-axis current error Δi q The absolute value of |Δi q | is the input, |Δi q |According to this, the initial value of the IP coefficient K IP0 Outputs.

[0039] The q-axis current controller 2 controls |Δi q It operates with the goal of |≈0. Therefore, K IP0 In the setting device 11, |Δi q If | is small, the initial IP coefficient K is set to improve disturbance suppression performance and stability. IP0 →1 is output. Conversely, |Δi q If | is large, aim to improve tracking performance, |Δi q In order to reduce |, K IP0 →0 is output.

[0040] K in this example IP0 In the setting device 11, |Δi qWhen the direction of the numerical change (increase or decrease) of | suddenly reverses, the initial value of the IP coefficient K IP0 To facilitate the smooth handling of sudden changes, the horizontal axis |Δi q |Above, we have four inflection points (er_minl≦er_minr≦er_maxl≦er_maxr), |Δi q The graph exhibits hysteresis characteristics in both the increasing and decreasing directions.

[0041] IP coefficient K IP0 is, K IP Filter 12 allows the time constant T to be filtered. KIP Filtered by a first-order lag process, the final IP coefficient K IP Outputs the time constant T. KIP This can take a few milliseconds to several tens of milliseconds, but it may be variable depending on the steepness of the position command value X commanded to the control device 1.

[0042] The q-axis gain setter 13 is set to the IP coefficient K IP (0≦K IP Depending on ≤1), the q-axis proportional gain G qp and q-axis integral gain G qi The result is calculated and output. In this example, the q-axis gain setter 13 is set to G qp and G qi In contrast, K is on an upward trend. IP - Gain characteristics are given, K IP = G at 0 o'clock qp_min ,G qi_min And, K IP = G at 1 o'clock qp_max ,G qi_max Set this in advance.

[0043] Figure 4 is a block diagram model for evaluating the characteristics of the q-axis current controller of the present invention. In Figure 4, the q-axis current controller 100 has a different relationship with respect to the q-axis current controller 2 in Figure 2. f →i qf * This is the version with converter 3 and coefficient / gain setter 10 removed. The controller output voltage Δe of the q-axis current controller 100. q In this case, voltage disturbance v rIPThis is added by the adder 101 and becomes the input voltage of the target plant model 102, which simulates the q-axis motor winding. The output of the target plant model 102 is the q-axis current detection value i q This is then fed back to the q-axis current controller 100.

[0044] Figures 5 and 6 are graphs showing the frequency characteristics for the block diagram model in Figure 4. Figure 5 shows the IP coefficient K. IP to K IP Command response when =0 is fixed (i q / i q * ) and disturbance suppression (i q / v rIP This is the frequency response of ). This example is equivalent to the frequency response obtained by PI control.

[0045] Figure 6 shows the IP coefficient K IP to K IP Command response when =1 is fixed (i q / i q * ) and disturbance suppression (i q / v rIP This is the frequency characteristic of ). In this case, the q-axis current command value i q * From this perspective, it becomes IP control, ensuring high stability, and thus the q-axis integral gain G qi You can set it to be large. Specifically, G qi_max =4·G qi_min And, K in Figure 5 IP = 4 times G compared to 0 o'clock qi This is set. Furthermore, the q-axis proportional gain G qp Regarding G qp_max =G qp_min Therefore, Figures 5 and 6 represent the same proportional gain.

[0046] In this invention, the proportional component of the q-axis current controller 2 is the acceleration q-axis current command value i qf * and q-axis current detection value i q The difference (i qf * -i q ) with a q-axis proportional gain G qp By amplifying and adding, KIP →Even if it becomes 1, i qf * For this purpose, it has high command tracking performance equivalent to PI control. Therefore, it is always K IP It can operate with approximately 1.

[0047] Figure 6(K) shows the control state according to the present invention. IP =1 o'clock) then 4 times G qi By setting this, disturbance suppression (i q / v rIP Regarding ), see Figure 5(K IP Compared to the 0 o'clock position, the disturbance response in the frequency band of several tens of Hz or less is reduced by -10 dB to -12 dB, meaning that the voltage disturbance suppression performance has improved.

[0048] In the PWM inverter (not shown) included in the dq→3-phase converter 69 in Figure 1, a dead time is provided to avoid short-circuit failure of the two power switching elements connecting the DC buses. When the current direction of each phase U, V, and W reverses, discontinuous voltage fluctuations caused by the dead time are applied to the motor windings. The improved voltage disturbance suppression performance according to the present invention can reduce fluctuations in the q-axis current due to the effects of the dead time.

[0049] On the other hand, Figure 6(K IP Command response (i = 1 o'clock) q / i q * ) is IP controlled, therefore 4 times G qi Even if this setting is enabled, stability will not be compromised, as shown in Figure 5(K IP Compared to the previous configuration (at 0 o'clock), it achieves a steep drop-off (cutoff) characteristic of -40 dB / dec in frequency bands of several hundred Hz or higher, such as -2 dB at 500 Hz, -8 dB at 1000 Hz, and -14 dB at 2000 Hz.

[0050] The q-axis current control loop functions as a torque control loop. Therefore, the loop gain of the speed control loop to which the present invention is applied reflects the high cutoff characteristics in the high-frequency range of the q-axis current control according to the present invention. As a result, mechanical resonance vibration suppression filters such as multi-stage notch filters that were inserted into conventional speed control loops can be reduced, and a simpler speed control loop can be constructed.

[0051] Figure 7 is a block diagram showing a second configuration example of the control device 20 for a three-phase AC motor according to the present invention. Since the parts excluding the speed controller 21 are the same as those in the example of the present invention shown in Figure 1 described above, the same numbers are used and their explanations are omitted.

[0052] The speed controller 21 according to the present invention uses a speed command value ω m * And the sub-speed command value ω f And, the speed detection value ω m With and as inputs, the controller output torque Δτ c The output is shown. Figure 8 is a block diagram showing an example configuration of the speed controller 21 of the present invention. Hereafter, the operation of the speed controller 21 in the present invention will be explained using Figure 8.

[0053] The speed controller 21 of the present invention has a configuration in which the q-axis current controller 2 of the present invention, as described in Figure 2, is replaced for speed control purposes, but the signal names and other aspects are different, so a general explanation will be given below. The speed controller 21 of the present invention shown in Figure 8 has a sub-speed command value ω f Amplifier 23 to K IP The value amplified by two, and Δω m (1-K) IP The value amplified by ) times and is added in adder 24. The output of adder 24 is used in proportional amplifier 25 to gain speed proportional gain G vp It is amplified by two. In other words, the speed controller 21 in Figure 8 controls the sub-speed command value ω f In addition, the IP coefficient K IP And, velocity-proportional gain G vp The value obtained by multiplying by Δτ is the controller output torque. c It is being added to.

[0054] On the other hand, the speed error Δωm This is the velocity integral gain G vi The signal is integrally amplified by the integral amplifier 26 and added to the output of the proportional amplifier 25 by the adder 27. Speed ​​detection value ω m At amplifier 28, K IP After being amplified twice, the proportional gain G is achieved by the proportional amplifier 29. vp It is amplified by two. The subtractor 30 subtracts the output of the proportional amplifier 29 from the output of the adder 27, and the controller output torque Δτ c Outputs.

[0055] The input-output relationship of the speed controller 21 can be expressed as equation (11).

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[0056] From equation (11), the output of the speed controller 21 of the present invention can be summarized as follows.

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[0057] Next, the configuration and operation of the coefficient / gain setter 31 shown in Figure 8 will be explained. The configuration of the coefficient / gain setter 31 is a direct replacement of the configuration of the coefficient / gain setter 10 shown in Figure 3 for speed control purposes, with the input being the speed error Δω m The absolute value of |Δω m |According to this, the initial value of the IP coefficient K IP0 Set K IP0 Setting device and K IP Filter and K IP Accordingly, the proportional gain G of the speed controller vp and integral gain G vi This configuration (not shown) consists of speed gain setters, each set to have an upward slope, connected in series.

[0058] The speed controller 21 is |Δω m It operates with the goal of |≈0. Therefore, as an example of the operation of the speed controller 21, similar to the q-axis current controller 2 of the present invention, |Δω mIf | is small, the IP coefficient K is used to aim for improved disturbance suppression performance and stability. IP →Operates in one direction, with an integral gain G vi |Δω m If | is large, the aim is to improve tracking performance, |Δω m | to decrease, K IP →Operates in the direction of 0, with integral gain G vi The value will decrease.

[0059] In the second configuration example of the present invention, the proportional component of the speed controller 21 is set to the speed command value ω f and velocity detection value ω m The difference (ω f -ω m ) proportional gain G vp By amplifying and adding, K IP →Even if it becomes 1, ω f For this purpose, it has high command tracking performance equivalent to PI control. Therefore, it is always K IP It can operate at approximately 1, improving disturbance suppression performance.

[0060] Since disturbances in speed controllers become torque disturbances, improving disturbance suppression performance can reduce speed impact drop due to load torque and improve speed regulator performance required for processes such as skiving. [Explanation of symbols]

[0061] 1 Control device (Example 1 of the invention), 2 q-axis current controller (Example 1 of the invention), 3 τf→i qf * Converter (Example 1 of Invention), 4 Amplifier (Example 1 of Invention), 5 Adder (Example 1 of Invention), 10 Coefficient / Gain Setter (Example 1 of Invention), 11 K IP0 Setting device (Example 1 of invention), 12 K IPFilter (Example 1 of Invention), 13 q-axis gain setter (Example 1 of Invention), 20 control device (Example 2 of Invention), 21 speed controller (Example 2 of Invention), 22 amplifier (Example 2 of Invention), 23 amplifier (Example 2 of Invention), 24 adder (Example 2 of Invention), 25 proportional amplifier (Example 2 of Invention), 26 integral amplifier (Example 2 of Invention), 27 adder (Example 2 of Invention), 28 amplifier (Example 2 of Invention), 29 proportional amplifier (Example 2 of Invention), 30 subtractor (Example 2 of Invention), 31 coefficient / gain setter (Example 2 of Invention), 50 subtractor, 51 differentiator, 52 position deviation amplifier, 53 adder, 54 differentiator, 55 subtractor, 56 differentiator, 57 speed controller (conventional), 58 adder, 59 multiplier, 60 differentiator, 61 dq axis control calculation unit, 62 Subtractor, 63 d-axis current controller, 64 subtractor, 65 q-axis current controller (conventional), 66 3-phase to dq converter, 67 adder, 68 adder, 69 dq to 3-phase converter, 70 U-phase current detection circuit, 71 V-phase current detection circuit, 80 amplifier, 81 proportional amplifier, 82 integral amplifier, 83 adder, 84 amplifier, 85 proportional amplifier, 86 subtractor, 100 q-axis current controller (block diagram model), 101 adder (block diagram model), 102 target plant model (block diagram model), 200 position control device (conventional), 300 motor, 301 position detector, 302 load

Claims

1. A control device for controlling the current of a three-phase AC motor by d-q axis control, The system includes a q-axis current controller that takes an acceleration torque command value calculated based on a position command value as one of its inputs and outputs a controller output voltage that is added to the q-axis voltage command value. The q-axis current controller is, Based on the acceleration torque command value, the acceleration q-axis current command value, which is the current command value corresponding to the acceleration torque command value, is calculated. The controller output voltage is added to the value obtained by multiplying the acceleration q-axis current command value by the IP coefficient, which indicates the switching ratio between PI control and I-P control, and the q-axis proportional gain. A control device for a three-phase AC motor, characterized by the following features.

2. A control device for a three-phase AC motor according to claim 1, The aforementioned IP coefficient is K ＩＰ In that case, The q-axis current controller further, The q-axis current error, which is the deviation between the q-axis current command value and the q-axis current detection value, is (1 - K ＩＰ The value multiplied by the q-axis proportional gain is amplified, and the amplified value is added to the controller output voltage. The q-axis current detection value K ＩＰ The doubled value is amplified by the q-axis proportional gain, and the amplified value is subtracted from the controller output voltage. A control device for a three-phase AC motor, characterized by the following features.

3. A control device for a three-phase AC motor according to claim 1, The q-axis current controller further, The initial value of the IP coefficient is determined from the absolute value of the q-axis current error, which is the deviation between the q-axis current command value and the q-axis current detection value. The initial value of the IP coefficient is subjected to a first-order lag process to output the IP coefficient. Based on the output IP coefficient, the q-axis proportional gain and q-axis integral gain are output. A control device for a three-phase AC motor, characterized by the following features.

4. A control device for a three-phase AC motor according to any one of claims 1 to 3, further, The speed controller includes a sub-speed command value obtained by time differentiation of the aforementioned position command value as one of its inputs, and outputs a controller output torque which is added to the torque command value. The speed controller is, The value obtained by multiplying the sub-speed command value by the IP coefficient and the speed-proportional gain is added to the controller output torque. A control device for a three-phase AC motor, characterized by the following features.

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