Synchronous machine control device

Abstract

The purpose of the present invention is to provide a synchronous machine control device that inhibits overcurrent or torque fluctuation when having switched from a three-phase open or short-circuit mode to a torque mode (PWM mode). In order to achieve the purpose, a synchronous machine control device according to the present invention controls a power converter that supplies power to a synchronous machine, the synchronous machine control device comprising: a first magnetic flux command calculation unit that calculates a first magnetic flux command value from a current command value of the synchronous machine; a magnetic flux estimation unit that estimates a magnetic flux value of the synchronous machine from a current detection value of the synchronous machine; and a second magnetic flux command calculation unit that uses an integration controller to calculate a second magnetic flux command value such that the first magnetic flux command value and the magnetic flux value match each other, wherein, when the power converter switches from a three-phase short-circuit to PWM control, a d-axis side of the integration controller is initialized with a d-axis magnetic flux value, and a q-axis side of the integration controller is initialized with a q-axis magnetic flux value.

Description

Technical Field

[0001] The present invention relates to a synchronous machine control device for driving a synchronous machine such as a synchronous motor.

Background Art

[0002] In order to miniaturize a synchronous motor, high performance such as high-speed rotation and high magnetic flux density of the motor has been advanced. Particularly, in electric vehicles such as electric automobiles, since the weight of the motor affects the power consumption, this tendency is remarkable.

[0003] As a control device capable of high performance of a motor, the control device described in Patent Document 1 is known. In the control device described in Patent Document 1, a first magnetic flux command value is calculated from a current command value of a synchronous machine, a magnetic flux value of the synchronous machine is estimated from a current detection value of the synchronous machine, and a voltage command value of a power converter is created so that the first magnetic flux command value and the magnetic flux value match.

Prior Art Documents

Patent Documents

[0004]

Patent Document 1

Summary of the Invention

Problems to be Solved by the Invention

[0005] In the control device described in Patent Document 1, in a PI controller that calculates a second d-axis magnetic flux command value, a difference between a first d-axis magnetic flux command value and a d-axis magnetic flux estimated value is calculated by an adder. Further, the difference calculation value is integrated by an integrator, and an integration gain (KI) is multiplied by the integration value. The difference calculation value multiplied by a proportional gain 97 and the integration value multiplied by the integration gain are added by an adder 99, and a second d-axis magnetic flux command value is calculated. Also, a second q-axis magnetic flux command value is calculated in the same manner.

[0006] In this case, when switching the power converter from three-phase open or short-circuit mode to torque mode (PWM mode), if the initial value of the integrator is incorrect, a difference may occur between the output voltage of the power converter and the voltage of the synchronous machine during the PWM mode switch, potentially causing overcurrent or torque fluctuations.

[0007] This invention has been made in view of these problems, and aims to provide a synchronous machine control device that suppresses overcurrent and torque fluctuations when switching from a three-phase open or short-circuit mode to a torque mode (PWM mode). [Means for solving the problem]

[0008] To solve the above problems, the synchronous machine control device according to the present invention is a synchronous machine control device that controls a power converter that supplies power to a synchronous machine, The aforementioned A first magnetic flux command calculation unit calculates the first magnetic flux command value from the current command value of the synchronous machine, The aforementioned From the current detection value of the synchronous machine The aforementioned Synchronous machine d-axis and q-axis A magnetic flux estimation unit that estimates the magnetic flux value, A voltage calculation unit that creates a voltage command value for the power converter such that the first magnetic flux command value and the magnetic flux value match, Using an integral controller, The aforementioned First magnetic flux command value and The aforementioned So that the magnetic flux values ​​match d-axis side and q-axis side The second magnetic flux command value each It comprises a second magnetic flux command calculation unit that performs calculations, The aforementioned The integral controller is, The aforementioned Power converter The drive state When switching, The aforementioned The d-axis side is the d-axis magnetic flux value. The aforementioned The q-axis side is initialized with the q-axis magnetic flux value. Furthermore, the value initialized differs depending on the drive state before the switchover; when the drive state before the switchover is three-phase open, the d-axis side is initialized with magnetic flux. . [Effects of the Invention]

[0009] According to the present invention, it is possible to suppress overcurrent and torque fluctuations when switching from a three-phase open or short-circuit mode to a torque mode (PWM mode).

[0010] Other issues, configurations, and effects not mentioned above will be clarified by the following description of the embodiments. [Brief explanation of the drawing]

[0011] [Figure 1] This is a block diagram showing the functional configuration of a synchronous machine control device according to an embodiment. [Figure 2] This is a block diagram showing the functional configuration of the PI controller in the second dq-axis magnetic flux command calculation unit 25. [Figure 3] The configuration of the voltage vector calculation unit 19, which is constructed based on the inverse model represented by equation (1), is shown. [Figure 4] An example of the relationship between magnetic flux and electric current is shown. [Figure 5] A block diagram showing the configuration of an integrator according to the embodiment. [Figure 6] This is a block diagram showing the configuration of an integrator according to a modified embodiment. [Figure 7] This figure shows the current waveform when switching from a three-phase short circuit to PWM mode in conventional control. [Figure 8] This figure shows the current waveform when switching from a three-phase short circuit to PWM mode in the control according to the embodiment. [Figure 9] This figure shows the current waveform when switching from three-phase open-circuit mode to PWM mode in conventional control. [Figure 10] This figure shows the current waveform when switching from three-phase open-circuit to PWM mode in the control according to the embodiment. [Modes for carrying out the invention]

[0012] Embodiments of the present invention will be described below with reference to the drawings. In each figure, elements with the same reference number represent the same or similar functional elements.

[0013] In the following description, the synchronous machine being controlled is a permanent magnet synchronous motor (hereinafter referred to as "PMSM" (an abbreviation for Permanent Magnet Synchronous Motor)).

[0014] [Overall structure] Figure 1 is a block diagram showing the functional configuration of a synchronous machine control device according to this embodiment. In this embodiment, a computer system such as a microcomputer functions as the synchronous machine control device shown in Figure 1 by executing a predetermined program.

[0015] In Figure 1, the power converter 2 converts DC power from a DC voltage source 9 (e.g., a battery) into AC power and outputs it to the PMSM1. The PMSM1 is driven to rotate by this AC power. The power converter 2 is equipped with an inverter main circuit consisting of semiconductor switching elements. The DC power is converted into AC power by the on / off control of the semiconductor switching elements by gate signals. For example, IGBTs (Insulated Gate Bipolar Transistors) are used as semiconductor switching elements.

[0016] The phase current detector 3 detects the three-phase motor currents flowing from the power converter 2 to the PMSM1, namely the U-phase current Iu, V-phase current Iv, and W-phase current Iw, and outputs them as the U-phase current detection value Iuc, the V-phase current detection value Ivc, and the W-phase current detection value Iwc, respectively. A Hall CT (Current Transformer) or similar device can be used as the phase current detector 3.

[0017] The magnetic pole position detector 4 detects the magnetic pole position of the PMSM1 and obtains magnetic pole position information θ * It outputs the following. A resolver or the like can be used as the magnetic pole position detector 4.

[0018] The frequency calculation unit 5 processes the magnetic pole position information θ output by the magnetic pole position detector 4. * Therefore, velocity information ω1 is obtained by time differentiation operations, etc. * Perform the calculation and output the result.

[0019] The coordinate transformation unit 7 converts the Iuc, Ivc, and Iwc output by the phase current detector into magnetic pole position information θ. * Accordingly, the dq axis current detection values ​​Idc and Iqc in the rotating coordinate system are converted and output as Idc and Iqc.

[0020] Based on the dq-axis current detection values Idc and Iqc output by the coordinate conversion unit 7, the dq-axis flux estimation unit 23 estimates the dq-axis flux estimation values φdc and φqc by referring to a look-up table (table data). The look-up table (table data) referred to by the dq-axis flux estimation unit 23 is table data representing the correspondence between Idc, Iqc and φdc, φqc, and is stored in a storage device (not shown) provided in the synchronous machine control device of the present embodiment. Note that, instead of the look-up table, a predetermined function (such as an approximation formula) may be used.

[0021] The first dq-axis flux command calculation unit 21 calculates the first dq-axis flux command values φd * , Iqc * by referring to a look-up table (table data) based on the dq-axis current command values Idc * , φq * given from a higher-level control device or the like, and outputs the calculated values. The look-up table (table data) referred to by the first dq-axis flux command calculation unit 21 is table data representing the correspondence between Idc * , Iqc * and φd * , φq * and is stored in a storage device (not shown) provided in the synchronous machine control device of the present embodiment. Note that, instead of the look-up table, a predetermined function (such as an approximation formula) may be used.

[0022] The second dq-axis flux command calculation unit 25 calculates the second dq-axis flux command values φd * , φq * by a proportional-integral (PI) controller so that the first dq-axis flux command values φd ** , φq ** match the dq-axis flux estimation values φdc and φqc, and outputs the calculated values.

[0023] [PI controller] FIG. 2 is a block diagram showing the functional configuration of the PI controller in the second dq-axis flux command calculation unit 25.

[0024] As shown in the upper part of Figure 2, the second d-axis magnetic flux command value φd ** In the PI controller that calculates the first d-axis magnetic flux command value φd, the adder / subtractor 81 calculates the first d-axis magnetic flux command value φd * The difference between the estimated d-axis magnetic flux φdc and (φd * -φdc) is calculated, and the difference calculation value is proportional gain 87(K P The difference value is multiplied by the integral gain 85(K). The difference value is then integrated by the integrator 83, and the integral value is multiplied by the integral gain 85(K). I The difference value multiplied by the proportional gain 87 and the integral value multiplied by the integral gain 85 are added by the adder 89 to obtain the second d-axis magnetic flux command value φd ** The calculation is performed.

[0025] As shown in the lower part of Figure 2, the second q-axis magnetic flux command value is φq ** In the PI controller that calculates the first q-axis magnetic flux command value φq, the adder / subtractor 91 calculates the first q-axis magnetic flux command value φq * The difference between the estimated q-axis magnetic flux φqc and (φq * -φqc) is calculated, and the difference is given a proportional gain of 97(K P The difference calculation value is multiplied by φq. The difference calculation value multiplied by the proportional gain 97 and the integral value multiplied by the integral gain 95 are added by the adder 99 to obtain the second d-axis magnetic flux command value φq. ** The calculation is performed.

[0026] [Voltage vector calculation] The voltage vector calculation unit 19 shown in Figure 1 generates voltage command values ​​using an inverse model of the motor model.

[0027] The inverse model of the motor model can be expressed by a voltage equation such as equation (1), where φd and φq are the d-axis and q-axis magnetic fluxes of the motor, respectively, Vd and Vq are the d-axis voltages and q-axis voltages of the motor, respectively, and ω1 is the motor speed.

[0028]

number

[0029] As will be discussed later, equation (1) takes into account the magnetic saturation of the motor.

[0030] Figure 3 shows the configuration of the voltage vector calculation unit 19, which is built based on the inverse model represented by equation (1). R, Ld, Lq, and Ke are the winding resistance, d-axis inductance, q-axis inductance, and magnetic flux of the PMSM1, respectively.

[0031] As shown in Figure 3, the differentiator 45 determines φd ** The derivative of is calculated. Also, the adder / subtractor 44 calculates φd ** The difference between and Ke (φd ** -Ke) is calculated, and this difference calculation value is multiplied by R / Ld(46). The differential calculation value from the differentiator 45 and the difference calculation value multiplied by the gain R / Ld(46) are added by the adder 47. Also, the multiplier 48 calculates ω1 * and φq ** and are multiplied. Furthermore, the adder / subtractor 49 calculates the difference between the sum calculated by the adder 47 and the multiplied value calculated by the multiplier 48, and Vd * This is created.

[0032] Furthermore, as shown in Figure 3, the differentiator 35 determines φq ** The derivative of is calculated. Also, φq ** This is multiplied by R / Lq(36). The differential value obtained by differentiator 35 and φq multiplied by R / Lq(36) is then given. ** These are added by the adder 37. Also, the multiplier 38 adds ω1 * and φd **and are multiplied. Furthermore, the adder 39 adds the sum of the sum calculated by adder 37 and the multiplicative sum calculated by multiplier 38, resulting in Vq * This is created.

[0033] In this way, a voltage vector calculation unit can be constructed based on a voltage equation that represents the inverse model of the motor model.

[0034] The coordinate transformation unit 11 shown in Figure 1 outputs the dq-axis voltage command value Vd for the power converter 2 from the voltage vector calculation unit 19. * ,Vq * The magnetic pole position information θ is detected by the magnetic pole position detector 4. * By performing a coordinate transformation using this method, the three-phase voltage command value Vu for the power converter 2 is obtained. * ,Vv * VW * Create and output the file.

[0035] The DC voltage detector 6 detects the voltage of the DC voltage source 9 and outputs DC voltage information Vdc.

[0036] The PWM controller 12 receives a three-phase voltage command value Vu from the voltage vector calculation unit 19. * ,Vv * VW * The PWM controller 12 receives the DC voltage information Vdc from the DC voltage detector 6 and, based on these, creates and outputs a gate signal to be supplied to the power converter 2 by pulse width modulation. For example, the PWM controller 12 uses a triangular wave as the carrier signal and creates the gate signal by pulse width modulation using the three-phase voltage command value as the modulated wave.

[0037] [Means for creating a voltage command receipt] The following describes the means for creating a voltage command value that takes into account the magnetic saturation of the PMSM1, which is used in the voltage vector calculation unit 19 of this embodiment.

[0038] First, if we consider the current (dq-axis current Id, Iq) as a state variable, and take magnetic saturation into account, the voltage equation can be expressed as shown in equation (2).

[0039]

number

[0040] Figure 4 shows an example of the relationship between magnetic flux and electric current. The solid lines on the vertical and horizontal axes in the diagram represent examples of the relationship between magnetic flux and electric current.

[0041] As shown in Figure 4, due to the effect of magnetic saturation, the rate of increase in the q-axis magnetic flux (φq) becomes slower as the q-axis current (Iq) increases. For this reason, dynamic inductance and static inductance are defined as follows: Dynamic inductance Lqh is the slope (dφq / dt) of the tangent line (dashed line in the figure) at a certain operating point (Iq, φq). Static inductance Lq is the slope (φq / Iq) of the straight line (dashed line in the figure) connecting the point where the q-axis current (Iq) is 0 and the operating point.

[0042] Although not shown in the diagram, the relationship between d-axis magnetic flux and d-axis current, dynamic inductance Ldh, and static inductance Ld are the same as in Figure 4.

[0043] In equation (2), the coefficient (matrix) in the current derivative term (second term on the right-hand side) is the dynamic inductance, and the coefficient (matrix) in the induced voltage term (third term on the right-hand side) is the static inductance.

[0044] Furthermore, when magnetic saturation is significant, mutual interference occurs between the control axes, i.e., between the dq axes. Such mutual interference is represented by the dynamic inductances Ldqh,Lqdh in the coefficients (matrix) of the current derivative term and the static inductances Ldq,Lqd in the coefficients (matrix) of the induced voltage term.

[0045] Based on equation (2), that is, when controlling the PMSM1 with current as a state variable and considering magnetic saturation, the eight types of inductances (Ldh, Lqh, Ldqh, Lqdh, Ld, Lq, Ldq, Lqd) described above are used. Therefore, in this case, the synchronous machine control device will have eight table data or functions (approximation formulas, etc.) that represent the correspondence between each inductance value and the current value (d-axis current value and q-axis current value).

[0046] Furthermore, considering the temperature dependence of these inductances, each of the table data or formulas (approximation formulas, etc.) becomes a three-variable table data or function, with the d-axis current value, the q-axis current value, and temperature as variables.

[0047] Furthermore, since the magnetic flux Ke in equation (2) depends on the q-axis current Iq and temperature T, the synchronous machine control device will have one two-variable table data or function (such as an approximation formula) that represents the relationship between the two variables Iq and T and Ke.

[0048] Thus, when controlling the PMSM1 with current as the state variable and considering magnetic saturation, the synchronous machine control device will have multiple multivariable table data or multivariable functions.

[0049] Therefore, as explained below, in this embodiment, by using magnetic flux as a state variable, similar to the inverse model of the motor model represented by equation (1) above, the total number of table data or functions (approximation formulas, etc.) used in the synchronous machine control device (9 when current is used as a state variable as described above) is reduced, while still considering magnetic saturation.

[0050] When magnetic flux (dq-axis magnetic flux φd, φq) is used as a state variable, and considering magnetic saturation, the voltage equation can be expressed as shown in equation (3).

[0051]

number

[0052] In this case, the synchronous machine control device includes table data or functions (such as approximation formulas) that represent the correspondence between the d-axis magnetic flux (φd) and the q-axis magnetic flux (φq) and the currents (d-axis current Id, q-axis current Iq). Therefore, the synchronous machine control device includes a total of two table data or functions.

[0053] In this way, by using magnetic flux as a state variable, the number of table data or functions used in motor control is reduced. As a result, the control system is simplified while still considering magnetic saturation, which reduces the computational load on the synchronous machine control device and shortens the parameter identification time.

[0054] Furthermore, in this embodiment, the second dq-axis magnetic flux command value φd is created by the second dq-axis magnetic flux command calculation unit 25 using the inverse model of the motor model. ** ,φq ** Based on this, the dq axis voltage command value Vd * ,Vq * This is how the second d-axis magnetic flux command value φdc is created. Therefore, even in the high-speed region, the estimated d-axis magnetic flux value φdc and the estimated q-axis magnetic flux value φqc are used to obtain the second d-axis magnetic flux command value φd ** and the second q-axis magnetic flux command value φq ** This allows for highly accurate synchronization. Therefore, the synchronous machine control device according to this embodiment enables high-speed rotation control of the PMSM1.

[0055] Furthermore, in this embodiment, the effect of the temperature dependence of the magnetic flux is mitigated by the PI controller or I controller provided in the second dq-axis magnetic flux command calculation unit 25. For this reason, the table data or function used to calculate the magnetic flux (φd, φq) may be a table data or function (such as an approximation formula) that does not include temperature as a variable and only uses current as a variable. This reduces the computational load on the synchronous machine control device and shortens the parameter identification time.

[0056] Furthermore, by using the same table data or function in the first dq-axis magnetic flux command calculation unit 21 and the dq-axis magnetic flux estimation unit 23, Idc and Iqc are, so to speak, transmitted via magnetic flux, respectively to Id * IQ * It is controlled to match this. In this case, it effectively constitutes a current control system.

[0057] Furthermore, Idc, Iqc, and Id are input to the first dq-axis magnetic flux command calculation unit 21 and the dq-axis magnetic flux estimation unit 23. * IQ * By using both, control that takes into account the mutual interference between axes becomes possible. In this case, the first dq-axis magnetic flux command calculation unit 21 and the dq-axis magnetic flux estimation unit 23 each calculate the dq-axis magnetic flux command value (φd * ,φq * ) and dq axis current command value (Id * IQ * Table data or functions representing the correspondence between (φdc, φqc) and (φdc, φqc), and table data or functions representing the correspondence between the estimated dq-axis magnetic flux (φdc, φqc) and the detected dq-axis current (Idc, Iqc) are used.

[0058] Furthermore, since the synchronous machine control device of this embodiment substantially takes into account the dynamic and static inductance of the motor, it is suitable for application to electric vehicles such as electric cars that use PMSMs, which are greatly affected by magnetic saturation, and for which accurate torque response is required.

[0059] Furthermore, the lookup table, table data, and functions (approximation formulas) mentioned above, which represent the correspondence between magnetic flux and current in PMSM1, can be set based on actual measurements, magnetic field analysis, etc.

[0060] [Integrator Initialization] Next, the initialization of the integrator according to this embodiment will be explained using Figures 5 to 8.

[0061] When switching power converter 2 from three-phase open mode or three-phase short-circuit mode to torque mode (PWM mode), it is necessary to set the initial values ​​of integrators 83 and 93. If the initial values ​​are incorrect, a difference will occur between the output voltage of power converter 2 and the voltage of PMSM1 when switching to PWM mode, which may cause overcurrent or torque fluctuations.

[0062] Figure 7 shows the current waveforms during the switching from a three-phase short circuit to PWM mode in conventional control. It shows the waveforms of the d-axis current, q-axis current, torque, and three-phase AC current, as well as the gate mode switching signal, when the d-axis integrator 83 is initialized to the magnetic flux Ke and the q-axis integrator 93 is initialized to 0. The dotted line in Figure 7 indicates the switching timing of the power converter 2 from a three-phase short circuit to PWM mode.

[0063] As shown in Figure 7, it can be seen that the fluctuations in the d-axis current value and q-axis current value become large after the switching timing. This is due to a difference between the output voltage of the power converter 2 and the motor voltage. Therefore, in order to prevent overcurrent, it is necessary to set appropriate initial values ​​for the integrators 83 and 93 at the switching timing.

[0064] Figure 5 is a block diagram of the integrator according to this embodiment. As shown in Figure 5, the integrator 83 receives a gate mode switching signal from a higher-level controller (not shown) or the like when the drive state (gate mode) of the power converter 2 is switched. In that case, the d-axis magnetic flux estimate φdc is used as the initial value for the d-axis integrator 83. Similarly, the q-axis magnetic flux estimate φqc is used as the initial value for the q-axis integrator 93. Alternatively, the values ​​obtained by applying a low-pass filter to the d-axis magnetic flux estimate φdc and the q-axis magnetic flux estimate φqc may be used as initial values.

[0065] In vector control using magnetic flux as the control variable, as in this embodiment, in the steady state, the integral value on the d-axis side matches the estimated d-axis magnetic flux value φdc. Similarly, the integral value on the q-axis side matches the estimated q-axis magnetic flux value φqc. Therefore, by initializing the d-axis integrator 83 with the estimated d-axis magnetic flux value φdc and the q-axis integrator 93 with the estimated q-axis magnetic flux value φqc, the voltage of the PMSM1 and the voltage of the power converter 2 match, thereby suppressing changes in current during switching.

[0066] Specifically, when power converter 2 is in a three-phase short circuit, the voltage becomes 0V, and a current flows through PMSM1 that results in a voltage of 0V. Therefore, when power converter 2 transitions from a three-phase short circuit to PWM mode, it should output a voltage that allows the current flowing during the three-phase short circuit to flow. In other words, by initializing the d-axis integrator 83 with the estimated d-axis magnetic flux φdc and the q-axis integrator 93 with the estimated q-axis magnetic flux φqc, the voltage of PMSM1 and the voltage of power converter 2 will match, thus suppressing the change in current during switching.

[0067] Figure 8 shows the change in current when switching from a three-phase short circuit to PWM mode in the control according to this embodiment. The waveforms of the d-axis current, q-axis current, torque, and three-phase AC current, as well as the gate mode switching signal, are shown when the d-axis integrator 83 is initialized with the estimated d-axis magnetic flux value φdc and the q-axis integrator 93 is initialized with the estimated q-axis magnetic flux value φqc. As shown in Figure 8, the control device according to this embodiment suppresses changes in the d-axis current and q-axis current after switching of the power converter 2. This makes it possible to suppress the occurrence of overcurrent.

[0068] Furthermore, when the power converter 2 is in a three-phase open state, the current in the power converter 2 becomes 0A (id=iq=0A), and only the induced voltage (Vq=ωKe) is output from the motor. Therefore, when the power converter 2 transitions from a three-phase open state to PWM mode, the power converter 2 only needs to output a voltage that cancels out the induced voltage of the PMSM1. In other words, the d-axis integrator 83 is initialized with the estimated d-axis magnetic flux φdc, and the q-axis integrator 93 is initialized with the estimated q-axis magnetic flux φqc=0. This makes it possible to suppress the generation of overcurrent even when transitioning from a three-phase short circuit to PWM mode.

[0069] [Differentiation] Next, a modified example of this embodiment will be described using Figure 6. Figure 6 shows an integrator according to a modified example of this embodiment.

[0070] As shown in the upper part of Figure 6, the initialization value of the d-axis integrator 183 differs depending on the state of the power converter 2 before the drive mode is switched. This allows the integrator 183 to be initialized with an appropriate value depending on the state of the power converter 2 before the drive mode is switched.

[0071] For example, in the configuration shown in Figure 6, if the drive state of the power converter 2 before switching is three-phase open, it is initialized with the magnetic flux Ke considering the rotor temperature, and if the drive state of the power converter 2 before switching is three-phase short, it is initialized with the estimated d-axis magnetic flux φdc. Since the estimated d-axis magnetic flux φdc changes depending on the rotor temperature of the PMSM1, more accurate control is possible by using the magnetic flux Ke considering the rotor temperature. The reason for considering the rotor temperature only when three-phase open is that in automobiles and the like, three-phase open is actively used during operation to reduce losses. Therefore, it is desirable to suppress torque fluctuations as much as possible when switching between three-phase open and PWM mode to prevent deterioration of ride comfort, etc. On the other hand, three-phase short is expected to occur under special conditions, such as when the power converter 2 fails. Therefore, it is sufficient if overcurrent is suppressed when switching between three-phase short and PWM mode. This makes it possible to reduce the labor required for adjusting the PMSM1.

[0072] Figure 9 shows the current waveforms during the switching from three-phase open circuit to PWM mode in conventional control. It shows the time variations of the d-axis current, q-axis current, torque, and three-phase AC current, as well as the gate mode switching signal, when the d-axis integrator 83 is initialized to the magnet flux Ke (without considering rotor temperature) and the q-axis integrator 93 is initialized to 0. The dotted line in Figure 9 indicates the switching timing of the power converter 2 from three-phase open circuit to PWM mode.

[0073] As shown in Figure 9, it can be seen that the fluctuations in the d-axis current value and q-axis current value become large after the switching timing. This is because the magnetic flux of the PMSM1 changes with the rotor temperature, resulting in a difference from the initial magnetic flux Ke set. As a result, there is a risk of torque fluctuations occurring due to the difference between the output voltage of the power converter 2 and the voltage of the PMSM1. Therefore, in order to prevent torque fluctuations, it is necessary to set appropriate initial values ​​for the integrators 83 and 93 at the switching timing.

[0074] Figure 10 shows the current waveform during switching from three-phase open to PWM mode in a modified control of this embodiment. It shows the time variation of the d-axis current, q-axis current, torque, and three-phase AC current, as well as the gate mode switching signal, when the d-axis integrator 83 is initialized with the magnetic flux Ke considering the rotor temperature, and the q-axis integrator 93 is initialized with the estimated q-axis magnetic flux φqc (=0). As shown in Figure 10, the control device according to this embodiment suppresses changes in the d-axis current and q-axis current after switching of the power converter 2. This makes it possible to suppress the occurrence of deterioration of ride comfort caused by sudden torque fluctuations.

[0075] Furthermore, the dq-axis magnetic flux estimation unit 23 may be configured to estimate the d-axis and q-axis magnetic flux values ​​of the PMSM1 based on a three-dimensional table that takes into account the rotor temperature of the PMSM1. Although this increases the motor calibration work, it allows for consideration of the rotor temperature, making it possible to more accurately prevent overcurrents and torque fluctuations caused by changes in the driving state of the power converter 2.

[0076] Furthermore, if rotor temperature information cannot be obtained, the integrator may be initialized with a predetermined rotor temperature. This prevents overcurrent and torque fluctuations caused by changes in the drive state of the power converter 2, even when the rotor temperature cannot be obtained.

[0077] The embodiment described above can be summarized as follows:

[0078] A synchronous machine control device for controlling a power converter that supplies power to a synchronous machine comprises: a first magnetic flux command calculation unit that calculates a first magnetic flux command value from the current command value of the synchronous machine; a magnetic flux estimation unit that estimates the magnetic flux values ​​of the d-axis and q-axis of the synchronous machine from the current detection value of the synchronous machine; a voltage calculation unit that creates a voltage command value for the power converter so that the first magnetic flux command value and the magnetic flux value match; and a second magnetic flux command calculation unit that uses an integrator to calculate second magnetic flux command values ​​for the d-axis side and the q-axis side, respectively, so that the first magnetic flux command value and the magnetic flux value match. The integrator is initialized with the d-axis magnetic flux value on the d-axis side and the q-axis magnetic flux value on the q-axis side when the drive state of the power converter changes. This prevents overcurrents and the like caused by changes in the drive state of the power converter.

[0079] Furthermore, the integrator controller may be configured such that the value initialized differs depending on the drive state before the power converter's drive state changes. This allows the integrator to be initialized with an appropriate value, thereby more accurately preventing torque fluctuations and other issues caused by changes in the power converter's drive state.

[0080] Furthermore, when the drive state before the switchover is three-phase open, the d-axis side may be initialized to a value determined based on the temperature of the synchronous machine's rotor. This prevents deterioration of ride comfort and other issues caused by sudden torque fluctuations when the power converter switches from three-phase open to PWM mode.

[0081] Furthermore, when the drive state before switching is a three-phase short circuit, the d-axis side is initialized with the d-axis magnetic flux value and the q-axis side with the q-axis magnetic flux value. When the drive state before switching is a three-phase open circuit, the d-axis side is initialized with the magnetic flux Ke considering the temperature of the synchronous machine's rotor, and the q-axis side is initialized with the q-axis magnetic flux value. This configuration allows for more accurate prevention of overcurrents and other issues caused by changes in the drive state of the power converter.

[0082] Furthermore, the magnetic flux estimation unit may be configured to estimate the magnetic flux values ​​of the d-axis and q-axis of the synchronous machine based on a three-dimensional table that takes into account the rotor temperature of the synchronous machine. This allows for consideration of the rotor temperature, making it possible to more accurately prevent overcurrents and torque fluctuations caused by changes in the drive state of the power converter.

[0083] Furthermore, if rotor temperature information cannot be obtained, the integrator may be initialized with a predetermined rotor temperature. This prevents overcurrents, torque fluctuations, etc., caused by changes in the drive state of the power converter, even when the rotor temperature cannot be obtained.

[0084] It should be noted that the present invention is not limited to the embodiments described above, and various modifications are included. For example, the embodiments described above are described in detail for the purpose of clearly illustrating the present invention, and are not necessarily limited to those having all the configurations described. Furthermore, it is possible to add, delete, or replace some of the configurations in each embodiment with other configurations.

[0085] For example, the synchronous machine to be controlled is not limited to PMSMs; it could also be a synchronous reluctance motor, a permanent magnet synchronous generator, a wound-field synchronous motor, a wound-field synchronous generator, or any other type of synchronous machine.

[0086] Furthermore, the PMSM may be either an embedded magnet type or a surface magnet type, and may be either an abducting type or an adduction type.

[0087] Furthermore, the semiconductor switching elements that make up the inverter main circuit are not limited to IGBTs; MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) and other types may also be used.

[0088] Furthermore, the synchronous machine control devices according to each of the above embodiments can be applied as control devices in various synchronous machine drive systems that include a synchronous machine, a power converter for driving the synchronous machine, and a control device for controlling the power converter. [Explanation of Symbols]

[0089] 1: PMSM, 2: Power converter, 3: Phase current detector, 4: Magnetic pole position detector, 5: Frequency calculation unit, 6: DC voltage detector, 7: Coordinate transformation unit, 10: Voltage vector addition unit, 11: Coordinate transformation unit, 15: DQ axis voltage command calculation unit, 19: Voltage vector calculation unit, 21: First DQ axis magnetic flux command calculation unit, 23: DQ axis magnetic flux estimation unit, 25: Second DQ axis magnetic flux command calculation unit, 35: Differentiator, 37: Adder, 38: Multiplier, 39: Adder, 44: Adder / Subtractor, 45: Differentiator, 47: Adder, 48: Multiplier, 49: Adder / Subtractor, 55: Multiplier, 57: Multiplier, 81: Adder / Subtractor, 83: Integrator, 85: Integral Gain, 87: Proportional Gain, 89: Adder, 91: Adder / Subtractor, 93: Integrator, 95: Integral Gain, 97: Proportional Gain, 98: Adder, 99: Adder, 183: Integrator, 193: Integrator.

Claims

1. A synchronous machine control device that controls a power converter that supplies power to a synchronous machine, A first magnetic flux command calculation unit that calculates a first magnetic flux command value from the current command value of the synchronous machine, A magnetic flux estimation unit that estimates the magnetic flux values ​​of the d-axis and q-axis of the synchronous machine from the current detection value of the synchronous machine, A voltage calculation unit that creates a voltage command value for the power converter such that the first magnetic flux command value and the magnetic flux value match, The system includes a second magnetic flux command calculation unit that uses an integral controller to calculate second magnetic flux command values ​​for the d-axis and q-axis sides, respectively, so that the first magnetic flux command value and the magnetic flux value match. The integral controller, when the drive state of the power converter is switched, The d-axis side is initialized with the d-axis magnetic flux value, and the q-axis side is initialized with the q-axis magnetic flux value, and the values ​​initialized differ depending on the drive state before switching. A synchronous machine control device characterized in that, when the drive state before the aforementioned switching is three-phase open, the d-axis side is initialized with magnetic flux.

2. A synchronous machine control device for controlling a power converter that supplies power to a synchronous machine, A first magnetic flux command calculation unit that calculates a first magnetic flux command value from the current command value of the synchronous machine, A magnetic flux estimation unit that estimates the magnetic flux values ​​of the d-axis and q-axis of the synchronous machine from the current detection value of the synchronous machine, A voltage calculation unit that creates a voltage command value for the power converter such that the first magnetic flux command value and the magnetic flux value match, The system includes a second magnetic flux command calculation unit that uses an integral controller to calculate second magnetic flux command values ​​for the d-axis and q-axis sides, respectively, so that the first magnetic flux command value and the magnetic flux value match. The integral controller, when the drive state of the power converter is switched, The d-axis side is initialized with the d-axis magnetic flux value, and the q-axis side is initialized with the q-axis magnetic flux value, and the values ​​initialized differ depending on the drive state before switching. When the drive state before the aforementioned switching is a three-phase short circuit, the d-axis side is initialized with the d-axis magnetic flux value, and the q-axis side is initialized with the q-axis magnetic flux value. A synchronous machine control device characterized in that, when the drive state before the aforementioned switching is three-phase open, the d-axis side is initialized with a magnetic flux that takes into account the rotor temperature of the synchronous machine, and the q-axis side is initialized with the q-axis magnetic flux value.

3. A synchronous machine control device according to claim 1, The magnetic flux estimation unit is based on a three-dimensional table that takes into account the rotor temperature of the synchronous machine, A synchronous machine control device characterized by estimating the magnetic flux values ​​of the d-axis and q-axis of the synchronous machine.

4. A synchronous machine control device according to claim 2 or 3, A synchronous machine control device characterized in that, if the rotor temperature information cannot be obtained, the integrating controller is initialized with the rotor temperature set to a predetermined value.

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