Motor control devices and electric drive vehicles

Abstract

To provide a motor control device and an electrically driven vehicle that can more easily suppress current oscillations caused by 6n-th order harmonic current components when performing control in which a control period is not synchronized with an inverter frequency.SOLUTION: A motor control device that outputs a voltage to be applied to an AC motor on the basis of an output voltage command value includes: a control period determination unit that determines a control period according to the fundamental frequency of an AC motor and the cutoff frequency of current control; a control interrupt unit that determines a control interrupt on the basis of the control period determined by the control period determination unit; a current control unit that controls an output voltage command value on the basis of the control interrupt determined by the control interrupt unit; and a PWM control unit that controls a voltage output by PWM according to the output voltage command value controlled by the current control unit.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] The present invention relates to a motor control device and an electrically driven vehicle. [Background Art]

[0002] In a control system for an alternating-current motor mounted in an electrically driven vehicle or the like, in order to improve the voltage utilization rate of an inverter, there are known technologies including a technology of operating the inverter in an overmodulation mode (a so-called overmodulation region) where an output voltage command of the inverter exceeds a maximum output level that can be output by a sine wave, and a technology such as synchronous PWM that outputs PWM pulses in synchronization with an inverter output frequency.

[0003] On the other hand, in an overmodulation region and synchronous PWM, due to switching of the inverter, a 6n-order harmonic current (where n is a natural number) is included in a current of a dq-axis coordinate system used in vector control, and there is a problem that current oscillation occurs due to this harmonic current.

[0004] As technologies related to suppression of such a harmonic current, those described in Patent Documents 1 and 2 are known, for example.

[0005] Patent Document 1 discloses a current detection device for a synchronous PWM power converter, comprising: a power conversion main circuit that applies a voltage for driving an AC power load; voltage command value generation means that respectively outputs a command value for a voltage modulation rate and a command value for a voltage phase angle; PWM control signal generation means that generates a PWM control signal based on the command value of the modulation rate and the command value of the phase angle, and outputs the PWM control signal to the power conversion main circuit; current detection means that detects a current supplied to the AC power load and outputs a current signal corresponding to the current; and sampling means that samples and holds a current detection value based on the current signal to output a first sample hold value, wherein the sampling means includes voltage phase angle setting means in which a predetermined phase angle serving as a sampling timing is set, and samples and holds the current detection value at a timing when the command value of the phase angle becomes equal to a set value of the predetermined phase angle.

[0006] Furthermore, Non-Patent Document 1 discloses a filter design method based on a filter approach, aimed at preventing degradation of control performance by reducing the sensitivity of the current control system to superimposed disturbance components in the overmodulation region. [Prior art documents] [Patent Documents]

[0007] [Patent Document 1] Japanese Patent Publication No. 2004-15949 [Non-patent literature]

[0008] [Non-Patent Document 1] Yosuke Nakayama, Shinji Michiki, "Design of a Band-Reject Filter Enabling PMSM Vector Control in the Inverter Overmodulation Region," Transactions of the Institute of Electrical Engineers of Japan, Vol. 138, No. 11, pp. 884-893 (2018). [Overview of the project] [Problems that the invention aims to solve]

[0009] In Patent Document 1, harmonic currents are suppressed by synchronizing the carrier frequency with the inverter output frequency and detecting the current at the timing when the harmonic current becomes zero. However, in applications such as automobiles where rapid acceleration and deceleration are required, the control period is frequently changed with frequency, making it difficult to synchronize the control period with the inverter frequency.

[0010] Furthermore, Non-Patent Document 1 attempts to remove the sixth-order component of the current in the overmodulation region, and this can be achieved with a constant control period. However, the band rejection filter used to remove harmonics needs to be changed according to the inverter output frequency, and a high-speed calculation period is required to remove the target frequency.

[0011] The present invention has been made in view of the above, and aims to provide a motor control device and an electric drive vehicle that can more easily suppress current oscillations due to 6th harmonic current components when performing control in which the control period is not synchronized with the inverter frequency. [Means for solving the problem]

[0012] The present invention includes multiple means for solving the above problems, but to give one example, a motor control device that outputs a voltage to be applied to an AC motor based on an output voltage command value, comprising: a control period determination unit that determines a control period according to the fundamental wave frequency of the AC motor and the cutoff frequency of current control; a control interrupt unit that determines a control interrupt based on the control period determined by the control period determination unit; a current control unit that controls the output voltage command value based on the control interrupt determined by the control interrupt unit; and a PWM control unit that controls the voltage output by PWM according to the output voltage command value controlled by the current control unit. [Effects of the Invention]

[0013] According to the present invention, when performing control that does not synchronize the control period with the inverter frequency, it is possible to more easily suppress current oscillations caused by 6th-order harmonic current components. [Brief explanation of the drawing]

[0014] [Figure 1] This is a functional block diagram showing the motor control device according to the first embodiment, along with its related configurations. [Figure 2] This figure shows the relationship between the three-phase fundamental wave current, the 6th harmonic current on the dq axis, and the reading timing. [Figure 3] This figure shows an example of a lookup table according to the first embodiment. [Figure 4] This figure shows a first modified example of a lookup table. [Figure 5] This figure shows a second modified example of the lookup table. [Figure 6]It is a diagram showing a third modification of a lookup table. [Figure 7] It is a functional block diagram showing a motor control device according to a second embodiment together with related configurations. [Figure 8] It is a diagram showing the relationship among three-phase fundamental wave currents, 6th-order harmonic currents on dq axes, and reading timing. [Figure 9] It is a diagram showing an example of a lookup table according to a second embodiment, [Figure 10] It is a diagram showing the relationship among three-phase fundamental wave currents, 6th-order harmonic currents on dq axes, and reading timing. [Figure 11] It is a functional block diagram showing a motor control device according to a third embodiment together with related configurations. [Figure 12] It is a functional block diagram showing processing contents of a control cycle calculation unit according to a third embodiment. [Figure 13] It is a diagram showing an example of a lookup table according to a third embodiment. [Figure 14] It is a functional block diagram showing a motor control device according to a fourth embodiment together with related configurations. [Figure 15] It is a functional block diagram showing processing contents of a control cycle calculation unit according to a fourth embodiment. [Figure 16] It is a functional block diagram showing a motor control device according to a fifth embodiment together with related configurations. [Figure 17] It is a functional block diagram showing processing contents of a control cycle calculation unit according to a fifth embodiment. [Figure 18] It is a functional block diagram showing a motor control device according to a sixth embodiment together with related configurations. [Figure 19] It is a diagram showing the relationship among three-phase fundamental wave currents, 6th-order harmonic currents on dq axes, and reading timing. [Figure 20] It is a diagram schematically showing, together with a control system, an extracted drive system of an electrically driven vehicle equipped with an AC motor controlled by the motor control device. DETAILED DESCRIPTION OF THE INVENTION

[0015] Embodiments of the present invention will be described below with reference to the drawings. In the following description, a permanent magnet synchronous motor (PMSM) of an AC motor is used as an example of the control target of the motor control device, but the present invention is not limited thereto. For example, the present invention can be applied to any AC machine such as a synchronous lulucantance motor, a permanent magnet synchronous generator, a wound synchronous machine, an induction motor, or an induction generator to obtain similar effects. Furthermore, while IGBTs are used as an example of semiconductor switching elements in the inverter device, the present invention is not limited thereto. For example, the present invention can be applied to any other power semiconductor element such as a MOSFET to obtain similar effects.

[0016] <First Embodiment> A first embodiment of the present invention will be described with reference to Figures 1 to 6 and Figure 20.

[0017] Figure 1 is a functional block diagram showing the motor control device according to this embodiment along with its related components. Figure 20 is a schematic diagram showing the drive system of an electric vehicle equipped with an AC motor controlled by the motor control device, along with the control system.

[0018] As shown in Figure 20, the motor control device 100 in this embodiment controls the power supplied to the PMSM1 from a DC voltage source 9 (for example, a battery) via a power converter 2 (inverter).

[0019] The PMSM1 is connected to the drive shaft 105 via the transmission 101 and differential gear 103, and supplies power to the wheels 107.

[0020] In this embodiment, the case in which power from the PMSM1 is transmitted via the transmission 101 is described as an example, but the present invention is not limited to this, and can also be applied to configurations in which the PMSM is directly connected to the differential gear, or in which the front wheels and rear wheels each have a PMSM and an inverter.

[0021] As shown in Figure 1, the motor control device 100 controls the power supplied to the PMSM1 from the DC voltage source 9 via the power converter 2, and is generally composed of a phase current detector 3, a magnetic pole position detector 4, a frequency calculation unit 5, a DC voltage detection device 6, coordinate transformation units 7 and 11, a current reading unit 8, a PWM controller 12, a control period calculation unit 15, a control interrupt generation unit 17, and a current control unit 19. The coordinate transformation units 7 and 11 and the current control unit 19 constitute the control calculation unit 13.

[0022] The power converter 2 converts DC power from a DC voltage source 9 (e.g., a battery) into AC power according to the gate signal from the PWM controller 12, and outputs it to drive the PMSM 1.

[0023] The phase current detector 3 is composed of, for example, a Hall CT (Current Transformer) and detects the current waveforms Iud, Ivd, and Iwd of the U-phase, V-phase, and W-phase of the three-phase current flowing from the power converter 2 to the PMSM1.

[0024] The magnetic pole position detector 4 is composed of, for example, a resolver, and detects the magnetic pole position of the PMSM1 and outputs it as magnetic pole position information θ*.

[0025] The frequency calculation unit 5 calculates the angular velocity from the magnetic pole position information θ* detected by the magnetic pole position detector 4, for example by differential calculation, and outputs it as velocity information ω1*.

[0026] The current reading unit 8 reads the current waveforms Iud, Ivd, and Iwd detected by the phase current detector 3 when the current reading signal cl from the control interrupt generation unit 17 turns ON, and outputs them as three-phase currents Iuc, Ivc, and Iwc to the coordinate transformation unit 7 of the control calculation unit 13.

[0027] The coordinate transformation unit 7 calculates and outputs the dq-axis current detection values ​​Idc and Iqc by performing a coordinate transformation on the three-phase currents Iuc, Ivc, and Iwc read by the current reading unit 8 based on the magnetic pole position information θ* detected by the magnetic pole position detector 4.

[0028] The current control unit 19 calculates and outputs dq-axis voltage command values ​​Vd* and Vq* so that the dq-axis current command values ​​Id* and Iq* match the dq-axis current detection values ​​Idc and Iqc from the coordinate transformation unit 7.

[0029] The coordinate transformation unit 11 calculates and outputs three-phase voltage command values ​​Vu*, Vv*, and Vw* by performing a coordinate transformation on the dq-axis voltage command values ​​Vd* and Vq* from the current control unit 19 based on the magnetic pole position information θ* detected by the magnetic pole position detector 4.

[0030] The control period calculation unit 15 calculates and outputs the control period tc based on the velocity information ω1* calculated by the frequency calculation unit 5.

[0031] The control interrupt generation unit 17 generates a control interrupt signal int and a current reading signal cl based on the control period fc from the control period calculation unit 15. In this embodiment, the control interrupt signal int and the current reading signal cl are synchronized and are identical signals.

[0032] The coordinate transformation unit 7, coordinate transformation unit 11, and current control unit 19, which constitute the control calculation unit 13, operate in response to the call of the control interrupt signal int from the control interrupt generation unit 17.

[0033] The DC voltage detection device 6 detects the voltage of the DC voltage source 9 and outputs the detection result as DC voltage information Vdc.

[0034] The PWM controller 12 calculates a duty cycle signal based on the three-phase voltage command values ​​Vu*, Vv*, and Vw* from the coordinate transformation unit 11 and the DC voltage information Vdc from the DC voltage detection device 6. It then generates and outputs a gate signal by comparing the calculated duty cycle signal with the carrier wave.

[0035] Now, let me explain the basic principles of this invention.

[0036] First, we will explain the principles of current oscillation generation in the overmodulation region of AC motors and in synchronous PWM control.

[0037] Figure 2 shows the relationship between the three-phase fundamental wave current, the 6th harmonic current on the dq axis, and the reading timing.

[0038] In AC motor control, it is known that a 6th-order component is generated in the dq-axis current in the overmodulation region (see, for example, Non-Patent Document 1), and current oscillation occurs when the control period fc approaches 6n times the inverter output frequency (where n is a natural number).

[0039] Figure 2 shows the three-phase fundamental current 61, the dq-axis sixth-harmonic current 63, the reading timing, and the influence of the sixth-order component (n=1) among the 6n-order components on the control. In this embodiment, the current reading unit 8 reads the current in synchronization with the control calculation unit 13 at the control period fc. Figure 2 shows the case where the control period fc is shifted by Δf from 6 times the inverter output frequency f0 (i.e., fc = 6f0 + Δf). At this time, the sixth-order harmonic component i'h6 detected at each point can be obtained by the following equations (Equation 1) to (Equation 4).

[0040]

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[0044] As shown in equations (1) to (4) above, the current is detected at frequency (6f0 + Δf), so when viewed in dq coordinates, the current appears to oscillate at frequency (-Δf). The amplitude at this time is the amplitude Ih of the harmonic current.

[0045] Here, if frequency Δf is within the response bandwidth of the current control, the current, which should actually be oscillating in the sixth order, is treated as a current oscillating at frequency Δf, and the current control acts to cancel out the oscillation at frequency Δf. Since this oscillation at frequency Δf does not normally occur, when the current control is applied, the current control itself generates an oscillation of Δf. This is the mechanism by which current oscillations are generated.

[0046] Therefore, in this embodiment, the generation of current oscillations is suppressed by controlling the control period calculation unit 15. The basic principle and details will be explained below.

[0047] The control period calculation unit 15 is composed of a lookup table that takes velocity information ω1* as input and outputs the control period fc.

[0048] Figure 3 shows an example of a lookup table used in the control period calculation unit, with velocity information ω1* on the horizontal axis and the control period fc on the vertical axis.

[0049] As shown in Figure 3, the relationship between velocity information ω1* and the control period fc (line 51) is selected in a way that avoids the exclusion band defined by the upper limit 53 and the lower limit 55. Here, we illustrate the case where the control period fc=f1 is set to the range where velocity information ω1* < angular velocity a, or angular velocity b ≤ velocity information ω1*, and the control period fc=f2 is set to the range where angular velocity a ≤ velocity information ω1* < angular velocity b. However, angular velocity a is the angular velocity when fc=f1 intersects with the upper limit 53 of the exclusion band, and angular velocity b is the angular velocity when fc=f1 intersects with the lower limit 55 of the exclusion region.

[0050] The upper limit 53 and lower limit 55 of the non-selection bandwidth are determined by the following (Equation 5).

[0051]

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[0052] Here, in equation (5) above, f0 represents the inverter output frequency, and facr represents the cutoff frequency of the current control unit 19. The variable m is a positive value, and its setting method is explained below.

[0053] The apparent current generated in the dq axis current falls within the response band of the current control, which is the cause of current oscillations. Therefore, in this embodiment, the control is performed so that the frequency Δf of the apparent dq axis current harmonic falls outside the response band of the current control. Generally, frequencies 10 times or more the current control cutoff frequency facr fall outside the response band of the current control. For example, if the number of current detections is 1 per control period, setting the variable m = 10 will make the frequency Δf outside the response band. If some current oscillations are considered acceptable, it can also be said that the variable m can be set to around 5. In this way, by setting the control period to avoid the selection exclusion band shown in (Equation 5) above, it is possible to avoid oscillations at low current frequencies even if a 6th-order component is generated in the dq axis current in the overmodulation region.

[0054] In this embodiment, the 6th-order component (i.e., the case where n=1) was used as an example for explanation, but the above principle applies to all 6n-order components (i.e., the case where n>1), such as the 12th, 18th, and 24th-order components.

[0055] Thus, when detecting the current used for current control at the same time as the control interrupt int, it is possible to avoid oscillations by avoiding integer multiples of 6 as a forbidden bandwidth to prevent oscillations caused by the 6th-order current.

[0056] Furthermore, although this embodiment illustrates the case where the control interrupt int and the current read signal cl are exactly the same, if the periods of these signals are the same, a slight timing difference is acceptable and a similar effect can be obtained.

[0057] Furthermore, although this embodiment illustrates changing the control period according to angular velocity (velocity information ω1*), the same effect can be obtained by applying the present invention when changing the frequency according to the inverter output frequency (which does not coincide with velocity in the case of induction motors) instead of angular velocity. Also, even when changing the frequency according to information other than angular velocity (for example, torque), the same effect can be obtained by setting the control period to avoid the prohibited band determined by angular velocity.

[0058] Furthermore, since automobiles are applications with demanding requirements for rapid acceleration / deceleration performance and vibration suppression, the effects of the present invention are more pronounced in these applications than in others. Similarly, railways, like automobiles, also have demanding requirements for rapid acceleration / deceleration performance and vibration suppression, making them applications where the effects of the present invention are easily apparent. In other words, by applying the present invention to "control that does not synchronize the control cycle with the inverter frequency," which is a method suitable for improving rapid acceleration / deceleration performance in automobiles and railways, vibrations caused by 6f can be suppressed, thereby improving the ride comfort of drivers or passengers.

[0059] In this embodiment, as shown in Figure 3, we have illustrated and explained the case where a lookup table is used to determine the control period while avoiding the exclusion band rectangularly and to the minimum extent possible. However, the embodiment is not limited to this, and for example, as shown in the first to third modifications below, the same effects as in the first embodiment can be obtained by using a lookup table in which the control period avoids the exclusion band.

[0060] Figure 4 shows a lookup table relating to the first modified example of this embodiment. In Figure 4, the relationship between velocity information ω1* and control period fc (line 51A) is set to determine the control period while avoiding the exclusion band rectangularly with sufficient margin. Here, the case where the control period fc = f1 is set in the range of velocity information ω1* < angular velocity a, or angular velocity b ≤ velocity information ω1*, and the control period fc = f2 is set in the range of angular velocity a ≤ velocity information ω1* < angular velocity b is illustrated as an example. However, the angular velocities a, b and control periods f1, f2 are set so that the control periods f1, f2 do not take values ​​above the upper limit 53 and lower limit 55 of the exclusion band, that is, they are values ​​with sufficient margin from the exclusion region.

[0061] Figure 5 shows a lookup table relating to a second modified example of this embodiment. In Figure 5, the relationship between velocity information ω1* and the control period fc (line 51B) is set to determine the control period by avoiding the exclusion band in a stepwise manner and to the minimum extent possible. Here, an example is shown where the control period fc=f1 in the range of velocity information ω1* < angular velocity a, and the control period fc=f2 in the range of angular velocity a ≤ velocity information ω1*. However, angular velocity a is the angular velocity when fc=f1 intersects with the upper limit 53 of the exclusion band.

[0062] Figure 6 shows a lookup table relating to a third modified example of this embodiment. In Figure 6, the relationship between velocity information ω1* and the control period fc (line 51C) is selected in a way that avoids the exclusion band defined by the upper limit 53 and the lower limit 55. Here, we illustrate the case where the control period fc = f1 in the range of velocity information ω1* < angular velocity a, or angular velocity b ≤ velocity information ω1*, and the control period fc is set to a value along the lower limit 55 in the range of angular velocity a ≤ velocity information ω1* < angular velocity b. However, angular velocity a is the angular velocity when fc = f1 intersects with the upper limit 53 of the exclusion band, and angular velocity b is the angular velocity when fc = f1 intersects with the lower limit 55 of the exclusion region.

[0063] <Second Embodiment> A second embodiment of the present invention will be described with reference to Figures 7 to 10.

[0064] This embodiment demonstrates a case where the current reading signal is controlled to turn ON at twice the frequency of the interrupt signal, and control is performed according to the difference between the two three-phase currents read in response to the current reading signal.

[0065] In this embodiment, only the differences from the first embodiment will be described. In the drawings used in this embodiment, the same reference numerals are used for components similar to those in the first embodiment, and their descriptions are omitted as appropriate.

[0066] Figure 7 is a functional block diagram showing the motor control device according to this embodiment, along with its related components.

[0067] As shown in Figure 7, the motor control device 100A controls the power supplied to the PMSM1 from the DC voltage source 9 via the power converter 2, and is generally composed of a phase current detector 3, a magnetic pole position detector 4, a frequency calculation unit 5, a DC voltage detection device 6, coordinate transformation units 7A and 11, a current reading unit 8A, a PWM controller 12, a control period calculation unit 15A, a control interrupt generation unit 17A, and a current control unit 19. Note that the coordinate transformation units 7A and 11 and the current control unit 19 constitute the control calculation unit 13A.

[0068] The control interrupt generation unit 17A generates and outputs a current reading signal cl such that it turns ON at twice the period of the interrupt signal int.

[0069] The current reading unit 8A reads the current waveforms Iud, Ivd, and Iwd of the three-phase current when the current reading signal cl is turned ON. At this time, it outputs the current at the reading timing as three-phase currents Iuc1, Ivc1, and Iwc1, and outputs the previously read current as three-phase currents Iuc2, Ivc2, and Iwc2 to the coordinate transformation unit 7A of the control calculation unit 13A.

[0070] The coordinate transformation unit 7A corrects the timing difference between the readings of the three-phase currents Iuc1, Ivc1, Iwc1 and Iuc2, Ivc2, Iwc2, respectively, into rotation angles and transforms them into dq coordinates. Then, it averages the transformed dq axis currents and outputs the detected dq axis current values ​​Idc and Iqc to the current control unit 19.

[0071] Here, we will explain the operation and effects of this embodiment configured as described above.

[0072] Figure 8 shows the relationship between the three-phase fundamental wave current, the dq-axis sixth harmonic current, and the reading timing. Figure 8 shows the three-phase fundamental wave current 61, the dq-axis sixth harmonic current 63, and the reading timing.

[0073] In Figure 8, the current reading unit 8A reads the current in synchronization with the control calculation unit 13A with a period of twice the normal rate, fcl = 2fc, and the case where the control period fc is shifted by Δf from 6 times the inverter output frequency f0 (i.e., fcl = 2fc = 12f0 + 2Δf) is shown. In this case, the 6th harmonic component i'h6 detected at each point can be obtained by the following equations (Equations 6) to (Equations 9).

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[0078] Furthermore, since the average value of two detected currents is used for current control, the sixth harmonic component i'h6_c included in the current used for current control can be obtained using the following equations (Equation 10) and (Equation 11).

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[0081] As shown in (Equations 10) and (Equation 11) above, the current is detected at frequency (12f0 + 2Δf), so when viewed in dq coordinates, the current appears to be oscillating at frequency (-2Δf). The amplitude at this time is the amplitude of the harmonic current (Ih × 2π × 0.5Δf ÷ (12f0 + 2Δf)). The amplitude is smaller than when detecting one current in one period of the control cycle (see the first embodiment), but as the frequency Δf increases, the amplitude also increases, so oscillation occurs similarly (however, the frequency and amplitude of the oscillation are different).

[0082] Therefore, in this embodiment, the current oscillation is suppressed by changing the control period fc from the control period calculation unit 15A according to the speed ω1*.

[0083] The control period calculation unit 15A is composed of a lookup table that takes speed information ω1* as input and outputs the control period fc.

[0084] Figure 9 shows an example of a lookup table used in the control period calculation unit, with velocity information ω1* on the horizontal axis and the control period fc on the vertical axis.

[0085] As shown in Figure 9, the relationship between velocity information ω1* and the control period fc (line 51D) is selected in a way that avoids the exclusion band defined by the upper limit 53A and the lower limit 55A, and the exclusion region defined by the upper limit 53B and the lower limit 55B. Here, we illustrate the case where the control period fc = f1 in the range of velocity information ω1* < angular velocity a, angular velocity b ≤ velocity information ω1* < angular velocity c, or angular velocity d ≤ velocity information ω1*, the control period fc = f2 in the range of angular velocity a ≤ velocity information ω1* < angular velocity b, and the control period fc = f3 in the range of angular velocity c ≤ velocity information ω1* < angular velocity d. However, angular velocity a is the angular velocity when fc=f1 intersects with the upper limit 53A of the exclusion zone, angular velocity b is the angular velocity when fc=f1 intersects with the lower limit 55A of the exclusion zone, angular velocity c is the angular velocity when fc=f1 intersects with the upper limit 53B of the exclusion zone, and angular velocity d is the angular velocity when fc=f1 intersects with the lower limit 55B of the exclusion zone.

[0086] The upper limit 53A and lower limit 55A of the non-selectable bandwidth are determined by the following (Equation 12).

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[0088] Here, in equation 12 above, f0 represents the inverter output frequency, and facr represents the cutoff frequency of the current control unit 19 (current control cutoff frequency). The variable m1 is a positive value, and its setting method is explained below.

[0089] If the current detection frequency is twice the control cycle, the frequency will be twice that of the single detection frequency. Therefore, setting the variable m1 to 5 will result in the current control response being outside the bandwidth at a frequency 10 times the current control cutoff frequency facr. However, if some current oscillation is considered acceptable, it can be said that setting the variable m1 to around 2.5 is also acceptable.

[0090] Furthermore, if the current is detected twice, similar oscillations occur even when the control cycle is close to three times the inverter output frequency. Therefore, a setting prohibition region is added, defined by the upper limit 53B and lower limit 55B determined by (Equation 13) below.

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[0092] Here, in the above equation (13), k is a positive value, and is basically set to k = m1 / 2.

[0093] Next, we will explain why similar vibrations occur when the control period is near three times the inverter output frequency.

[0094] Figure 10 shows the relationship between the three-phase fundamental wave current, the 6th harmonic current on the dq axis, and the reading timing. Figure 10 shows the three-phase fundamental wave current 61, the 6th harmonic current on the dq axis 63, and the reading timing.

[0095] In Figure 10, the current reading unit 8A reads the current in synchronization with the control calculation unit 13A with a period of twice the normal rate, fcl = 2fc, and the case where the control period fc is shifted by Δf from three times the inverter output frequency f0 (i.e., fcl = 2fc = 6f0 + 2Δf) is shown. At this time, the sixth harmonic component i''h6 detected at each point can be obtained by the following equations (Equations 14) to (Equations 17).

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[0100] Furthermore, since the average value of two detected currents is used for current control, the sixth harmonic component i''h6_c included in the current used for current control can be obtained using the following equations (Equations 18 and 19).

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[0103] As shown in (Equations 18) and (Equation 19) above, the current is detected at frequency (6f0 + 2Δf), so when viewed in dq coordinates, the current appears to be oscillating at frequency 4Δf. The amplitude at this time is the amplitude Ih of the harmonic current. The amplitude is the same as when detecting one current in one period of the control cycle (see the first embodiment), and oscillation occurs in the same way. However, since the frequency is doubled, the forbidden band for suppressing oscillation can be halved.

[0104] The other configurations are the same as in the first embodiment.

[0105] In this embodiment configured as described above, the same effects as in the first embodiment can be obtained.

[0106] Furthermore, if the current used for current control is detected multiple times at equal intervals within the control cycle and its average value is taken, then it is necessary to avoid integer multiples of (6 ÷ number of current detections) as a forbidden band to prevent oscillations caused by the 6th-order current. In this embodiment, an example of two detections is shown, but oscillations can be avoided by setting the same parameters even if the number of detections increases. Also, if the period of the control interrupt int and the current read signal cl is doubled, the same effect can be obtained even if the phase is shifted. In addition, if the lookup table is set to set the control period fc in a way that avoids the forbidden band, the same effect can be obtained even if it differs from the lookup table exemplified in Figure 9 in this embodiment.

[0107] <Third Embodiment> A third embodiment of the present invention will be described with reference to Figures 11 to 13.

[0108] This embodiment demonstrates a case where the control period is controlled according to the modulation rate calculated from the dq-axis voltage command value and the DC voltage.

[0109] In this embodiment, only the differences from the first embodiment will be described. In the drawings used in this embodiment, the same reference numerals are used for components similar to those in the first embodiment, and their descriptions are omitted as appropriate.

[0110] Figure 11 is a functional block diagram showing the motor control device according to this embodiment, along with its related components.

[0111] As shown in Figure 11, the motor control device 100B controls the power supplied to the PMSM1 from the DC voltage source 9 via the power converter 2, and is generally composed of a phase current detector 3, a magnetic pole position detector 4, a frequency calculation unit 5, a DC voltage detection device 6, coordinate transformation units 7 and 11, a current reading unit 8, a PWM controller 12, a control period calculation unit 15B, a control interrupt generation unit 17, a modulation rate calculation unit 18, and a current control unit 19. Note that the coordinate transformation units 7 and 11, the modulation rate calculation unit 18, and the current control unit 19 constitute the control calculation unit 13B.

[0112] The modulation rate calculation unit 18 calculates the modulation rate Mf based on the dq-axis voltage command values ​​Vd*,Vq* from the current control unit 19 and the DC voltage Vdc from the DC voltage detection device 6, and outputs it to the control period calculation unit 15B.

[0113] Figure 12 is a functional block diagram showing the processing details of the control period calculation unit.

[0114] As shown in Figure 12, the control period calculation unit 15B calculates the control period fc based on the speed information ω1* from the frequency calculation unit 5 and the modulation rate Mf from the modulation rate calculation unit 18, and outputs it to the control interrupt generation unit 17. It is roughly composed of a control period selection unit (normal) 21, a control period selection unit 23 (overmodulation), a modulation mode determination unit 25, and a switching unit 22.

[0115] The modulation mode determination unit 25 determines whether or not the control of the PMSM1 is being performed in the overmodulation region based on the modulation rate Mf.

[0116] The control period selection unit (during overmodulation) 23 selects the control period fc to be used when the system is controlled in the overmodulation region based on a lookup table. The control period selection unit (during overmodulation) 23 uses, for example, the lookup table shown in Figure 2 of the first embodiment. This selects a control period fc that avoids current oscillations that are a concern when controlling in the overmodulation region.

[0117] The control period selection unit (normal operation) 21 selects the control period fc to be used when control is not in the overmodulation region, according to the lookup table shown in Figure 13. Figure 13 is a diagram showing an example of a lookup table used by the control period calculation unit, with velocity information ω1* on the horizontal axis and the control period fc on the vertical axis. When control is performed in a range that is not the overmodulation region, there is no need to consider the prohibited bandwidth, so as shown in Figure 13, a lookup table is used in which the relationship between velocity information ω1* and the control period fc (line 51E) is such that the control period fc = f1 regardless of the velocity information ω1*. In this case, the control period f1 can be set high because there is no need to consider the suppression of current oscillations.

[0118] If the modulation mode determination unit 25 determines that the system is being controlled in an overmodulation region, the switching unit 22 switches the output to the control interrupt generation unit 17 to the control period fc selected by the control period selection unit (overmodulation) 23. If the modulation mode determination unit 25 determines that the system is not being controlled in an overmodulation region, the switching unit 22 switches the output to the control interrupt generation unit 17 to the control period fc selected by the control period selection unit (normal) 21 (for example, fc=f1).

[0119] The other configurations are the same as in the first embodiment.

[0120] In this embodiment configured as described above, the same effects as in the first embodiment can be obtained.

[0121] Furthermore, in this embodiment, considering that 6f vibration does not occur in the dq axis current in the normal region (when not in the overmodulation region), the control period fc is selected to avoid the prohibited band only in the overmodulation region where vibration occurs. As a result, vibration can be suppressed during overmodulation, and the control period fc in normal conditions can be set higher, thereby improving control performance.

[0122] <Fourth Embodiment> A fourth embodiment of the present invention will be described with reference to Figures 14 and 15.

[0123] This embodiment shows a case where the method for calculating the control period and the method for controlling the PWM are changed depending on whether the control is synchronous or asynchronous PWM.

[0124] In this embodiment, only the differences from the first embodiment will be described. In the drawings used in this embodiment, the same reference numerals are used for components similar to those in the first embodiment, and their descriptions are omitted as appropriate.

[0125] Figure 14 is a functional block diagram showing the motor control device according to this embodiment, along with its related components.

[0126] As shown in Figure 14, the motor control device 100C controls the power supplied to the PMSM1 from the DC voltage source 9 via the power converter 2, and is generally composed of a phase current detector 3, a magnetic pole position detector 4, a frequency calculation unit 5, a DC voltage detection device 6, coordinate transformation units 7 and 11, a current reading unit 8, a PWM controller 12A, a control period calculation unit 15C, a synchronous PWM signal generation unit 16, a control interrupt generation unit 17, and a current control unit 19. Note that the coordinate transformation units 7 and 11, the synchronous PWM signal generation unit 16, and the current control unit 19 constitute the control calculation unit 13C.

[0127] The synchronous PWM signal generation unit 16 calculates the firing angles αu*, αv*, and αw* used for synchronous PWM based on the dq-axis voltage command values ​​Vd* and Vq* from the current control unit 19 and the DC voltage Vdc from the DC voltage detection device 6, and outputs them to the PWM controller 12A.

[0128] Furthermore, the synchronous PWM signal generation unit 16 calculates the modulation rate based on the dq-axis voltage command values ​​Vd*,Vq* from the current control unit 19 and the DC voltage Vdc from the DC voltage detection device 6. Based on the calculated modulation rate, it determines whether it is synchronous PWM control (hereinafter referred to as synchronous PWM mode) or asynchronous PWM control (hereinafter referred to as asynchronous PWM mode), and outputs a PWM mode signal indicating which mode is being used to the control period calculation unit 15C and the PWM controller 12A.

[0129] When the PWM mode signal from the synchronous PWM signal generation unit 16 indicates synchronous PWM mode, the PWM controller 12A generates a gate signal by comparing the firing angles αu*, αv*, αw* from the synchronous PWM signal generation unit 16 with the magnetic pole position θ* from the magnetic pole position detector 4. On the other hand, when the PWM mode signal from the synchronous PWM signal generation unit 16 indicates asynchronous PWM mode, the PWM controller 12A calculates a duty cycle signal based on the three-phase voltage command values ​​Vu*, Vv*, Vw* from the coordinate transformation unit 11 and the DC voltage information Vdc from the DC voltage detection device 6, generates a gate signal by comparing the calculated duty cycle signal with the carrier wave, and outputs it to the power converter 2.

[0130] Figure 15 is a functional block diagram showing the processing details of the control period calculation unit.

[0131] As shown in Figure 15, the control period calculation unit 15C calculates the control period fc based on the speed information ω1* from the frequency calculation unit 5 and the modulation rate Mf from the modulation rate calculation unit 18, and outputs it to the control interrupt generation unit 17. It is roughly composed of a control period selection unit (normal operation) 21, a control period selection unit 24 (synchronous PWM operation), and a switching unit 22.

[0132] The control period selection unit (synchronous PWM) 24 selects the control period fc to be used in synchronous PWM mode based on a lookup table. The control period selection unit (synchronous PWM) 24 uses, for example, the lookup table shown in Figure 2 of the first embodiment. This selects a control period fc that avoids current oscillations that are a concern when controlling with synchronous PWM.

[0133] The control period selection unit (normal operation) 21 selects the control period fc to be used in asynchronous PWM mode according to the lookup table. In the control period selection unit (normal operation), for example, the lookup table shown in Figure 13 of the third embodiment is used. This allows the control period fc to be set higher in asynchronous PWM mode, where there is no need to consider current oscillation suppression and no need to consider the prohibited band.

[0134] If the PWM mode signal from the synchronous PWM signal generation unit 16 indicates synchronous PWM mode, the switching unit 22 switches the output to the control interrupt generation unit 17 to the control period fc selected by the control period selection unit (synchronous PWM) 24. If the PWM mode signal from the synchronous PWM signal generation unit 16 indicates asynchronous PWM mode, the switching unit 22 switches the output to the control interrupt generation unit 17 to the control period fc selected by the control period selection unit (normal) 21.

[0135] The other configurations are the same as in the first embodiment.

[0136] In this embodiment configured as described above, the same effects as in the first embodiment can be obtained.

[0137] Furthermore, in this embodiment, considering that 6f oscillations do not occur in the dq-axis current in the normal region, the control period fc is selected to avoid the prohibited band only in the overmodulation region where oscillations occur. As a result, oscillations can be suppressed during synchronous PWM, and the control period fc in normal conditions can be set higher, thereby improving control performance.

[0138] <Fifth Embodiment> A fifth embodiment of the present invention will be described with reference to Figures 16 and 17.

[0139] This embodiment shows a case where the gain of current control is changed according to the cutoff frequency calculated based on speed information.

[0140] In this embodiment, only the differences from the third embodiment will be described. In the drawings used in this embodiment, the same reference numerals are used for components similar to those in the third embodiment, and their descriptions are omitted as appropriate.

[0141] Figure 16 is a functional block diagram showing the motor control device according to this embodiment, along with its related components.

[0142] As shown in Figure 16, the motor control device 100D controls the power supplied to the PMSM1 from the DC voltage source 9 via the power converter 2, and is generally composed of a phase current detector 3, a magnetic pole position detector 4, a frequency calculation unit 5, a DC voltage detection device 6, coordinate transformation units 7 and 11, a current reading unit 8, a PWM controller 12, a current control cutoff frequency calculation unit 14, a control period calculation unit 15D, a control interrupt generation unit 17, a modulation rate calculation unit 18, and a current control unit 19A. Note that the coordinate transformation units 7 and 11, the modulation rate calculation unit 18, and the current control unit 19A constitute the control calculation unit 13D.

[0143] The current control cutoff frequency calculation unit 14 calculates and modifies the current control cutoff frequency facr according to the speed information ω1* from the frequency calculation unit 5, and outputs it to the current control unit 19.

[0144] The current control unit 19A changes the gain of the current control according to the current control cutoff frequency facr from the current control cutoff frequency calculation unit 14.

[0145] Figure 17 is a functional block diagram showing the processing details of the control period calculation unit.

[0146] As shown in Figure 17, the control period calculation unit 15D calculates the control period fc based on the speed information ω1* from the frequency calculation unit 5, the modulation rate Mf from the modulation rate calculation unit 18, and the current control cutoff frequency facr from the current control cutoff frequency calculation unit 14, and outputs it to the control interrupt generation unit 17. It is roughly composed of a modulation mode determination unit 25 and control period selection units 31 and 33.

[0147] The modulation mode determination unit 25 determines whether or not the control of the PMSM1 is being performed in the overmodulation region based on the modulation rate Mf.

[0148] The control period selection unit 31 selects the control period fc according to, for example, the lookup table shown in Figure 13. The lookup table shown in Figure 13 sets the control period fc when it is not necessary to consider the prohibited bandwidth.

[0149] If the modulation mode determination unit 25 determines that the system is being controlled in an overmodulation region, the control period selection unit 33 changes the control period (before restriction) from the control period selection unit 31 based on the current control cutoff frequency facr from the current control cutoff frequency calculation unit 14 so that it does not fall into the prohibited band (see (Equation 5) and Figure 13 in the first embodiment, etc.), and outputs it as the control period fc to the control interrupt generation unit 17.

[0150] The other configurations are the same as in the third embodiment.

[0151] In this embodiment configured as described above, the same effects as those of the third embodiment can be obtained.

[0152] Furthermore, in this embodiment, it is possible to counter even when the current control cutoff frequency is variable, and current oscillations can be suppressed even if the current control cutoff frequency facr changes according to the speed information ω1*.

[0153] In this embodiment, a configuration in which the current control cutoff frequency is changed according to the speed information ω1* was described as an example, but the embodiment is not limited to this, and the same effects as in this embodiment can be obtained by changing the current control cutoff frequency according to other information, such as the torque command value.

[0154] <Sixth Embodiment> A sixth embodiment of the present invention will be described with reference to Figures 18 and 19.

[0155] This embodiment shows a case where the current reading signal is controlled to turn ON with twice the period of the interrupt signal, and control is performed according to the difference between two three-phase currents read in response to the current reading signal, and the gain of the current control is changed according to the cutoff frequency calculated based on the speed information.

[0156] In this embodiment, only the differences from the second embodiment will be described. In the drawings used in this embodiment, the same reference numerals are used for components similar to those in the second embodiment, and their descriptions are omitted as appropriate.

[0157] Figure 18 is a functional block diagram showing the motor control device according to this embodiment, along with its related components.

[0158] As shown in Figure 18, the motor control device 100E controls the power supplied to the PMSM1 from the DC voltage source 9 via the power converter 2, and is generally composed of a phase current detector 3, a magnetic pole position detector 4, a frequency calculation unit 5, a DC voltage detection device 6, coordinate transformation units 7B and 11, a current reading unit 8A, a PWM controller 12, a current control cutoff frequency calculation unit 14, a control period calculation unit 15E, a control interrupt generation unit 17A, and a current control unit 19A. Note that the coordinate transformation units 7B and 11 and the current control unit 19A constitute the control calculation unit 13E.

[0159] The control interrupt generation unit 17A generates and outputs a current reading signal cl such that it turns ON at twice the period of the interrupt signal int.

[0160] The current control cutoff frequency calculation unit 14 calculates and modifies the current control cutoff frequency facr according to the speed information ω1* from the frequency calculation unit 5, and outputs it to the current control unit 19.

[0161] The current control unit 19A changes the gain of the current control according to the current control cutoff frequency facr from the current control cutoff frequency calculation unit 14.

[0162] The control period calculation unit 15E selects the control period fc according to, for example, the lookup table shown in Figure 13. The lookup table shown in Figure 13 sets the control period fc when it is not necessary to consider the prohibited bandwidth.

[0163] The coordinate transformation unit 7B determines whether the control period fc falls within the prohibited band (see equations 12 and 13 in the second embodiment, Figure 9, etc.) based on the velocity information ω1* from the frequency calculation unit 5, the control period fc from the control period calculation unit 15E, and the current control cutoff frequency facr from the current control cutoff frequency calculation unit 14.

[0164] If the coordinate transformation unit 7B determines that the control period fc is not within the prohibited band, it corrects the timing difference between the readings of the three-phase currents Iuc1, Ivc1, Iwc1 and Iuc2, Ivc2, Iwc2 into rotation angles and transforms them into dq coordinates. It then averages the transformed dq axis currents and outputs the detected dq axis currents Idc and Iqc to the current control unit 19A.

[0165] Furthermore, if the coordinate transformation unit 7B determines that the control period fc is in the prohibited band, it switches to a method of alternately calculating the dq axis current detection values ​​Idc and Iqc, and outputs the calculated dq axis current detection values ​​Idc and Iqc to the current control unit 19A.

[0166] Figure 19 shows the relationship between the three-phase fundamental wave current, the dq-axis sixth harmonic current, and the reading timing. Figure 19 shows the three-phase fundamental wave current 61C, the dq-axis sixth harmonic current 63C, and the reading timing.

[0167] As shown in Figure 19, in the method of alternately calculating the dq axis current detection values ​​Idc and Iqc, the value of (n) is used in interrupt 1, the value of (n+3) is used in interrupt 2, and the value of (n+4) is used in interrupt 3. As a result, the 6th-order component appears as a harmonic component on the dq axis, making it possible to move it outside the response band with current control. In other words, the control period can be kept constant, and the control response can be improved.

[0168] The other configurations are the same as in the second embodiment.

[0169] In this embodiment configured as described above, the same effects as in the second embodiment can be obtained.

[0170] <Note> It should be noted that the present invention is not limited to the embodiments described above, and includes various modifications and combinations that do not depart from the spirit of the invention. Furthermore, the present invention is not limited to having all the configurations described in the embodiments described above, and includes those in which some of the configurations have been omitted.

[0171] For example, the configurations of the third and fourth embodiments may be combined to avoid the prohibited bandwidth only during synchronous PWM or in the overmodulation region. A common characteristic between synchronous PWM and the overmodulation region is that the switching period is not constant. Therefore, considering both synchronous PWM and the overmodulation region, it can be said that the prohibited bandwidth should only be avoided in the region where the switching period is not constant. [Explanation of Symbols]

[0172] 2...Power converter, 3...Phase current detector, 4...Magnetic pole position detector, 5...Frequency calculation unit, 6...DC voltage detection device, 7,7A,7B...Coordinate transformation unit, 8,8A...Current reading unit, 9...DC voltage source, 11...Coordinate transformation unit, 12,12A...PWM controller, 13,13A,13B,13C,13D,13E...Control calculation unit, 14...Current control cutoff frequency calculation unit, 15,15A,15B,15C,15D,15E...Control period calculation unit, 16...Synchronized PWM signal generation unit, 17,17A... Control interrupt generation unit, 18... Modulation rate calculation unit, 19, 19A... Current control unit, 21, 23, 24, 33... Control period selection unit, 22... Switching unit, 25... Modulation mode determination unit, 31... Control period selection unit, 61, 61C... Three-phase fundamental wave current, 63, 63C... dq-axis 6th harmonic current, 100, 100A, 100B, 100C, 100D, 100E... Motor control device, 101... Transmission, 103... Differential gear, 105... Drive shaft, 107... Wheel

Claims

1. A motor control device that outputs a voltage to be applied to an AC motor based on an output voltage command value, A control period calculation unit that determines the control period according to the fundamental frequency of the AC motor and the cutoff frequency of the current control, A control interrupt generation unit that determines a control interrupt based on the control period determined by the control period calculation unit, Based on the control interrupt determined by the control interrupt generation unit, a current control unit controls the output voltage command value, A PWM controller that controls the voltage output by PWM according to the output voltage command value controlled by the current control unit. Equipped with, The motor control device is characterized in that the control period calculation unit determines the control period to be greater than the range of the 6th order of the fundamental wave frequency ± 10 times the cutoff frequency of the current control.

2. In the motor control device according to claim 1, The motor control device is characterized in that the control period calculation unit determines the control period to be greater than the range of 6nth order of the fundamental wave frequency ± 10 times the cutoff frequency of the current control only when the switching period of the PWM controller is not constant.

3. In the motor control device according to claim 1, The motor control device is characterized in that the control period calculation unit determines the control period according to the fundamental frequency of the AC motor, the cutoff frequency of the current control, and the modulation rate.

4. In the motor control device according to claim 1, The motor control device is characterized in that the current control unit changes the gain of current control based on the control interrupt determined by the control interrupt generation unit and the cutoff frequency.

5. An electric vehicle equipped with an AC motor, An electric drive vehicle characterized by comprising a motor control device according to any one of claims 1 to 4, which outputs a voltage to be applied to the AC motor.

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