Electric power conversion device

Abstract

This electric power conversion device (50) comprises: a switching circuit (4) in which a plurality of legs (41) are connected in parallel, each of the plurality of legs (41) having an upper arm (S1) and a lower arm (S2) connected in series; a magnetic coupling reactor (3) in which a number of windings (31, 32, 33, 34) equal to the number of legs (41) are magnetically coupled to each other; and a control circuit (7) that performs PWM control at a phase shift of 180 degrees and drives the plurality of legs (41). In the magnetic coupling reactor (3): one end of each of the windings (31, 32, 33, 34) is connected to an input terminal (18a); the other end of each of the windings (31, 32, 33, 34) is connected to a connection point (Na) between the upper arm (S1) and the lower arm (S2) in the corresponding leg (41); and the degree of magnetic coupling between windings (31, 33 and 32, 34) connected to legs (41) having the same PWM phase, which is the phase of the PWM control, is higher than the degree of magnetic coupling between windings connected to legs having different PWM phases.

Description

Power Conversion Device

[0001] The present application relates to a power conversion device.

[0002] There is a demand for higher output and smaller size of power conversion devices, and there is a particularly strong demand for smaller magnetic components and capacitors, which are passive components that occupy a large volume of the device. For example, Patent Document 1 discloses a four-phase tightly coupled non-isolated boost DC-DC converter. The DC-DC converter in Patent Document 1 includes a boost inductor and a four-phase tightly coupled inductor, and by using a tightly coupled inductor with four windings wound around a magnetic core, the number of inductors or reactors, which are magnetic components for boosting, is reduced.

[0003] JP 2017-195726 A

[0004] The DC-DC converter in Patent Document 1 is an example of a multi-parallel power converter. To miniaturize a multi-parallel power converter equipped with multiple switching elements that energize multiple phase windings at different phases, or a multi-parallel power converter in which multiple windings and switching elements energized at different phases are connected in parallel, miniaturization of magnetic components and passive components such as capacitors, which occupy a large volume of the device, is necessary. In particular, when power converters such as multi-parallel power converters and multi-parallel power converters are increased in output, the main circuit current increases, making it necessary to increase the cross-sectional area of ​​the magnetic material to prevent magnetic saturation. For this reason, these power converters wind the windings around larger magnetic cores, which inevitably increases winding loss. Furthermore, when the current flowing through a capacitor increases, the amount of heat generated increases due to the equivalent series resistance (ESR) within the capacitor. Therefore, these power converters require large-capacity capacitors that reduce the ESR within the capacitor and suppress heat generation.

[0005] The DC-DC converter of Patent Document 1 has four windings wound around a magnetic core, and discloses that the mutual inductance between the windings increases inductance to reduce the current amplitude (current ripple) in the winding of the tightly coupled inductor. However, the DC-DC converter of Patent Document 1 has a boost inductor connected to the input side of the tightly coupled inductor, and the tightly coupled inductor and the boost inductor form a boost reactor. In this case, the current amplitude in the winding is reduced by the combined inductance of the tightly coupled inductor and the boost inductor. Therefore, the current amplitude in the winding of the tightly coupled inductor does not necessarily decrease if there is no boost inductor. Furthermore, the DC-DC converter of Patent Document 1 does not disclose a technology for reducing the current amplitude in the winding by utilizing the magnitude relationship of the magnetic coupling between the windings, i.e., the magnitude relationship of the degree of coupling.

[0006] The technology disclosed in this specification aims to increase the inductance and reduce the current amplitude in the windings by using the magnitude relationship of the degree of coupling between the windings.

[0007] The power conversion device disclosed in this specification as an example converts a DC or AC voltage input between two input terminals into DC power of a predetermined voltage and outputs the DC power, and includes: a switching circuit in which a plurality of legs are connected in parallel, each of which has an upper arm and a lower arm, each having a switching element, connected in series; a magnetic coupling reactor in which the same number of windings as the number of legs are magnetically coupled to each other; and a control circuit that drives the plurality of legs by performing PWM control with phases shifted by 180 degrees. One end of each winding of the magnetic coupling reactor is connected to the input terminal, and the other end of each winding is connected to a connection point between the upper arm and the lower arm of the corresponding leg, and the degree of coupling between the windings connected to legs having the same PWM phase, which is the phase of the PWM control, is higher than the degree of coupling between the windings connected to legs having different PWM phases.

[0008] An example of a power conversion device disclosed in the present specification includes a magnetic coupling reactor on the input side of a switching circuit, and the degree of coupling between windings connected to legs with the same PWM phase, which is the phase of PWM control, is higher than the degree of coupling between windings connected to legs with different PWM phases. Therefore, the magnitude relationship of the degree of coupling between the windings can be used to increase inductance and reduce the current amplitude in the windings.

[0009] 1 is a diagram illustrating a configuration of a power conversion apparatus according to a first embodiment. FIG. 1 is a diagram illustrating a configuration of a leg of FIG. 1. FIG. 2 is a diagram illustrating a voltage of the magnetic coupling reactor of FIG. 1. FIG. 3 is a diagram illustrating a DC magnetic flux of the magnetic coupling reactor of FIG. 1. FIG. 4 is a diagram illustrating a first example of AC magnetic flux in the magnetic coupling reactor of FIG. 1. FIG. 5 is a diagram illustrating a second example of AC magnetic flux in the magnetic coupling reactor of FIG. 1. FIG. 6 is a diagram illustrating a first example of operation timing of a lower arm in the switching circuit of FIG. 1. FIG. 7 is a diagram illustrating a second example of operation timing of the lower arm in the switching circuit of FIG. 1. FIG. 8 is a diagram illustrating a first example of a gradient of a current occurring in the winding of FIG. 1. FIG. 9 is a diagram illustrating a second example of a gradient of a current occurring in the winding of FIG. 1. FIG. 10 is a diagram illustrating a third example of a gradient of a current occurring in the winding of FIG. 1. FIG. 11 is a diagram illustrating a fourth example of a gradient of a current occurring in the winding of FIG. 1. FIG. 12 is a diagram illustrating a relationship between an upper limit value of the degree of coupling between windings and a duty ratio according to the first embodiment. FIG. 13 is a diagram illustrating a first example of a control signal waveform for analyzing a current amplitude according to the first embodiment. FIG. 14 is a diagram illustrating a first example of a current waveform of a comparative example. FIG. 15 is a diagram illustrating a first example of a current waveform according to the first embodiment. FIG. 16 is a diagram illustrating a first example of a total current waveform according to the first embodiment. 27 and 28. FIG. 28 is a diagram showing a second example of a control signal waveform for analyzing a current amplitude according to the first embodiment. FIG. 29 is a diagram showing a second example of a current waveform of a comparative example. FIG. 29 is a diagram showing a second example of a current waveform according to the first embodiment. FIG. 30 is a diagram showing a second example of a total current waveform according to the first embodiment. FIG. 31 is a diagram showing a configuration of a magnetic coupling reactor according to the first embodiment. FIG. 32 is a diagram showing a DC magnetic flux corresponding to the magnetic coupling reactor of FIG. 22. FIG. 33 is a diagram showing the current amplitude of the power conversion device according to the first embodiment. FIG. 34 is a diagram showing an example of a hardware configuration that realizes the function of the control circuit of FIG. 1. FIG. 35 is a diagram showing a first example of a configuration of a magnetic coupling reactor according to the second embodiment. FIG. 36 is a diagram showing a second example of a configuration of a magnetic coupling reactor according to the second embodiment. FIG. 37 is a diagram showing the DC magnetic flux corresponding to the magnetic coupling reactor of FIGS. 26 and 27. FIG. 38 is a diagram showing a configuration of a power conversion device according to the third embodiment. FIG. 39 is a diagram showing the operation of the control circuit according to the third embodiment. FIG. 39 is a diagram showing a first example of a DC magnetic flux of the magnetic coupling reactor when the output current according to the third embodiment is equal to or greater than a set value. FIG. 39 is a diagram showing a first example of a DC magnetic flux of the magnetic coupling reactor when the output current according to the third embodiment is less than a set value.FIG. 40 is a diagram showing a second example of the DC magnetic flux of the magnetic coupling reactor when the output current is equal to or greater than a set value according to embodiment 3. FIG. 41 is a diagram showing a second example of the DC magnetic flux of the magnetic coupling reactor when the output current is less than a set value according to embodiment 3. FIG. 42 is a diagram showing a first example of the AC magnetic flux of the magnetic coupling reactor when the output current is equal to or greater than a set value according to embodiment 3. FIG. 43 is a diagram showing a first example of the AC magnetic flux of the magnetic coupling reactor when the output current is less than a set value according to embodiment 3. FIG. 44 is a diagram showing a second example of the AC magnetic flux of the magnetic coupling reactor when the output current is equal to or greater than a set value according to embodiment 3. FIG. 45 is a diagram showing a second example of the AC magnetic flux of the magnetic coupling reactor when the output current is less than a set value according to embodiment 3. FIG. 46 is a diagram showing a configuration of a power conversion device according to embodiment 4. FIG. 47 is a diagram showing a configuration of a power conversion device according to embodiment 5. FIG. 48 is a diagram showing the voltage of the magnetic coupling reactor of FIG. 40.

[0010] Embodiment 1. FIG. 1 is a diagram showing the configuration of a power conversion apparatus according to embodiment 1, and FIG. 2 is a diagram showing the configuration of a leg of FIG. 1. FIG. 3 is a diagram showing the voltage of the magnetic coupling reactor of FIG. 1. FIG. 4 is a diagram showing the DC magnetic flux of the magnetic coupling reactor of FIG. 1. FIG. 5 is a diagram showing a first example of the AC magnetic flux in the magnetic coupling reactor of FIG. 1, and FIG. 6 is a diagram showing a second example of the AC magnetic flux in the magnetic coupling reactor of FIG. 1. FIG. 7 is a diagram showing a first example of the operation timing of the lower arm in the switching circuit of FIG. 1, and FIG. 8 is a diagram showing a second example of the operation timing of the lower arm in the switching circuit of FIG. 1. FIG. 9 is a diagram showing a first example of the gradient of the current generated in the winding of FIG. 1, and FIG. 10 is a diagram showing a second example of the gradient of the current generated in the winding of FIG. 1. FIG. 11 is a diagram showing a third example of the gradient of the current generated in the winding of FIG. 1, and FIG. 12 is a diagram showing a fourth example of the gradient of the current generated in the winding of FIG. 1. FIG. 13 is a diagram showing the relationship between the upper limit value of the degree of coupling between windings and the duty ratio according to embodiment 1. FIG. 14 is a diagram showing a first example of a control signal waveform for analyzing current amplitude according to embodiment 1. FIG. 15 is a diagram showing a first example of a current waveform of a comparative example, FIG. 16 is a diagram showing a first example of a current waveform according to embodiment 1, and FIG. 17 is a diagram showing a first example of a total current waveform according to embodiment 1. FIG. 18 is a diagram showing a second example of a control signal waveform for analyzing current amplitude according to embodiment 1. FIG. 19 is a diagram showing a second example of a current waveform of a comparative example, FIG. 20 is a diagram showing a second example of a current waveform according to embodiment 1, and FIG. 21 is a diagram showing a second example of a total current waveform according to embodiment 1. FIG. 22 is a diagram showing the configuration of a magnetic coupling reactor according to embodiment 1, and FIG. 23 is a diagram showing DC magnetic flux corresponding to the magnetic coupling reactor of FIG. 22. FIG. 24 is a diagram showing the current amplitude of a power conversion device according to embodiment 1, and FIG. 25 is a diagram showing an example of a hardware configuration that realizes the function of the control circuit of FIG. 1.

[0011] The power conversion device 50 shown in Fig. 1 is an example of a four-phase non-isolated boost DC-DC converter that outputs DC power by boosting an input DC voltage to a predetermined voltage. Fig. 1 also illustrates a DC voltage source 1 and connected device 6 along with the power conversion device 50. The power conversion device 50 includes input terminals 18a and 18b to which DC power is input, an input capacitor 2, a magnetic coupling reactor 3, a switching circuit 4, an output capacitor 5, a control circuit 7 that controls the switching circuit 4, and output terminals 19a and 19b that output the boosted DC power. The DC voltage source 1 is connected to the input terminals 18a and 18b, and connected devices 6, such as a load and a chargeable / dischargeable DC voltage source, are connected to the output terminals 19a and 19b. In Fig. 1, the reference numeral 50 for the power conversion device is underlined with leading lines within a dashed frame that indicates the switching circuit 4, which is one of the components of the power conversion device.

[0012] The magnetic coupling reactor 3 has the same number of windings as the legs 41, which are magnetically coupled to each other. Here, an example will be described in which four windings 31, 32, 33, and 34 are wound around a magnetic core 35. The switching circuit 4 has a plurality of legs 41, each of which has an upper arm S1 and a lower arm S2 connected in series between the positive wiring 15 and the negative wiring 16 on the output side. The plurality of legs 41 are connected in parallel. That is, the switching circuit 4 has a plurality of legs 41a, 41b, 41c, and 41d connected in parallel. FIG. 1 shows the switching circuit 4 having four phase legs, namely, A phase, B phase, C phase, and D phase. The A phase leg 41a and the B phase leg 41b are referred to as a first circuit 48a, and the C phase leg 41c and the D phase leg 41d are referred to as a second circuit 48b. The first circuit 48a and the second circuit 48b are connected in parallel on the side of the output terminals 19a, 19b to which DC power is output. In FIGS. 1 and 2, an example is shown in which the upper arm S1 and the lower arm S2 each have one switching element. The switching element connected to the positive side wiring 15 is referred to as the upper arm S1, and the switching element connected to the negative side wiring 16 is referred to as the lower arm S2. When the upper arm and the lower arm are referred to simultaneously, they are referred to as the upper and lower arms. Here, the reference numeral S1 is also used for the switching element of the upper arm S1, and the reference numeral S2 is also used for the switching element of the lower arm S2. The A-phase leg 41a includes an upper-arm switching element Sa1 and a lower-arm switching element Sa2, and the B-phase leg 41b includes an upper-arm switching element Sb1 and a lower-arm switching element Sb2. The C-phase leg 41c includes an upper arm switching element Sc1 and a lower arm switching element Sc2, and the D-phase leg 41d includes an upper arm switching element Sd1 and a lower arm switching element Sd2.

[0013] Four-phase legs 41a, 41b, 41c, and 41d of the switching circuit 4 are connected in parallel to the output capacitor 5 and connected device 6. The legs are generally designated by 41, with 41a, 41b, 41c, and 41d used when distinguishing between them. The upper arm and its switching elements are generally designated by S1, with Sa1, Sb1, Sc1, and Sd1 used when distinguishing between them. The lower arm and its switching elements are generally designated by S2, with Sa2, Sb2, Sc2, and Sd2 used when distinguishing between them.

[0014] The control circuit 7 outputs a control signal sig1 to the switching circuit 4. The control signal sig1 includes a control signal sigc1 output to the switching element S1 of each phase and a control signal sigc2 output to the switching element S2 of each phase. In the case of the switching circuit 4 including the four-phase legs 41, the control signal sig1 includes A-phase control signals sigc1 and sigc2, B-phase control signals sigc1 and sigc2, C-phase control signals sigc1 and sigc2, and D-phase control signals sigc1 and sigc2.

[0015] As shown in FIG. 2 , leg 41 includes a positive-side switching element S1 and a negative-side switching element S2 connected in series between positive-side wiring 15 and negative-side wiring 16. In FIGS. 1 and 2 , switching elements S1 and S2 are illustrated as MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors). The switching elements S1 and S2 each include a drain terminal d, a source terminal s, and a gate terminal g. Each switching element S1 and S2 includes a transistor Tr and a diode Di. The diode Di may be a separate element from the transistor Tr or may be a parasitic diode. A control signal sigc1 output from the control circuit 7 is input to the gate terminal g of switching element S1, and a control signal sigc2 output from the control circuit 7 is input to the gate terminal g of switching element S2. The drain terminal d of switching element S1 is connected to the positive-side wiring 15, the source terminal s of switching element S1 is connected to the drain terminal d of switching element S2, and the source terminal s of switching element S2 is connected to the negative-side wiring 16. A winding of the magnetic coupling reactor 3 is connected to a connection point where the source terminal s of the switching element S1 and the drain terminal d of the switching element S2 are connected, i.e., an arm connection point Na. The connection points between the upper arm and the lower arm in the leg are generally designated by the symbol Na, and when they need to be distinguished, Naa, Nab, Nac, and Nad are used.

[0016] The switching elements S1 and S2 perform an opening / closing operation, i.e., an on / off operation including an on operation and an off operation, in response to control signals sigc1 and sigc2 output from the control circuit 7, and output the boosted DC power between the positive side wiring 15 and the negative side wiring 16. Note that although the switching elements Sa1, Sa2, Sb1, Sb2, Sc1, Sc2, Sd1, Sd2, S1, and S2 are MOSFETs, this is not a limitation and switching elements other than MOSFETs, such as IGBTs (Insulated Gate Bipolar Transistors) and thyristors, may also be used.

[0017] The magnetic coupling reactor 3 includes a magnetic core 35 and a plurality of windings 31, 32, 33, and 34 wound around the magnetic core 35. One ends of the windings 31, 32, 33, and 34 are connected to one another at a winding connection point Ne, which is connected to an input terminal 18a connected to the positive side of the DC voltage source 1 via a positive side wiring 14 on the input side. An input terminal 18b connected to the negative side of the DC voltage source 1 is connected to an output terminal 19b via a negative side wiring 16. The terminal of the upper arm of each phase of the switching circuit 4 opposite the arm connection point Na, i.e., the drain terminal d of the switching element S1, is connected to the output terminal 19a via a positive side wiring 15. The input capacitor 2 is connected between the positive side wiring 14 and the negative side wiring 16 and is connected in parallel to the DC voltage source 1. The output capacitor 5 is connected between the positive side wiring 15 and the negative side wiring 16 on the output terminals 19a and 19b side and is connected in parallel to the connected device 6.

[0018] The other end of the winding 31 opposite to the winding connection point Ne is connected to an arm connection point Naa of the switching elements Sa1 and Sa2 of the A-phase leg 41 a in the switching circuit 4. Similarly, the other end of the winding 32 opposite to the winding connection point Ne is connected to an arm connection point Nab of the switching elements Sb1 and Sb2 of the B-phase leg 41 b in the switching circuit 4. The other end of the winding 33 opposite to the winding connection point Ne is connected to an arm connection point Nac of the switching elements Sc1 and Sc2 of the C-phase leg 41 c in the switching circuit 4. The other end of the winding 34 opposite to the winding connection point Ne is connected to an arm connection point Nad of the switching elements Sd1 and Sd2 of the D-phase leg 41 d in the switching circuit 4.

[0019] The power conversion device 50 includes a voltage sensor 11 that measures an input voltage Vin, which is the voltage across the input capacitor 2 or the voltage between the input terminals 18a and 18b, and a voltage sensor 12 that measures an output voltage Vout, which is the voltage across the output capacitor 5 or the voltage between the output terminals 19a and 19b. The power conversion device 50 also includes a current sensor 13a that measures a winding current i1, which is the current through the winding 31, a current sensor 13b that measures a winding current i2, which is the current through the winding 32, a current sensor 13c that measures a winding current i3, which is the current through the winding 33, a current sensor 13d that measures a winding current i4, which is the current through the winding 34, and a current sensor 20 that measures an output current Iout, which is the current output from the output terminal 19a to the connected device 6. The winding currents i1, i2, i3, and i4 of the windings 31 to 34 of the magnetic coupling reactor 3 are collectively referred to as a winding current Iw as appropriate. 1 shows an example of voltage sensor 11 that measures the voltage across input capacitor 2, and voltage sensor 12 that measures the voltage across output capacitor 5. Also shown is an example of current sensors 13a to 13d that measure the current in the wiring that connects the other ends of windings 31, 32, 33, and 34 to arm connection points Naa, Nab, Nac, and Nad, respectively, and current sensor 20 that measures the current in positive side wiring 15 between the positive side of output capacitor 5 and output terminal 19a.

[0020] The control circuit 7 generates a control signal sig1 that controls the switching circuit 4 based on the input voltage Vin, output voltage Vout, output current Iout, and winding current Iw. The control circuit 7 controls the switching circuit 4 using the control signal sig1 so that the output voltage Vout becomes a target voltage value. Specifically, the control circuit 7 adjusts the duty ratio D according to equation (1) to perform PWM (Pulse Width Modulation) control, maintaining the voltages of the output capacitor 5 and connected device 6 at predetermined target voltages while controlling the currents flowing through the windings 31-34, i.e., the winding currents i1, i2, i3, and i4, so that they flow evenly. Here, "evenly flowing winding currents" means that the differences between the winding currents i1, i2, i3, and i4 are small. In other words, the differences between the winding currents are within a predetermined tolerance. The control circuit 7 controls the lower arms Sa2, Sb2, Sc2, and Sd2 of the legs 41a to 41d for phases A to D, i.e., the switching elements Sa2, Sb2, Sc2, and Sd2, and the upper arms Sa1, Sb1, Sc1, and Sd1 of the legs 41a to 41d for phases A to D, i.e., the switching elements Sa1, Sb1, Sc1, and Sd1, so that they are complementarily turned on and off without being turned on simultaneously with the lower arms of each phase. A dead time is provided in the control signal sig1 to prevent the lower arms and upper arms of each phase from being turned on simultaneously. The duty ratio D represents the ratio of the on time to the period T of the control signal sig1. Vout on the left side of equation (1) represents the target value of the output voltage Vout.

[0021]

[0022] The control circuit 7 includes an operating leg determination unit 25, a control signal generation unit 26, and a drive circuit 28. The operating leg determination unit 25 determines which leg to operate from phases A to D of the switching circuit 4 based on the input voltage Vin, output voltage Vout, output current Iout, and winding current Iw, and generates operating leg information sigm, which is information about the leg to operate. The control signal generation unit 26 generates a digital control signal sig1 that controls the operating leg determined based on the operating leg information sigm. The drive circuit 28 converts the digital control signal sig1 into predetermined high and low voltages and outputs the control signal sig1 to the switching circuit 4. For example, a high voltage corresponding to a digital signal "1" and a low voltage corresponding to a digital signal "0" are set according to the switching elements S1 and S2 of the leg 41. Note that the functions of the operating leg determination unit 25 and the control signal generation unit 26, which are functional blocks of the control circuit 7, may be realized by a processor 98 and a memory 99 shown in FIG. 25 . 25 is a diagram showing an example of a hardware configuration that realizes the functional blocks of the control circuit 7. In this case, the operating leg determination unit 25 and the control signal generation unit 26 are realized by a processor 98 executing a program stored in a memory 99. Furthermore, a plurality of processors 98 and a plurality of memories 99 may cooperate to execute each function.

[0023] Next, the operation of the power conversion device 50 of the first embodiment will be described in detail. In the power conversion device 50 of the first embodiment, the control circuit 7 performs PWM control with phases shifted by 180° to drive the multiple legs 41a to 41d. The PWM phase, which is the phase of PWM control, is shifted by 180° between the A phase and the B phase, and similarly between the C phase and the D phase. For example, the A phase to the D phase legs 41a to 41d constituting the switching circuit 4 are driven such that the PWM phase for the A phase is 0°, the B phase is 180°, the C phase is 0°, and the D phase is 180°, respectively. Where appropriate, the 0° PWM phase will be referred to as the 0° phase, and the 180° PWM phase will be referred to as the 180° phase.

[0024] Next, the configuration of the magnetic coupling reactor 3 will be described using Fig. 4. Fig. 4 is a diagram showing the direction of DC magnetic flux generated in the magnetic core 35 when DC current flows through each of the windings 31 to 34 of the magnetic coupling reactor 3. The direction of the DC current is from the winding connection point Ne, which is the connection point on the DC voltage source 1 side, to the arm connection points Naa to Nad of the legs 41a to 41d. The direction of the DC magnetic flux is also the direction of the magnetomotive force caused by the DC current.

[0025] The DC magnetic fluxes generated in the windings 31 to 34 are designated Φdc1 to Φdc4, respectively. The DC magnetic fluxes Φdc1 and Φdc3 are connected by forward coupling, in which the DC magnetic fluxes reinforce each other, i.e., add to each other. Similarly, the DC magnetic fluxes Φdc2 and Φdc4 are connected by forward coupling, in which the DC magnetic fluxes reinforce each other, i.e., add to each other. Furthermore, the pair of DC magnetic fluxes Φdc1 and Φdc3, and the pair of DC magnetic fluxes Φdc2 and Φdc4 are in an inverse coupling relationship, in which the same polarities, i.e., positive polarities, or negative polarities, face each other, via magnetic resistances R1 and R2 due to gaps g1 and g2 (see FIG. 22) provided in the magnetic core 35. In an inverse coupling relationship, the DC magnetic fluxes weaken each other. That is, windings 31 and 33 are wound to form a first forward coupling, and windings 32 and 34 are wound to form a second forward coupling that is opposite to the first forward coupling. When windings 31 to 34 are wound in this manner around magnetic core 35 having gaps g1 and g2, the DC magnetic flux generated in magnetic core 35 acts to cancel out.

[0026] Next, using Figures 5 and 6, we will explain the AC magnetic flux, which is the circulating magnetic flux generated in the magnetic core 35 of the magnetic coupling reactor 3 when the switching circuit 4 is operating. Figure 5 shows the state of magnetic flux circulating clockwise through the magnetic core 35, and Figure 6 shows the state of magnetic flux circulating counterclockwise. Consider a case where PWM control is performed with the A-phase leg 41a and the C-phase leg 41c of the switching circuit 4 in a 0° phase relationship, and the B-phase leg 41b and the D-phase leg 41d in a 180° phase relationship. If the AC magnetic fluxes generated by the windings 31 to 34 when the switching circuit 4 is operating are designated Φac1 to Φac4, respectively, the magnetic fluxes circulate clockwise and counterclockwise as shown in Figures 5 and 6. The direction of the AC magnetic flux is also the direction of the magnetomotive force caused by the AC current.

[0027] FIG. 5 corresponds to a state in which the switching elements Sa2 and Sc2 of the lower arms of A phase and C phase are on, i.e., the on state, and the switching elements Sb2 and Sd2 of the lower arms of B phase and D phase are off, i.e., the off state, and FIG. 6 corresponds to a state in which the switching elements Sa2 and Sc2 of the lower arms of A phase and C phase are off, and the switching elements Sb2 and Sd2 of the lower arms of B phase and D phase are on.

[0028] Next, we will explain the gradient of the current in the windings 31 to 34 that occurs when the windings 31 to 34 are wound so as to generate magnetic flux in accordance with the DC magnetic flux described above and the switching circuit 4 is operated by PWM control, i.e., when PWM operation is performed. The gradient of the current will be referred to as the current gradient where appropriate. The winding currents that flow through the windings 31 to 34 are designated i1, i2, i3, and i4, respectively. The self-inductances of the windings 31 to 34 are designated L1, L2, L3, and L4. The mutual inductance between the windings 31 to 34 is as follows: Let M12 be the mutual inductance between windings 31 and 32, M13 be the mutual inductance between windings 31 and 33, M14 be the mutual inductance between windings 31 and 34, M23 be the mutual inductance between windings 32 and 33, M24 be the mutual inductance between windings 32 and 34, and M34 be the mutual inductance between windings 33 and 34. As shown in FIG. 3 , if the voltages across windings 31 to 34 are defined as voltages from switching element arm connection points Naa to Nad to winding connection point Ne, and are respectively designated V1, V2, V3, and V4, then equations (2) to (5) can be obtained from Kirchhoff's law. Note that dashed line 21 indicates the potential level at winding connection point Ne. Where appropriate, the voltages across the windings will be referred to as the voltages across the windings.

[0029]

[0030] The voltages V1 to V4 across the windings 31 to 34 when the lower arm switching elements Sa2, Sb2, Sc2, and Sd2 are turned on are referred to as lower arm on-voltages V ＯＮ The lower arm on-voltage V ＯＮ is the input voltage Vin input from the DC voltage source 1, and is expressed as in equation (6). Also, the voltages V1 to V4 across the windings 31 to 34 when the lower arm switching elements Sa2, Sb2, Sc2, and Sd2 are turned off are defined as the lower arm off voltages V ＯＦＦ The lower arm off voltage V ＯＦＦ A difference voltage Vin-Vout occurs between the input voltage Vin input from the DC voltage source 1 and the output voltage Vout, which is the voltage output to the connected device 6, and is expressed as in equation (7).

[0031]

[0032] 7 shows a first example of the operation timing of the lower arm in the switching circuit 4, that is, the operation timing and the on / off states of the lower arm switching elements Sa2 to Sd2 when the duty ratio D is less than 0.5 (D<0.5). FIG. 8 shows a second example of the operation timing of the lower arm in the switching circuit 4, that is, the operation timing and the on / off states of the lower arm switching elements Sa2 to Sd2 when the duty ratio D is greater than 0.5 (D>0.5). Specifically, the diagram shows the on / off states of the lower arm switching elements Sa2 and Sc2 in the A-phase leg 41a and the C-phase leg 41c, which are at 0° phase, and the lower arm switching elements Sb2 and Sd2 in the B-phase leg 41b and the D-phase leg 41d, which are at 180° phase, when the duty ratio D in FIG. 7 is less than 0.5 (D<0.5) and when the duty ratio D in FIG. 8 is greater than 0.5 (D>0.5). As mentioned above, the upper arm and lower arm perform on / off operations in a complementary manner, so the explanation will be given for the state of the lower arm. The winding currents shown in Figures 9 to 12 are currents when the lower arm is on, and in this case, the current can be called the lower arm on-current.

[0033] There are four states, i.e., four modes MD1 to MD4, depending on the switching state, i.e., on / off state, of the switching elements Sa2 to Sd2 of the switching circuit 4. Mode MD1 occurs when only switching elements Sa2 and Sc2 are on, and mode MD2 occurs when only switching elements Sb2 and Sd2 are on. Mode MD3 occurs when all switching elements Sa2, Sb2, Sc2, and Sd2 of the lower arm are off, and mode MD4 occurs when all switching elements Sa2, Sb2, Sc2, and Sd2 of the lower arm are on.

[0034] The gradients of the currents flowing through the windings 31 to 34 in the respective modes MD1 to MD4, i.e., the current gradients di1 / dt, di2 / dt, di3 / dt, and di4 / dt, can be obtained by applying equations (6) and (7), which are relational equations for the winding voltages, to equations (2) to (5).

[0035] To realize a simple magnetic coupling reactor 3, we will consider a simplified design. The self-inductances L1 to L4 of the windings 31 to 34 are all equal, with a value of L, and the degree of coupling K between the windings 31 and 33, through which currents of the same phase flow under PWM control, and the degree of coupling K between the windings 32 and 34, are high and ideally close to 1. The degree of coupling K is expressed as Mab / √(La × Lb), where La and Lb are the self-inductances of the windings, and Mab is the mutual inductance between the two windings. The mutual inductances M12, M13, M14, M23, M24, and M34 between the windings 31 to 34 are calculated by multiplying the product of the self-inductances of each winding to the 1 / 2 power, i.e., the root operation, by the degree of coupling K. Therefore, when the degree of coupling K is ideally close to 1, the mutual inductance M13 between the windings 31 and 33 and the mutual inductance M24 between the windings 32 and 34 are approximately equal to L. That is, M13=M24≈L.

[0036] In PWM control, mutual inductances M12, M14, M23, and M34 between windings 31 and 32, between windings 31 and 34, between windings 32 and 33, and between windings 33 and 34, are approximately equal. If the value of mutual inductances M12, M14, M23, and M34 is M, then they are all approximately equal to M. That is, M12 ≒ M14 ≒ M23 ≒ M34 = M (< L). The degree of coupling K between windings through which currents with a 180° phase shift flow in PWM control is smaller than the degree of coupling K between windings through which currents of the same phase flow in PWM control, so M < L.

[0037] Generally, when a magnetic path is formed between core parts that make up a magnetic core around which windings are wound, such as through magnetic resistance due to a gap, the magnetic flux generated by the winding leaks through the gap space, reducing the magnetic flux that interlinks with the other winding, thereby reducing the degree of coupling K between these windings.

[0038] Therefore, if the coupling degrees K between windings 31 and 33, which form magnetic paths not passing through gaps g1 and g2, and between windings 32 and 34 are K13 and K24, and the coupling degrees K between windings 31 and 32, 31 and 34, 32 and 33, and 33 and 34, which form magnetic paths via gaps g1 and g2, are K12, K14, K23, and K34, respectively, the relationship between these is expressed by equation (8).

[0039]

[0040] When there are no gaps g1, g2, the coupling degrees K13, K24 are approximately equal to each other. When there are gaps g1, g2, the coupling degrees K12, K14, K23, K34 are also approximately equal to each other. If the coupling degrees K13, K24 are defined as first coupling degrees and the coupling degrees K12, K14, K23, K34 are defined as second coupling degrees, the first coupling degree is greater than the second coupling degree.

[0041] Due to the circuit symmetry of the windings 31 to 34 of the magnetic coupling reactor 3 and the legs 41a to 41d of the switching circuit 4 in FIG. 1, the gradient of the current in mode MD1 is derived as an example, assuming that i3=i1 and i4=i2.

[0042] In mode MD1, the A-phase lower arm switching element Sa2 is turned on. ＯＮ By applying equation (2) to equation (6) and further applying L1=L, M12=M, M13=L, M14=M, i3=i1, and i4=i2, the voltage V can be expressed as equation (9). In mode MD1, the B-phase lower arm switching element Sb2 is turned off. The B-phase lower arm off-voltage V ＯＦＦ By applying equation (3) to equation (7) and further applying M12=M, L2=L, M23=M, M24=L, i3=i1, and i4=i2, equation (10) can be obtained.

[0043]

[0044] Adding equation (10) to equation (9) gives equation (11), and subtracting equation (10) from equation (9) gives equation (12).

[0045] From equations (11) and (12), equations (13) and (14) are obtained.

[0046] Although the derivation for the other modes MD2 to MD4 is omitted, the equations showing the current gradients can be obtained by using a similar procedure. The equations showing the current gradients in modes MD1 to MD4 are summarized below.

[0047] Mode MD1

[0048] Mode MD2

[0049] Mode MD3

[0050] Mode MD4

[0051] What is common to all modes MD1 to MD4 is that the inductance components (L - M) and (L + M) that determine the slope of the current are multiplied by a coefficient of 2. In the case of two-phase magnetic coupling, the inductance components (L + M) and (L - M) that determine the slope of the current are multiplied by a coefficient of 1. This means that in the case shown here in which four-phase windings 31 to 34 are magnetically coupled, the inductance components are twice as large as in the case of two-phase magnetic coupling, and this indicates that the magnitude of the slope of the current can be reduced by using the inductance of the other phase compared to the case of two-phase magnetic coupling.

[0052] The current gradients di1 / dt, di2 / dt, di3 / dt, and di4 / dt in modes MD1 and MD2, where there is no overlapping period of the on-states of the lower arms, are expressed as the sum of a first term including 2(L - M) and a second term including 2(L + M). The first term is the gradient of the current obtained by dividing the voltage by twice the difference between the self-inductance L and the mutual inductance M between the windings that are 180 degrees out of phase, and is the current gradient component due to the DC magnetic flux shown in Figure 4. The second term is the gradient of the current ripple obtained by dividing the voltage by twice the sum of the self-inductance L and the mutual inductance M between the windings that are 180 degrees out of phase, and is the current gradient component due to the AC magnetic flux circulating within the magnetic core 35 shown in Figures 5 and 6, where the direction of the magnetic flux changes due to the switching operation of legs 41a to 41d of each phase.

[0053] In other words, by controlling the value of the mutual inductance M between the 0° phase winding and the 180° phase winding, the magnitude of the current gradient due to the DC magnetic flux and AC magnetic flux can be adjusted.

[0054] In mode MD3 where both lower arms are in the OFF state and mode MD4 where both are in the ON state, the component due to AC magnetic flux becomes zero, as shown in equations (19) and (20), respectively.

[0055] First and second examples of the current gradients di1 / dt and di2 / dt in the windings 31 and 32 when the duty ratio D is D<0.5 are shown schematically in Figures 9 and 10. Third and fourth examples of the current gradients di1 / dt and di2 / dt in the windings 31 and 32 when the duty ratio D is D>0.5 are shown schematically in Figures 11 and 12. In Figures 9 to 12, the current gradients di1 / dt and di2 / dt are shown separated into components due to DC magnetic flux and components due to AC magnetic flux.

[0056] 9 and 11 show the case where the current gradient component due to DC magnetic flux is larger than the current gradient component due to AC magnetic flux, while Fig. 10 and 12 show the case where the current gradient component due to AC magnetic flux is larger than the current gradient component due to DC magnetic flux. In other words, Fig. 9 and 11 show the case where the value of M is adjusted so that the first term of Equations (15) to (18) is larger than the second term, while Fig. 10 and 12 show the case where the value of M is adjusted so that the second term of Equations (15) to (18) is larger than the first term. Note that the comparison of the magnitude relationship between the first and second terms is performed excluding the signs of the terms, i.e., the absolute values ​​of the first and second terms are compared.

[0057] 9 to 12, the operation timing of the lower arm is shown in the upper part, and the current gradient component due to DC magnetic flux, the current gradient component due to AC magnetic flux, and the current gradient are shown depending on the on / off state of the lower arm. The operation timing of the lower arm in FIGS. 9 and 10 is the operation timing shown in FIG. 7, and the operation timing of the lower arm in FIGS. 11 and 12 is the operation timing shown in FIG. 8. Current gradient characteristics 61, 62, 71, and 72 in FIGS. 9 to 12 are current gradient components due to DC magnetic flux. Current gradient characteristics 63j, 65j, 73j, and 75j in FIGS. 9 to 12 are current gradient components due to AC magnetic flux of winding current i1. Current gradient characteristics 63k, 65k, 73k, and 75k in FIGS. 9 to 12 are current gradient components due to AC magnetic flux of winding current i2. 9 to 12 represent the slope di1 / dt of the winding current i1. Current slope characteristics 64k, 66k, 74k, and 76k represent the slope di2 / dt of the winding current i2. Note that although the A-phase and B-phase winding currents i1 and i2 are used for the explanation in FIGS. 9 to 12, if the C-phase and D-phase are operated in the same manner as the A-phase and B-phase, the C-phase winding current i3 will be the same as the A-phase winding current i1, and the D-phase winding current i4 will be the same as the B-phase winding current i2.

[0058] 9 and 11, the current slopes di1 / dt and di2 / dt of winding current i1 and winding current i2, i.e., current slope characteristics 64j, 64k, 74j, and 74k, show that the slopes of winding current i1 and winding current i2 have the same trend, that is, both increase and both decrease, in all of modes MD1, MD2, MD3, and MD4, which are the entire operating period of cycle T. Input capacitor 2 on the input side of winding connection point Ne generates a current amplitude that is a combination of winding currents i1 and i2, so there is no cancellation effect due to winding current i1 and winding current i2, and the current amplitude of input capacitor 2 is large.

[0059] In contrast, looking at the current slopes di1 / dt and di2 / dt of winding current i1 and winding current i2 in Figures 10 and 12, i.e., current slope characteristics 66j, 66k, 76j, and 76k, we see that in inner modes MD1 and MD2 of the mode with period T, the slopes of winding current i1 and winding current i2 have opposite trends, i.e., one increases and the other decreases. Input capacitor 2 on the input side of winding connection point Ne generates a current amplitude that is a combination of winding currents i1 and i2, resulting in a cancellation effect between winding current i1 and winding current i2, thereby reducing the current amplitude of input capacitor 2 compared to the states in Figures 9 and 11. Winding current i1 is the lower-arm on-current that flows through winding 31 when A-phase leg 41a (e.g., the first leg), whose PWM phase is 0°, is driven. The winding current i2 is the lower-arm on-current that flows through the winding 32 when the B-phase leg 41b (e.g., the second leg), whose PWM phase is the other 180°, is driven. Therefore, it can be said that the slope of the lower-arm on-current flowing through the winding connected to the first leg is opposite to the slope of the lower-arm on-current flowing through the winding connected to the second leg. Here, the first leg is the leg driven by one PWM phase, and the second leg is the leg driven by the other PWM phase different from the first leg. If the first leg is the B-phase leg 41b, the second leg is the A-phase leg 41a.

[0060] The conditions for reducing the current amplitude of input capacitor 2 are summarized below. First, consider the case where duty ratio D is D<0.5. In mode MD2, the sum of the current gradients due to the DC magnetic flux component and AC magnetic flux component of winding current i1 is negative, meaning that current gradient characteristic 66j tends to decrease, i.e., because di1 / dt<0 in equation (17) is satisfied, equation (21) is obtained. In mode MD1, the sum of the current gradients due to the DC magnetic flux component and AC magnetic flux component of winding current i2 is negative, meaning that current gradient characteristic 66k tends to decrease, i.e., because di2 / dt<0 in equation (16) is satisfied, equation (21) is also obtained.

[0061] Equation (21) is rearranged using the relationship in equation (1) to obtain equation (22).

[0062] Next, consider the case where duty ratio D is D>0.5. In mode MD1, the sum of the current gradients due to the DC magnetic flux component and AC magnetic flux component of winding current i1 is positive, that is, the current gradient characteristic 76j tends to increase, i.e., the condition di1 / dt>0 in equation (15) is satisfied, so equation (23) is obtained. In mode MD2, the sum of the current gradients due to the DC magnetic flux component and AC magnetic flux component of winding current i2 is positive, that is, the condition di2 / dt>0 in equation (18) is satisfied, so equation (23) is obtained as well.

[0063] Equation (23) is rearranged using the relationship in equation (1) to obtain equation (24).

[0064] The coefficient mk1 applied to the self-inductance L on the right-hand side of equation (22) is D / (1-D), and the coefficient mk2 applied to the self-inductance L on the right-hand side of equation (24) is (1-D) / D. When considering that D<0.5, the coefficient mk1 becomes 0<mk1<1. When considering that D>0.5, the coefficient mk2 becomes 0<mk2<1. Here, when the self-inductances L1 and L2 are both L, the degree of coupling K is M / L, so equation (22) becomes equation (25), and equation (24) becomes equation (26).

[0065] The coefficients mk1 and mk2 correspond to the upper limit of the degree of coupling K. Therefore, the mutual inductance M between the 0° winding for PWM control, i.e., the windings 31 and 33, and the 180° winding for PWM control, i.e., the windings 32 and 34, can be set so as to satisfy equations (25) and (26). In the case of PWM control, when both duty ratios D for D<0.5 and D>0.5 are used, the mutual inductance M between the 0° winding for PWM control, i.e., the windings 31 and 33, and the 180° winding for PWM control, i.e., the windings 32 and 34, can be set using the upper limit of the degree of coupling K that satisfies equation (27).

[0066] 13 shows the relationship between the duty ratio D on the horizontal axis and the upper limit of coupling, which is the upper limit of coupling K, on ​​the vertical axis. When the duty ratio D is D<0.5, the upper limit of coupling is coefficient mk1, and when the duty ratio D is D>0.5, the upper limit of coupling is coefficient mk2.

[0067] Examples of calculations of current and current amplitude using analysis software are shown in Figures 14 to 21. Figure 14 shows the operation timing of the lower-arm switching elements Sa2 and Sb2 when the duty ratio D is 0.3 (<0.5). Figures 15 to 17 show current waveforms including the current amplitude of the windings 31 and 32 corresponding to the operation timing shown in Figure 14 when the duty ratio D is 0.3 (<0.5). Figure 18 shows the operation timing of the lower-arm switching elements Sa2 and Sb2 when the duty ratio D is 0.8 (>0.5). Figures 19 to 21 show current waveforms including the current amplitude of the windings 31 and 32 corresponding to the operation timing shown in Figure 18 when the duty ratio D is 0.8 (>0.5). The setting conditions are: input voltage Vin from DC voltage source 1 is 100 V, self-inductance L of windings 31 to 34 is 100 μH, PWM frequency is 70 kHz, dead time of upper and lower arms is 500 ns, load of connected device 6 is 20 Ω, and duty ratio D is D = 0.3, 0.8. The setting conditions for the degree of coupling K of the windings are as follows: The degree of coupling K of the windings between different PWM phases is K12 = K14 = K23 = K34 = 0.2, which satisfies the relationships of equations (22) and (24) to reduce the current amplitude of input capacitor 2, and the degree of coupling K of the windings between the same PWM phases is K13 = K24 = 0.99.

[0068] To demonstrate the effect of reducing the current amplitude of the input capacitor 2 by the power conversion device 50 of the first embodiment, a current waveform including an interleaved current amplitude in which the windings 31 to 34 are not magnetically coupled is also shown as a comparative example. FIG. 14 shows the operation timing of the lower-arm switching elements Sa2 and Sb2 when the duty ratio D is D=0.3. The horizontal axis represents time, and the vertical axis represents digital values ​​of 0 and 1. Where appropriate, a digital value of 0 is represented as "0" and a digital value of 1 is represented as "1." The switching element Sa2 is turned on and off in response to the control signal waveform 81a, and the switching element Sb2 is turned on and off in response to the control signal waveform 81b. In FIGS. 14 to 19, the periods from t0 to t13 represent the timings at which the control signal waveforms 81a and 81b change. The control signal waveform 81b changes to "1" at time t0, to "0" at time t1, to "1" at time t4, and to "0" at time t5. Thereafter, control signal waveform 81b changes to "1" at time t8, to "0" at time t9, to "1" at time t12, and to "0" at time t13. Control signal waveform 81a changes to "1" at time t2, to "0" at time t3, to "1" at time t6, to "0" at time t7, to "1" at time t10, and to "0" at time t11.

[0069] FIG. 15 shows the current waveforms of windings 31 and 32 of the comparative example, i.e., the current waveforms of windings 31 and 32 when there is no coupling. The horizontal axis represents time, and the vertical axis represents current. FIG. 16 shows the currents of windings 31 and 32 in power conversion device 50 of embodiment 1. The horizontal axis represents time, and the vertical axis represents current. FIG. 17 shows the total current waveform of windings 31 and 32. The horizontal axis represents time, and the vertical axis represents total current. Current waveform 82a is the waveform of winding current i1 in the comparative example in response to control signal waveform 81a, and current waveform 82b is the waveform of winding current i2 in the comparative example in response to control signal waveform 81b. Current waveform 83a is the waveform of winding current i1 in embodiment 1 in response to control signal waveform 81a, and current waveform 83b is the waveform of winding current i2 in embodiment 1 in response to control signal waveform 81b. A total current waveform 84 is the total current waveform of the winding current i1 and the winding current i2 in the first embodiment, and a total current waveform 85 is the total current waveform of the winding current i1 and the winding current i2 in the comparative example.

[0070] FIG. 18 shows the operation timing of the lower arm switching elements Sa2 and Sb2 when the duty ratio D is 0.8. The horizontal axis represents time, and the vertical axis represents digital values ​​of 0 and 1. Switching element Sa2 turns on and off in response to control signal waveform 86a, and switching element Sb2 turns on and off in response to control signal waveform 86b. In FIGS. 18 to 21, times t0 to t13 represent the timings at which the control signal waveforms 86a and 86b change. Control signal waveform 86b changes to "1" at time t0, "0" at time t3, "1" at time t4, "0" at time t7, "1" at time t8, "0" at time t11, and "1" at time t12. The control signal waveform 86a changes to "0" at time t1, changes to "1" at time t2, changes to "0" at time t5, changes to "1" at time t6, changes to "0" at time t9, changes to "1" at time t10, and changes to "0" at time t13.

[0071] FIG. 19 shows current waveforms of windings 31 and 32 of a comparative example, i.e., current waveforms of windings 31 and 32 when there is no coupling. The horizontal axis represents time, and the vertical axis represents current. FIG. 20 shows current waveforms of windings 31 and 32 in power conversion device 50 of embodiment 1. The horizontal axis represents time, and the vertical axis represents current. FIG. 21 shows the total current waveform of windings 31 and 32. The horizontal axis represents time, and the vertical axis represents total current. Current waveform 87a is the waveform of winding current i1 in the comparative example in response to control signal waveform 86a, and current waveform 87b is the waveform of winding current i2 in the comparative example in response to control signal waveform 86b. Current waveform 88a is the waveform of winding current i1 in embodiment 1 in response to control signal waveform 86a, and current waveform 88b is the waveform of winding current i2 in embodiment 1 in response to control signal waveform 86b. A total current waveform 89 is the total current waveform of the winding current i1 and the winding current i2 in the first embodiment, and a total current waveform 90 is the total current waveform of the winding current i1 and the winding current i2 in the comparative example.

[0072] Each current waveform corresponds to a period corresponding to modes MD1 and MD2, and it can be seen that the slope of the current waveforms of windings 31 and 32, i.e., the tendency of the current slope to increase or decrease, is reversed. The obtained current amplitude values ​​are summarized in FIG. 24. The first row, where numerical values ​​are listed, is the winding currents i1 and i2 without coupling, which are the winding currents i1 and i2 of the comparative example. The second row is the winding currents i1 and i2 with coupling, which are the winding currents i1 and i2 of embodiment 1. The third row is the sum of the winding currents i1 and i2 without coupling, which is the sum of the winding currents i1 and i2 of the comparative example. The fourth row is the sum of the winding currents i1 and i2 with coupling, which is the sum of the winding currents i1 and i2 of embodiment 1. The +Apk column is the positive peak current value, the -Apk column is the negative peak current value, and the App column is the current value range between the positive and negative peak current values. The column for reduction rate shows the percentage of App1 / App2 in the current value range, where App1 is the current value range in the first embodiment and App2 is the current value range in the comparative example.

[0073] The sum of the winding currents i1 and i2 in the third and fourth rows of Figure 24 is the sum of the two windings, and therefore represents half of the current amplitude value flowing through input capacitor 2. The sum of winding currents i1 and i2 in embodiment 1 can reduce the current amplitude, i.e., ripple current, generated in windings 31 and 32 compared to the sum of winding currents i1 and i2 in the comparative example. Unlike the DC-DC converter of Patent Document 1, power conversion device 50 of embodiment 1 does not have a boost reactor between input capacitor 2 and magnetic coupling reactor 3. Therefore, the current amplitude, i.e., ripple current, generated in windings 31 and 32 flows directly into input capacitor 2, thereby reducing the current amplitude, i.e., ripple current, generated in input capacitor 2. Winding currents i3 and i4 are also similar to winding currents i1 and i2, and therefore the current amplitude, i.e., ripple current, generated in windings 31 to 34 and the current amplitude, i.e., ripple current, generated in input capacitor 2 can be reduced.

[0074] In the power conversion device 50 of the first embodiment, the degree of coupling K between the windings of the magnetic coupling reactor 3 satisfies the relationship shown in equation (8), i.e., the degrees of coupling K13, K24 between the windings of the same PWM phase are higher than the degrees of coupling K12, K14, K23, K34 between the windings of different PWM phases, thereby making it possible to reduce the current amplitude, i.e., ripple current, generated in the windings 31 to 34 and the input capacitor 2. Making the degrees of coupling K13, K24 between the windings of the same PWM phase higher than the degrees of coupling K12, K14, K23, K34 between the windings of different PWM phases (first degree of coupling relationship) also increases the inductance of the magnetic coupling reactor 3. The first degree of coupling relationship indicates that a magnitude relationship is established between the degrees of coupling between the windings of the magnetic coupling reactor 3. Therefore, in power conversion device 50 of the first embodiment, the inductance is increased using the magnitude relationship of the degree of coupling between the windings of magnetic coupling reactor 3, and therefore the current amplitude of windings 31 to 34 can be reduced.

[0075] Next, the structure of the magnetic coupling reactor 3 that satisfies the first degree of coupling relationship will be described with reference to Figures 22 and 23. Figure 22 shows a specific structure of the windings 31 to 34 and gaps g1 and g2 provided on the magnetic core 35. Figure 23 shows the direction of DC magnetic flux when a DC current is passed through the windings 31 to 34 from the winding connection point Ne to the arm connection points Naa to Nad in the state where the magnetic coupling reactor 3 is configured as shown in Figure 22.

[0076] The magnetic coupling reactor 3 shown in Figure 22 includes four reactor components 39a, 39b, 39c, and 39d. In Figure 22, the magnetic core 35 is composed of four core components 36a, 36b, 36c, and 36d. Core components 36a and 36c, which form the first core portion 37a, are connected without a gap to form a magnetic path. Core component 36a is provided with windings 31 and 33, while core component 36c is not provided with a winding. Similarly, core components 36b and 36d, which form the second core portion 37b, are connected without a gap to form a magnetic path. Core component 36b is provided with windings 32 and 34, while core component 36d is not provided with a winding. Each reactor component 39a, 39b, 39c, and 39d is as follows: The reactor component 39a has a core component 36a and windings 31 and 33, and the reactor component 39b has a core component 36b and windings 32 and 34. The reactor component 39c has a core component 36c, and the reactor component 39d has a core component 36d.

[0077] A gap g2 is provided between core parts 36a and 36d, and a gap g1 is provided between core parts 36b and 36c. These gaps g1 and g2 are positioned symmetrically with respect to the center of the magnetic coupling reactor 3, i.e., the reactor center 38. The reactor center 38 can also be described as the center of the magnetic coupling reactor 3 located inside the closed magnetic circuit formed by the magnetic core 35. The arrangement of gaps g1 and g2 makes it possible to equalize the degrees of coupling K13 and K24 between multiple windings of the same PWM phase and to equalize the degrees of coupling K12, K14, K23, and K34 between windings of different PWM phases. The reactor center 38 is the midpoint between the gap center 24a of gap g1 and the gap center 24b of gap g2. The magnetic core 35 is configured in an annular shape and extends in a direction perpendicular to the plane of FIG. 22 . Therefore, the reactor center 38 and the gap centers 24a and 24b can be considered to be points on a plane perpendicular to the extension direction of the magnetic core 35. Gap g1 is the space between dashed lines 22a and 22b, and gap g2 is the space between dashed lines 22c and 22d. For example, gap center 24a is located on dashed line 23a that passes through the center of core part 36b, and gap center 24b is located on dashed line 23b that passes through the center of core part 36a.

[0078] When the magnetic core 35 and the windings 31 to 33 are configured in this manner, there are no gaps in the magnetic paths between the windings, such as between windings 31 and 33 and between windings 32 and 34. In other words, the magnetic coupling reactor 3 of embodiment 1 has a magnetic core 35 around which the multiple windings 31 to 33 are wound and which forms a closed magnetic path. The magnetic core 35 has gaps g1 and g2 between the windings connected to legs with different PWM phases, and no gaps g1 and g2 between the windings connected to legs with the same PWM phase. In this case, ideally, the magnetic flux generated in a winding interlinks with the other winding, i.e., the mating winding, without leakage, and therefore the degrees of coupling K13 and K24 can be configured to be high. Furthermore, if a gap is provided between the first core portion 37a and the second core portion 37b, i.e., if gaps g1 and g2 are provided in the magnetic path between the windings, leakage magnetic flux occurs from the gaps g1 and g2, reducing the amount of magnetic flux in the windings that link with each other, and making it possible to configure the degrees of coupling K12, K14, K23, and K34 to be lower than in a magnetic path without a gap.

[0079] Figure 23 shows that the DC magnetic flux between windings of the same PWM phase, i.e., the DC magnetic flux between windings 31 and 33 and the DC magnetic flux between windings 32 and 34, reinforce each other. Figure 23 also shows that the DC magnetic flux between windings of different PWM phases, i.e., the DC magnetic flux between windings 31 and 32, the DC magnetic flux between windings 31 and 34, the DC magnetic flux between windings 32 and 33, and the DC magnetic flux between windings 33 and 34, are cancelled out. Therefore, windings 31 to 34 are wound around magnetic core 35 so that the DC magnetic flux between windings of the same PWM phase reinforce each other and the DC magnetic flux between windings of different PWM phases cancel each other out.

[0080] 22 and 23, the degrees of coupling K13, K24 between windings of the same PWM phase are high, and the degrees of coupling K12, K14, K23, K34 between windings of different PWM phases can be set low to or below a predetermined set value. The predetermined set value is a value that satisfies any one of equations (25) to (37). The degrees of coupling K12, K14, K23, K34 between windings of different PWM phases can be controlled by adjusting the gaps g1, g2, which affect the amount of leakage magnetic flux, so that a magnetic coupling reactor 3 that can suppress current amplitude can be configured. In the power conversion device 50 of the first embodiment, when configuring a parallel converter having reactors and switching circuits in a power conversion device such as a multi-parallel power conversion device, a multiplex power conversion device, or a multiplex-multiple-parallel power conversion device, the number of reactors, which increases when simply paralleling or multiplexing, can be reduced by using the magnetic coupling reactor 3, and the current amplitude of the windings 31 to 34 of the magnetic coupling reactor 3 and the current amplitude of the input capacitor 2 can be reduced.

[0081] As described above, the power conversion device 50 of the first embodiment is a power conversion device that converts a DC voltage input between the two input terminals 18a, 18b into DC power of a predetermined voltage and outputs the converted DC power, and includes: a switching circuit 4 in which a plurality of legs 41a, 41b, 41c, 41d are connected in parallel, each of which has an upper arm S1 and a lower arm S2 each having a switching element connected in series; a magnetic coupling reactor 3 in which the same number of windings 31, 32, 33, 34 as the legs 41a, 41b, 41c, 41d are magnetically coupled to each other; and a control circuit 7 that performs PWM control with phases shifted by 180 degrees to drive the plurality of legs 41a, 41b, 41c, 41d. In the magnetic coupling reactor 3, one end of each of the windings 31, 32, 33, and 34 is connected to the input terminal 18a, and the other end of each of the windings 31, 32, 33, and 34 is connected to a connection point (arm connection point Na) between the upper arm S1 and the lower arm S2 in the corresponding leg 41a, 41b, 41c, and 41d. The degree of coupling K between the windings 31, 33 and between the windings 32, 34 connected to the legs 41a, 41c and 41b, 41d having the same PWM phase, which is the phase of PWM control, is higher than the degree of coupling K between the windings connected to the legs having different PWM phases. With this configuration, the power conversion device 50 of the first embodiment includes the magnetic coupling reactor 3 on the input side of the switching circuit 4, and the degree of coupling K between the windings 31, 33 and between the windings 32, 34 connected to the legs 41 a, 41 c and 41 b, 41 d having the same PWM phase, which is the phase of PWM control, is higher than the degree of coupling K between the windings connected to the legs having different PWM phases. Therefore, it is possible to increase the inductance and reduce the current amplitude of the windings by utilizing the magnitude relationship of the degree of coupling between the windings.

[0082] Embodiment 2. FIG. 26 is a diagram showing a first example of the configuration of a magnetic coupling reactor according to embodiment 2, and FIG. 27 is a diagram showing a second example of the configuration of a magnetic coupling reactor according to embodiment 2. FIG. 28 is a diagram showing DC magnetic flux corresponding to the magnetic coupling reactors of FIGS. 26 and 27. Embodiment 2 shows another structure of a magnetic coupling reactor 3 that satisfies the first degree of coupling relationship described above. FIG. 26 shows a first example of a specific configuration of the windings 31 to 34 and gaps g1 and g2 provided on the magnetic core 35, and FIG. 27 shows a second example of a specific configuration of the windings 31 to 34 and gaps g1 and g2 provided on the magnetic core 35. FIG. 28 shows the direction of DC magnetic flux when a DC current is passed through the windings 31 to 34 from the winding connection point Ne to the arm connection points Naa to Nad in the magnetic coupling reactor 3 configured as shown in FIGS. 26 and 27.

[0083] The magnetic coupling reactor 3 shown in Figures 26 and 27 includes four reactor components 39a, 39b, 39c, and 39d, as shown in Figure 22 of the first embodiment. The magnetic coupling reactor 3 of the second embodiment shown in Figures 26 and 27 differs from the magnetic coupling reactor 3 of the first embodiment shown in Figure 22 in that each of the core components 36a, 36b, 36c, and 36d is provided with one winding. The reactor component 39a includes the core component 36a and the winding 31, and the reactor component 39b includes the core component 36b and the winding 32. The reactor component 39c includes the core component 36c and the winding 33, and the reactor component 39d includes the core component 36d and the winding 34. In a first example of the magnetic coupling reactor 3 of the second embodiment, the core components 36a, 36b, 36c, and 36d of the magnetic core 35 are arranged in a substantially square shape. In the second example of the magnetic coupling reactor 3 of the second embodiment, the core parts 36a, 36b, 36c, and 36d of the magnetic core 35 are arranged in a rectangular shape. The difference between the first example of the magnetic coupling reactor 3 of the second embodiment and the second example of the magnetic coupling reactor 3 of the second embodiment is the arrangement of the core parts 36a, 36b, 36c, and 36d. The following mainly describes the parts that differ from the magnetic coupling reactor 3 of FIG.

[0084] 26 and 27, the magnetic core 35 is made up of four core parts 36a, 36b, 36c, and 36d, and is the same in that it has windings 31 to 34 and two gaps g1 and g2, and that a magnetic path is formed by connecting core parts wound with windings of the same PWM phase where the degree of coupling K is to be increased without providing a gap, and a magnetic path is formed by providing a gap between core parts wound with windings of different PWM phases where the degree of coupling K is to be reduced to a predetermined set value or less.The operating principle for suppressing the current amplitude is also the same as in embodiment 1.

[0085] An A-phase winding 31 and a C-phase winding 33 are wound around core parts 36a and 36c of the first core portion 37a, respectively, and a B-phase winding 32 and a D-phase winding 34 are wound around core parts 36b and 36d of the second core portion 37b, respectively. A gap g2 is provided between core parts 36a and 36d, and a gap g1 is provided between core parts 36b and 36c. These gaps g1 and g2 are positioned symmetrically with respect to the center of the magnetic coupling reactor 3, i.e., reactor center 38. Gap g1 is the space between dashed lines 22a and 22b, and gap g2 is the space between dashed lines 22c and 22d. In Figure 26, gap center 24a is located on dashed line 23a passing through the center of core part 36c, and gap center 24b is located on dashed line 23b passing through the center of core part 36d. 27, the gap center 24a is located on the dashed line 23a that passes through the center of the core part 36b, and the gap center 24b is located on the dashed line 23b that passes through the center of the core part 36a. Note that the dashed lines 23a and 23b are auxiliary lines for explaining the gap centers 24a and 24b, and do not represent the center lines of any particular core parts.

[0086] The magnetic coupling reactor 3 in Figures 26 and 27 is configured by combining pairs of core components 36a, 36b, 36c, and 36d, each with a winding wound thereon, i.e., a first reactor section 40a and a second reactor section 40b. Because the first reactor section 40a and the second reactor section 40b are paired and have the same configuration, a uniform magnetic flux is generated in the closed magnetic circuit. Therefore, a power conversion device 50 equipped with a magnetic coupling reactor 3 using a magnetic core 35 with a uniform magnetic flux has the advantage of achieving stable characteristics. The first reactor section 40a includes the core components 36a and 36c of the first core section 37a, and the windings 31 and 33. The second reactor section 40b includes the core components 36b and 36d of the second core section 37b, and the windings 32 and 34.

[0087] In the magnetic coupling reactor 3 of FIG. 26 , the reactor components 39 a, 39 b, 39 c, and 39 d have core components 36 a, 36 b, 36 c, and 36 d of the same shape. When core components of the same shape are used in the magnetic coupling reactor 3, the performance of the core components is consistent. Therefore, a power conversion device 50 including a magnetic coupling reactor 3 using core components of the same shape has the advantage of obtaining stable characteristics. In the magnetic coupling reactor 3 of FIG. 27 , the first reactor portion 40 a and the second reactor portion 40 b have first core portions 37 a and second core portions 37 b of the same shape, respectively. When the magnetic coupling reactor 3 uses the first core portions 37 a and second core portions 37 b of the same shape, the performance of the first core portions 37 a and second core portions 37 b is consistent. Therefore, a power conversion device 50 including a magnetic coupling reactor 3 using the first core portions 37 a and second core portions 37 b of the same shape also has the advantage of obtaining stable characteristics. The magnetic coupling reactor 3 in FIG. 26 also has a first core portion 37a and a second core portion 37b of the same shape.

[0088] As described above, in the magnetic coupling reactor 3 of Fig. 26, the core parts 36a, 36b, 36c, and 36d of the magnetic core 35 are arranged in a substantially square shape. In the magnetic coupling reactor 3 of Fig. 26, when the first reactor part 40a and the second reactor part 40b are moved and brought into contact so that gaps g1 and g2 are eliminated, the core parts 36a, 36b, 36c, and 36d are arranged in a square shape. When the first reactor part 40a and the second reactor part 40b are moved from this state so that gaps g1 and g2 are formed, the arrangement of the core parts 36a, 36b, 36c, and 36d changes from the square shape to a substantially square shape. The substantially square shape refers to a shape that is shifted from the square shape by gaps g1 and g2.

[0089] The DC magnetic fluxes Φdc1 to Φdc4 shown in Fig. 28 differ from the DC magnetic fluxes Φdc1 to Φdc4 shown in Fig. 23 in that the C-phase Φdc3 has moved toward gap g1 and the D-phase Φdc4 has moved toward gap g2. The magnetic path along which the DC magnetic fluxes Φdc1 to Φdc4 shown in Fig. 28 are generated is substantially the same as the magnetic path along which the DC magnetic fluxes Φdc1 to Φdc4 shown in Fig. 23 are generated.

[0090] As described above, the power conversion device 50 including the magnetic coupling reactor 3 of embodiment 2 achieves the same effects as the power conversion device 50 of embodiment 1. Furthermore, the power conversion device 50 including the magnetic coupling reactor 3 of embodiment 2 has the first core portion 37a and the second core portion 37b of the same shape, and therefore can achieve more stable characteristics than the power conversion device 50 of embodiment 1.

[0091] Embodiment 3. Fig. 29 is a diagram showing the configuration of a power conversion device according to embodiment 3, and Fig. 30 is a diagram showing the operation of a control circuit according to embodiment 3. Fig. 31 is a diagram showing a first example of DC magnetic flux in a magnetic coupling reactor when the output current according to embodiment 3 is equal to or greater than a set value, and Fig. 32 is a diagram showing a first example of DC magnetic flux in a magnetic coupling reactor when the output current according to embodiment 3 is less than the set value. Fig. 33 is a diagram showing a second example of DC magnetic flux in a magnetic coupling reactor when the output current according to embodiment 3 is equal to or greater than a set value, and Fig. 34 is a diagram showing a second example of DC magnetic flux in a magnetic coupling reactor when the output current according to embodiment 3 is less than the set value. Fig. 35 is a diagram showing a first example of AC magnetic flux in a magnetic coupling reactor when the output current according to embodiment 3 is equal to or greater than a set value, and Fig. 36 is a diagram showing a first example of AC magnetic flux in a magnetic coupling reactor when the output current according to embodiment 3 is less than the set value. FIG. 37 is a diagram showing a second example of the AC magnetic flux of the magnetic coupling reactor when the output current according to embodiment 3 is equal to or greater than the set value, and FIG. 38 is a diagram showing a second example of the AC magnetic flux of the magnetic coupling reactor when the output current according to embodiment 3 is less than the set value.

[0092] The power conversion device 50 of the third embodiment is an example in which the number of operating legs 41 constituting the switching circuit 4 varies depending on the output current Iout output to the connected device 6. In FIG. 29 , the DC voltage source 1 and the connected device 6 are also illustrated along with the power conversion device 50. The reference numeral 50 of the power conversion device is indicated in the same manner as in FIG. 1 . The power conversion device 50 of the third embodiment differs from the power conversion device 50 of the first embodiment in that a leg determination unit 27 is added to the control circuit 7, and the switching circuit 4 is operated on a first operating leg or a second operating leg having fewer operating legs than the first operating legs, as determined by the operating leg determination unit 25 based on a leg determination signal sig2 generated by the leg determination unit 27. For example, the first operating legs are all legs. The second operating legs are legs having one or more sets each having one of all the PWM phases to be used, and having a smaller number than the first operating legs. In the case of A to D phases, each of which has two PWM phases of 0° and 180°, the first operating legs are all the legs, and the second operating legs are half the legs.

[0093] In addition, in the case of six phases with three PWM phases each, i.e., A to D, E, and F, where the PWM phases are 0° and 180°, the first operating legs are all legs, and the second operating legs are four or two legs. When the first operating legs have a first set, a second set, and a third set with PWM phases of 0° and 180°, the second operating legs having four legs include two sets, either the first set, the second set, or the third set. Similarly, the second operating legs having two legs include one set, either the first set, the second set, or the third set. In the case of six PWM phases, the magnetic coupling reactor 3 has six windings, and the switching circuit 4 includes six legs. This example will mainly describe the differences from the power conversion device 50 of embodiment 1.

[0094] The magnetic coupling reactor 3 will be described as an example in which the one shown in Embodiments 1 and 2 is applied. The control circuit 7 includes an operating leg determination unit 25, a control signal generation unit 26, a leg determination unit 27, and a drive circuit 28. As in Embodiment 1, the PWM phases of phases A to D, i.e., the PWM phases of legs 41a to 41d, will be described as 0°, 180°, 0°, and 180°, respectively. The operating leg determination unit 25 determines a first operating leg, which is a leg to be operated from phases A to D of the switching circuit 4, based on the input voltage Vin, the output voltage Vout, the output current Iout, and the winding current Iw, and generates operating leg information sigm, which is information on the first operating leg. When the output current Iout is equal to or greater than a predetermined set value Ios, the leg determination unit 27 determines to maintain the first operating leg determined by the operating leg determination unit 25 and generates a leg determination signal sig2 indicating the first operating leg. Furthermore, when the output current Iout is less than a predetermined set value Ios, the leg determination unit 27 determines a second operating leg, which is half the number of the first operating legs determined by the operating leg determination unit 25, and generates a leg determination signal sig2 indicating the second operating leg. For example, the leg determination signal sig2 indicating the first operating leg is at a high level, i.e., a digital signal of "1," and the leg determination signal sig2 indicating the second operating leg is at a low level, i.e., a digital signal of "0."

[0095] The operation of the control circuit 7 will be described using FIG. 30. In step ST1, the control circuit 7 acquires input information, including the input voltage Vin, the output voltage Vout, the output current Iout, and the winding current Iw (input information acquisition step). In step ST2, the operating leg determination unit 25 determines the operating leg, i.e., the first operating leg, and generates operating leg information sigm, which is information about the first operating leg (operating leg determination step). In step ST3, the leg determination unit 27 determines whether the output current Iout is equal to or greater than a predetermined set value Ios, and generates a leg determination signal sig2 indicating either the first operating leg or the second operating leg (leg determination step). In step ST3, if the output current Iout is equal to or greater than the set value Ios, the leg determination unit 27 generates a leg determination signal sig2 indicating the first operating leg. Then, the process proceeds to step ST4. In step ST3, if the output current Iout is less than the set value Ios, the leg determination unit 27 generates a leg determination signal sig2 indicating the second operating leg. Then, the process moves to step ST5.

[0096] In step ST4, the control signal generator 26 generates a digital control signal sig1 for controlling all of the first operating legs, i.e., a digital control signal sig1 for PWM-operating all of the legs determined in step ST2, based on the leg determination signal sig2 indicating the first operating leg and the operating leg information sigm, which is information about the first operating leg (first operating leg control step). The drive circuit 28 converts the digital control signal sig1 into predetermined high and low voltages, and outputs the control signal sig1 to the switching circuit 4. The control circuit 7 ends the first operating leg control step of step ST4.

[0097] In step ST5, the control signal generator 26 generates a digital control signal sig1 for controlling the second operating legs (a smaller half of the first operating legs) based on the leg determination signal sig2 indicating the second operating leg and the operating leg information sigm, which is information about the first operating leg. This digital control signal sig1 controls PWM operation on half of the legs selected by a predetermined selection method from the legs determined in step ST2 (second operating leg control step). The predetermined selection method selects one or more pairs of legs each containing one of all the PWM phases to be used, and the number of legs is fewer than the first operating legs. In this case, the number of second operating legs is half that of the first operating legs. Details of the selection method will be described later. The drive circuit 28 converts the digital control signal sig1 into predetermined high and low voltages and outputs the converted control signal sig1 to the switching circuit 4. The control circuit 7 then terminates the second operating leg control step of step ST5.

[0098] FIG. 29 illustrates an example in which the switching circuit 4 includes four legs 41a to 41d. In the operating leg determination process of step ST2, which of the multiple legs is to be operated is determined. Here, an example will be described in which the four legs 41a to 41d are operated at the operation timings shown in FIGS. 7 and 8 of the first embodiment. The operating leg determination unit 25 determines phases A to D of the switching circuit 4 as first operating legs, which are the legs to be operated. The operating leg determination process also includes determining the operation timing associated with a change in duty ratio D. For example, if there are two operation timings based on two duty ratios D, the operating leg determination process also includes determining either a first control that operates the first operating leg at the first operation timing or a second control that operates the first operating leg at the second operation timing. For example, the first operation timing is the operation timing shown in FIG. 7, and the second operation timing is the operation timing shown in FIG. 8.

[0099] The control circuit 7 of the power conversion device 50 shown in FIG. 29 controls all of the legs 41a to 41d of phases A to D in the switching circuit 4 to operate at the first operating timing or the second operating timing when the output current Iout is equal to or greater than the set value Ios, i.e., during non-light-load operation. Furthermore, the control circuit 7 of the power conversion device 50 shown in FIG. 29 stops the switching operation of two legs with different PWM phases selected as a set that includes all of the PWM phases to be used, each set including one of the PWM phases to be used. In the case of a four-phase system, there are two sets of legs with different PWM phases, a first set P1 and a second set P2. Therefore, the control circuit 7 of the power conversion device 50 shown in FIG. 29 stops the switching operation of one of the first set P1 and the second set P2 when the output current Iout is less than the set value Ios, i.e., during light-load operation.

[0100] As in the first embodiment, the PWM phases of the A-phase to D-phase legs 41a to 41d are 0°, 180°, 0°, and 180°, respectively. For example, if the first set P1 is the A-phase leg 41a and the B-phase leg 41b so that current flows through the windings 31 and 32, the second set P2 is the C-phase leg 41c and the D-phase leg 41d so that current flows through the windings 33 and 34 (see FIG. 31). If the first set P1 is the A-phase leg 41a and the D-phase leg 41d so that current flows through the windings 31 and 34, the second set P2 is the B-phase leg 41b and the C-phase leg 41c so that current flows through the windings 32 and 33 (see FIG. 33). The predetermined selection method described above selects either the first set P1 or the second set P2. It should be noted that the PWM phase of the AC magnetic flux is also shown for the DC magnetic flux shown in FIGS.

[0101] By operating the switching circuit 4 in this manner, during light load operation where the ratio of loss to transmitted power increases, it is possible to reduce losses that occur when the switching circuit 4 operates or conduction losses that occur in the current path of the conductor, thereby improving the efficiency of the power conversion device 50.

[0102] FIG. 31 shows the state of DC magnetic flux when all legs 41a to 41d are operating with the output current Iout equal to or greater than the set value Ios. FIG. 32 shows the state of DC magnetic flux when one of the first group P1 and the second group P2 of different PWM phases selected as a group having all the PWM phases to be used is stopped, i.e., when half of the legs are stopped. The state of DC magnetic flux is shown in the same format as in FIG. 4. The example shown in FIG. 32 is an example in which the C-phase leg 41c and the D-phase leg 41d are stopped, and the DC magnetic flux generated by the winding 31 connected to the A-phase leg 41a and the DC magnetic flux generated by the winding 32 connected to the B-phase leg 41b act to cancel each other out within the magnetic core 35.

[0103] The selection of the first set P1 and the second set P2 in the third embodiment is not limited to the first example of the DC magnetic flux of the magnetic coupling reactor shown in Figures 31 and 32, and may be the second example of the DC magnetic flux of the magnetic coupling reactor shown in Figures 33 and 34. A first selection example, which is a selection of the first set P1 and the second set P2 corresponding to the first example of the DC magnetic flux, is a case where the first set P1 is the A-phase leg 41a and the B-phase leg 41b, and the second set P2 is the C-phase leg 41c and the D-phase leg 41d. A second selection example, which is a selection of the first set P1 and the second set P2 corresponding to the second example of the DC magnetic flux, is a case where the first set P1 is the A-phase leg 41a and the D-phase leg 41d, and the second set P2 is the B-phase leg 41b and the C-phase leg 41c. Figure 33 shows the state of DC magnetic flux in the second selection example when all legs 41a to 41d are operated with the output current Iout equal to or greater than the set value Ios. Figure 34 shows the state of DC magnetic flux in the second selection example when one of the first group P1 and the second group P2, each having all the PWM phases to be used, is stopped, i.e., when half of the legs are stopped. The state of DC magnetic flux is shown in the same format as in Figure 4. The example shown in Figure 34 is an example in which the C-phase leg 41c and the B-phase leg 41b are stopped, and the DC magnetic flux generated by the winding 31 connected to the A-phase leg 41a and the DC magnetic flux generated by the winding 34 connected to the D-phase leg 41d act to cancel each other out within the magnetic core 35.

[0104] Figures 35 and 37 show the state of AC magnetic flux when all legs 41a to 41d are operated with the output current Iout equal to or greater than the set value Ios in the first and second selection examples, respectively. Figures 36 and 38 show the state of AC magnetic flux when one of the first and second selection examples P1 and P2, which are selected as groups each containing all the PWM phases to be used, is stopped, i.e., when half of the legs are stopped, in the first and second selection examples, respectively. The state of AC magnetic flux in Figures 35 and 36 is shown in the same format as in Figure 5, and the state of AC magnetic flux in Figures 37 and 38 is shown in the same format as in Figure 6. In the first selection example of Figure 36, the AC magnetic flux generated by the winding 31 connected to the A-phase leg 41a and the AC magnetic flux generated by the winding 32 connected to the B-phase leg 41b act constructively within the magnetic core 35. In the second selection example of Figure 38, the AC magnetic flux generated by the winding 31 connected to the A-phase leg 41a and the AC magnetic flux generated by the winding 34 connected to the D-phase leg 41d act to reinforce each other within the magnetic core 35.

[0105] In the power conversion device 50 of the third embodiment, even when the PWM operation of half of the legs is stopped in order to reduce loss during light load operation, the principle operation of the magnetic coupling reactor 3, which is established by the relationship between the DC magnetic flux and the AC magnetic flux described in the first embodiment, remains unchanged. Therefore, in the modes MD1 and MD2 in which the lower arm is turned on as shown in the first embodiment, the power conversion device 50 of the third embodiment can achieve an effect in which the slopes of the winding currents between the windings with different PWM phases tend to be opposite to each other, and can reduce the current amplitude of the windings 31 to 34 of the magnetic coupling reactor 3 and the current amplitude of the input capacitor 2.

[0106] The functions of the leg determining unit 27 may be realized by the processor 98 and memory 99 described above. In this case, the leg determining unit 27 is realized by the processor 98 executing a program stored in the memory 99. Furthermore, a plurality of processors 98 and a plurality of memories 99 may cooperate to perform the functions of the leg determining unit 27. The functions of the operating leg determining unit 25 and the control signal generating unit 26, together with the leg determining unit 27, may also be realized by the processor 98 and memory 99.

[0107] An example of the E and F phases will be described below in the case where the PWM phases are 0° and 180° and there are six phases, i.e., three phases each, i.e., A to D, E, and F. As shown in FIGS. 31 to 38, when the PWM phases of A to D are 0°, 180°, 0°, and 180°, respectively, the E and F phases are 0°, 180°, or 180°, 0°, respectively. When the PWM phases of E and F are 0° and 180°, respectively, the E phase leg is disposed in the magnetic path of A and C phases that does not pass through magnetic resistances R1 and R2, and the F phase leg is disposed in the magnetic path of B and D phases that does not pass through magnetic resistances R1 and R2. There are multiple patterns of legs that are stopped by operating the second operating leg, but the legs are each a pair of PWM phases of 0° and 180°.

[0108] Although the power conversion device 50 of the third embodiment differs in the configuration of the control circuit 7 and the method of driving the switching circuit 4, the other configurations are similar to those of the power conversion device 50 of the first embodiment, and therefore the same effects as those of the first embodiment can be achieved. Furthermore, in the power conversion device 50 of the third embodiment, the switching circuit 4 includes a first set P1 and a second set P2 having two legs with PWM phases shifted by 180 degrees, and the control circuit 7 drives the first set P1 and the second set P2 when the output current Iout output by the switching circuit 4 is equal to or greater than a predetermined set value Ios, and drives one of the first set P1 or the second set P2 when the output current Iout is less than the predetermined set value Ios. Therefore, the power conversion device 50 of the third embodiment can switch between non-light-load operation and light-load operation, thereby reducing loss during light-load operation and improving the efficiency of the power conversion device 50.

[0109] Fourth Embodiment. FIG. 39 is a diagram illustrating the configuration of a power conversion device according to a fourth embodiment. The power conversion device 50 of the fourth embodiment is an example of a two-parallel interleaved totem-pole PFC (Power Factor Correction) converter. In FIG. 39, an AC voltage source 8 and a connected device 6 are also illustrated along with the power conversion device 50. The reference numeral 50 of the power conversion device is illustrated in the same manner as in FIG. 1. The two-parallel interleaved totem-pole PFC converter is a power converter in which a first circuit 48a including two sets of reactors and totem-pole-connected legs 41 in a totem-pole PFC, one for phase 0° and one for phase 180°, and a second circuit 48b similar to the first circuit 48a, are connected in parallel. The power conversion device 50 of the fourth embodiment is configured such that the magnetic coupling reactor 3 of the first or second embodiment is applied to the four reactors of a two-parallel interleaved totem-pole PFC converter that converts an input AC voltage into DC power of a predetermined voltage and outputs the converted DC power. The predetermined voltage is, for example, a voltage higher than the input voltage Vin.

[0110] The power conversion device 50 of the fourth embodiment differs from the power conversion device 50 of the first embodiment in the following manner. Instead of DC power from the DC voltage source 1, AC power from the AC voltage source 8 is input to the input terminals 18a, 18b, there is no input capacitor 2, and series-connected diode series bodies 10a, 10b are connected in parallel between the output terminals 19a, 19b connected to the connected device 6. The diode series body 10a has diodes 9a, 9b connected in series, and the diode series body 10b has diodes 9c, 9d connected in series. The diode series body 10a, the A-phase leg 41a, and the B-phase leg 41b form a first circuit 48a, and the diode series body 10b, the C-phase leg 41c, and the D-phase leg 41d form a second circuit 48b. A first circuit 48a of the fourth embodiment differs from the first circuit 48a of the first embodiment in that a diode series body 10a is added, and a second circuit 48b of the fourth embodiment differs from the second circuit 48b of the first embodiment in that a diode series body 10b is added. The first circuit 48a and the second circuit 48b are connected in parallel on the side of output terminals 19a and 19b from which DC power is output.

[0111] The input terminal 18a, to which one end of the AC voltage source 8 is connected, is connected by a first wiring 43 to a diode connection point Nf1, which is a connection point between the diodes 9a and 9b in the diode series body 10a, and to a diode connection point Nf2, which is a connection point between the diodes 9c and 9d in the diode series body 10b. The input terminal 18b, to which the other end of the AC voltage source 8 is connected, is connected by a second wiring 44 to a winding connection point Ne, which is a connection point connecting the input sides of the windings 31 to 34 of the magnetic coupling reactor 3. The terminal of the diode 9a in the diode series body 10a opposite the diode connection point Nf1 is connected to a positive wiring 15. The terminal of the diode 9b in the diode series body 10a opposite the diode connection point Nf1 is connected to a negative wiring 16. The terminal of the diode 9c in the diode series body 10b opposite the diode connection point Nf2 is connected to the positive wiring 15. The terminal of the diode 9d in the diode series body 10b opposite the diode connection point Nf2 is connected to the negative wiring 16. The voltage sensor 11 measures the input voltage Vin, which is the voltage between the input terminals 18a and 18b. The configuration on the output side of the magnetic coupling reactor 3 relative to the winding connection point Ne and the control circuit 7 are the same as those of the power conversion device 50 of the first embodiment. The following mainly describes the parts that are different from the power conversion device 50 of the first embodiment.

[0112] The operation of two parallel interleaved totem-pole PFC converters will be described. The current path when the voltage between input terminals 18a and 18b connected to AC voltage source 8 is positive, with input terminal 18b as the reference, will be described using phase A as a representative example. When switching element Sa1 is on and switching element Sa2 is off, the current path is through input terminal 18a, diode 9a, switching element Sa1, winding 31, and input terminal 18b. When switching element Sa1 is off, the current path is through input terminal 18a, diode 9a, output capacitor 5, the diode of switching element Sa2, winding 31, and input terminal 18b. When switching element Sa1 is on, the magnetic energy stored in winding 31 serves to boost the voltage of output capacitor 5. The boosted voltage between the terminals of output capacitor 5 is supplied to connected device 6 from output terminals 19a and 19b.

[0113] The current path when the voltage between input terminals 18a and 18b connected to AC voltage source 8 is negative relative to input terminal 18b will be described below, taking phase A as an example. When switching element Sa2 is on and switching element Sa1 is off, the reverse current path is the path from input terminal 18a, diode 9b, switching element Sa2, winding 31, and input terminal 18b. When switching element Sa2 is off, the reverse current path is the path from input terminal 18a, diode 9b, output capacitor 5, the diode of switching element Sa1, winding 31, and input terminal 18b. When switching element Sa2 is on, the magnetic energy stored in winding 31 serves to boost the voltage of output capacitor 5. The boosted voltage between the terminals of output capacitor 5 is supplied to connected device 6 from output terminals 19a and 19b.

[0114] Although the explanation has been given using phase A as an example, the operating phases of the legs 41a to 41d of phases A to D that make up the switching circuit 4 are as shown in embodiment 1, with phases A and B performing PWM operation with a phase difference of 180°, and phases C and D performing PWM operation with a phase difference of 180° as well. Phases A and C perform PWM operation with an in-phase relationship, and phases B and D perform PWM operation with an in-phase relationship. The upper and lower arms may perform synchronous rectification operation by providing dead time to prevent short-circuit operation. In the case of synchronous rectification operation, the upper and lower arms operate complementarily.

[0115] In this way, control circuit 7 selectively operates the arms of switching circuit 4 depending on the voltage phase of AC voltage source 8, i.e., the phase of the polarity change of the voltage between input terminals 18a and 18b connected to AC voltage source 8 with input terminal 18b as the reference, to maintain output voltage Vout, which is the voltage between output terminals 19a and 19b, in a boosted state higher than the peak voltage of input voltage Vin, while balancing the currents in windings 31 to 34 and matching the current phase of winding currents i1 to i4 to the voltage phase of AC voltage source 8, i.e., the voltage phase of input voltage Vin, to achieve high power factor operation.

[0116] In the power conversion device 50 of the fourth embodiment, the configuration of the magnetic coupling reactor 3 can be the same as that shown in the first and second embodiments. Therefore, in the power conversion device 50 of the fourth embodiment, even in a PFC converter, when configuring a parallel converter having reactors and switching circuits in a power conversion device such as a multi-parallel power conversion device, a multiplex power conversion device, or a multiplex-multi-parallel power conversion device, the number of reactors that increases when simply parallelized or multiplexed can be reduced by using the magnetic coupling reactor 3, and the current amplitude of the windings 31 to 34 of the magnetic coupling reactor 3 and the harmonic current superimposed on the AC voltage source 8 can be reduced.

[0117] As described above, the power conversion device 50 of the fourth embodiment is a power conversion device that converts an AC voltage input between two input terminals 18a, 18b into DC power of a predetermined voltage and outputs the DC power, and includes: a switching circuit 4 in which a plurality of legs 41a, 41b, 41c, 41d are connected in parallel, each of which has an upper arm S1 and a lower arm S2 each having a switching element connected in series; a magnetic coupling reactor 3 in which the same number of windings 31, 32, 33, 34 as the legs 41a, 41b, 41c, 41d are magnetically coupled to each other; a control circuit 7 that performs PWM control with a phase difference of 180 degrees to drive the plurality of legs 41a, 41b, 41c, 41d; and two diode series bodies 10a, 10b in which two diodes 9a, 9b or 9c, 9d are connected in series. The two diode series bodies 10a, 10b are connected in parallel to the multiple legs 41a, 41b, 41c, and 41d and are arranged closer to the input terminals 18a and 18b than the multiple legs 41a, 41b, 41c, and 41d. There are four legs. One diode series body 10a and the two legs 41a and 41b with different PWM phases are defined as a first circuit 48a, and the other diode series body 10b and the two legs 41c and 41d with different PWM phases from the two legs 41a and 41b in the first circuit 48a are defined as a second circuit 48b. The first circuit 48a and the second circuit 48b are connected in parallel on the side of the output terminals 19a and 19b from which DC power is output. The ends of the windings 31, 32, 33, and 34 opposite to the legs 41 a, 41 b, 41 c, and 41 d are connected to one input terminal 18 b. In each of the diode series bodies 10 a, 10 b, the connection points (diode connection points Nf1, Nf2) of the two series-connected diodes 9 a, 9 b or 9 c, 9 d are connected to the other input terminal 18 a. The other ends of the windings 31, 32, 33, and 34 of the magnetic coupling reactor 3 are connected to the connection points (arm connection points Na) of the upper arms S1 and lower arms S2 of the corresponding legs 41 a, 41 b, 41 c, and 41 d. The degree of coupling K between the windings 31, 33 and 32, 34 connected to the legs 41 a, 41 c and 41 b, and 41 d having the same PWM phase, which is the phase of PWM control, is higher than the degree of coupling K between the windings connected to legs having different PWM phases.With this configuration, the power conversion device 50 of the fourth embodiment includes the magnetic coupling reactor 3 on the input side of the switching circuit 4, and the degree of coupling K between the windings 31, 33 and between the windings 32, 34 connected to the legs 41 a, 41 c and 41 b, 41 d having the same PWM phase, which is the phase of PWM control, is higher than the degree of coupling K between the windings connected to the legs having different PWM phases. Therefore, it is possible to increase the inductance and reduce the current amplitude of the windings by utilizing the magnitude relationship of the degree of coupling between the windings.

[0118] Fifth Embodiment. FIG. 40 is a diagram showing the configuration of a power conversion device according to a fifth embodiment, and FIG. 41 is a diagram showing the voltage of the magnetic coupling reactor of FIG. 40. The power conversion device 50 of the fifth embodiment is an example of a two-parallel interleaved totem-pole PFC converter, similar to the power conversion device 50 of the fifth embodiment, but is an example that can reduce the high-frequency current to the input terminals 18 a, 18 b more than the power conversion device 50 of the fourth embodiment. In FIG. 40, an AC voltage source 8 and a connected device 6 are also shown together with the power conversion device 50. The reference numeral 50 of the power conversion device is shown in the same manner as in FIG. 1. The power conversion device 50 of the fifth embodiment differs from the power conversion device 50 of the fourth embodiment in that the circuits connected to the input terminals 18 a, 18 b to which the AC voltage source 8 is connected are made equal, and the impedance with respect to the AC voltage source 8 is matched. The following mainly describes the differences from the power conversion device 50 of the fourth embodiment.

[0119] As in the fourth embodiment, the diode series body 10a, the A-phase leg 41a, and the B-phase leg 41b form a first circuit 48a, and the diode series body 10b, the C-phase leg 41c, and the D-phase leg 41d form a second circuit 48b. The first circuit 48a and the second circuit 48b are connected in parallel on the side of output terminals 19a and 19b from which DC power is output. The circuit connected to input terminal 18a connected to one end of AC voltage source 8 and the circuit connected to input terminal 18b connected to the other end of AC voltage source 8 will be described. The circuit connected to input terminal 18a connected to one end of AC voltage source 8 is the diode series body 10a and windings 33 and 34 of the magnetic coupling reactor 3 via first wiring 43. The input terminal 18a is connected by a first wiring 43 to a diode connection point Nf1, which is the connection point between the diode 9a and the diode 9b in the diode series body 10a, and to a winding connection point Ne2 of the windings 33, 34, which are connected to legs 41c, 41d of the C phase and the D phase, whose PWM phases of the magnetic coupling reactor 3 are 0° and 180°.

[0120] The circuit connected to input terminal 18b, which is connected to the other end of AC voltage source 8, is diode series body 10b and windings 31, 32 of magnetic coupling reactor 3, via second wiring 44. Input terminal 18b is connected, via second wiring 44, to diode connection point Nf2, which is the connection point between diode 9c and diode 9d in diode series body 10b, and to winding connection point Ne1 of windings 31, 32, which are connected to legs 41a, 41b between A phase and B phase, whose PWM phases of magnetic coupling reactor 3 are 0° and 180°. When the circuits connected to input terminals 18a, 18b are configured in this manner, the circuits connected to input terminals 18a, 18b to which AC voltage source 8 is connected, are identical, and the common-mode current flowing from input terminals 18a, 18b to AC voltage source 8 can be reduced, ideally to zero.

[0121] In the fifth embodiment, the windings 31 and 33 of the magnetic coupling reactor 3, which have the same PWM phase, are not connected, and the windings 32 and 34, which have the same PWM phase, are not connected, so the voltages across the windings 31 to 34 are as shown in FIG. 41 . The voltage across the winding 31 is a voltage V1 from the arm connection point Naa of the switching element to the winding connection point Ne1. The voltage across the winding 32 is a voltage V2 from the arm connection point Nab of the switching element to the winding connection point Ne1. The voltage across the winding 33 is a voltage V3 from the arm connection point Nac of the switching element to the winding connection point Ne2. The voltage across the winding 34 is a voltage V4 from the arm connection point Nad of the switching element to the winding connection point Ne2. Note that dashed line 29a indicates the potential level at the winding connection point Ne1, and dashed line 29b indicates the potential level at the winding connection point Ne2. The voltage between the dashed lines 29a and 29b is the input voltage Vin.

[0122] The operation of the switching circuit 4 of embodiment 5 will be described for phases A and C. When the voltage between input terminals 18a and 18b connected to the AC voltage source 8 with input terminal 18b as the reference is positive, for example, the upper arm switching element Sa1 is turned on with phase A, and the lower arm switching element Sc2 is turned on with phase C. At this time, the lower arm switching element Sa2 for phase A and the upper arm switching element Sc1 for phase C are turned off. When the voltage between input terminals 18a and 18b connected to the AC voltage source 8 with input terminal 18b as the reference is negative, for example, the lower arm switching element Sa2 is turned on with phase A, and the upper arm switching element Sc2 is turned on with phase C. At this time, the upper arm switching element Sa1 for phase A and the lower arm switching element Sc2 for phase C are turned off.

[0123] The operation of the switching circuit 4 of embodiment 5 differs from that of embodiments 1 and 4 in that, when the voltage polarity is reversed as in the case of the AC voltage source 8, and the windings 31 and 33 of the magnetic coupling reactor 3 having the same PWM phase are not connected, and the windings 32 and 34 of the magnetic coupling reactor 3 having the same PWM phase are not connected, the upper and lower arms are reversed between the legs for operation even if the PWM phases are the same, if the legs are different. Note that, when the voltage polarity is reversed as in the case of the AC voltage source 8, and the windings 31 and 33 of the magnetic coupling reactor 3 having the same PWM phase are not connected, and the windings 32 and 34 of the magnetic coupling reactor 3 having the same PWM phase are not connected, it can also be said that the current path for the AC voltage source 8 is divided into an outward path and a return path. Note that the outward path is, for example, a current path from the input terminal 18a through the first wiring 43, and the return path is a current path that passes through the second wiring 44 to the input terminal 18b. If the current path from the input terminal 18b through the second wiring 44 is the outward path, the return path is the current path that passes through the first wiring 43 and reaches the input terminal 18a.

[0124] In the switching circuit 4 of the fifth embodiment, the interlocked switching is as follows: In the legs 41a and 41c with a PWM phase of 0°, the switching elements Sa1 and Sc2 are interlocked, and the switching elements Sa2 and Sc1 are interlocked. In the legs 41b and 41d with a PWM phase of 180°, the switching elements Sb1 and Sd2 are interlocked, and the switching elements Sb2 and Sd1 are interlocked.

[0125] Although the magnetic coupling reactor 3 of the fifth embodiment has a different connection structure, i.e., connection points, for the windings 31 to 34 on the input terminals 18a and 18b side, the structure and principle of operation of the magnetic coupling reactor 3 are the same as those shown in the first to fourth embodiments. Therefore, the power conversion device 50 of the fifth embodiment achieves the same effects as the power conversion device 50 of the fourth embodiment, and furthermore, can reduce the common mode current, which is a high-frequency current to the input terminals 18a and 18b, more than the power conversion device 50 of the fourth embodiment. Therefore, the power conversion device 50 of the fifth embodiment can achieve a PFC converter with lower noise than the power conversion device 50 of the fourth embodiment.

[0126] Similarly to the power conversion device 50 of the fourth embodiment, the power conversion device 50 of the fifth embodiment is also a PFC converter, and when configuring a parallel converter having reactors and switching circuits in a power conversion device such as a multi-parallel power conversion device, a multiplex power conversion device, or a multiplex-multiple-parallel power conversion device, the number of reactors that would increase if simply paralleled or multiplexed can be reduced by using the magnetic coupling reactor 3, and the current amplitude of the windings 31 to 34 of the magnetic coupling reactor 3 and the harmonic current superimposed on the AC voltage source 8 can be reduced. Furthermore, the power conversion device 50 of the fifth embodiment can achieve a PFC converter with lower noise than the power conversion device 50 of the fourth embodiment.

[0127] As described above, the power conversion device 50 of the fifth embodiment is a power conversion device that converts an AC voltage input between two input terminals 18a, 18b into DC power of a predetermined voltage and outputs the DC power. The power conversion device 50 includes: a switching circuit 4 in which a plurality of legs 41a, 41b, 41c, 41d are connected in parallel, each of which has an upper arm S1 and a lower arm S2 each having a switching element connected in series; a magnetic coupling reactor 3 in which the same number of windings 31, 32, 33, 34 as the legs 41a, 41b, 41c, 41d are magnetically coupled to each other; a control circuit 7 that performs PWM control with phases shifted by 180 degrees to drive the plurality of legs 41a, 41b, 41c, 41d; and a first diode series body 10a and a second diode series body 10b in which two diodes 9a, 9b or two diodes 9c, 9d are connected in series. The first diode series body 10a and the second diode series body 10b are connected in parallel to the multiple legs 41a, 41b, 41c, and 41d and are arranged closer to the input terminals 18a and 18b than the multiple legs 41a, 41b, 41c, and 41d. There are four legs. The first diode series body 10a and the two legs 41a and 41b with different PWM phases form a first circuit 48a, and the second diode series body 10b and the two legs 41c and 41d with different PWM phases from the two legs 41a and 41b in the first circuit 48a form a second circuit 48b. The first circuit 48a and the second circuit 48b are connected in parallel on the side of the output terminals 19a and 19b from which DC power is output. In the first diode series body 10a, the junction point (diode junction point Nf1) of the two series-connected diodes 9a and 9b is connected to a first input terminal 18a, which is one of the input terminals. In the second diode series body 10b, the junction point (diode junction point Nf2) of the two series-connected diodes 9c and 9d is connected to a second input terminal 18b, which is the other of the input terminals. In the magnetic coupling reactor 3, one end of each of the windings 31, 32, 33, and 34 is connected to the input terminal 18a or 18b, and the other end of each of the windings 31, 32, 33, and 34 is connected to the junction point (arm junction point Na) between the upper arm S1 and the lower arm S2 of the corresponding leg 41a, 41b, 41c, and 41d. The connections of the windings 31, 32, 33, and 34 on the input terminal 18a and 18b sides are as follows.The two windings 33, 34 connected to the two legs 41c, 41d in the second circuit 48b have ends opposite to the legs 41c, 41d connected to the first input terminal 18a, and the two windings 31, 32 connected to the two legs 41a, 41b in the first circuit 48a have ends opposite to the legs 41a, 41b connected to the second input terminal 18b. In the magnetic coupling reactor 3, the degree of coupling K between the windings 31, 33 and between the windings 32, 34 connected to the legs 41a, 41c and 41b, 41d having the same PWM phase, which is the phase of PWM control, is higher than the degree of coupling K between the windings connected to legs having different PWM phases. With this configuration, the power conversion device 50 of the fifth embodiment includes the magnetic coupling reactor 3 on the input side of the switching circuit 4, and the degree of coupling K between the windings 31, 33 and between the windings 32, 34 connected to the legs 41 a, 41 c and 41 b, 41 d having the same PWM phase, which is the phase of PWM control, is higher than the degree of coupling K between the windings connected to the legs having different PWM phases. Therefore, it is possible to increase the inductance and reduce the current amplitude of the windings by utilizing the magnitude relationship of the degree of coupling between the windings.

[0128] Although the example in which the upper arm S1 and the lower arm S2 each have one switching element has been shown, the upper arm S1 and the lower arm S2 may each have a plurality of switching elements connected in series.

[0129] Furthermore, although various exemplary embodiments and examples are described in this application, the various features, aspects, and functions described in one or more embodiments are not limited to the application of a particular embodiment, but may be applied to the embodiments alone or in various combinations. Therefore, countless modifications not illustrated are contemplated within the scope of the technology disclosed in this specification. For example, this includes cases where at least one component is modified, added, or omitted, or where at least one component is extracted and combined with components of another embodiment.

[0130] 3...magnetic coupling reactor, 4...switching circuit, 7...control circuit, 9a, 9b, 9c, 9d...diodes, 10a, 10b...diode series body, 18a, 18b...input terminals, 19a, 19b...output terminals, 24a, 24b...gap centers, 31, 32, 33, 34...windings, 35...magnetic core, 37a...first core portion, 37b...second core portion, 40a...first reactor portion, 40b...second reactor portion, 41, 41a, 41b, 41c, 41d...legs, 48a...first circuit, 48b...second circuit Path, 50...power conversion device, di1 / dt, di2 / dt, di3 / dt, di4 / dt...current gradient, g1, g2...gap, i1, i2, i3, i4...winding current, Iout...output current, Ios...set value, Na, Naa, Nab, Nac, Nad...arm connection points, Nf1, Nf2...diode connection points, P1...first set, P2...second set, S1, Sa1, Sb1, Sc1, Sd1...upper arm (switching elements), S2, Sa2, Sb2, Sc2, Sd2...lower arm (switching elements)

Claims

1. A power conversion device that converts a DC or AC voltage input between two input terminals into DC power of a predetermined voltage and outputs the converted DC power, a switching circuit in which a plurality of legs, each of which is connected in parallel, are connected in series to an upper arm and a lower arm, each of which has a switching element; a magnetic coupling reactor in which the same number of windings as the number of legs are magnetically coupled to each other; a control circuit for driving the plurality of legs by performing PWM control with phases shifted by 180 degrees; The magnetic coupling reactor includes: one end of each of the windings is connected to the input terminal, and the other end of each of the windings is connected to a connection point between the upper arm and the lower arm in the corresponding leg, a degree of coupling between the windings connected to the legs having the same PWM phase, which is a phase of PWM control, is higher than a degree of coupling between the windings connected to the legs having different PWM phases; Power conversion equipment.

2. a current flowing through the winding when the lower arm is on is defined as a lower arm on-current, The leg driven by one of the PWM phases is designated as a first leg, The leg driven by the other PWM phase different from the first leg is defined as a second leg, The magnetic coupling reactor includes: a slope of the lower-arm on-current flowing through the winding connected to the first leg is reverse to a slope of the lower-arm on-current flowing through the winding connected to the second leg; The power converter according to claim 1 .

3. A magnetic flux generated when a direct current is applied to the winding from the input terminal side is defined as a direct current magnetic flux, The magnetic coupling reactor includes: The DC magnetic fluxes of the windings connected to the legs having the same PWM phase reinforce each other, The DC magnetic fluxes generated by the windings connected to the legs having different PWM phases weaken each other. The power converter according to claim 1 .

4. A magnetic flux generated when a direct current is applied to the winding from the input terminal side is defined as a direct current magnetic flux, The magnetic coupling reactor includes: The DC magnetic fluxes of the windings connected to the legs having the same PWM phase reinforce each other, The DC magnetic fluxes generated by the windings connected to the legs having different PWM phases weaken each other. The power converter according to claim 2.

5. The magnetic coupling reactor has a magnetic core around which the plurality of windings are wound and which forms a closed magnetic circuit, The magnetic core is A gap is provided between the windings connected to the legs having different PWM phases; the gap is not provided between the windings connected to the legs having the same PWM phase; The power converter according to claim 1 .

6. The magnetic coupling reactor has a magnetic core around which the plurality of windings are wound and which forms a closed magnetic circuit, The magnetic core is A gap is provided between the windings connected to the legs having different PWM phases; the gap is not provided between the windings connected to the legs having the same PWM phase; The power converter according to claim 2.

7. The magnetic coupling reactor has a magnetic core around which the plurality of windings are wound and which forms a closed magnetic circuit, The magnetic core is A gap is provided between the windings connected to the legs having different PWM phases; the gap is not provided between the windings connected to the legs having the same PWM phase; The power converter according to claim 3.

8. The magnetic coupling reactor has a magnetic core around which the plurality of windings are wound and which forms a closed magnetic circuit, The magnetic core is A gap is provided between the windings connected to the legs having different PWM phases; the gap is not provided between the windings connected to the legs having the same PWM phase; The power converter according to claim 4.

9. The magnetic core is provided with two of the gaps, a center of the magnetic coupling reactor located in a space on the inner side of the closed magnetic circuit formed by the magnetic core is defined as a reactor center; The two gaps are arranged such that the gap centers, which are intermediate positions of the gaps, are point-symmetric with respect to the center of the reactor. The power converter according to claim 5.

10. The magnetic core has two of the gaps, a center of the magnetic coupling reactor located in a space on the inner side of the closed magnetic circuit formed by the magnetic core is defined as a reactor center; The two gaps are arranged such that the gap centers, which are intermediate positions of the gaps, are point-symmetric with respect to the center of the reactor. The power converter according to claim 6.

11. The magnetic core is provided with two of the gaps, a center of the magnetic coupling reactor located in a space on the inner side of the closed magnetic circuit formed by the magnetic core is defined as a reactor center; The two gaps are arranged such that the gap centers, which are intermediate positions of the gaps, are point-symmetric with respect to the center of the reactor. The power converter according to claim 7.

12. The magnetic core has two of the gaps, a center of the magnetic coupling reactor located in a space on the inner side of the closed magnetic circuit formed by the magnetic core is defined as a reactor center; The two gaps are arranged such that the gap centers, which are intermediate positions of the gaps, are point-symmetric with respect to the center of the reactor. The power converter according to claim 8.

13. the magnetic core has a first core portion and a second core portion separated by the gap, the magnetic coupling reactor includes a first reactor portion in which two of the windings are wound around the first core portion, and a second reactor portion in which the other two windings are wound around the second core portion, The first reactor unit and the second reactor unit are paired with each other and have the same configuration. The power converter according to claim 5.

14. The magnetic core has a first core portion and a second core portion separated by the gap, the magnetic coupling reactor includes a first reactor portion in which two of the windings are wound around the first core portion, and a second reactor portion in which the other two windings are wound around the second core portion, The first reactor unit and the second reactor unit are paired with each other and have the same configuration. The power converter according to claim 6.

15. The magnetic core has a first core portion and a second core portion separated by the gap, the magnetic coupling reactor includes a first reactor portion in which two of the windings are wound around the first core portion, and a second reactor portion in which the other two windings are wound around the second core portion, The first reactor unit and the second reactor unit are paired with each other and have the same configuration. The power converter according to claim 7.

16. The magnetic core has a first core portion and a second core portion separated by the gap, the magnetic coupling reactor includes a first reactor portion in which two of the windings are wound around the first core portion, and a second reactor portion in which the other two windings are wound around the second core portion, The first reactor unit and the second reactor unit are paired with each other and have the same configuration. The power converter according to claim 8.

17. The magnetic core has a first core portion and a second core portion separated by the gap, the magnetic coupling reactor includes a first reactor portion in which two of the windings are wound around the first core portion, and a second reactor portion in which the other two windings are wound around the second core portion, The first reactor unit and the second reactor unit are paired with each other and have the same configuration. The power converter according to claim 9.

18. The magnetic core has a first core portion and a second core portion separated by the gap, the magnetic coupling reactor includes a first reactor portion in which two of the windings are wound around the first core portion, and a second reactor portion in which the other two windings are wound around the second core portion, The first reactor unit and the second reactor unit are paired with each other and have the same configuration. The power converter according to claim 10.

19. The magnetic core has a first core portion and a second core portion separated by the gap, the magnetic coupling reactor includes a first reactor portion in which two of the windings are wound around the first core portion, and a second reactor portion in which the other two windings are wound around the second core portion, The first reactor unit and the second reactor unit are paired with each other and have the same configuration. The power converter according to claim 11.

20. The magnetic core has a first core portion and a second core portion separated by the gap, the magnetic coupling reactor includes a first reactor portion in which two of the windings are wound around the first core portion, and a second reactor portion in which the other two windings are wound around the second core portion, The first reactor unit and the second reactor unit are paired with each other and have the same configuration. The power converter according to claim 12.

21. the switching circuit includes a first set and a second set having two legs with the PWM phases shifted by 180 degrees; The control circuit includes: When an output current output by the switching circuit is equal to or greater than a predetermined set value, the first set and the second set are driven; driving one of the first set and the second set when the output current is less than a predetermined set value; 21. The power conversion device according to claim 1 .

22. A DC voltage is input between the two input terminals, One end of each of the windings is connected to the positive input terminal.

21. The power conversion device according to claim 1 .

23. A DC voltage is input between the two input terminals, One end of each of the windings is connected to the positive input terminal.

22. The power converter of claim 21.

24. An AC voltage is input between the two input terminals, The diode series circuit includes two diodes connected in series, the two diode series bodies are connected in parallel to the plurality of legs and are disposed closer to the input terminal than the plurality of legs; The number of legs is four, one of the diode series circuits and the two legs having different PWM phases from each other are defined as a first circuit; a second circuit includes the other diode series circuit and two legs different from the two legs in the first circuit and having PWM phases different from each other; the first circuit and the second circuit are connected in parallel on an output terminal side from which the DC power is output, Each of the windings has an end opposite to the leg connected to one of the input terminals, In each of the diode series elements, a node between two of the diodes connected in series is connected to the other of the input terminals.

21. The power conversion device according to claim 1 .

25. An AC voltage is input between the two input terminals, The diode series circuit includes two diodes connected in series, the two diode series bodies are connected in parallel to the plurality of legs and are disposed closer to the input terminal than the plurality of legs; The number of legs is four, one of the diode series circuits and the two legs having different PWM phases from each other are defined as a first circuit; a second circuit includes the other diode series circuit and two legs different from the two legs in the first circuit and having PWM phases different from each other; the first circuit and the second circuit are connected in parallel on an output terminal side from which the DC power is output, Each of the windings has an end opposite to the leg connected to one of the input terminals, In each of the diode series elements, a node between two of the diodes connected in series is connected to the other of the input terminals.

22. The power converter of claim 21.

26. An AC voltage is input between the two input terminals, The power supply circuit includes a first diode series body and a second diode series body in which two diodes are connected in series, the first diode series body and the second diode series body are connected in parallel to the plurality of legs and are disposed closer to the input terminal than the plurality of legs, The number of legs is four, the first diode series circuit and the two legs having different PWM phases are defined as a first circuit; a second circuit is formed by including the second diode series circuit and two legs that are different from the two legs in the first circuit and have PWM phases different from each other; the first circuit and the second circuit are connected in parallel on an output terminal side from which the DC power is output, the first diode series body has a connection point of two of the diodes connected in series connected to a first input terminal which is one of the input terminals, the second diode series body has a connection point between two of the diodes connected in series and connected to a second input terminal which is the other of the input terminals, the two windings connected to the two legs in the second circuit have ends opposite to the legs connected to the first input terminal; The two windings connected to the two legs in the first circuit have ends opposite to the legs connected to the second input terminal.

21. The power conversion device according to claim 1 .

27. ​​An AC voltage is input between the two input terminals, The power supply circuit includes a first diode series body and a second diode series body in which two diodes are connected in series, the first diode series body and the second diode series body are connected in parallel to the plurality of legs and are disposed closer to the input terminal than the plurality of legs, The number of legs is four, the first diode series circuit and the two legs having different PWM phases are defined as a first circuit; a second circuit is formed by including the second diode series circuit and two legs that are different from the two legs in the first circuit and have PWM phases different from each other; the first circuit and the second circuit are connected in parallel on an output terminal side from which the DC power is output, the first diode series body has a connection point of two of the diodes connected in series connected to a first input terminal which is one of the input terminals, the second diode series body has a connection point between two of the diodes connected in series and connected to a second input terminal which is the other of the input terminals, the two windings connected to the two legs in the second circuit have ends opposite to the legs connected to the first input terminal; The two windings connected to the two legs in the first circuit have ends opposite to the legs connected to the second input terminal.

22. The power converter of claim 21.