Device and method for thermal distribution switching control in resonant bidirectional converter

Abstract

The present invention relates to a heat dissipation switching control device and method, and more specifically, to a heat dissipation switching control device and method for a resonant bidirectional converter. According to one embodiment of the present invention, the present invention can evenly distribute switch losses, lower the maximum temperature applied to the switch, extend the lifespan of the power semiconductor switch, and simplify the design of the heat sink of the switch.

Description

Technology Field

[0001] The present invention relates to a thermal dissipation switching control device and method, and more specifically, to a thermal dissipation switching control device and method for a resonant bidirectional converter. Background Technology

[0003] A resonant bidirectional converter can be implemented by utilizing four switches on the primary side of the high-frequency transformer, four switching elements on the secondary side of the high-frequency transformer, a resonant inductor, a resonant capacitor, and an external inductor for zero-voltage switching.

[0004] In a resonant bidirectional converter, the switch located on the primary full-bridge can be turned on with zero-voltage switching in forward buck mode, and the switch located on the secondary full-bridge can be turned on with zero-voltage switching in forward boost mode. Since the resonant bidirectional converter has a symmetrical structure, zero-voltage switching is possible in reverse boost and buck modes as well as in forward boost and buck modes. The resonant bidirectional converter enables power conversion with high efficiency under high-frequency switching. The resonant bidirectional converter can generally be utilized in various applications, such as battery chargers and dischargers.

[0005] However, in a resonant bidirectional converter, the switching loss that occurs when the two primary switches are turned off is different from the switching loss that occurs when the other two switches are turned off, so there is a problem where the thermal balance between the switches is disrupted during high-frequency switching.

[0006] In addition, since the dead time required for zero-voltage switching is different for two pairs of switches in the resonant bidirectional converter, there is a problem in that two of the four switches do not switch to zero voltage when the same dead time is applied.

[0007] To solve these problems, it is necessary to provide a thermal dissipation switching control device and method for a resonant bidirectional converter.

[0008] The background technology of the present invention is disclosed in Korean Registered Patent No. 10-2335940. The problem to be solved

[0010] The present invention provides a thermal dissipation switching control device and method for a resonant bidirectional converter that continuously switches the roles between primary and secondary switches. means of solving the problem

[0012] According to one aspect of the present invention, a thermal dissipation switching control device for a resonant bidirectional converter is provided.

[0013] A thermal dissipation switching control device of a resonant bidirectional converter according to one embodiment of the present invention may include a first full-bridge section that generates an AC voltage by alternately performing switching based on an input voltage, an auxiliary inductor section that receives the AC voltage and adjusts it according to resonant characteristics, an LC resonant tank section that forms resonance based on the adjusted AC voltage and transmits a resonant voltage, a transformer section that receives the resonant voltage and steps up or steps down the AC voltage according to a winding ratio, a magnetic inductor section that receives and adjusts the stepped-up or stepped-down AC voltage, and a second full-bridge section that rectifies the adjusted AC voltage by performing switching and generates a DC voltage.

[0014] According to another aspect of the present invention, a thermal dissipation switching control method for a resonant bidirectional converter is provided.

[0015] A thermal dissipation switching control method for a resonant bidirectional converter according to one embodiment of the present invention may include the steps of: generating an AC voltage by alternately performing switching on a switch based on an input voltage; receiving the AC voltage and adjusting the AC voltage to match resonant characteristics; forming resonance based on the adjusted AC voltage and transmitting a resonant voltage; receiving the resonant voltage and stepping up or stepping down the AC voltage according to a winding ratio; receiving and adjusting the stepped-up or stepped-down AC voltage; and rectifying the adjusted AC voltage by performing switching and generating a DC voltage. Effects of the invention

[0017] According to one embodiment of the present invention, the present invention can evenly distribute switch losses.

[0018] According to one embodiment of the present invention, the present invention can lower the maximum temperature applied to the switch and extend the lifespan of the power semiconductor switch.

[0019] According to one embodiment of the present invention, the present invention can simplify the design of a heat sink for a switch.

[0020] The effects of the present invention are not limited to the effects described above, and should be understood to include all effects that can be inferred from the composition of the invention described in the description or claims of the present invention. Brief explanation of the drawing

[0022] FIG. 1 is a drawing showing a thermal dissipation switching control device of a resonant bidirectional converter according to one embodiment of the present invention. FIG. 2 is a flowchart illustrating a thermal dissipation switching control method of a resonant bidirectional converter according to one embodiment of the present invention. FIG. 3 is a circuit diagram of a resonant bidirectional converter according to one embodiment of the present invention. FIG. 4 is a diagram showing the theoretical waveform of a resonant bidirectional converter in forward buck mode according to one embodiment of the present invention. FIG. 5 is a diagram showing the sequence of operations in a forward buck mode according to an embodiment of the present invention. FIG. 6 is a diagram showing the theoretical waveform of a resonant bidirectional converter in a forward boost mode according to one embodiment of the present invention. FIG. 7 is a diagram showing the sequence of operations in a forward boost mode according to an embodiment of the present invention. FIG. 8 is a curve showing the dead time required for the switch of a resonant bidirectional converter according to one embodiment of the present invention. FIGS. 9 and FIGS. 10 are drawings showing simulation waveforms of a resonant bidirectional converter according to one embodiment of the present invention. FIG. 11 is a diagram showing the turn-off switching loss in an upper switch and a lower switch according to an embodiment of the present invention. FIG. 12 shows a prototype of a resonant DC / DC converter according to one embodiment of the present invention. FIG. 13 shows the design specifications and parameters of a converter according to one embodiment of the present invention. FIG. 14 is a diagram showing an experimental waveform in a forward buck mode according to one embodiment of the present invention. FIG. 15 is a diagram showing an experimental waveform in a forward boost mode according to one embodiment of the present invention. FIG. 16 is a diagram showing an experimental waveform during load fluctuation according to one embodiment of the present invention. FIG. 17 is a diagram showing the efficiency curves of the buck mode and boost mode in the forward and reverse directions according to one embodiment of the present invention. FIG. 18 is a diagram comparing the temperature difference between an upper switch and a lower switch in buck mode or boost mode according to one embodiment of the present invention. FIG. 19 is a diagram showing constant current mode and constant voltage mode operation through PSIM simulation according to one embodiment of the present invention. Specific details for implementing the invention

[0023] The present invention is susceptible to various modifications and may have various embodiments; therefore, specific embodiments are illustrated in the drawings and described in detail. However, this is not intended to limit the present invention to specific embodiments, and it should be understood that it includes all modifications, equivalents, and substitutions that fall within the spirit and scope of the invention. In describing the present invention, detailed descriptions of related prior art are omitted if it is determined that such detailed descriptions may unnecessarily obscure the essence of the invention. Furthermore, singular expressions used in this specification and claims should generally be interpreted as meaning "one or more" unless otherwise stated.

[0024] Throughout the specification, when it is stated that a part is "connected (connected, in contact, combined)" with another part, this includes not only cases where they are "directly connected," but also cases where they are "indirectly connected" with other members interposed between them. Furthermore, when it is stated that a part "includes" a certain component, this means that, unless specifically stated otherwise, it does not exclude other components but rather allows for the inclusion of additional components.

[0025] The terms used herein are merely for describing specific embodiments and are not intended to limit the invention. Singular expressions include plural expressions unless the context clearly indicates otherwise. In this specification, terms such as “comprising” or “having” are intended to indicate the presence of the features, numbers, steps, actions, components, parts, or combinations thereof described in the specification, and should be understood as not precluding the existence or addition of one or more other features, numbers, steps, actions, components, parts, or combinations thereof.

[0026] The present invention will be described below with reference to the attached drawings. However, the present invention may be implemented in various different forms and is not limited to the embodiments described herein. Furthermore, in order to clearly explain the present invention in the drawings, parts unrelated to the explanation have been omitted, and similar parts throughout the specification have been given similar reference numerals.

[0027] FIG. 1 is a drawing showing a thermal dissipation switching control device of a resonant bidirectional converter according to one embodiment of the present invention.

[0028] Referring to FIG. 1, the thermal dissipation switching control device of a resonant bidirectional converter includes a first full-bridge section (110), an auxiliary inductor section (120), an LC resonant tank section (130), a transformer section (140), a magnetic inductor section (150), and a second full-bridge section (160).

[0029] Resonant bidirectional converters have two operating modes: buck mode and boost mode. Buck mode is a step-down mode that transfers power from the primary side to the secondary side when the primary side voltage is higher and the secondary side voltage is lower. Boost mode is a step-up mode that transfers power from the secondary side to the primary side when the secondary side voltage is higher and the primary side voltage is lower.

[0030] Referring to FIG. 1, in buck mode, the thermal dissipation switching control device operates in the order of the first full bridge section (110), auxiliary inductor section (120), LC resonant tank section (130), transformer section (140), magnetic inductor section (150), and second full bridge section (160).

[0031] The first full bridge section (110) switches the input power to supply power to the resonant circuit.

[0032] The first full bridge section (110) is an input voltage , input capacitor , switch , , , , parallel capacitor , , , , diode , , , Includes

[0033] The first full bridge section (110) is an input voltage Switch based on , , , Alternately performing switching to AC voltage Creates.

[0034] The first full bridge section (110) is a switch , , , switch pair and switch pair It is classified as follows.

[0035] For example, the gate driving signal is a switch pair main duty ratio and complementary duty ratio It consists of, and the main duty and complementary duty are divided into a fixed duty of 0.5.

[0036] That is, switching pair The gate is driven by a gate driving signal in which the main duty and complementary duty are divided into a fixed duty of 0.5.

[0037] Switch pair silver switch pair It is driven by the same gate driving signal, but is driven by applying a 360° phase difference.

[0038] Since the asymmetric current flows through all switches during two switching cycles, all switches depend on the magnitude of the Zero Voltage Switching (ZVS) current or Dead time is required.

[0039] The switch requires basic dead time and additional dead time when having a weak ZVS current, and only basic dead time when having a strong ZVS current.

[0040] The first full bridge section (110) converts power through switching and supplies voltage to nodes a and b, which are the primary side input terminals of the transformer. For example, the switch , When it is turned on, between the primary transformer input terminals a and b The voltage is formed and the switch , When it turns on A voltage is formed.

[0041] The first full bridge section (110) is an alternating voltage generated through switching. It outputs and transmits to the auxiliary inductor section (120).

[0042] The first full bridge section (110) minimizes switching losses by utilizing ZVS.

[0043] The auxiliary inductor section (120) is an auxiliary inductor Includes

[0044] The auxiliary inductor section (120) is the AC voltage generated in the first full-bridge section (110). It receives input, adjusts it according to the resonance characteristics, and transmits it to the LC resonance tank section (130).

[0045] The auxiliary inductor (120) complements the operation of the resonant circuit and induces the resonant current to flow in an optimal form.

[0046] The auxiliary inductor (120) regulates the current flowing during switching to mitigate excessive voltage changes and helps the switch achieve ZVS over a wider range.

[0047] The LC resonant tank section (130) is a resonant capacitor and resonant inductor Includes

[0048] The LC resonant tank section (130) forms resonance based on the AC voltage regulated by the auxiliary inductor section (120) and transmits a stable resonant voltage to the transformer section (140).

[0049] The LC resonant tank section (130) is and It induces natural resonance to smoothly regulate changes in voltage and current, thereby satisfying the ZVS condition during switching and minimizing switching losses.

[0050] The LC resonant tank section (130) optimizes the voltage and current waveforms in the resonant circuit so that the energy transferred to the transformer section (140) can be efficiently converted.

[0051] The transformer section (140) is a high-frequency transformer Includes

[0052] The transformer section (140) receives the resonant voltage generated in the LC resonant tank section (130) and, according to the winding ratio (turns ratio), increases or decreases the AC voltage and transmits it to the magnetic inductor section (150).

[0053] The transformer section (140) provides electrical insulation between the primary side and the secondary side to ensure the stability of the system and allows the power conversion formed in the resonant circuit to be efficiently transferred to the secondary side.

[0054] The self-inductor part (150) is a self-inductance Includes

[0055] The magnetic inductor section (150) receives the AC voltage boosted or stepped down from the transformer section (140), stably adjusts it, strengthens the DC component, and transmits it to the second full bridge section (160).

[0056] The magnetic inductor section (150) reduces the ripple of the output current through a magnetic storage function and increases power conversion efficiency by compensating for the energy generated during the switching process.

[0057] The second full bridge section (160) is an output voltage , output capacitor , switch , , , , parallel capacitor , , , , diode , , , Includes

[0058] The second full bridge section (160) receives the voltage converted from the magnetic inductor section (150), performs switching, converts the output power into a form suitable for the load, and finally delivers it.

[0059] The second full bridge section (160) is a switch , , , switch pair and switch pair It is classified as follows.

[0060] For example, the gate driving signal is a switch pair main duty ratio and complementary duty ratio It consists of, and the main duty and complementary duty are divided into a fixed duty of 0.5.

[0061] That is, switching pair The gate is driven by a gate driving signal in which the main duty and complementary duty are divided into a fixed duty of 0.5.

[0062] Switch pair silver switch pair It is driven by the same gate driving signal, but is driven by applying a 360° phase difference.

[0063] Since the asymmetric current flows through all switches during two switching cycles, all switches depend on the magnitude of the Zero Voltage Switching (ZVS) current or Dead time is required.

[0064] The switch requires basic dead time and additional dead time when having a weak ZVS current, and only basic dead time when having a strong ZVS current.

[0065] The second full bridge section (160) rectifies the AC voltage through switching to generate a DC voltage, and converts the voltage into a stable DC voltage by smoothing it through an output capacitor.

[0066] The second full bridge section (160) operates to stably maintain the DC voltage on the output side and performs optimal switching control according to load conditions.

[0067] Referring to FIG. 1, in boost mode, the heat dissipation switching control device operates in the order of the second full bridge section (160), the magnetic inductor section (150), the transformer section (140), the LC resonant tank section (130), the auxiliary inductor section (120), and the first full bridge section (110).

[0069] FIG. 2 is a flowchart illustrating a thermal dissipation switching control method of a resonant bidirectional converter according to one embodiment of the present invention.

[0070] In step S210, the heat dissipation switching control device generates an AC voltage by having the switches alternately perform switching based on the input voltage.

[0071] For example, a switch , , , and, the above switch is a switch pair and switch pair It is classified as follows.

[0072] The above switch pair The main duty and complementary duty are driven by a gate driving signal in which the main duty and complementary duty are divided by a fixed duty of 0.5, and the switch pair The above switch pair It is driven by the same gate driving signal as above, but is driven by applying a 360° phase difference.

[0073] The switch requires basic dead time and additional dead time when having a weak ZVS current, and only basic dead time when having a strong ZVS current.

[0074] In step S220, the heat dissipation switching control device receives an AC voltage and adjusts the AC voltage to match the resonance characteristics.

[0075] In step S230, the heat dissipation switching control device forms a resonance based on the regulated AC voltage and transmits a resonant voltage.

[0076] In step S240, the heat dissipation switching control device receives a resonant voltage and steps up or steps down the AC voltage according to the winding ratio.

[0077] In step S250, the heat dissipation switching control device receives and regulates the boosted or stepped-down AC voltage.

[0078] In step S260, the heat dissipation switching control device performs switching to rectify the regulated AC voltage and generate a DC voltage.

[0079] The above switch , , , and, the above switch is a switch pair and switch pair It is classified as follows.

[0080] The above switch pair The main duty and complementary duty are driven by a gate driving signal in which the main duty and complementary duty are divided by a fixed duty of 0.5, and the switch pair The above switch pair It is driven by the same gate driving signal as above, but is driven by applying a 360° phase difference.

[0081] The switch requires basic dead time and additional dead time when having a weak ZVS current, and only basic dead time when having a strong ZVS current.

[0083] FIG. 3 is a circuit diagram of a resonant bidirectional converter according to one embodiment of the present invention.

[0084] Referring to FIG. 3, the switching waveform and loss applied during resonant bidirectional converter and thermal dissipative switching control are shown.

[0085] The heat dissipation switching control of the present invention can be implemented in both analog and digital ways.

[0086] Referring to FIG. 3, FIG. 3 (a) is the result of applying a conventional duty modulation method, and FIG. 3 (b) is the result of applying the divided complementary duty modulation method of the present invention.

[0087] Referring to FIG. 3(a), when a resonant bidirectional converter operates in forward buck mode using a conventional duty modulation method, a switch pair class They have a 180° phase difference from each other and the same duty cycle Wow, Dead Time It is individually controlled by two PWM signals having.

[0088] Conventional duty modulation methods have a problem in that high heat is generated at the upper switch on the primary side, whereas low heat is generated at the lower switch on the primary side.

[0089] To solve this problem of switching loss imbalance, the divided complementary duty modulation method of the present invention is applied to a resonant bidirectional converter.

[0090] Referring to FIG. 3(b), the gate driving signal is a switch pair main duty ratio and complementary duty ratio It consists of, and the main duty and complementary duty are divided into a fixed duty of 0.5.

[0091] That is, switching pair The gate is driven by a gate driving signal in which the main duty and complementary duty are divided into a fixed duty of 0.5.

[0092] Referring to FIG. 3(b), the switch pair silver switch pair It is driven by the same gate driving signal, but is driven by applying a 360° phase difference.

[0093] Referring to FIG. 3(b), since an asymmetric current flows through all switches during two switching cycles, all switches [depending on] the magnitude of the Zero Voltage Switching (ZVS) current or Dead time is required.

[0094] Referring to FIG. 3(b), the divided complementary duty modulation method of the present invention can solve the heat generation problem by evenly distributing the switching loss on the primary side.

[0095] Referring to FIG. 3(a), when a resonant bidirectional converter operates in forward boost mode using a conventional duty modulation method, a switch pair class They have a 180° phase difference from each other and the same duty cycle Wow, Dead Time It is individually controlled by two PWM signals having.

[0096] Conventional duty modulation methods have a problem in that high heat is generated in the lower switch on the secondary side, whereas low heat is generated in the upper switch on the secondary side.

[0097] To solve this problem of switching loss imbalance, the divided complementary duty modulation method of the present invention is applied to a resonant bidirectional converter.

[0098] Referring to FIG. 3(b), the gate driving signal is a switch pair main duty ratio and complementary duty ratio It consists of, and the main duty and complementary duty are divided into a fixed duty of 0.5.

[0099] That is, switching pair The gate is driven by a gate driving signal in which the main duty and complementary duty are divided into a fixed duty of 0.5.

[0100] Referring to FIG. 3(b), the switch pair silver switch pair It is driven by the same gate driving signal, but is driven by applying a 360° phase difference.

[0101] Referring to FIG. 3(b), since an asymmetric current flows through all switches during two switching cycles, all switches [depending on] the magnitude of the Zero Voltage Switching (ZVS) current or Dead time is required.

[0102] Referring to FIG. 3(b), the divided complementary duty modulation method of the present invention can solve the heat generation problem by evenly distributing the switching loss on the secondary side.

[0103] Power can be transmitted in both forward and reverse directions. However, the switching frequency and theoretical waveform are identical in bidirectional power flow due to the symmetry of circuit operation.

[0104] Therefore, the present invention is explained by considering only the buck mode and boost mode in the forward direction.

[0106] FIG. 4 is a diagram showing the theoretical waveform of a resonant bidirectional converter in forward buck mode according to one embodiment of the present invention.

[0107] Referring to FIG. 4, the switching method of the present invention can generate a switching signal in which a signal with a pulse width of 0.5 is additionally inserted in the middle by comparing it with a conventional primary side switching signal in a forward step-down mode.

[0108] In the switching method of the present invention, the loss occurring when the primary side switch is turned off is distributed evenly to each switch during two switching cycles.

[0110] FIG. 5 is a diagram showing the sequence of operations in a forward buck mode according to an embodiment of the present invention.

[0111] Referring to FIG. 5, FIG. 5 (a) represents section 1, FIG. 5 (b) represents section 2, FIG. 5 (c) represents section 3, FIG. 5 (d) represents section 4, FIG. 5 (e) represents section 5, and FIG. 5 (f) represents section 6.

[0112] Referring to FIG. 5(a), section 1 is [ , ]am.

[0113] Referring to FIG. 5 (a), Previously, the switch , , and This is turned on, and the primary side voltage of the transformer It is fixed at 0V, and the transformer's secondary voltage silver It is maintained as.

[0114] Referring to FIG. 5 (a), switch at class It turns off.

[0115] Referring to FIG. 5(a), the magnetizing inductor current is a switch and It flows in the negative (-) direction through, and the diode and He attains enlightenment.

[0116] Referring to Fig. 5(b), section 2 is [ , ]am.

[0117] Referring to Fig. 5(b), switch at and It turns on with this ZVS, but the diode during the dead time and This was because he was in enlightenment.

[0118] Referring to Fig. 5(b), Is It is switched to, silver It increases linearly in proportion to.

[0119] Referring to Fig. 5(c), section 3 is [ , ]am.

[0120] Referring to Fig. 5 (c), switch at It turns off.

[0121] Referring to FIG. 5(c), the auxiliary inductor current It flows in the negative (-) direction, and the diode He attains enlightenment. In this process Is It is switched to Is While maintaining, Is It increases linearly in proportion to.

[0122] Referring to Fig. 5(d), section 4 is [ , ]am.

[0123] Referring to Fig. 5 (d), switch at It turns on with this ZVS, diode This was because he was in enlightenment.

[0124] Referring to FIG. 5(d), a resonant inductor is a resonant capacitor It continuously resonates with.

[0125] Referring to Fig. 5(e), section 5 is [ , ]am.

[0126] Referring to Fig. 5 (e), switch at It turns off.

[0127] Referring to FIG. 5(e), the resonant inductor current and is a switch Since it flows through, After attaining enlightenment Discharges.

[0128] Referring to Fig. 5(e), It is fixed at 0V and Is Maintains.

[0129] Referring to (f) in Fig. 5, section 6 is [ , ]am.

[0130] Referring to (f) of Fig. 5, switch at It turns on with this ZVS, diode This was because he was in enlightenment.

[0131] Referring to (f) of Fig. 5, and is a switch class It circulates freely through.

[0133] FIG. 6 is a diagram showing the theoretical waveform of a resonant bidirectional converter in a forward boost mode according to one embodiment of the present invention.

[0134] Referring to FIG. 6, the switching method of the present invention can generate a switching signal in which a signal with a pulse width of 0.5 is additionally inserted in the middle by comparing it with a conventional secondary switching signal in a forward boost mode.

[0135] In the switching method of the present invention, the loss occurring when the secondary side switch is turned off is distributed evenly to each switch during two switching cycles.

[0137] FIG. 7 is a diagram showing the sequence of operations in a forward boost mode according to an embodiment of the present invention.

[0138] Referring to FIG. 7, FIG. 7 (a) represents section 1, FIG. 7 (b) represents section 2, FIG. 7 (c) represents section 3, FIG. 7 (d) represents section 4, FIG. 7 (e) represents section 5, and FIG. 7 (f) represents section 6.

[0139] Referring to FIG. 7(a), section 1 is [ , ]am.

[0140] Referring to Fig. 7(a), switch at and This turns off.

[0141] Referring to Fig. 7(a), is a switch class Since it flows through, the diode class He attains enlightenment.

[0142] Referring to Fig. 7(a), Is It is switched to It is maintained at 0V.

[0143] Referring to Fig. 7(b), section 2 is [ , ]am.

[0144] Referring to Fig. 7(b), switch at class It turns on with ZVS, the diode class This is because he was in a state of enlightenment.

[0145] Referring to Fig. 7(b), It increases linearly.

[0146] Referring to Fig. 7(c), section 3 is [ , ]am.

[0147] Referring to Fig. 7 (c), Previously, the switch , , and This turns on, Is is It is fixed at 0V.

[0148] Referring to Fig. 7 (c), switch at This turns off.

[0149] Referring to Fig. 7 (c), is a switch Since it flows through, the diode He attains enlightenment.

[0150] Referring to Fig. 7 (c), Is It is switched to, Is It increases linearly in proportion to.

[0151] Referring to Fig. 7(d), section 4 is [ , ]am.

[0152] Referring to (d) of Fig. 7, switch at It turns on with ZVS, the diode This was because he was in enlightenment.

[0153] Referring to (d) of Fig. 7, Is It continuously resonates with.

[0154] Referring to Fig. 7(e), section 5 is [ , ]am.

[0155] Referring to Fig. 7 (e), switch at This turns off.

[0156] Referring to Fig. 7 (e), Since it flows in the positive (+) direction, the diode He attains enlightenment.

[0157] Referring to Fig. 7 (e), It switches to 0V. Is Maintains.

[0158] Referring to (f) in Fig. 7, section 6 is [ , ]am.

[0159] Referring to (f) of Fig. 7, switch at It turns on with ZVS, the diode This was because he was in enlightenment.

[0161] FIG. 8 is a curve showing the dead time required for the switch of a resonant bidirectional converter according to one embodiment of the present invention.

[0162] Referring to FIG. 8, FIG. 8 (a) shows the minimum dead time according to the ZVS current of the primary switch in buck mode, and FIG. 8 (b) shows the minimum dead time according to the ZVS current of the secondary switch in boost mode.

[0163] Referring to Fig. 8, the x-axis represents Power [W], and the y-axis represents the Auxiliary Inductor. [μH] or Magnetizing Inductor [μH], and the z-axis represents the minimum dead-time [ns].

[0164] Referring to Fig. 8, the dead time required for a switch with a weak ZVS current and the dead time required for a switch with a strong ZVS current are different for the primary side and the secondary side, respectively.

[0165] Conventional switching methods use different dead times, but the switching method of the present invention enables ZVS by naturally applying different dead times.

[0166] Referring to Fig. 8, the basic dead time is the dead time that is essentially required for the primary switch and the secondary switch having a strong ZVS current.

[0167] Referring to Fig. 8, the additional dead time is the additional dead time required for the primary switch and the secondary switch having a weak ZVS current.

[0168] The divided complementary duty modulation method of the present invention During this period, the average current value is the same as the average current value when using the conventional phase shift modulation method.

[0169] That is, the resulting voltage gain in buck mode and boost mode is the same as the conventional method.

[0170] Here, the voltage gain of buck mode is represented by Equation 1 below, and the voltage gain of boost mode is represented by Equation 2 below.

[0171] [Mathematical Formula 1]

[0172]

[0174] [Mathematical Formula 2]

[0175]

[0176] Here, is the primary side voltage, is the secondary voltage, is the primary duty cycle in buck mode, represents the secondary duty ratio in boost mode.

[0177] Referring to Fig. 8(a), the dead time required to completely discharge the parasitic capacitance of the primary side switch having a weak ZVS current in buck mode can be expressed as Equation 3 below.

[0178] [Mathematical Formula 3]

[0179]

[0180] Here, is default dead time, is additional dead time, is the parasitic capacitance of the primary side switch, is the primary side voltage, is the secondary voltage, is the primary duty cycle in buck mode, is the switching period, is an auxiliary inductor, is the resonant impedance, represents output power.

[0181] Referring to Fig. 8(a), the dead time required for the primary switch having a strong ZVS current in buck mode can be expressed as Equation 4 below.

[0182] [Mathematical Formula 4]

[0183]

[0184] Referring to Fig. 8(b), the dead time required to completely discharge the parasitic capacitance of the secondary switch having a strong ZVS current in boost mode can be expressed as Equation 5 below.

[0185] [Mathematical Formula 5]

[0186]

[0187] Here, is default dead time, is the parasitic capacitance of the secondary side switch, is the secondary duty cycle in boost mode, represents the current of the resonant inductor.

[0188] Referring to Fig. 8(b), the dead time required for the secondary switch having a weak ZVS current in boost mode can be expressed as Equation 6 below.

[0189] [Mathematical Formula 6]

[0190]

[0191] Here, is additional dead time, represents a magnetizing inductor.

[0193] FIGS. 9 and FIGS. 10 are drawings showing simulation waveforms of a resonant bidirectional converter according to one embodiment of the present invention.

[0194] Referring to Fig. 9, This shows a resonant bidirectional converter applying the divided complementary duty modulation method of the present invention at the full load when V is 480V.

[0195] Referring to Fig. 9, the simulation signal is a gate signal and , resonant inductor current , drain-source voltage and , switch current and There is.

[0196] Referring to FIG. 9, FIG. 9(a) shows the case where there is no parasitic resistance of the resonant inductor and resonant capacitor. and is 0Ω, and FIG. 9(b) shows the case where the actual measured parasitic resistance value is applied. is 0.2Ω, is 0.002 Ω, and Fig. 9(c) shows the case where the parasitic resistance is very large, and is 1.5Ω, is 0.08Ω.

[0197] Referring to Figure 9(a), the ideal shape of the resonant current waveform is shown by assuming that the resonant tank has no parasitic resistance.

[0198] Referring to Fig. 9, the value measured experimentally to determine the effect of the parasitic resistance of the resonant tank on the ZVS of the switch is the DC resistance of the resonant inductor. The equivalent series resistance of this 0.2Ω resonant capacitor This is 0.002Ω.

[0199] Referring to Fig. 9(b), for the LC tank is 0.2Ω, After setting it to 0.002 Ω, run the PSIM simulation.

[0200] Referring to Figure 9(b), the resonant current waveform is slightly distorted, but the weak ZVS current is almost the same as in Figure 9(a) when there is no parasitic resistance.

[0201] Referring to FIG. 9(c), compared to FIG. 9(a) and FIG. 9(b) class By setting it to a much larger value, the weak ZVS current becomes very low. Therefore, the circuit may lose its ZVS function.

[0202] That is, referring to Fig. 9, it can be seen that ZVS operation is maintained at an appropriate parasitic resistance value, but ZVS may become impossible if the resistance is excessively large.

[0203] Therefore, it is important to minimize the parasitic resistance of the resonance tank, and this effect must be considered during the design.

[0205] Referring to Fig. 10 (a), The simulation results of a resonant bidirectional converter applying the divided complementary duty modulation method of the present invention at the full load when V is 1000V are shown.

[0206] Referring to Fig. 10(a), the gate signal is and , the resonant inductor current is , the primary side voltage is , the secondary voltage is , the drain-source voltage is , the switch current is am.

[0207] Referring to Fig. 10 (b), The simulation results of a resonant bidirectional converter applying the divided complementary duty modulation method of the present invention at full load when V is 240V are shown.

[0208] Referring to Fig. 10(b), the gate signal is and , the resonant inductor current is , the primary side voltage is , the secondary voltage is , the drain-source voltage is , the switch current is am.

[0209] Referring to FIG. 10, a resonant bidirectional converter applying the divided complementary duty modulation method of the present invention is used to determine the ZVS function over a very wide voltage range. PSIM simulation is performed under conditions where V is 1000V and V is 240V.

[0210] Referring to FIG. 10, the switching modulation method of the present invention It enables ZVS turn-on at all switches under conditions of 1000V and 240V.

[0211] In other words, the present invention can effectively maintain ZVS even over a wide voltage range.

[0213] FIG. 11 is a diagram showing the turn-off switching loss in an upper switch and a lower switch according to an embodiment of the present invention.

[0214] Referring to FIG. 11, FIG. 11 (a) shows a forward buck mode, and FIG. 11 (b) shows a forward boost mode.

[0215] Referring to Fig. 11, 200W, 600W, 800W, 1000W, 1200W, and 1600W are shown.

[0216] Referring to Fig. 11, the conventional switching method has a problem in that switching losses occur significantly in two switches on the primary or secondary side.

[0217] In conventional switching methods, the current flowing through the upper switch and the current flowing through the lower switch are different. Consequently, there is a difference in current stress at the time of turn-off, and particularly, switching losses increase in the upper switch, which intensifies heat generation in a specific switch.

[0218] Referring to FIG. 11, it can be seen that the switching method of the present invention has the same switching loss occurring in all switches located on the primary or secondary side.

[0219] The divided complementary duty modulation method of the present invention alternately changes the roles of the upper switch and the lower switch of a resonant bidirectional converter during every switching cycle.

[0220] In the method of the present invention, the loss of the primary side switch is evenly distributed in buck mode, and the loss of the secondary side switch is evenly distributed in boost mode.

[0221] In other words, switching losses are not concentrated on a specific switch, and the overall heat distribution is balanced.

[0222] In buck mode, the turn-off voltage is the primary side voltage It is identical to, and the turn-off loss in buck mode can be expressed as Equation 7 below.

[0223] [Mathematical Formula 7]

[0224]

[0225] Here, is a switch It is a turnoff loss of, and is a switch It is the current at the cutoff point, and is the mathematical formula 8 below, It can be expressed as shown in mathematical formula 9 below.

[0226] [Mathematical Formula 8]

[0227]

[0229] [Mathematical Formula 9]

[0230]

[0232] In boost mode, the turn-off voltage is the secondary voltage It is identical to, and the turn-off loss in boost mode can be expressed as Equation 10 below.

[0233] [Mathematical Formula 10]

[0234]

[0235] Here, is the mathematical formula 11 below, It can be expressed as shown in mathematical formula 12 below.

[0236] [Mathematical Formula 11]

[0237]

[0239] [Mathematical Formula 12]

[0240]

[0242] FIG. 12 shows a prototype of a resonant DC / DC converter according to one embodiment of the present invention.

[0243] Referring to FIG. 12, a prototype of the proposed converter, a DC source, a control board, an interface board, and a load of the converter of the present invention are shown.

[0244] Referring to FIG. 12, to verify the operation of the divided complementary duty modulation method of the present invention , , Experiments are performed using a prototype under the conditions of.

[0245] A switch with a weak ZVS current when α is 250μH and in buck mode , , , minimum dead time A switch with a strong ZVS current and less than 50ns , , , minimum dead time It is less than 30ns.

[0246] A switch with a weak ZVS current when α is 250μH and in boost mode , , , minimum dead time A switch with a strong ZVS current and less than 75ns , , , minimum dead time It is less than 25ns.

[0248] FIG. 13 shows the design specifications and parameters of a converter according to one embodiment of the present invention.

[0249] Referring to Fig. 13, is the primary-side voltage, is the secondary-side voltage is output power, is the switching frequency, is the resonant frequency, is the transformer turns ratio, is a magnetizing inductor, is an auxiliary inductor, is a resonant inductor, is a resonant capacitor, , is a switch , , , Dead time, , is a switch , , , Represents the dead time.

[0250] Referring to FIG. 13, the present invention experiments by applying a longer dead time to the switch to account for unpredictable errors. Here, unpredictable errors may reduce the ZVS current due to changes in the parasitic capacitance of the switch depending on the drain-source voltage and ambient temperature, or due to non-ideality of the resonant tank.

[0251] The digital control algorithm is developed in the Code Composer Studio program and executed by the TMS320F28384D digital signal processor, which transmits control signals to embedded control peripherals to drive the system.

[0253] FIG. 14 is a diagram showing an experimental waveform in a forward buck mode according to one embodiment of the present invention.

[0254] Referring to Fig. 14, is 480V, and It is 1.6kW.

[0255] Referring to FIG. 14, FIG. 14 (a) is the resonant inductor current , resonant capacitor voltage , primary side voltage , secondary side voltage It represents, and Fig. 14 (b) is the gate signal and , drain-source voltage , switch current It represents, and (c) of FIG. 14 is the gate signal and , drain-source voltage , switch current It represents.

[0256] Referring to FIG. 14(a), the primary side voltage It has three voltage levels, and the secondary voltage It has two voltage levels. and It is aligned to the center and applied to the LC tank. This structure generates resonant current and voltage waveforms.

[0257] Referring to FIG. 14(b), the divided complementary duty modulation method of the present invention is a switch class It is applied to. The switching signal generates an asymmetric current waveform during two consecutive switching cycles.

[0258] having a weak ZVS current It requires additional dead time in addition to the basic dead time and has a strong ZVS current. Only the default dead time is applied.

[0259] It is turned off with near-zero current switching (ZCS) and hard switching during two consecutive switching cycles.

[0260] Referring to FIG. 14(c), the gate driving signal has a fixed duty cycle of 0.5 and and It applies to. It turns on with ZVS.

[0262] FIG. 15 is a diagram showing an experimental waveform in a forward boost mode according to one embodiment of the present invention.

[0263] Referring to Fig. 15, is 320V, and It is 1.6kW.

[0264] Referring to FIG. 15, FIG. 15(a) is the resonant inductor current , resonant capacitor voltage , primary side voltage , secondary side voltage It represents, and Fig. 15 (b) is the gate signal and , drain-source voltage , switch current It represents, and (c) of FIG. 14 is the gate signal and , drain-source voltage , switch current It represents.

[0265] Referring to Fig. 15, the waveform of the boost mode is similar to the buck mode waveform of Fig. 14.

[0267] FIG. 16 is a diagram showing an experimental waveform during load fluctuation according to one embodiment of the present invention.

[0268] Referring to FIG. 16, FIG. 16 (a) is a load fluctuation waveform in buck mode and FIG. 16 (b) is a load fluctuation waveform in boost mode.

[0269] Referring to Fig. 16, is the output current, is the secondary voltage, is the primary side voltage, represents the resonant inductor current.

[0270] Referring to FIG. 16, the divided complementary modulation method of the present invention is tested under output power fluctuation conditions, where the output power fluctuation condition decreases from 1.3kW to 0.3kW or increases from 0.3kW to 1.3kW.

[0271] Referring to Fig. 16, the output current accurately tracks the expected value without excessive overshoot or undershoot and maintains a stable response.

[0272] In other words, reliability regarding output power fluctuations can be proven.

[0274] FIG. 17 is a diagram showing the efficiency curves of the buck mode and boost mode in the forward and reverse directions according to one embodiment of the present invention.

[0275] Referring to Fig. 17, the horizontal axis represents power [Kw], and the vertical axis represents efficiency [%].

[0276] Referring to Fig. 17, the power conversion efficiency curve is measured for bidirectional power flow under various primary voltage and power conditions.

[0277] For example, the power conversion efficiency curve is Experiment the results when the voltage is 320V, 360V, 400V, 440V, and 480V.

[0278] Referring to FIG. 17, the present invention achieves a peak efficiency of up to 97.3%.

[0279] Referring to Fig. 17, the effect of additional dead time is analyzed by conducting an experiment without applying additional dead time to a switch with a weak ZVS current. When tested without additional dead time at 320V or 480V, the efficiency at full load decreases by 0.5%. In other words, it can be seen that additional dead time improves ZVS operation and reduces switching losses.

[0280] Referring to FIG. 17, it can be seen that the efficiency is almost the same in the forward direction and the backward direction, so the modulation method of the present invention provides balanced performance in both directions.

[0282] FIG. 18 is a diagram comparing the temperature difference between an upper switch and a lower switch in buck mode or boost mode according to one embodiment of the present invention.

[0283] Referring to FIG. 18, it can be seen that the temperature between the primary switches and the secondary switches in the switching method of the present invention is maintained at almost the same level compared to the conventional method.

[0284] Referring to Fig. 18, an unbalanced heat distribution means an unbalanced heat distribution, and a balanced heat distribution means a balanced heat distribution.

[0285] Referring to FIG. 18, FIG. 18 (a) is a thermal image with a conventional modulation method applied in buck mode, FIG. 18 (b) is a thermal image with the modulation method of the present invention applied in buck mode, FIG. 18 (c) is a thermal image with a conventional modulation method applied in boost mode, and FIG. 18 (d) is a thermal image with the modulation method of the present invention applied in boost mode.

[0286] Referring to FIG. 18(a) and FIG. 18(c), when applying a conventional modulation method, the temperature difference between the primary side upper switch and lower switch is 13.2 °C in buck mode, and the temperature difference between the secondary side upper switch and lower switch is 14.6 °C in boost mode. In other words, the conventional modulation method causes excessive heat generation at specific switches, making thermal management difficult.

[0287] Referring to FIG. 18(b) and FIG. 18(d), when the modulation method of the present invention is applied, the temperature difference in buck mode is reduced to 3.8 °C, and the temperature difference in boost mode is maintained at nearly 0.1 °C. That is, the modulation method of the present invention can solve the heat generation problem by balancing the loss distribution between switches.

[0288] The modulation method of the present invention can simplify thermal management tasks and increase the reliability of the converter by evenly adjusting the loss distribution among all switches.

[0290] FIG. 19 is a diagram showing constant current mode and constant voltage mode operation through PSIM simulation according to one embodiment of the present invention.

[0291] Referring to FIG. 19, the circuit operates in constant current (CC) mode when the secondary voltage is less than a predetermined voltage of 370V, and switches to constant voltage (CV) mode when it reaches 370V.

[0292] In other words, the present invention can be seen through PSIM simulation that it switches from a stable CC mode to a CV mode.

[0294] The present invention described above may be implemented as computer-readable code on a computer-readable medium. The computer-readable recording medium may be, for example, a removable recording medium (CD, DVD, Blu-ray disc, USB storage device, removable hard disk) or a fixed recording medium (ROM, RAM, computer-equipped hard disk). The computer program recorded on the computer-readable recording medium may be transmitted to another computing device via a network such as the Internet and installed on the other computing device, thereby allowing it to be used on the other computing device.

[0295] Although it has been described above that all components constituting an embodiment of the present invention are combined or operate as a single unit, the present invention is not necessarily limited to such an embodiment. That is, within the scope of the purpose of the present invention, all components may be selectively combined in one or more ways to operate.

[0296] Although operations are depicted in a specific order in the drawings, it should not be understood that the operations must necessarily be executed in the specific order depicted or in a sequential order, or that all depicted operations must be executed to obtain the desired result. In certain situations, multitasking and parallel processing may be advantageous. Furthermore, the separation of various configurations in the embodiments described above should not be understood as necessarily required, and it should be understood that the described program components and systems can generally be integrated together into a single software product or packaged into multiple software products.

[0297] The present invention has been described above with reference to its embodiments. Those skilled in the art will understand that the present invention may be embodied in modified forms without departing from the essential characteristics of the invention. Therefore, the disclosed embodiments should be considered in an illustrative rather than a restrictive sense. The scope of the invention is defined by the claims, not by the foregoing description, and all variations within the scope of the claims should be interpreted as being included in the invention. Explanation of the symbols

[0299] 110: 1st Fullbridge Section 120: Auxiliary inductor section 130: LC Resonance Tank Section 140: Transformer section 150: Magnetic inductor section 160: 2nd Fullbridge Section

Claims

Claim 1 A heat dissipation switching control device for a resonant bidirectional converter, comprising: a first full-bridge section that generates an AC voltage by alternately performing switching based on an input voltage; an auxiliary inductor section that receives the AC voltage and adjusts it according to resonance characteristics; an LC resonance tank section that forms resonance based on the adjusted AC voltage and transmits a resonance voltage; a transformer section that receives the resonance voltage and steps up or steps down the AC voltage according to a winding ratio; a magnetic inductor section that receives and adjusts the stepped-up or stepped-down AC voltage; and a second full-bridge section that performs switching to rectify the adjusted AC voltage and generate a DC voltage. Claim 2 In claim 1, the first full bridge portion is the switch , , , and, the above switch is a switch pair and switch pair A thermal dissipation switching control device for a resonant bidirectional converter that further includes distinguishing as. Claim 3 In paragraph 2, the switch pair The main duty and complementary duty are driven by a gate driving signal in which the main duty and complementary duty are divided by a fixed duty of 0.5, and the switch pair The above switch pair A thermal dissipation switching control device for a resonant bidirectional converter that is driven by the same gate driving signal as above, but further includes being driven by applying a 360° phase difference. Claim 4 A thermal dissipation switching control device for a resonant bidirectional converter, wherein, in paragraph 2, the switch requires a basic dead time and an additional dead time when having a weak ZVS current, and only a basic dead time when having a strong ZVS current. Claim 5 In claim 1, the second full bridge portion is the switch , , , and, the above switch is a switch pair and switch pair A thermal dissipation switching control device for a resonant bidirectional converter that further includes distinguishing as. Claim 6 In paragraph 5, the above switch pair The main duty and complementary duty are driven by a gate driving signal in which the main duty and complementary duty are divided by a fixed duty of 0.5, and the switch pair The above switch pair A thermal dissipation switching control device for a resonant bidirectional converter that is driven by the same gate driving signal as above, but further includes being driven by applying a 360° phase difference. Claim 7 A thermal dissipation switching control device for a resonant bidirectional converter, wherein, in paragraph 5, the switch requires a basic dead time and an additional dead time when having a weak ZVS current, and only a basic dead time when having a strong ZVS current. Claim 8 A thermal dissipation switching control method for a resonant bidirectional converter, comprising: a step of generating an AC voltage by alternately performing switching on a switch based on an input voltage; a step of receiving the AC voltage and adjusting the AC voltage to match resonance characteristics; a step of forming resonance based on the adjusted AC voltage and transmitting a resonant voltage; a step of receiving the resonant voltage and stepping up or stepping down the AC voltage according to a winding ratio; a step of receiving and adjusting the stepped-up or stepped-down AC voltage; and a step of rectifying the adjusted AC voltage by performing switching and generating a DC voltage. Claim 9 In claim 8, the step of generating an AC voltage by having the switch alternately perform switching based on the input voltage is that the switch , , , and, the above switch is a switch pair and switch pair A thermal dissipation switching control method for a resonant bidirectional converter that further includes distinguishing as follows. Claim 10 In Clause 9, the above switch pair The main duty and complementary duty are driven by a gate driving signal in which the main duty and complementary duty are divided by a fixed duty of 0.5, and the switch pair The above switch pair A thermal dissipation switching control method for a resonant bidirectional converter, which is driven by the same gate driving signal as above but further includes driving by applying a 360° phase difference. Claim 11 A thermal dissipation switching control method for a resonant bidirectional converter according to claim 9, further comprising the fact that the switch requires a basic dead time and an additional dead time when having a weak ZVS current, and only a basic dead time when having a strong ZVS current. Claim 12 In claim 8, the step of rectifying the AC voltage regulated by performing the switching and generating a DC voltage is that the switch , , , and, the above switch is a switch pair and switch pair A thermal dissipation switching control method for a resonant bidirectional converter that further includes distinguishing as follows. Claim 13 In Clause 12, the above switch pair The main duty and complementary duty are driven by a gate driving signal in which the main duty and complementary duty are divided by a fixed duty of 0.5, and the switch pair The above switch pair A thermal dissipation switching control method for a resonant bidirectional converter, which is driven by the same gate driving signal as above but further includes driving by applying a 360° phase difference. Claim 14 A thermal dissipation switching control method for a resonant bidirectional converter, wherein, in paragraph 12, the switch requires a basic dead time and an additional dead time when having a weak ZVS current, and only requires a basic dead time when having a strong ZVS current.

Images