AUDIO DECORRELER, PROCESSING SYSTEM AND METHOD FOR DECORRELATING AN AUDIO SIGNAL

Abstract

A decorrelator comprises a plurality of delay units, where each delay unit is configured to receive a portion of a frequency representation based on an audio signal, and where each delay unit is configured to delay the received portion to provide a delayed portion. The decorrelator comprises an envelope shaper configured to receive and combine signals based on the delayed portions of the frequency representation. The envelope shaper receives the frequency representation of the audio signal and is configured to adjust the energy of the delayed portions relative to the frequency representation of the audio signal. The envelope shaper is configured to provide a combined shaped frequency representation. Transient signal portions are handled by means of an adapted operation of the decorrelator.

Description

AUDIO DECORRELER, PROCESSING SYSTEM AND METHOD FOR DECORRELATING AN AUDIO SIGNAL FIELD OF INVENTION The present invention relates to a decorrelator for an audio signal, a processing system having such a decorrelator, a decorrelation method, and a computer program product. In particular, the present invention relates to an audio signal decorrelator. BACKGROUND OF THE INVENTION In perceptual audio coding, decorrelators are an important component for parametric spatial audio coding. Well-known solutions are referred to as parametric spatial audio decorrelators, such as parametric stereophonic or MPEG surround. Decorrelators, as described in [1] or [2], utilize computationally expensive time-domain reverb filters with a long impulse response. Decorrelators, as described in [3] or [4], require the use of a substantial quadrature mirror filter bank (QMF) and computationally expensive grid filters. Therefore, there is a need for a decorrelator, a processing system that has such a decorrelator, and a method for correlating portions of an audio signal that allows for low processing delay and / or low computational complexity decorrelation. BRIEF DESCRIPTION OF THE INVENTION An objective of the present invention is to provide what is necessary for a decorrelator, a processing system, and a method for decorrelation that allow low processing delay and / or decorrelation with low complexity and high perceptual quality, especially in the processing of signals containing transients. This objective is achieved through the material in question as defined in the independent claims. A key feature of the present invention is that dividing a frequency representation into multiple parts for processing—that is, delaying each part with a separate delay unit—allows for low processing latency, since the computation of the different parts can be performed in parallel. At the same time, such frequency-domain operations require low computational complexity. According to one embodiment, a decorrelator comprises a plurality of delay units, wherein each delay unit is configured to receive a portion of a frequency representation based on an audio signal, and each delay unit is configured to delay the received portion to provide a delayed portion. The decorrelator comprises an envelope shaper configured to receive a combination of signals based on the delayed portions of the frequency representation, to receive the representation of CQor / n / eznz / q / Yi audio signal frequency, to adjust the energy of the delayed parts with respect to the frequency representation of the audio signal and to provide a combined frequency representation. According to one approach, different parts of the frequency representation comprise an equal or different number of frequency bins. Where the same number of frequency bins may allow for the same processing time, a different number of frequency bins may allow for adaptation to application requirements. According to one embodiment, the decorrelator comprises a phase shifter configured to change the phase of the frequency representation of the audio signal, or to change the phase of the audio signal in a time domain to obtain a phase-shifted audio signal. The phase shift can allow for the perception of reverberation and thus high audio quality. According to one embodiment, the phase shifter is configured to change the phase of the audio signal's frequency representation and comprises a plurality of all-pass filters, where each all-pass filter is configured to change the phase of an associated portion of the audio signal's frequency representation. That is, the all-pass filter can be associated with and adapted to the respective portion of the audio signal, which can allow for high overall audio quality. According to one embodiment, a total-pass filter of the plurality of total-pass filters comprises a set of total-pass filter structures connected in series with each other, i.e., using Schroeder IIR filters. The total-pass filter structures are adapted to provide different time delays. Alternatively or additionally, the total-pass filter structures comprise a nested total-pass filter structure. According to one modality, a number of total-pass filter structures and / or the circuitry of the total-pass filter structure may be equivalent or different between different total-pass filters. This allows for high flexibility of the decorrelator. In one approach, the different time delays are based on a prime multiple of a local sampling frequency used to obtain the frequency representation of the audio signal. This allows for the perception of high audio quality. According to one embodiment, the set of full-pass filter structures comprises four full-pass filter structures, each designed to provide a delay of 1, 2, 3, and 5 time units. Such a time unit can be based on a block size of the conversion in the frequency domain. For example, using a block size of 256 with a 50% overlap, one time unit might result in 128 samples at 48 kHz = 2.7 ms. Other reasonable time units could be, for example, 32 or 64 samples, or other values. Preferably, the time units are short enough to allow sufficient time resolution in the subsequent time / frequency envelope shaping. In an alternative solution, a delay of 1, 3, 5, and 7 is provided by the four full-pass filter structures. This avoids overlaps in the time domain. CQor / n / eznz / q / Yi According to one configuration, the gain factor of the full-pass filter is adjusted to a value with a magnitude (i.e., positive or negative) of 0.7 within a tolerance range. The tolerance range is, for example, 20%, 10%, or 5%. According to one embodiment, the phase shifter is configured to change the phase of the audio signal in the time domain, wherein the phase shifter comprises a set of full-pass filter structures connected in series with each other, wherein the full-pass filter structures are adapted to provide different time delays. Alternatively or additionally, the full-pass filter structures comprise a nested full-pass filter structure. According to one approach, the different total step time delays are based on a prime multiple of the reciprocal of a sampling frequency used to obtain the frequency representation of the audio signal. As in the frequency domain, a corresponding advantage can also be obtained in the time domain. In the time domain, different time delays can be based on a prime number obtained by multiplying each of a set of minimal prime numbers, for example, 1, 2, 3, and 5 as one example set, or 1, 3, 5, and 7 as another example set, with a resolution reduction factor used to generate the parts of the audio signal's frequency representation to obtain an intermediate result, and then using a subsequent prime number relative to that intermediate result.Like a next prime number, a closer distance can be understood, for example, to obtain the next largest or next smallest prime value. In the given example, the values ​​131, 257, 383, and 641 can be obtained for the first set, and 131, 383, 641, and 907 can be obtained for the second set. Here, a unit of time can be 1 sample. The sample can refer to a sampling frequency, for example, 48 kHz. In other modes, the sampling frequency can also be 44.1 kHz, 32 kHz, or other values. According to one embodiment, the decorrelator comprises a first conversion unit to obtain the frequency representation of the audio signal for the envelope shaper, and a second conversion unit to obtain a frequency representation of the reverberated audio signal, wherein parts of the frequency representation form parts of the frequency representation of the reverberated audio signal. This allows the signal used to be generated directly in the decorrelator. According to one modality, the decorrelator is adapted to additionally implement an equal and predefined delay for a subset of all parts of the frequency representation. That is, a delay that is equal for the respective parts, or delay lines can also be commonly applied in a common delay module that allows simple delay units in the respective delay lines for an associated part. According to one modality, the delay units associated with one spectral portion of the plurality of delay units are configured to delay the associated portion of the frequency representation differently when compared to delay units associated with other spectral portions. This allows for high perceived quality by treating different frequency portions differently. According to one modality, the delay unit is configured to delay parts of the performance CQor / n / eznz / q / Yi of frequency comprising lower frequencies with a higher time delay when compared to parts of the frequency representation comprising higher frequencies. Depending on the modality, the relationship between different time delays is linear, logarithmic, and / or based on rounding in sub-band samples. This allows for the perception of high quality. According to one embodiment, the decorrelator comprises a conversion unit for receiving and converting the audio signal, or a reverberated version of the audio signal, into parts by performing a time-block Discrete Fourier Transform (DFT) or a Short-Time Fourier Transform (STFT), wherein the conversion unit is configured to convert blocks that have a 50% overlap within a tolerance range. Such block conversion allows for short delays for each respective part to be obtained and for parallel processing of the different parts. According to one modality, the envelope shaper is configured to operate in a sub-band domain and with a temporal resolution of less than 4 milliseconds. According to one model, the decorrelator comprises a signal processing stage configured to receive a signal based on the combined conformal frequency representation, for example, as a monaural signal, and to process the monaural signal as at least a stereophonic signal. This allows for an enhanced perception by the listener. According to one modality, the decorrelator comprises a signal processing stage configured to process the combined conformal frequency representation at least in a stereophonic signal and for source extension modeling based on said stereophonic signal at least, for example, in the frequency domain. According to one modality, a processing system comprises a decorrelator as described in this document and a processing stage to transform a mid / side decomposed signal into a left / right decomposed signal. Depending on the configuration, the processing system can perform transient suppression to eliminate echoes, such as pre-echoes and post-echoes caused by a transient. This transient handling may involve muting the output of a decorrelator and, correspondingly, amplifying the output of a delay compensation unit. This unit provides the necessary signal for a portion of the left / right decomposed signal, is parallel to the decorrelator, and is connected to the processing stage. According to one embodiment, a method comprises receiving a plurality of parts of a frequency representation based on an audio signal, delaying each of the received parts to provide a plurality of delayed parts, and receiving and combining signals based on the delayed parts of the frequency representation. The method comprises receiving the frequency representation of the audio signal and adjusting the energy of the delayed parts relative to the frequency representation of the audio signal. A combined conformal frequency representation is provided. According to one modality, a computer program or software product is provided CQor / n / eznz / q / Yi computer or non-transient storage medium that stores in it instructions to carry out respective instructions to execute the method, when executed on a computer. BRIEF DESCRIPTION OF THE DRAWINGS Additional convenient modalities are defined in the dependent claims. The appropriate modalities are described in greater detail by reference to the accompanying drawings, in which: Figure 1 shows a schematic block diagram of a modality-according decorrelator. Figure 2 shows a schematic block diagram of a decorrelator comprising a conversion unit for generating a frequency representation of a signal in the time domain according to a modality. Figure 3 shows a schematic block diagram of a decorrelation that further comprises a pre-delay according to a modality. Figure 4 shows a schematic block diagram of a one-way total pass filter. Figure 5 shows a schematic block diagram of a nested total-pass filter structure according to a modality. Figure 6 shows a schematic block diagram of a decorrelator comprising a phase changer configured to operate in the time domain according to a modality. Figure 7 shows a schematic block diagram of a decorrelator that connects to a source extension model according to a modality. Figure 8 shows a schematic block diagram of a processing system according to a modality. Figure 9 shows a schematic block diagram of a processing system configured for transient handling according to a modality. Figure 10 shows a schematic block diagram of a method according to a modality. DETAILED DESCRIPTION OF THE INVENTION Equal or equivalent elements or elements with equal or equivalent functionality are denoted in the following description by means of equal or equivalent reference numbers even if they are presented in different figures. The following description sets forth a plurality of details to provide a more complete explanation of embodiments of the present invention. However, it will be evident to those experienced in the art that embodiments of the present invention can be practiced without these specific details. In other cases, well-known structures and devices are shown in block diagram form rather than in detail. CQor / n / eznz / q / Yi in order to avoid obscuring the embodiments of the present invention. Additionally, the features of the different embodiments described herein can be combined with each other, unless specifically stated otherwise. Figure 1 shows a schematic block diagram of a decorrelator 10 according to a modality. The decorrelator 10 comprises at least two delay units 12i to 12n, with n > 1. Although Figure 1 illustrates two delay units 12, the number is preferably larger, for example, 4, 8, 16, or other values ​​obtained with a power of 2, where the modalities are not limited to such numbers. That is, the modalities may also comprise 3, 5, 7, or 9 delay units 12. Each delay unit is configured to receive an associated portion 14i to 14 of a frequency representation 14 that is based on an audio signal. For example, the frequency representation 14 may be or may comprise a spectrum obtained by means of a Fourier transform such as a discrete Fourier transform, DFT, or a short-time Fourier transform, STFT.The portions 14i to 14nse can be obtained, for example, as a subband of the spectrum, that is, a part of the representation in the frequency domain. For example, such a portion 14i to 14nse can be obtained using an appropriate window. Each delay unit 12i to 12n is configured to delay the received 14i to 14n part to provide a delayed 14j to 14n part, i.e., to have a delay in the time domain. The decorrelator 10 further comprises an envelope shaper 16 configured to receive signals based on the delayed portions 14j to 14'n. Such signals may be the delayed portions 14j to 14'n themselves or processed variants thereof. The envelope shaper 16 is configured to combine the received signals. Additionally, the envelope shaper is configured to receive the frequency representation 14 of the audio signal. The envelope shaper 16 is configured to adjust the energy of the delayed portions 14j to 14'n relative to the frequency representation 14 of the audio signal. The envelope shaper 16 is configured to provide a combined shaped frequency representation 18.In the combined conformal frequency representation 18, the respective parts 14i to 14n, signals resulting from the same respectively, can be correlated respectively with each other and / or with respect to the frequency representation 14. Although the envelope shaper 16 is illustrated to receive the combined frequency representation 14, as an alternative, the envelope shaper 16 can receive the respective information by receiving the possibly undelayed or commonly treated parts 14i to 14n. Figure 2 shows a schematic block diagram of a decorrelator 20 according to a modality. The decorrelator 20 is configured to receive an audio signal 22. The decorrelator 20 may comprise a conversion unit 24 configured to generate the frequency representation 14 shown in Figure 1. The conversion unit 24 may provide what is necessary for parts 14i to 14ιθ to be obtained by means of, for example, an STFT. For example, the frequency representation may comprise a total of 129 frequency bins. Alternatively, 128 bins may be used. For example, two types of CQor / n / eznz / q / Yi Digital Fourier Transforms (DFTs), the so-called “uniformly stacked” and an “unequally stacked” version. For example, as a “standard” DFT, the uniformly stacked version can be considered to have, in the example provided, 129 bands (127 complex, one real, and one imaginary). The unevenly stacked version can comprise 128 (complex) bands. Both transforms can be used in the modalities described in this document. Parts 14i to 14ιθ can comprise, partially or completely, an equal or different number of bins. For example, part 14i can comprise bins 1 through 9, e.g., 9 bins. Part 142 comprises, e.g., bins 10 through 19, and thus ten bins.The adaptation or selection with respect to the number of bins can be based on the sampling frequency, which in the illustrated example is 48 kHz, the overlap, which is, for example, 50%, and / or a number of 14i to 14w parts to be generated. The 14i to 14i6 parts can comprise an equal or different number of frequency bins, such that some or all of the 14i to 14w parts can also be generated to comprise an equal number of frequency bins. The decorrelator 20 further comprises a delay section 25 having delay lines 12i to 12w. Each delay line 12i to 12i6 is associated with a specific spectral part 14i to 14w and configured to receive that part, or a processed version thereof. The delay units 12i to 12w may be associated with a respective spectral part 14i to 14ιθ. Such a delay unit 12i to 12w may be configured to delay the associated part of the frequency representation 14 differently compared to the delay units associated with other spectral parts. Alternatively or additionally, the relationship between different time delays may be linear, logarithmic, and / or based on rounding in superband samples. The decorrelator 20 further comprises a phase shifter 26 coupled to the delay section 25, the phase shifter 26 being configured to receive the delayed portions 14j to 14je. The phase shift employed by the phase shifter 26 can introduce reverberation into the signal portions. However, depending on the configuration, the sequence of the delay section 25 and the reverberation section 26 can also be altered so that a respective portion 14i to 14w can first be filtered through a reverberation filter and subsequently delayed. The phase shifter 26 can be configured to shift the phase of the frequency representation 14 of the audio signal, a processed, for example, delayed version thereof. Phase shifting can also be performed before converting the audio signal 22 to the frequency domain; a corresponding phase shifter can be configured to shift the phase of the audio signal 22 in the time domain to obtain a phase-shifted audio signal. In the short configuration, where the phase shifter 26 is configured to shift the phase of the frequency representation of the audio signal 14, the delayed version thereof, the phase shifter can comprise a plurality of full-pass filters 28i to 28ie.In the example shown, the full-pass filters 28i to 28i6 are configured to receive the delayed parts 14'ia and 14'i6. The term "full-pass filter" should be understood to mean that the frequency range to be passed corresponds to the frequency range of the respective part 14i to 14w. This may include examples where each of the full-pass filters 28i to 28iβ passes the entire frequency range provided in the frequency representation. The passbands of different total pass filters 28i to 28i6 may also differ from each other based on the different frequency bins contained in the respective parts 14i to 14ιβ. Each of the 28i to 28ie total pass filters is configured to change the phase of an associated part of the frequency representation of the audio signal. That is, a number of total-pass filter structures and / or a circuitry of the total-pass filter structure can be the same, i.e., equal or comparable, or it can, alternatively, be different between different total-pass filters 28i to 28ιβ· The time delay provided by delay lines 12i to 12w can be the same or different for different parts, 14i to 14ib. As shown in Figure 2, parts of the frequency representation comprising lower frequencies can be delayed with a greater time delay compared to parts of the frequency representation comprising higher frequencies. From bin 1 to higher bins, the represented frequency can increase. As represented in the Z domain, the time delay can decrease with increasing frequencies. Signals 32i to 32w may comprise a result of delay and phase shifting, for example, as an output of full-pass filters 28i to 28i6. Envelope shaper 16 may be configured to receive signals 321 to 32ie and an unfiltered or undelayed version of them, i.e., parts 14i to 14w, i.e., the frequency representation of audio signal 22. Parts 14i to 14ie may be understood as sub-bands. Envelope shaper 16 may be configured to operate in a sub-band domain. For example, the time resolution of envelope shaper 16 may be at most or less than 4 milliseconds, e.g., 4 milliseconds, 3.5 milliseconds, 3 milliseconds, or less. The decorrelator 20 may comprise another conversion unit 35 that can provide what is necessary for an inverse operation when compared to the conversion unit 24. For example, the conversion unit 34 can perform an inverse Short Term Fourier Transform (STFT). The combined shape frequency representation 18 can comprise information regarding the frequency domain present in each of the bins such that the combined shape frequency representation 18 can be processed accordingly for the output of the conversion unit 24. That is, the conversion unit 34 can receive the processed versions of parts 14i to 14w of the frequency representation 14 and synthesize a synthesized signal 36 from the processed versions 14j to 14'i6 based on, for example, an aggregation-overlap procedure.Signal 36 can be provided, for example, on an interface 38 of the decorrelator 20. The envelope shaper 16 can be configured to shape spectral bins in time and / or frequency. Shaping can be performed by the envelope shaper 26 for individual bins and / or for groups of bins, for example, by implementing interdependent common shaping processing or at least by groups. CQor / n / eznz / q / Yi Referring again to conversion unit 24, it can be configured to receive and convert audio signal 22 or a reverberated version of it in parts 14i to 14ie, where 16 is just an example. The reverberated version of audio signal 22 can be an input if phase shifter 26 operates in the time domain and can therefore be processed upstream of conversion unit 24. Conversion unit 24 can perform a time-block discrete Fourier transform (DFT) or a short-time Fourier transform (STFT). The conversion unit can be configured to convert blocks that have an overlap of, for example, 50% within a tolerance range. For example, the tolerance range could be 0% as far as possible, at most 5%, at most 10%, at most 15%, or more. The blocks may comprise a block length of, for example, 128 samples, 256 samples, or 512 samples, where a value of 256 may be preferable. Figure 3 shows a schematic block diagram of a decorrelation 30. Compared to decorrelator 20, decorrelator 30 may additionally include a pre-delay 42, where the term "pre-delay" does not limit the delay to being implemented directly before or after any specific block. The pre-delay 42 can be located at any stage prior to the envelope shaper 16, preferably, when operating in the frequency domain, after the conversion unit 24. That is, for example, a sequence between the reverb full-pass filters or phase shifter 26 and the pre-delay 42 can be interchanged when compared to the illustration in Figure 3. The pre-delay 42, or the delay block 42, can be configured to further implement an equal and predefined delay for a subset or all of the parts 14i to 14w of the frequency representation.This may allow the implementation of the same delay for each part 14i to 1416 or a group of them to combine processing at this stage and use delay lines 12i to 12w to add a probably individual delay that differs from the common delay implemented in block 42. For example, pre-delay 42 is set up to allow a constant pre-delay for all spectral bands. Figure 4 shows a schematic block diagram of a full-pass filter 40 according to a modality that can be operated at least as part of one of the filters 28i to 28i6 of the decorrelator 20 and / or 30. The full-pass filter 40 may comprise a Schroeder IIR filter structure, for example, and may comprise a forward branch 46 in combination with a backward branch 48 in combination with a delay block 52 to provide a respective output signal 54 based on an input signal 44 of the full-pass filter 40. A full-pass filter 28 of the decorrelator 20 and / or 30 may comprise one or more such full-pass filters 40 connected in series with each other. To provide different time delays in different full-pass filters 28i to 28i6, a different number of full-pass filter structures 14 may be connected in series. In other words, Figure 4 shows a full-pass filter stage. Figure 5 shows a schematic block diagram of a 50-way total-pass filter structure, which is a nested total-pass filter structure. Alternatively or in addition to a total-pass filter structure CQor / n / eznz / q / Yi 40, one or more full-pass filter structures 50 may form at least part of a full-pass filter 28i to 28i6 of the decorrelator 20 and / or 30. Although all delay blocks 52i and 522 are shown, a different and especially larger number of delay blocks 52 may be present, possibly resulting in a greater number of forward branches 46 and / or backward branches 48. In addition, gains gV-gi and / or g2 / -g2 may be adopted. When considering, for example, connecting delay blocks 52 in series within one or more full-pass filter structures 40 and / or one or more full-pass filter structures 50, different full-pass filters 28i to 28" can be implemented to comprise a different time delay when compared to other full-pass filters. For example, the different delays of different full-pass filter structures and / or full-pass filter structure circuitry can be based on a prime multiple of a total sampling frequency, e.g., 48 kHz, used to obtain the frequency representation 14 of the audio signal 22. For example, a set of full-pass filter structures that form at least part of a full-pass filter can comprise four full-pass filter structures, e.g., full-pass filter structures 40.The different delay blocks within them can be adapted to provide delays of 1, 2, 3, and 5. For example, four full-pass filter structures can provide delays of 1, 3, 5, and 7 units in the Z domain. These values ​​can form a set of prime values; that is, a number of 2, 3, 4, 5, or more prime values ​​can be grouped together. When this modality is transferred, the sets of prime values, respectively, to the possible operations of full-pass filters in the time domain, the time delays are based on a prime multiple of the reciprocal of a sampling frequency used to obtain the frequency representation of the audio signal in a modality. For example, different time delays can be based on a prime number obtained by multiplying each of a set of prime numbers mentioned, e.g., 1, 2, 3, and 5 or 1, 3, 5, and 7, by a resolution reduction factor used to generate the parts of the frequency representation of the audio signal to obtain an intermediate result. Instead of the intermediate result, a further prime number relative to the intermediate result can be used.For example, when referring to the resolution reduction factor of 128 and considering the sets of prime numbers above, such a result could be the delay of 131, 257, 383, and 641 on one hand and 131, 383, 641, and 907 on the other, where each delay can refer to a multiplication by 1 sample at the sampling frequency, which, for a sampling frequency of 48 kHz, is approximately 20.8 ps. Other sets of prime numbers are possible without limitation. When referring, for example, to Figure 4, the gain factor g of the full-pass filter can be adjusted to a value of 0.7 within a tolerance range of, for example, ±20%, ±10%, or ±5%. However, the gain value can also be negative, for example, -0.7 within the aforementioned tolerance range. That is, the gain factor can be adjusted to a value within a magnitude of 0.7 within the tolerance range. In other words, in addition to the series-pass configuration of Figure 4, also a CQor / n / eznz / q / Yi is a nested configuration in which the delay element of a Schroeder total-pass filter is replaced by another inner total-pass configuration, or a combination of both configurations can be implemented. Figure 5 shows a simple nested total-pass filter stage. Figure 6 shows a schematic block diagram of a decorrelator 60 according to one modality. The decorrelator 60 comprises the phase shifter 26 configured to operate in the time domain. A full-pass filter structure 28' can be configured to use the following respective prime numbers when compared with the sets of prime numbers described in relation to the decorrelator 20 and / or 30. To ensure accurate operation of the decorrelator 60, it can comprise conversion units 24i and 242. While conversion unit 24i can provide what is necessary for the frequency representation of the audio signal, conversion unit 242 can receive the reverberated or phase-shifted audio signal 22' provided by the phase shifter 28'.The resulting 14”1 to 14”ie portions can be delayed by means of delay units 12i to 12w, resulting in a comparable input for the envelope shaper 16 when compared with the decorrelator 20 and / or 30, while allowing time-domain-based reverberation. In other words, the frequency representation portions can form parts of the frequency representation of the reverberated audio signal 22’. According to the modalities, a decorrelator as described in this document can be combined with additional functionality, i.e., the output signal can be further processed. In other words, Figure 6 shows an alternative implementation of a decorrelator with respect to Figure 2. Furthermore, inventive decorrelators can be combined with transient handling processing. Transients can introduce artifacts into the uncorrelated stereo signal, such as after-echoes or unwanted panning effects. To mitigate this, transient handling can be combined with the decorrelator described herein. The transient handling can mute the decorrelator output to preserve the initial direct waveform and suppress the after-echo caused by the pre-delay. Figure 7 shows a schematic block diagram of a decorrelator 70 according to one modality. The decorrelator 70 comprises at least a portion of the decorrelator 10, wherein alternatively or additionally at least portions of the decorrelators 20, 30, and / or 60 may be arranged. The decorrelator 70 may comprise a signal processing stage 56 configured to process the combined conformal frequency representation 18 or a signal based thereon. The combined conformal frequency representation 18 may be considered as a monaural signal, i.e., it may represent a single channel. From the received monaural signal, the processing stage may provide at least the signals 58i and 582, which represent a stereophonic signal. A source extender 58 that models the perceptual effect of a spatially extended sound source from a monaural signal from a point source and a correlated version thereof can be coupled to the decorrelator 70. The source extender 58 may comprise filters 64i to 642 that allow modeling of CQor / n / eznz / q / Yi source extension based on the stereo signal containing signals 58i and 582. Source extension modeling can be performed, for example, in the frequency domain and may result in stereo output signals 64i (e.g., left channel) and 642 (e.g., right channel). Note that source extender 58 may also be part of decorrelator 70. In other words, Figure 7 shows a schematic block diagram of the source extension processing. Figure 8 shows a schematic block diagram of a processing system 80 according to one modality. The processing system 80 may include the decorrelator 10. Alternatively or additionally, the decorrelators 20, 30, 60, and / or 70 may be provided. The processing system 80 includes a processing stage 66 configured to transform a mid / side decomposed signal 68 into a left / right decomposed signal 72. That is, the mid / side decomposed signal 68 may comprise at least a first signal 74i, for example, representing one of the mid / intermediate or side portions of a second signal 742 representing the other portion. The processing stage 66 may be configured to transform the signals 74i to 742 and possibly additional signals into at least the signals 76i to 762 representing a left channel and a right channel.One channel, for example, the left channel (L), can be obtained by adding the mid component (M) and the side component (M+S); while the other, for example, the right channel, can be obtained by subtracting one component from the other, for example, MS. According to a different approach, both channels can be obtained using 50% or a factor of 0.5 of each other, that is, 0.5(M+S) and 0.5(MS). Other factors and / or rules of determination are possible. According to one modality, signal 74i is provided by the decorrelator of the processing system 80. The other signal, 742, can be provided by a delay compensation unit 78, which is connected in parallel to the decorrelator 10 and is configured to also receive the audio signal 22. Therefore, the delay compensation unit 78 is connected to the processing stage 66. The delay compensation unit 78 can be configured to provide a time delay that is comparable to that of the decorrelator. Preferably, for modalities in the frequency domain, the delay is equal to the processing delay introduced by the STFT analysis / synthesis of the decorrelator. However, the decorrelator 10 can provide the necessary additional signal processing that leads to a decorrelation such that signal 742 exhibits a similar delay when compared to signal 74i.According to one modality, the 742 signal can be unprocessed except for the time delay. The decorrelator 10 in the processing system 80 can provide the combined conformal frequency representation as at least a portion of the mid / side decomposed signal to the processing stage 66. The processing stage 66 can transform the combined conformal frequency representation, along with the delay signal 742, into the left / right decomposed signal in the frequency domain. The output of the processing stage 66 can be an L / R signal 72. The decorrelator 10 itself can produce a monaural signal S (side, component 18), insofar as it is only a portion of it. With transient handling, the direct portion CQor / n / eznz / q / Yi M (742; 74'2) and the decorrelator output S (signal 18) can be tightly coupled, since signal S will be muted and “replaced” by an amplified signal M (signal 745). Consequently, both units, the decorrelator and the “channel separation unit” 66, are tightly coupled, and thus processing stage 66 finally provides the uncorrelated stereo signal. If the decorrelator were to be operated independently with monaural output, for example, without processing stage 66, then the direct, unscaled, delay-compensated signal would be added directly to the monaural output to fill the muted space and provide a “full” signal. In other words, Figure 8 shows a decorrelator in M / S to L / R configuration, with monaural input delay compensation (average signal). Figure 9 shows a schematic block diagram of a processing system 90 according to one modality. When compared to the processing system 80, the processing system 90 comprises a transient suppressor 82 configured to detect a transient in the audio signal 22 or its frequency representation 14 at a decorrelator input. The transient suppressor may comprise a transient detection unit 84 configured to receive the audio signal 22 or its frequency representation. The transient detection unit 84 can detect a transient in the audio signal, for example, by processing the audio signal 22. The transient suppressor 82 may further comprise a muting unit 86 configured to receive the combined conformal frequency representation 18 and mute it based on a control signal.However, it should be noted that an equal or comparable effect can also be achieved by controlling the decorrelator 10 or the decorrelator contained in the processing system 90 to mute the decorrelator output. That is, the muting unit 86 can also be part of the decorrelator. However, the signal 74i, which forms the input to the processing stage 66, can be muted based on a transient detected in the audio signal 22. The transient suppressor 82 can be configured to temporarily mute the portion provided by the decorrelator to suppress echoes in the processing stage 66, where the echoes may be related to pre-echoes and / or post-echoes. When operating in the time domain, a window for soft muting can be used to prevent additional transients from being introduced by the muting process.If done in the frequency domain, the STFT window, described in relation to the 20, 30 and 60 decorrelators, can provide what is needed for this purpose automatically, i.e., in a synergistic way. With respect to the processing stage 66, muting the output of the decorrelator 10 could lead to an undesirable change in the input power of the signal processing stage 66. To avoid the negative effects, an amplifier 82 can be connected between the delay compensation unit 78 and the signal processing stage 66 to temporarily amplify signal 74 to obtain the amplified signal 743. The amplification of signal 742 can be conditional upon muting the output of the decorrelator 10. That is, the transient suppressor 82 can be configured to amplify the portion of the delay compensation unit 78 that corresponds to muting the portion of the decorrelator. An amplification level can be fixed or it can be controlled. According to an example, the amplification factor of amplifier 82 can be a factor of CQor / n / eznz / q / Yi V2 when compared to an unmuted portion of the decorrelator. That is, when the decorrelator output is muted, amplifier 88 can amplify signal 742 by V2 while the signal is not amplified 742 during times when muting is off, i.e., g = 1. Optionally, to avoid unwanted effects during transient suppression, the transient suppressor 82 can be configured to suppress a detected transient in the audio signal and to suppress a subsequent transient no sooner than a predefined hold time. For example, the transient suppressor 82 can comprise a control unit 92 configured to control and / or apply a hold time, hysteresis, and / or hold time. For example, the hold time can be shorter than the hold time. The hold time can refer to the time during which the output of the decorrelator 10 is muted in response to a detected transient, i.e., a property determined by the transient detection unit 84. The hold time can be longer than the hold time to avoid unwanted effects.For example, the wait time, i.e., the time to silence, can be 1, 2, 4, 6, 7 or 8 blocks, while the inhibit time can be at least twice the time, for example, at least 14, at least 20, at least 30 or 56 blocks or any other duration of time. According to one example, control unit 92 can also provide hysteresis to mitigate the on / off switching of transient suppression for audio signals such as low-frequency pulse trains. That is, the inhibit time provided by control unit 92 can be a first inhibit time. Transient suppressor 82 can be configured to reset the inhibit time as a second inhibit time, which is longer than the first inhibit time, if a transient occurs during the first inhibit time. In other words, even if the wait time has elapsed, but the inhibit time has not yet elapsed, and if a new transient is detected (regardless of whether the wait time has elapsed or not), the inhibit timer can be reset.Optionally, the reset inhibit timer can be longer than the canceled inhibit timer. In other words, when a transient is first detected, a wait counter and an inhibit counter are started. The transient can be muted until the wait counter reaches its stop count, for example, 8 blocks. After that, the wait counter can be reset and muting stopped. The inhibit counter can reach its stop / reset count much later, for example, 56 blocks. If a new transient is detected during this ongoing inhibit count, the inhibit counter is reset, but with a higher stop count, for example, 64 blocks. This implements hysteresis through conditional switching and modifications to the stop count.In other words, during the execution of the inhibition counter, a new activation of transient suppression or silencing can be deactivated. Transient suppressor 82 can be configured to operate in the frequency domain. Alternatively or additionally, transient suppressor 82 can be configured to mute the decorrelator portion for a longer time compared to a decorrelator pre-delay. That is, if a transient is detected in audio signal 22, the muting should still be in effect when the transient reaches the decorrelator output. In other words, modality-compliant decorrelators operate in the short-time Fourier transform (STFT) domain on overlapping transform blocks of short duration. This allows for a small processing delay of a few milliseconds, for example, 2.7 milliseconds assuming a transform size of 256 and a sampling frequency of 48 kHz, in contrast to the high delay of the PS / MDS decorrelator as described in [2] or [3], which can reach a delay time of 13.3 milliseconds at a sampling frequency of 48 kHz.Furthermore, the described decorrelators can be implemented using very low computational total-pass filters and can therefore be computationally much more efficient than time-domain decorrelation as described in [1] or [2]. If further spectral processing is required or desired, for example, source extension modeling, the described decorrelators can be made to interface directly with this processing stage in the STFT domain to achieve low computational complexity. The decorrelators described herein can therefore provide what is necessary for short processing latency and moderate computational complexity. These decorrelators can be combined with additional low-processing to model audio objects that have a spatial dimension, called spatially extended sound sources (SESS), with a perceptual property of "source extension." In other words, Figure 2 and Figure 9 show preferred embodiments of the present invention. First, the input signal or audio signal (sound from a point source, for example) can be fed into the decorrelator 20 comprising a time-block DFT with, for example, a block length of 256 samples and, for example, a 50% overlap. Then, the spectral bins of the DFT are time-delayed for a frequency-dependent duration, where lower frequencies may have a longer delay and higher frequencies may have a shorter delay. For example, the delay may be 16 sub-band samples (42.7 milliseconds at 48 kHz) for lower frequencies and may decrease to 1 sub-band sample for higher bins, i.e., z1. The decrease in delay over time may be linear, logarithmic, or otherwise with rounding to whole numbers of sub-band samples.Each bin is then sent through a full-pass filter, preferably comprising a simple chain of full-pass filters or a nested structure of full-pass filters. An exemplary full-pass filter is shown in Figure 4. A different structure is shown in Figure 5. With respect to Figure 4, a possible chain may comprise or consist of four such full-pass filters. The parameter g can be chosen to be, for example, 0.7, and the delays Mi can be prime numbers. CQor / n / eznz / q / Yi Note that Figure 4 shows the first part of the chain, i.e., Mi. Since these filters can operate in spectral bands of resolution reduction, for example, a resolution reduction factor of 128, the delays can be very small, for example, the prime numbers 1, 2, 3, and 5, or, as another example, 1, 3, 5, and 7. A time / frequency envelope shaping process can then be applied. The input signals to the envelope shaping process can be the DFT bins directly and their delayed and filtered versions. Finally, an IDFT with added overlay can synthesize the output signal. The output signal can be further processed in the time domain to obtain a left / right stereo signal from a monaural input signal in a configuration as shown in Figure 8.Alternatively, the left / right stereo signal can be assembled in the DFT frequency domain and further processed in the frequency domain, for example, for source extension / SESS modeling by means of fast convolution, if this is beneficial for overall computational efficiency. A setup for source extension modeling is shown in Figure 7. In contrast to other modalities, the alternative modality having delays Mi can be chosen as prime numbers that are approximately 128 times (corresponding to the resolution reduction factor mentioned above) larger than those chosen in the sub-band domain, for example, 131, 257, 383, and 641 (for the set of prime values ​​1, 2, 3, and 5) or 131, 383, 641, and 907 (for the set of prime values ​​1, 3, 5, and 7).For different sets of prime values ​​with a different number of prime numbers and / or different prime numbers, corresponding values ​​can be chosen. Additionally, the alternative mode may require a STFT to obtain the direct signal input to the time / frequency envelope shaper. Figure 9 shows an exemplary decorrelator in an M / S to L / R configuration with transient handling processing. Aspects of these modes are: • A transient detection detects the presence of an isolated transient. • If a transient is detected, the unrelated sound is muted for a “time out” and the compensated delay direct signal is amplified accordingly. To compensate for the coherent addition effect, a factor of 2 / V² is applied to amplify the direct signal where it replaces the unrelated signal. • To avoid triggering fast pulse trains, which are perceived as tones, an inhibition prevents triggering by the next transient for a certain “inhibition time”; the inhibition time is reset for each new transient detection during the “wait time”. • Hysteresis prevents alternating the detection of transients (for example, by increasing the “inhibition time” in case of reactivated inhibition). • Transient detection, muting, direct sound amplification, detection inhibition, and hysteresis can be conveniently implemented in the STFT domain: o The STFT block overlap provides a smooth, gradual transition. or The silencing time is greater than the decorrelator pre-delay. or The mute block counter to mute the unrelated signal and amplify CQor / n / eznz / q / Yi the direct signal. or Inhibit the block counter to inhibit transient detection. or Hysteresis to avoid alternating in the detection of transients. The embodiments of the present invention relate to: An apparatus / method for decorrelating an audio signal. • Decorator, which includes: or A pair of DFT / IDFTs (optional, if they interact directly with SESS processing in the frequency domain). or Delays in the sub-band domain; preferably lower frequencies have a higher delay and higher frequencies have a lower delay; delay distribution along the frequency: linear, logarithmic, etc. or Full-pass filters in the sub-band domain; optionally: low frequencies can have a higher delay / order and high frequencies have a lower delay / order; higher-order full-pass filters can be implemented by means of a lower-order full-pass filter stage. Short Schroeder IIR filters in the DFT (resolution reduction) sub-band domain using small integer delay prime numbers in combination with vanishing frequency delays. or T / F envelope adjuster with high time resolution (<4 ms) that works in the sub-band domain; measure the energy before and after full delay / step processing; adjust the sub-band signal energy to (as much as possible) match the original sub-band signal energy. • Low-delay decorrelator as part of the “source extension” modeling / processing (as opposed to the MPEG envelope decorrelator). • Interface for source extension processing under the time or frequency domain of DFT for computational efficiency. • Alternative implementation: Full-pass filters before delays (“post-delays’j. Figure 10 shows a schematic block diagram of a method 1000 according to a modality that can be implemented, for example, by a decorrelator described in this document. The method 1000 comprises a step 1010 in which a plurality of parts based on an audio signal are received. In 1020, each of the received parts is delayed to provide a plurality of delayed parts. 1030 comprises receiving and combining signals based on the delayed parts of the frequency representation. 1040 comprises receiving the frequency representation of the audio signal. 1050 comprises adjusting the energy of the delayed parts relative to the frequency representation of the audio signal. 1060 comprises providing a combined conformal frequency representation, for example, using the envelope shaper 16. Although some aspects have been described in the context of an apparatus, it is clear that these aspects also represent a description of the corresponding method, where a block or device corresponds to a method step or a feature of a method step. Similarly, the aspects described in the context of a method step also represent a description of a corresponding block, element, or feature of an apparatus. CQor / n / eznz / q / Yi corresponding. The inventive encoded audio signal can be stored on a digital storage medium or transmitted on a transmission medium such as a wireless transmission medium or a wired transmission medium such as the Internet. Depending on certain implementation requirements, the embodiments of the invention can be implemented in hardware or software. Implementation can be carried out using a digital storage medium, for example, a floppy disk, DVD, CD, ROM, PROM, EPROM, EEPROM, or flash memory, having electronically readable control signals stored therein, which cooperates (or is capable of cooperating) with a programmable computer system in such a way as to carry out the respective method. Some embodiments according to the invention comprise a data carrier having electronically readable control signals, which is capable of cooperating with a programmable computer system in such a way as to carry out each of the methods described in this document. Generally, the embodiments of the present invention can be implemented as a computer program product with program code, the program code being operative to carry out each of the methods when the computer program product is executed on a computer. The program code can be stored, for example, on a machine-readable medium. Other modalities include the computer program to carry out each of the methods described in this document, stored on a machine-readable carrier. In other words, a form of inventive method is, therefore, a computer program that has program code to carry out each of the methods described in this document, when the computer program is executed on a computer. An additional modality of inventive methods is, therefore, a data carrier (or a digital storage medium, or a computer-readable medium) comprising, graduated therein, the computer program for carrying out each of the methods described in this document. An additional embodiment of the inventive method is, therefore, a data stream or a sequence of signals representing the computer program for carrying out each of the methods described in this document. The data stream or the sequence of signals can be configured, for example, to be transmitted via a data communication connection, such as the Internet. An additional modality comprises a processing means, for example a computer, or a programmable logic device, configured or adapted to carry out each of the methods described in this document. An additional modality comprises a computer that has installed on it the computer program to carry out each of the methods described in this document. In some modalities, a programmable logic device (e.g., a field-programmable gate array) can be used to carry out some or all of the functionalities of the methods described in this CQor / n / eznz / q / Yi document. In some embodiments, a field-programmable gate array can cooperate with a microprocessor to carry out one or more of the methods described in this document. Generally, the methods are preferably carried out by means of any hardware device. The embodiments described above are merely illustrative of the principles of the present invention. It is understood that modifications and variations of the arrangements and details described herein will be evident to others skilled in the art. It is therefore intended that the scope of the imminent patent claims be limited only by the specific details presented herein for the purpose of describing and explaining the embodiments. References [1] W. Gomen, E. Schuijers, B. den Brinker, and J. Breebaart, “Advances in Parametric Coding for HighQuality Audio”, Paper 5852, (March 2003). [2] J. Breebaart, S. van de Par, A. Kohlrausch, and E. Schuijers, “High-quality Parametric Spatial Audio Coding at Low Bitrates”, Paper 6072, (May 2004). PS in the QMF domain: [3] H. Purnhagen, J. Engdegard, J. Roden, and L. Liljeryd, “Synthetic Ambience in Parametric Stereo Coding”, Paper 6074, (May 2004). [4] J. Herre, K. Kjórling, J. Breebaart, C. Faller, S. Disch, H. Purnhagen, J. Koppens, J. Hilpert, J. Ródén, W. Comen, K. Linzmeier, and KO. SE. Chong, “MPEG Surround-The ISO / MPEG Standard for Efficient and Compatible Multichannel Audio Coding”, (MPEG surround: the ISO / MPEG standard for efficient and compatible multichannel audio coding), J. Audio Eng. Soc., vol. 56, no. 11, pp. 932-955, (November 2008).

Claims

1. A decorrelator, comprising: a plurality of delay units (12), wherein each delay unit (12) is configured to receive a portion (14i-14n) of a frequency representation based on an audio signal (22); wherein each delay unit (12) is configured to delay the received portion (14i-14n) to provide a delayed portion (14i-14n); and an envelope shaper (16) configured to receive and combine signals based on the delayed portions (14i-14n) of the frequency representation; to receive the frequency representation of the audio signal (22); to adjust an energy of the delayed portions (14i-14n) relative to the frequency representation of the audio signal (22); and to provide a combined shaped frequency representation.

2. The decorrelator according to claim 1, wherein different parts (14i-14n) of the frequency representation comprise an equal or different number of frequency bins.

3. The decorrelator according to claim 1 or 2, further comprising a phase shifter (26) configured to shift the phase of the frequency representation (14) of the audio signal (22); or to shift the phase of the audio signal (22) in a time domain to obtain a phase-shifted audio signal (22).

4. The decorrelator according to claim 3, wherein the phase shifter (26) is configured to shift the phase of the frequency representation of the audio signal (22) and comprises a plurality of full-pass filters, wherein each full-pass filter (28) is configured to shift the phase of an associated part (14i-14n) of the frequency representation of the audio signal (22).

5. The decorrelator according to claim 4, wherein a full-pass filter (28) of the plurality of full-pass filters comprises a set of full-pass filter structures (40; 50) such as Schroeder IIR filters, connected in series with each other; wherein the full-pass filter structures (40; 50) are adapted to provide different time delays; or wherein the full-pass filter structures (40; 50) comprise a nested structure of full-pass filters.

6. The decorrelator according to claim 5, wherein a number of total pass filter structures (40; 50) and / or a circuitry of the total pass filter structure is equal or different between different total pass filters (28).

7. The decorrelator according to claim 5 or 6, wherein the different time delays are based on a prime number multiple of a local sampling frequency used to obtain the frequency representation of the audio signal (22).

8. The decorrelator according to any one of claims 5 to 7, wherein the set of full-pass filter structures (40; 50) comprises four full-pass filter structures (40; 50) and are adapted to provide a delay of 1, 2, 3 and 5 or 1, 3, 5 and 7, respectively. CQor / n / eznz / q / Yi 9. The decorrelator according to any one of claims 4 to 8, wherein a full-pass filter gain factor (28) is tailored to a value having a magnitude of 0.7 within a tolerance range of, for example, 20%.

10. The decorrelator according to claim 3, wherein the phase shifter (26) is configured to shift the phase of the audio signal (22) in a time domain; wherein the phase shifter (26) comprises a set of full-pass filter structures (40; 50) such as Schroeder IIR filters, connected in series with each other; wherein the full-pass filter structures (40; 50) are adapted to provide different time delays; or wherein the full-pass filter structures (40; 50) comprise a nested structure of full-pass filters.

11. The decorrelator according to claim 10, wherein the different total step time delays are based on a prime number multiple of a reciprocal of a sampling frequency used to obtain the frequency representation of the audio signal (22).

12. The decorrelator according to claim 10 or 11, wherein the different time delays are based on a prime number obtained by multiplying each of a set of minimal prime numbers, for example, 1, 2, 3 and 5; or 1, 3, 5 and 7, with a resolution reduction factor used to generate the (14i-14n) parts of the audio signal frequency representation (22) to obtain an intermediate result; and to use a subsequent prime number with respect to the intermediate result, for example, such as 131, 257, 383, 641 or 131, 383, 641, 907.

13. The decorrelator according to any one of claims 10 to 12, comprising a first conversion unit (24) for obtaining the frequency representation of the audio signal (22) from the audio signal (22) for the envelope shaper (16); and comprising a second conversion unit 34 for obtaining a frequency representation from the reverberated audio signal (22); wherein the parts (14i-14n) of the frequency representation form the parts (14i-14n) of the frequency representation from the reverberated audio signal (22).

14. The decorrelator according to one of the preceding claims, wherein the parts (14i-14n) of the frequency representation comprise an equal or different number of frequency bins.

15. The decorrelator according to one of the preceding claims, which is adapted to obtain a number of 16 parts (14i-14n) of the frequency representation.

16. The decorrelator according to one of the preceding claims, which is adapted to obtain the frequency representation with a number of 128 or 129 frequency bins.

17. The decorrelator of one of the preceding claims, wherein the decorrelator is adapted to further implement an equal and predefined delay for a subset or all of the (14i-14n) parts of the frequency representation.

18. The decorrelator according to one of the preceding claims, wherein the delay units (12) associated with a spectral (14i-14n) part of the plurality of delay units (12) are configured to delay the associated (14i-14n) part of the frequency representation differently when compared to delay units (12) associated with other spectral (14i-14n) parts.

19. The decorrelator according to one of the preceding claims, wherein the plurality of delay units (12) are configured to delay portions (14i-14n) of the frequency representation comprising lower frequencies by a greater time delay when compared to portions (14i-14n) of the frequency representation comprising higher frequencies.

20. The decorrelator according to claim 19, wherein a relationship between different time delays is one of linear, logarithmic and / or based on rounding in sub-band samples.

21. The decorrelator according to any one of the preceding claims, comprising a conversion unit (24) for receiving and converting the audio signal (22) or a reverberated version of the audio signal (22) in parts (14i-14n) by performing a time-block discrete Fourier transform, DFT, or a short-time Fourier transform, STFT; wherein the conversion unit (24) is configured to convert blocks having a 50% overlap within a tolerance range.

22. The decorrelator according to any of the preceding claims, comprising a conversion unit (24) for receiving and converting the audio signal (22) or a reverberated version of the audio signal (22) into parts (14i-14n) by performing a time-block discrete Fourier transform, DFT, or a short-time Fourier transform, STFT; wherein the blocks comprise a block length of 256 samples.

23. The decorrelator according to one of the preceding claims, comprising an inverse conversion unit (34) for receiving processed versions of parts of the frequency representation (14) and for synthesizing a synthesized signal from the processed versions based on an added superposition procedure.

24. The decorrelator according to one of the preceding claims, wherein the envelope shaper (16) is configured to operate in a sub-band domain and with a time resolution of less than 4 ms.

25. The decorrelator according to one of the preceding claims, comprising an interface (38) for providing a signal (36) based on the combined conformal frequency representation.

26. The decorrelator according to one of the preceding claims, wherein the envelope shaper (16) is for shaping spectral bins in time and frequency individually or as a group, for example, by implementing interdependent common shaping processing or at least by groups.

27. The decorrelator according to one of the preceding claims, comprising a signal processing stage (66) configured to receive a signal based on the combined conformal frequency representation as a monaural signal and to process the monaural signal at least as a stereophonic signal.

28. The decorrelator according to one of the preceding claims, comprising a signal processing stage (66) configured to process the combined conformal frequency representation at least CQor / n / eznz / q / Yi as a stereo audio signal; and to model the source extent based on said at least one stereo signal, for example, in the frequency domain.

29. A processing system, comprising: a decorrelator according to one of the preceding claims; and a processing stage (66) for transforming a mid / side decomposed signal to a left / right decomposed signal.

30. The processing system according to claim 29, wherein a portion (74i) of the decomposed mean / side signal is provided by the decorrelator and the other portion (742) is provided by a delay compensation unit (78) connected in parallel with the decorrelator and connected to the processing stage (66).

31. The processing system according to claim 30, comprising a transient suppressor (82) configured to detect a transient in the audio signal (22) or the frequency representation (14) thereof at an input of the decorrelator; wherein the transient suppressor (82) is configured to temporarily silence the portion (74i) provided by the decorrelator to suppress echoes in the processing stage.

32. The processing system according to claim 31, wherein the transient suppressor (82) is configured to amplify the portion of the delay compensation unit that corresponds to silencing the decorrelator portion.

33. The processing system according to claim 32, wherein the transient suppressor (82) is configured to amplify the delay compensation unit portion by a factor of V2 when compared to an unsilenced portion of the decorrelator.

34. The processing system according to any one of claims 31 to 33, wherein the transient suppressor (82) is configured to suppress a detected transient and to suppress a subsequent transient not before a predefined inhibition time.

35. The processing system according to any one of claims 31 to 34, wherein the inhibition time is a first inhibition time; wherein the transient suppressor (82) is configured to reset the inhibition time as a second inhibition time that is longer than the first inhibition time in case a transient occurs during the first inhibition time.

36. The processing system according to any one of claims 31 to 35, wherein the transient suppressor (82) is configured to operate in the frequency domain.

37. The processing system according to any one of claims 31 to 36, wherein the transient suppressor (82) is configured to silence the decorrelator portion for a longer time compared to a pre-delay of the decorrelator. CQor / n / eznz / q / Yi 38. The processing system according to any one of claims 29 to 37, wherein the decorrelator is for providing the combined conformal frequency representation as part of the mid / side decomposed signal to the processing stage; and the processing stage is for transforming the combined conformal frequency representation and a delayed version of the audio signal (22) into the left / right decomposed signal 5 in the frequency domain.

39. A method comprising: receiving (1010) a plurality of parts of a frequency representation based on an audio signal; delaying (1020) each of the received parts to provide a plurality of delayed parts; and receiving (1030) and combining signals based on the delayed parts of the frequency representation; receiving (1040) the frequency representation of the audio signal; adjusting (1050) an energy of the delayed parts relative to the frequency representation of the audio signal; and providing (1060) a combined conformal frequency representation.

40. The method according to claim 39, further comprising: 15 detecting a transient in the audio signal (22) or the frequency representation (14) thereof; temporarily muting a portion (74i) provided by a decorrelator to suppress echoes in a processing stage.

41. A computer-readable means for providing a combined conformal frequency representation comprising the method according to claim 39 or 40.

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