Magnetic component device

Description

[0001] The utility model relates to the field of electrical engineering, in particular to the designs of the magnetic component of a transformer and can be used to reduce losses in transformer windings.

[0002] A high-voltage transformer device is known (Patent RU 2435242, published on 27.11.2011), which contains a primary flat winding, a secondary winding of the litz wire type, a core and a coil having a plurality of slots in which the litz wire winding is wound, wherein the surfaces of the flat windings rest against the flat surfaces of the core.

[0003] This solution ensures effective cooling of the windings, but has a low specific power, since the additional losses caused by eddy currents in the transformer windings are not taken into account when creating this device. Optimization and consideration of these losses when forming the device allows for an increase in the specific power and a reduction in the overall dimensions of the product.

[0004] A pulse transformer design (Patent RU 2547809, published July 20, 2014) is known, comprising a magnetic core, coils, windings, and conductive connections. The proposed technical solution is aimed at minimizing additional losses in the windings at high frequencies and ensuring a high utilization factor of the magnetic core window with a minimum turn length.

[0005] The disadvantage of the proposed device is the low accuracy of optimization of high-frequency losses caused by the proximity effect in magnetic windings made with alternating layers, as a result of which the specific power decreases and the weight and dimensions of the final device increase.

[0006] The technical result of the utility model consists in creating a magnetic component device that has high manufacturability due to high specific power while reducing the weight and dimensions of the device.

[0007] The technical result is achieved in that in the device of a magnetic component, including a core, primary and secondary windings, characterized in that the layers of the primary and secondary windings are arranged alternately between each other, while the thickness of the conductor of each winding corresponds to the minimum value of losses and is set according to the minimum value of dissipated power in the windings for each layer (i), and the field strength coefficients on the internal (n 1i ) and external (n 2i ) the boundaries of the layers are set as follows:

[0008] n 11 =0 - for the outer layer of the winding (i=1);

[0009] n 1i =n 2(i-1) - for all other layers (i>1);

[0010] n 2i =n 1i +q i - for the primary winding;

[0011] n 2i =n 1i -q i N1 / k tr - for the secondary winding;

[0012] q i - number of turns in a layer;

[0013] k tr=N2 / N1 - transformation ratio;

[0014] N1 - number of turns of the primary winding;

[0015] N2 - number of turns of the secondary winding.

[0016] The essence of the proposed utility model is explained by drawings, where

[0017] Fig. 1 shows the field strength at the boundary of the winding layer,

[0018] Fig. 2 shows the transformation of a round cross-section conductor into a flat conductor with an equivalent cross-section,

[0019] Fig. 3 shows the PPSS topology transformer and the field strength diagram at the layer boundaries,

[0020] Fig. 4 shows the diagrams of currents in the windings of a forward transformer,

[0021] Fig. 5 shows the PSSP topology transformer and the field strength diagram at the layer boundaries,

[0022] Fig. 6 shows a graph of the dependence of losses in the windings of a forward transformer of the PPSS topology on the conductor diameter,

[0023] Fig. 7 shows a graph of the dependence of dissipated power on the conductor diameter in the windings of a forward transformer of the PSSP topology.

[0024] The further representation of the magnetic component device is presented under the condition of simultaneous current flow in the windings.

[0025] The preliminary stages of constructing the magnetic component device are the selection of the required number of turns of the primary and secondary windings N1 and N2, the transformation ratio k tr =N2 / N1 based on the core parameters and the required product parameters, as well as the choice of the initial topology of the transformer turns.

[0026] To reduce the field strength modulus between the windings, the layers of the primary and secondary windings are made alternately.

[0027] Description of the implementation of a method for optimizing the topology of transformer turns with simultaneous current flow in the windings:

[0028] Determining the thickness of the conductor for alternating layers of the primary and secondary windings begins with setting the field strength coefficients on the internal (n 1i ) and external (n 2i ) layer boundaries according to the following rules:

[0029] n 11 =0 - for the outer layer of the winding (i=1);

[0030] n 1i =n 2(i-1) - for all other layers (i>1);

[0031] n 2i =n 1i +q i - for the primary winding;

[0032] n 2i = n 1i -q i / k tr - for the secondary winding.

[0033] where q i - the number of turns in layer i,

[0034] The field strength coefficients show how many times the field at the boundaries of layer i differs from the field,

[0035] ,

[0036] created by the outer layer of the winding,

[0037] where I is the current in the first layer of the winding;

[0038] h - core window height.

[0039] The field strength coefficients at the layer boundaries increase with each layer of the primary winding and decrease with each layer of the secondary winding. Thus, the magnetic field magnitude between layers is reduced by alternating windings. This reduces the influence of the proximity effect on winding losses during the formation of the magnetic component windings. Therefore, the field strength coefficients influence the dissipated power, which is necessary for selecting the conductor thickness.

[0040] The thickness of the conductor of each winding corresponds to the minimum loss value and is set according to the minimum value of dissipated power in the windings.

[0041] Conversion of stress coefficients n 1i and n 2i for the secondary winding by multiplying them by the transformation ratio k tr (n 1i =k tr n 1i ; n 2i =k tr n 1i.) is necessary for the subsequent calculation of losses caused by the proximity effect in each layer and understanding the thickness of the turns that make up the magnetic component device.

[0042] By default, winding layers are considered, which are rectangular cross-section conductors (Fig. 2), located along the entire height of the window, and each winding layer contains only one conductor (q i =1 for all layers). If the winding layer consists of round cross-section conductors or several flat conductors, then the variables d0 and σ0 are replaced by equivalent values ​​[2], i.e. for a conductor with a round cross-section:

[0043] ,

[0044] where d is the layer thickness,

[0045] d0 - diameter of a round conductor,

[0046] w - coil width,

[0047] q - number of turns in a layer,

[0048] h - core window height,

[0049] σ0 - specific conductivity of copper,

[0050] σ - reduced conductivity.

[0051] For flat conductors:

[0052] .

[0053] For the final selection of the thickness of each layer (i), it is necessary to represent the volume of dissipated power, which is calculated based on the formula:

[0054] ,

[0055]

[0056] where P ACi - AC losses in layer i,

[0057] P in - the power dissipated in layer i for the n-th harmonic,

[0058] ,

[0059] f - operating frequency of the magnetic component,

[0060] l i - the length of the turn of layer i, h - the height of the core window,

[0061] I n - the root-mean-square current of the n-th harmonic in the corresponding (primary or secondary) winding,

[0062] n 1i - field strength coefficient at the outer boundary of layer i,

[0063] n 2i - field strength coefficient at the inner boundary of layer i,

[0064] q i - the number of turns in layer i,

[0065] I DCi - the constant component of the current in the winding corresponding to layer i,

[0066] R DCi - DC resistance of layer i;

[0067] The graph of the dissipated power P generated based on the results of construction i (Fig. 6) When setting the required specified voltage coefficients for each layer, it is possible to obtain a layer thickness d corresponding to the minimum value of losses in the windings, taking into account the limitations of the core geometry. For a circular conductor, the inverse transformation of the layer thickness d into the diameter d0 is performed.

[0068] The conductor thickness is selected based on the obtained dependences of dissipated power on the conductor thickness (diameter) (Fig. 6). The core window height h is taken into account, limiting the maximum conductor diameter that can be wound according to a given turn topology (Figs. 3, 5).

[0069] The selected value of conductor thickness depending on the dissipated power in the windings P is optimal in terms of minimizing the proximity effect for a given transformer topology, which affects the increase in the final efficiency of the product.

[0070] The following is an example of the implementation of a device with an alternate arrangement of turns of the primary and secondary windings when selecting the optimal diameter of the transformer wire.

[0071] For a forward transformer made on an ER7.5 core, based on the required parameters of the final product and the permissible induction of the core, 26 turns of the primary winding d0=0.16 mm and 20 turns of the secondary winding d0=0.2 mm were selected. Transformation ratio

[0072] .

[0073] Based on the core geometry (window height h), it is proposed to wind 13 turns in two layers for the primary winding and 10 turns in two layers for the secondary winding (Fig. 3)

[0074] Set the values ​​of the field strength coefficients on the internal (n 1i ) and external (n 2i ) layer boundaries. Their values ​​are presented in Table 1. The diagrams of the currents flowing in the transformer windings are shown in Fig. 4.

[0075]

[0076] To minimize the field strength at the boundaries of the winding layers, we place the secondary winding between two layers of the primary winding, obtaining a PSSP-type topology (Fig. 5). The coefficient values ​​are presented in Table 2.

[0077]

[0078] Sum of squares of coefficients n 1i , n 2i , for the PSSP topology is smaller (676) than for the PPSS topology (2048). This implies that this arrangement during construction will be more optimal for minimizing the proximity effect in the transformer windings.

[0079] To select the optimal conductor diameter and calculate the losses in the winding, we perform a transformation of the voltage coefficients n 1i , n 2i for the secondary winding.

[0080] Let's compare the selected conductor thickness d depending on the dissipated power and plot a graph of the dependence of the dissipated power on the wire diameter for each winding.

[0081] Thus, we obtain graphs of dissipated power for the PPSS (Fig. 6) and PSSP (Fig. 7) topologies, from which we can determine the amount of losses in the transformer windings, as well as ensure the selection of the optimal wire diameter for each winding.

[0082] The graph for the non-interleaved topology (PPSS) shows local loss minima corresponding to d0 = 0.16 mm for the primary winding and d0 = 0.17 mm for the secondary winding. However, the graph for the interleaved winding topology (PSSP) does not show local minima, allowing the selection of the largest possible wire diameter (0.16 mm for the primary winding and 0.2 mm for the secondary winding, respectively). The dissipated power values ​​are significantly lower than in the non-interleaved case.

[0083] The values ​​of losses in windings for the selected wire diameters for two topologies are presented in Table 3.

[0084]

[0085] Thus, the magnetic component design has lower losses caused by eddy currents in the windings, which allows for a reduction in its overall dimensions and an increase in the efficiency of the final product.

Claims

A magnetic component device comprising a core, primary and secondary windings, characterized in that the layers of the primary and secondary windings are arranged alternately between each other, wherein the thickness of the conductor of each winding corresponds to the minimum value of losses and is set according to the minimum value of dissipated power in the windings for each layer (i), and the field strength coefficients on the internal (n 1i ) and external (n 2i ) the boundaries of the layers are set as follows: n 11 =0 - for the outer layer of the winding (i=1); n 1i =n 2(i-1) - for all other layers (i>1); n 2i =n 1i +q i - for the primary winding; n 2i =n 1i -q i N1 / k tr - for the secondary winding; q i - number of turns in a layer; k tr =N2 / N1 - transformation ratio; N1 - number of turns of the primary winding; N2 - number of turns of the secondary winding.

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