Bandgap reference circuit with leakage current compensation

Abstract

A bandgap reference voltage generation circuit includes: first and second bipolar junction transistors (BJTs) biased with first and second current densities, whereby the base-emitter voltages of the first and second BJTs have a base-emitter voltage difference, wherein the first BJT is used to determine a negative temperature coefficient signal; a differential voltage sensing resistor connected in series with the second BJT from ground potential to form a sub-branch, the differential voltage sensing resistor being located near the ground potential side; and a feedback circuit for controlling the sub-branch to have the same cross voltage as the first BJT, whereby the cross voltage across the differential voltage sensing resistor includes the base-emitter voltage difference, thereby determining a positive temperature coefficient signal; the feedback circuit further generates a reference voltage with a temperature coefficient of 0 based on the positive and negative temperature coefficient signals.

Description

Technical Field

[0001] This invention relates to a bandgap reference voltage generation circuit, and more specifically to a bandgap reference voltage generation circuit with leakage current compensation. Prior Technology

[0002] Figure 1 shows a prior art 1.2V bandgap reference voltage circuit. This circuit utilizes bipolar junction transistors (BJTs) Q1 and Q2 to generate a base-emitter voltage difference (ΔVBE), and through a combination of a negative temperature coefficient current Ictat1 and a positive temperature coefficient current Iptat1, generates a stable reference voltage Vbgp. The formula for the reference voltage Vbgp is:

[0003] Vbgp = (R2 / R3) * VT * ln(N) + VBE1.

[0004] Where VT is the thermal voltage, and ln(N) represents the logarithm of the area ratio of bipolar junction transistors Q2 and Q1. This circuit can provide a stable 1.2V reference voltage, but its supply voltage needs to be higher than 1.2V, making it difficult to support applications with lower supply voltages. Furthermore, in applications where the quiescent current of the bandgap reference voltage circuit is extremely low, as the temperature rises, the leakage current of the P-type substrate (e.g., the leakage current Iq2sl from the parasitic collector-body diodes D1 and D2 of Q1 and Q2) will affect the accuracy of the negative temperature coefficient current Ictat1 and the positive temperature coefficient current Iptat1, thus affecting the temperature coefficient of the reference voltage.

[0005] Figure 2 shows another prior art bandgap reference voltage circuit. Figure 2 is an improved version of Figure 1, aiming to achieve a bandgap reference voltage below 1V, suitable for low-voltage applications. This design folds the feedback resistors R1 and R2 to generate a current I2 with zero temperature coefficient, and uses a current mirror to generate a current I3 flowing through the output resistor R4 to generate the reference voltage Vbgp, thereby further scaling the reference voltage Vbgp to below 1V. The output voltage formula for this circuit structure is modified as follows:

[0006] Vbgp = (R4 / R2) * [(VBE1) + (R2 / R3) * VT * ln(N)].

[0007] As shown in the formula, by adjusting the ratio of resistors R4 to R2, the reference voltage Vbgp can be effectively reduced to below 1V. However, similar to Figure 1, this design is still affected by the leakage current of the P-type substrate. Especially under high temperature conditions, the leakage current will cause a shift in the negative temperature coefficient current Ictat1 and the positive temperature coefficient current Iptat1, further affecting the temperature coefficient of the reference voltage Vbgp.

[0008] In view of this, the present invention addresses the shortcomings of the prior art by proposing a bandgap reference voltage generation circuit that can effectively reduce the impact of leakage current. Summary of the Invention

[0009] In one viewpoint, the present invention provides a bandgap reference voltage generation circuit, comprising: a first bipolar junction transistor biased with a first current density to determine a negative temperature coefficient signal; a second bipolar junction transistor biased with a second current density, wherein the first current density is greater than the second current density, thereby creating a base-emitter voltage difference between the base-emitter voltage of the first bipolar junction transistor and the base-emitter voltage of the second bipolar junction transistor; and a differential voltage sensing resistor. A sub-branch is formed by connecting a supply potential in series with the second bipolar junction transistor, wherein the differential voltage sensing resistor is located close to the supply potential side; and a feedback circuit is used to control the sub-branch to have the same cross voltage as the first bipolar junction transistor, thereby the cross voltage on the differential voltage sensing resistor includes the base-emitter voltage difference, thereby determining a positive temperature coefficient signal; wherein the feedback circuit further generates a reference voltage with a temperature coefficient of 0 based on the positive temperature coefficient signal and the negative temperature coefficient signal.

[0010] In a preferred embodiment, by arranging the differential voltage sensing resistor close to the supply potential side, the voltage across the differential voltage sensing resistor does not include a first leakage current between the collector and body of the second bipolar junction transistor, wherein the base of the second bipolar junction transistor is coupled to the supply potential.

[0011] In a preferred embodiment, the bandgap reference voltage generation circuit further includes a third bipolar junction transistor with its base biased off and its collector coupled to the reference voltage, wherein a second leakage current between the collector and body of the third bipolar junction transistor is used to compensate for the component of the first leakage current in the reference voltage.

[0012] In a preferred embodiment, the bandgap reference voltage generation circuit further includes a fourth bipolar junction transistor and is coupled to the first bipolar junction transistor. The base of the fourth bipolar junction transistor is biased to be off, and a third leakage current between the collector and body of the fourth bipolar junction transistor is used to compensate for the first leakage current.

[0013] In a preferred embodiment, the area ratio of the first bipolar junction transistor to the second bipolar junction transistor is 1:N, wherein the area ratio of the first bipolar junction transistor to the fourth bipolar junction transistor is 1:(N-1), and N is greater than 1.

[0014] In a preferred embodiment, the feedback circuit includes: a first feedback resistor coupled to the first bipolar junction transistor to form a first branch; a second feedback resistor coupled to the sub-branch to form a second branch, the second feedback resistor being used to determine a negative temperature coefficient current based on the base-emitter voltage of the first bipolar junction transistor, corresponding to the negative temperature coefficient signal; an output resistor forming a third branch; and a controlled current mirror circuit that generates a first bias current, a second bias current, and a third bias current mirrored to each other based on an error amplification signal. A current is used to bias the first to third branches respectively to generate the reference voltage in the third branch; and an amplifier is used to generate the error amplification signal based on the voltage difference between the first branch and the second branch to adjust the first to third currents; wherein the voltage difference sensing resistor determines a positive temperature coefficient current based on the base-emitter voltage difference, which corresponds to the positive temperature coefficient signal; thereby, the error amplification signal controls the temperature coefficients of the positive temperature coefficient current and the second negative temperature coefficient current included in the third current to cancel each other out, thereby making the temperature coefficient of the reference voltage 0.

[0015] The following detailed description through specific embodiments will make it easier to understand the purpose, technical content, features and effects achieved by this invention. Simple Explanation of the Diagram

[0016] Figure 1 shows a circuit diagram of a prior art bandgap reference voltage circuit.

[0017] Figure 2 shows a circuit diagram of another prior art bandgap reference voltage circuit.

[0018] Figure 3A is a circuit diagram of a bandgap reference voltage generation circuit according to one embodiment of the present invention.

[0019] Figure 3B shows a circuit diagram of a specific bandgap reference voltage generation circuit according to one embodiment of the present invention.

[0020] Figure 3C shows a circuit diagram of a bandgap reference voltage generation circuit with a complete compensation structure according to one embodiment of the present invention.

[0021] Figure 4 shows the operating waveforms of the relevant signals of the bandgap reference voltage generation circuit according to one embodiment of the present invention.

[0022] Figure 5 shows the operating waveforms of the relevant signals of the bandgap reference voltage generation circuit according to one embodiment of the present invention. Implementation

[0023] The diagrams in this invention are all schematic and are mainly intended to show the coupling relationship between circuits and the relationship between signal waveforms. The circuits, signal waveforms and frequencies are not drawn to scale.

[0024] Figure 3A illustrates the basic structure of a bandgap reference voltage generation circuit according to one embodiment of the present invention. As shown in Figure 3A, the bandgap reference voltage generation circuit 400 includes bipolar junction transistors Q1 and Q2, and a feedback circuit 401. The bipolar junction transistors Q1 and Q2 are biased with different current densities, thereby generating a base-emitter voltage difference ΔVBE. In one embodiment, the bases of each bipolar junction transistor Q1 and Q2 are coupled to their respective collectors. In one embodiment, since the area ratio of bipolar junction transistors Q1 and Q2 is 1:N, where N is greater than 1, and their bias currents are, for example, equal, the current density of bipolar junction transistor Q1 is greater than the current density of bipolar junction transistor Q2. As shown in Figure 3A, the differential pressure sensing resistor R3 is connected in series with the bipolar junction transistor Q2 from the supply potential to form a sub-branch 430. In this embodiment, the supply potential is the ground potential. The differential pressure sensing resistor R3 is positioned close to the supply potential (ground potential), and the bipolar junction transistor Q2 is stacked on top of the differential pressure sensing resistor R3 and further coupled to the feedback circuit 401. The voltage across the differential pressure sensing resistor R3 includes the base-emitter voltage difference ΔVBE, thereby generating a positive temperature coefficient current Iptat2. In addition, the bipolar junction transistor Q1 is also coupled to the feedback circuit 401.

[0025] From one perspective, the present invention configures the bipolar junction transistors Q1 and Q2 as follows: a differential voltage sensing resistor R3 is connected in series with the emitter of the bipolar junction transistor Q2 to form a sub-branch 430 between the supply potential and the feedback circuit 401. The emitter and collector of the bipolar junction transistor Q1 are respectively coupled to the supply potential and the feedback circuit 401. The body (substrate) of the bipolar junction transistors Q1 and Q2 are also coupled to the supply potential. On the other hand, the collectors of the bipolar junction transistors Q1 and Q2 have related, for example, equal voltages through a virtual short circuit in the feedback circuit 401, so that the voltage across the bipolar junction transistor Q1 and the sub-branch 430 is equal. Thus, the base-emitter voltage difference ΔVBE of the bipolar junction transistors Q1 and Q2 is presented on the differential voltage sensing resistor R3. Due to the aforementioned configuration, the voltage across the differential pressure sensing resistor R3 does not include the leakage current Iq2sl of the parasitic diode D2 between the collector and body of the bipolar junction transistor Q2. Therefore, the leakage current Iq2sl can be effectively compensated, the details of which are described later.

[0026] Furthermore, it should be noted that the bipolar junction transistors Q1 and Q2 shown in the figures are both NPN transistors formed in a P-type substrate, and the supply potential coupled to the substrate is ground or a negative voltage. In other embodiments, they can also be PNP transistors formed in an N-type substrate, and the supply potential coupled to the substrate is ground or a negative voltage. The embodiments of PNP transistors can still achieve the same effect according to the configuration of the present invention described above, and the details can be deduced by those skilled in the art.

[0027] Feedback circuit 401 includes feedback amplifier circuit 404, which controls the sub-branch 430 and the bipolar junction transistor Q1 to have the same cross voltage through feedback control, so that the cross voltage across the differential voltage sensing resistor R3 corresponds to the base-emitter voltage difference ΔVBE, thereby generating a positive temperature coefficient signal Sptat and a negative temperature coefficient signal Scatt. Feedback circuit 401 further superimposes the positive temperature coefficient signal Sptat and the negative temperature coefficient signal Scatt to generate a reference voltage Vbg with a temperature coefficient of 0.

[0028] Please refer to Figure 3B. Figure 3B corresponds to Figure 3A and shows a specific bandgap reference voltage generation circuit. As shown in Figure 3B, the feedback circuit 401 includes an amplifier 410, a controlled current mirror circuit 420, and feedback resistors R1 and R2. Feedback resistor R1 is connected in parallel with bipolar junction transistor Q1 to form a first branch 431. Feedback resistor R1 is used to generate a negative temperature coefficient current Ictat1 based on the base-emitter voltage of bipolar junction transistor Q1. Feedback resistor R2 is connected in parallel with sub-branch 430 to form a second branch 432. Feedback resistor R2 is used to generate a negative temperature coefficient current Ictat2 based on the base-emitter voltage of bipolar junction transistor Q1. The positive temperature coefficient current Iptat2 corresponds to the positive temperature coefficient signal Sptat, and the negative temperature coefficient current Ictat2 corresponds to the negative temperature coefficient signal Scatt. Output resistor R4 forms a third branch 433.

[0029] Amplifier 410 generates an error amplification signal EAO based on the difference between the trans-voltages VA and VB of the first branch 431 and the second branch 432. The controlled current mirror circuit 420 generates mutually mirrored currents I30b, I31b, and I32b based on the error amplification signal EAO, which are used to bias the first branch 431, the second branch 432, and the third branch 433, respectively, to generate trans-voltages VA and VB and a reference voltage Vbg, respectively. The trans-voltages VA and VB are virtually short-circuited under the negative feedback control of amplifier 410 and have the same voltage.

[0030] Based on the appropriate design of the ratio of the feedback resistor R2 to the differential pressure sensing resistor R3, the bandgap reference voltage generation circuit 400 adjusts the controlled current mirror circuit 420 through the feedback mechanism of the amplifier 410, so that the temperature coefficients of the negative temperature coefficient current Ictat2 and the positive temperature coefficient current Iptat2 on the differential pressure sensing resistor R3 cancel each other out, generating currents I30b, I31b and I32b with zero temperature coefficients, thereby making the reference voltage Vbg generated on the output resistor R4 have zero temperature coefficients.

[0031] Please refer to Figure 3C. Figure 3C shows a bandgap reference voltage generation circuit with a complete compensation structure. This embodiment is similar to the embodiment in Figure 3B, except that a bipolar junction transistor Q3 is added as a compensation element. In this embodiment, the third branch 433 further includes the bipolar junction transistor Q3, whose base is grounded and is in a bias off state, and whose collector is coupled to the reference voltage Vbg.

[0032] Referring back to Figure 3B, the bandgap reference voltage generation circuit 400 in Figure 3B, through the aforementioned configuration, ensures that the voltage across the differential sensing resistor R3 does not include the leakage current Iq2sl of the parasitic diode D2 between the collector and body of the bipolar junction transistor Q2, but only reflects the base-emitter voltage difference ΔVBE. Therefore, the positive temperature coefficient current Iptat2 flowing through the differential sensing resistor R3 can have a purely positive temperature coefficient. However, the total current I31b still includes the leakage current Iq2sl of the parasitic diode D2, and the leakage current Iq2sl is also reflected in the current I32b and the reference voltage Vbg. Therefore, Figure 3C further utilizes the collector-body leakage current Iq3sl of the bipolar junction transistor Q3 to compensate for the leakage current Iq2sl component in the current I32b, further reducing the influence of the P-type substrate leakage current on the reference voltage Vbg.

[0033] In one embodiment, the first branch 431 in FIG3C further includes a bipolar junction transistor Q4, which is connected in parallel with the bipolar junction transistor Q1. The base of the bipolar junction transistor Q4 is also biased to be off. Although the currents I30b and I31b of the first branch 431 and the second branch 432 have a preset ratio (e.g., 1:1), since the areas of the bipolar junction transistors Q1 and Q2 are different, their leakage currents Iq1sl and Iq2sl are also different. In the embodiment shown in FIG3B, this will cause the current density of the bipolar junction transistors Q1 and Q4 to deviate from the preset ratio. Therefore, the bipolar junction transistor Q4 in FIG3C can be used to compensate for the difference between the leakage currents Iq2sl and Iq1sl. Specifically, in a preferred embodiment, the area of ​​the bipolar junction transistor Q4 is the difference between the areas of bipolar junction transistors Q2 and Q3. That is, if the area ratio of bipolar junction transistors Q1 to Q2 is 1:N, then the area ratio of bipolar junction transistors Q1 to Q4 is set to 1:(N-1).

[0034] Figure 4 shows an operational waveform diagram according to one embodiment of the present invention. The reference voltage Vbg is the reference voltage generated by the bandgap reference voltage generation circuit 400 in Figure 4, and the reference voltage Vbgp is the reference voltage generated by the prior art bandgap reference voltage generation circuit (as shown in Figure 2). As can be seen from Figure 4, since the current I3 flowing through the output resistor R4 in the prior art is proportional to Ictat2 + Iptat2 + Iq2sl, it will fluctuate greatly at high temperatures (as shown by the dashed circle). Therefore, the reference voltage Vbgp also exhibits a large temperature coefficient change at high temperatures. On the other hand, in the bandgap reference voltage generation circuit 400 in Figure 4, the current I32b flowing through the output resistor R4 is proportional to Ictat2 + Iptat2, and it only fluctuates slightly even at high temperatures. Therefore, the reference voltage Vbg is relatively smoother than the reference voltage Vbgp.

[0035] As can be seen from the foregoing embodiments, the present invention can effectively overcome the influence of substrate leakage current in applications with extremely low bias current, and maintain a low temperature coefficient of reference voltage Vbg in the range of -40°C to 150°C. Compared with the phenomenon of a sharp drop in the reference voltage Vbgp curve at high temperatures in the prior art, the present invention, through the adjustment of the position of the differential pressure sensing resistor R3 and the introduction of bipolar junction transistors Q3 and Q4, can accurately eliminate the influence of P-type substrate leakage current, and achieve more accurate and linear temperature compensation performance.

[0036] Figure 5 shows an operational waveform diagram according to one embodiment of the present invention. As shown in Figure 5, the nearly horizontal line represents the reference voltage Vbg generated by the bandgap reference voltage generation circuit with bipolar junction transistor Q3, while the line that increases upward with temperature in the high-temperature region represents the reference voltage Vbgp generated by the bandgap reference voltage generation circuit without bipolar junction transistor Q3. As can be seen from Figure 5, the reference voltage Vbg generated by the bandgap reference voltage generation circuit with bipolar junction transistor Q3 is much flatter than the reference voltage Vbgp generated by the bandgap reference voltage generation circuit without bipolar junction transistor Q3. In this embodiment, when the bipolar junction transistor Q3 is used to compensate for the leakage current of the bipolar junction transistor Q2, the formula for the reference voltage Vbg can be expressed as:

[0037] Vbg = R4 * (Ictat2 + Iptat2 + Iq2sl – Iq3sl)

[0038] As shown in the above formula, the leakage current Iq3sl of the bipolar junction transistor Q3 can accurately compensate for the leakage current Iq2sl of the bipolar junction transistor Q2, and the position design of the differential voltage sensing resistor R3 ensures that the base-emitter voltage difference ΔVBE is no longer affected by the leakage current. Therefore, this invention effectively maintains the stability of the reference voltage Vbg within the temperature range, making it particularly suitable for applications requiring high precision, low power consumption, and a wide temperature range.

[0039] The present invention has been described above with reference to preferred embodiments. However, the above description is only intended to facilitate understanding of the invention by those skilled in the art and is not intended to limit the broadest scope of the invention. The described embodiments are not limited to individual application and can also be combined. For example, two or more embodiments can be used in combination, and a component of one embodiment can replace a corresponding component in another embodiment. Furthermore, within the same spirit of the invention, those skilled in the art can conceive of various equivalent changes and combinations. For example, the phrase "processing or calculating based on a signal or generating an output result" in the present invention is not limited to the signal itself, but also includes, when necessary, performing voltage-to-current conversion, current-to-voltage conversion, and / or proportional conversion on the signal, and then processing or calculating based on the converted signal to generate an output result. Therefore, within the same spirit of the invention, those skilled in the art can conceive of various equivalent changes and combinations, and there are many combinations, which will not be listed here. Therefore, the scope of the present invention should cover the above and all other equivalent changes.

[0040] 400: Bandgap reference voltage generation circuit 401: Feedback Circuit 404: Feedback amplifier circuit 410: Amplifier 420: Controlled Current Mirror Circuit 430: Sub-branch 431: First Branch Road 432: Second branch road 433: Third Branch Road 440: Feedback Amplifier Circuit D2:Q2 Parasitic Diode Between Collector and Bulk Electrodes EAO: Error Amplification Signal I30b, I31b, I32b: Mirror bias current I32: Mirror current Ictat1, Ictat2: Negative temperature coefficient currents Iptat1, Iptat2: Positive temperature coefficient currents Iq1sl, Iq2sl, Iq3sl: Leakage current N: Area ratio parameter between Q1 and Q2 Q1, Q2, Q3, Q4: Bipolar junction transistors R1, R2: Feedback resistors R3: Differential pressure sensing resistor R4: Output resistor Sctat: Negative Temperature Coefficient Signal Sptat: Positive Temperature Coefficient Signal VA, VB: Trans-voltage Vbg, Vbgp: Reference voltage VT: Thermoelectric Voltage

Claims

1. A bandgap reference voltage generation circuit, comprising: a first bipolar junction transistor biased with a first current density to determine a negative temperature coefficient signal; a second bipolar junction transistor biased with a second current density, wherein the first current density is greater than the second current density, thereby having a base-emitter voltage difference between the base-emitter voltage of the first bipolar junction transistor and the base-emitter voltage of the second bipolar junction transistor, and the second bipolar junction transistor having a first leakage current between its collector and body; and a differential voltage sensing resistor connected in series with the second bipolar junction transistor from a supply potential to form a sub-branch, wherein the differential voltage sensing resistor is located close to the supply potential side; A feedback circuit is used to control the sub-branch to have the same cross voltage as the first bipolar junction transistor, thereby the cross voltage across the differential sensing resistor includes the base-emitter voltage difference, which in turn determines a positive temperature coefficient signal. The feedback circuit also generates a reference voltage based on the positive temperature coefficient signal and the negative temperature coefficient signal. A third bipolar junction transistor has its base coupled to the supply potential in a biased off state and its collector coupled to the reference voltage. A second leakage current between the collector and body of the third bipolar junction transistor is used to compensate for the first leakage current component in the reference voltage.

2. The bandgap reference voltage generating circuit as claimed in claim 1, wherein by arranging the differential voltage sensing resistor close to the supply potential side, the voltage across the differential voltage sensing resistor does not include the first leakage current between the collector and body of the second bipolar junction transistor, wherein the body of the second bipolar junction transistor is coupled to the supply potential.

3. The bandgap reference voltage generating circuit as described in claim 1 or 2, wherein the bandgap reference voltage generating circuit further includes a fourth bipolar junction transistor coupled to the first bipolar junction transistor, the base of the fourth bipolar junction transistor being biased to be off, wherein a third leakage current between the collector and body of the fourth bipolar junction transistor is used to compensate for the first leakage current.

4. The bandgap reference voltage generating circuit as described in claim 3, wherein the area ratio of the first bipolar junction transistor to the second bipolar junction transistor is 1:N, and the area ratio of the first bipolar junction transistor to the fourth bipolar junction transistor is 1:(N-1), wherein N is greater than 1.

5. The bandgap reference voltage generation circuit as described in claim 1, wherein the feedback circuit comprises: A first feedback resistor is connected in parallel to the first bipolar junction transistor to form a first branch; A second feedback resistor, coupled in parallel to the sub-branch to form a second branch, the second feedback resistor being used to determine a negative temperature coefficient current based on the base-emitter voltage of the first bipolar junction transistor, corresponding to the negative temperature coefficient signal; an output resistor forming a third branch; a controlled current mirror circuit generating a first bias current, a second bias current, and a third bias current mirrored to each other based on an error amplification signal, respectively used to bias the first to third branches to generate the reference voltage in the third branch; An amplifier is provided to generate the error amplification signal based on the voltage difference between the first branch and the second branch, so as to adjust the first to third bias currents; wherein the differential voltage sensing resistor determines a positive temperature coefficient current based on the base-emitter voltage difference, which corresponds to the positive temperature coefficient signal; and wherein the feedback circuit superimposes the positive temperature coefficient signal and the negative temperature coefficient signal.

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