Control method and control apparatus for resonant DC-DC converter, and switching power supply

Abstract

Disclosed in the present invention are a control method and control apparatus for a resonant DC-DC converter, and a switching power supply. The control method comprises: a resonant sampling step of obtaining a first voltage signal that represents the magnitude of a resonant cavity current of a resonant DC-DC converter; a signal processing step of converting the first voltage signal into a resonant charge signal that can be used for loop control; a feedback collection step of acquiring a first feedback signal UFB that represents the magnitude of an output voltage of the resonant DC-DC converter; a first compensation step of obtaining a second voltage signal that represents a switching period TSW of the resonant DC-DC converter, and using the second voltage signal to compensate the first feedback signal UFB, so as to obtain a second feedback signal UFB2, wherein the second feedback signal UFB2 satisfies the relational expression UFB2=k×UFB×TSW, k being a constant; and a pulse width generation step of comparing the resonant charge signal with the second feedback signal, and on the basis of the comparison result, generating a driving pulse width signal. The present invention eliminates the adverse effect of a switching period on a control loop.

Description

A control method, control device and switching power supply for a resonant DC-DC converter Technical Field

[0001] This invention relates to the field of switching converter control, and particularly to a control method, control device, and switching power supply for a resonant DC-DC converter. Background Technology

[0002] Traditional LLCs can only operate near the resonant point, resulting in a narrow input / output range. They typically employ PFM control with a single output voltage loop, which suffers from poor transient response and weak start-up capability. With technological advancements, the market demand for wider input / output voltage ranges has increased, leading to dual-loop control of resonant current and output voltage becoming the mainstream technology for LLC loop control. Dual-loop control offers advantages such as fast transient response and strong start-up capability. However, because the resonant current needs to be converted into an integral quantity for control, the control quantity varies with frequency. In wide input and wide output applications, this frequency variation range is significant, leading to larger control errors and problems such as instability under light loads, abnormal mode switching, and large differences in over-power protection points.

[0003] Research and development of the latest domestic and international control technologies revealed that existing technologies can only compensate for control quantity changes caused by frequency variations from the resonant point to the high-gain portion. However, significant differences remain between the resonant point and the low-gain portion. Although the latest international technologies have introduced input voltage feedforward compensation, which can optimize control errors caused by increased input voltage, it cannot compensate for errors caused by decreased output voltage. Furthermore, existing technologies all require the addition of analog-to-digital (A / D) and digital-to-analog (D / A) conversion circuits. Adding A / D and D / A conversion modules within the chip's internal circuitry increases circuit complexity, reduces the reliability of the switching power supply system and the loop response speed, and significantly increases cost.

[0004] With increasingly fierce market competition, it is of great significance to research a new type of resonant topology control technology that can be applied to a wide range of inputs and outputs and has a lower cost. Summary of the Invention

[0005] The technical problem to be solved by this invention is to provide a control method, control device and switching power supply for a resonant DC-DC converter. While retaining the advantage of fast loop response speed of charge-type resonant topology control, it eliminates the need to add input voltage feedforward compensation circuit. By using pulse width compensation, it solves the control problem of resonant topology in wide input and output range applications, thereby enabling wide application ranges of both high and low input voltages and wide output voltages. This results in smaller changes in overpower protection point and light load mode switching point under wide input and output voltages, reducing ripple and audio noise. Furthermore, it provides more flexible control, better compensation effect, and a simpler circuit with lower cost.

[0006] As a first aspect of the present invention, the embodiment of the resonant DC-DC converter control method is as follows:

[0007] A method for controlling a resonant DC-DC converter, used to generate a drive pulse width signal to control the high-side and low-side switching transistors of the resonant DC-DC converter, comprising:

[0008] The resonant sampling step obtains a first voltage signal characterizing the magnitude of the resonant cavity current of the resonant DC-DC converter;

[0009] The signal processing step converts the first voltage signal into a resonant charge signal that can be used for loop control;

[0010] The feedback acquisition step obtains a first feedback signal U, which characterizes the magnitude of the output voltage of the resonant DC-DC converter. FB ;

[0011] The first compensation step involves obtaining the operating period T, which characterizes the resonant DC-DC converter. SW The second voltage signal is used to adjust the first feedback signal U. FB The second feedback signal U is obtained by performing compensation. FB2 And the second feedback signal U FB2 Satisfying relation U FB2 =k×U FB ×T SW where k is a constant;

[0012] The pulse width generation step involves comparing the resonant charge signal with the second feedback signal and generating the driving pulse width signal based on the comparison result.

[0013] Furthermore, slope compensation is required in the signal processing steps to avoid the formation of resonant current large and small waves.

[0014] Preferably, the compensation amplitude of the slope compensation is 30-60% of the total resonant charge signal at maximum power.

[0015] Furthermore, the signal processing step increases the resonant charge signal through a second compensation step when the driving pulse width signal decreases; or the first compensation step decreases the second voltage signal through a third compensation step when the driving pulse width signal decreases.

[0016] Furthermore, the second compensation step and the third compensation step are two-segment oblique line compensations.

[0017] Preferably, the turning point of the two diagonal lines is located within a set range centered on the resonant frequency.

[0018] As a second aspect of the present invention, the technical solution of the provided resonant DC-DC converter control device is as follows:

[0019] A resonant DC-DC converter control device is used to generate a drive pulse width signal to control the high-side and low-side switching transistors of the resonant DC-DC converter, wherein the device includes:

[0020] A resonant sampling circuit is used to obtain a first voltage signal characterizing the magnitude of the resonant cavity current of the resonant DC-DC converter;

[0021] A signal processing circuit is used to convert the first voltage signal into a resonant charge signal that can be used for loop control;

[0022] The feedback acquisition circuit is used to acquire a first feedback signal U, which characterizes the magnitude of the output voltage of the resonant DC-DC converter. FB ;

[0023] The first compensation circuit is used to obtain the operating period T of the resonant DC-DC converter. SW The second voltage signal is used to adjust the first feedback signal U. FB The second feedback signal U is obtained by performing compensation. FB2 And the second feedback signal U FB2 Satisfying relation U FB2 =k×U FB ×T SW where k is a constant;

[0024] A pulse width generation circuit is used to compare the resonant charge signal with the second feedback signal and generate the driving pulse width signal based on the comparison result.

[0025] Preferably, the first compensation circuit includes:

[0026] The pulse width acquisition circuit is used to obtain the operating period T, which characterizes the resonant DC-DC converter. SW The second voltage signal;

[0027] A multiplier is used to convert the first feedback signal U FB The second voltage signal is used to perform calculations, thereby enabling the first feedback signal U to be processed using the second voltage signal. FB The second feedback signal U is obtained by performing compensation. FB2 .

[0028] Furthermore, the signal processing circuit further includes a second compensation circuit for increasing the resonant charge signal when the driving pulse width signal decreases; or the pulse width acquisition circuit further includes a third compensation circuit for decreasing the second voltage signal when the driving pulse width signal decreases.

[0029] Preferably, the first compensation circuit includes:

[0030] The pulse width acquisition circuit is used to obtain the operating period T, which characterizes the resonant DC-DC converter. SW The second voltage signal;

[0031] The feedback integrator circuit is used to integrate the first feedback signal U. FB The second voltage signal is used to perform calculations, thereby enabling the first feedback signal U to be processed using the second voltage signal. FB The second feedback signal U is obtained by performing compensation. FB2 .

[0032] Furthermore, the signal processing circuit further includes a second compensation circuit for increasing the resonant charge signal when the driving pulse width signal decreases; or the feedback integration circuit further includes a third compensation circuit for decreasing the second voltage signal when the driving pulse width signal decreases.

[0033] As a third aspect of the present invention, the technical solution of the provided switching power supply embodiment is as follows:

[0034] A switching power supply includes a resonant DC-DC converter, wherein: it further includes a resonant DC-DC converter control device as described in any of the second aspects above.

[0035] The advantages of this invention compared to the prior art are as follows:

[0036] 1. The technical solution of this invention includes a first compensation step / circuit to obtain a second voltage signal characterizing the operating cycle of the resonant DC-DC converter. The second voltage signal is used to compensate the first feedback signal to obtain a second feedback signal, and the second feedback signal satisfies the relationship U. FB2 =k×U FB ×T SW Where k is a constant, the influence of the duty cycle on the control loop is eliminated, so that the ratio of the average output current of the resonant DC-DC converter to the second feedback signal is a constant. The magnitude and change of the first feedback signal can be used to more accurately reflect the magnitude and change of the output load. That is, the control of the resonant topology with a wide range of applications can be realized by real-time compensation through the duty cycle / pulse width. Compared with the existing technology that uses input and output voltage feedforward compensation, the circuit is simpler.

[0037] 2. Compared with existing technical solutions on the market, the technical solution of this invention eliminates the working cycle T after introducing pulse width compensation. SWRegarding the impact on the control loop, the ratio of the average output current to the second feedback signal is a constant. The magnitude and change of the first feedback signal can be used to more accurately reflect the magnitude and change of the output load, and it is independent of the input voltage and output voltage. Therefore, in addition to compensating for the wide range of input high voltage and low voltage, it can also achieve wide range of output high voltage and low voltage compensation, making it more widely applicable. Attached Figure Description

[0038] Figure 1 is a first circuit block diagram of the resonant DC-DC converter control device according to a second embodiment of the present invention;

[0039] Figure 2 is a specific implementation circuit diagram of the resonant sampling circuit and signal processing circuit in Figure 1;

[0040] Figure 3 is a specific implementation circuit diagram of the feedback acquisition circuit in Figure 1;

[0041] Figure 4 is a specific implementation circuit diagram of the pulse width acquisition circuit in Figure 1;

[0042] Figure 5 is a specific implementation circuit diagram of the multiplier in Figure 1;

[0043] Figure 6 is a specific implementation circuit diagram of the pulse width generation circuit in Figure 1;

[0044] Figure 7 shows an example waveform of pulse width compensation implemented using the multiplier scheme in Figure 1.

[0045] Figure 8 is a specific implementation circuit diagram of adding a second compensation circuit to Figure 2;

[0046] Figure 9 is a specific implementation circuit diagram of adding a third compensation circuit to Figure 4;

[0047] Figure 10 is a second circuit block diagram of the resonant DC-DC converter control device according to the second embodiment of the present invention;

[0048] Figure 11 is a specific implementation circuit diagram of the pulse width compensation circuit in Figure 10;

[0049] Figure 12 is a specific implementation circuit diagram of the feedback integrator circuit in Figure 10;

[0050] Figure 13 shows an example waveform of pulse width compensation implemented using the feedback integration scheme in Figure 10;

[0051] Figure 14 is a specific implementation circuit diagram of adding a third compensation circuit to Figure 12. Detailed Implementation

[0052] It should be noted that, unless otherwise specified, the embodiments and features described in this application can be combined with each other.

[0053] To enable those skilled in the art to better understand the present application, the technical solutions in the embodiments of the present application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present application, and not all embodiments. Based on the embodiments in the present application, all other embodiments obtained by those of ordinary skill in the art without creative effort should fall within the scope of protection of the present application.

[0054] It should be noted that the terms "first," "second," etc., in the specification, claims, and accompanying drawings of this application are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence. It should be understood that such data can be used interchangeably where appropriate for the purposes of describing embodiments of this application herein. Furthermore, the terms "comprising" and "having," and any variations thereof, are intended to cover non-exclusive inclusion; for example, a process, method, system, product, or apparatus that comprises a series of steps or units is not necessarily limited to those steps or units explicitly listed, but may include other steps or units not explicitly listed or inherent to such processes, methods, products, or apparatus.

[0055] It should be understood that in the specification, claims, and drawings, when a step is described as continuing into another step, the step may directly continue into that other step or be continued into that other step through a third step; when an element / unit is described as "continuing" into another element / unit, the element / unit may be "directly connected" to that other element / unit or "connected" to that other element / unit through a third element / unit.

[0056] Furthermore, the accompanying drawings are merely illustrative of this disclosure and are not necessarily drawn to scale. The same reference numerals in the drawings denote the same or similar parts, and therefore repeated descriptions thereof will be omitted. Some block diagrams shown in the drawings are functional entities and do not necessarily correspond to physically or logically independent entities. These functional entities can be implemented in software, in one or more hardware modules or integrated circuits, or in different network and / or processor devices and / or microcontroller devices.

[0057] Traditional LLCs can only operate near the resonant point, have a narrow input-output range, and generally employ PFM control with a single output voltage loop, resulting in poor transient response and weak start-up capability. With technological advancements, the market demands increasingly wider input-output voltage ranges and faster response speeds. Resonant topologies have seen the emergence of single-cycle control techniques based on resonant current. Since the controlled object is the integral of the resonant current over a single cycle or half-cycle, it is also known as charge-mode control. Charge-mode control possesses unipolar transfer function characteristics, resulting in high loop bandwidth and simplified control loop compensation. However, it is computationally complex and requires slope compensation to achieve better unipolar characteristics.

[0058] Acquiring resonant current or voltage signals requires a high sampling rate. To improve system stability and simplify the sampling circuit, charge-type resonant controllers are typically placed near the primary side of the resonant circuit, while the feedback signal is usually on the secondary side. The output voltage is transmitted to the primary controller through the feedback circuit. When the switching converter operates under steady-state conditions, this feedback signal is proportional to the output power or output current, effectively reflecting the output load. However, when the input or output voltage changes, the resonant converter frequency changes, and the magnitude of the charge integral of the resonant current also changes, causing the comparison feedback signal to change accordingly. Therefore, it cannot reflect the output load. Furthermore, changes in transformer parameters can also significantly alter the feedback signal and the resonant current integral signal of the charge-type control.

[0059] In wide input or wide output applications, with high-voltage input or low-voltage output, the half-cycle resonant current exhibits a large negative current at the start of integration. This current becomes positive after rectification on the secondary side, causing the ratio of the primary-side resonant current integral to the secondary-side current to be nonlinear and smaller. Furthermore, the lower the gain of the resonant converter, the greater the deviation. With low-voltage input or high-voltage output, the half-cycle resonant current contains an excitation current before the integration ends. However, by this time, energy transfer from the primary to the secondary side has ceased, resulting in a non-zero integral of the half-cycle excitation current. This introduces an error and causes the ratio of the primary-side resonant current integral to the secondary-side current to be nonlinear and larger.

[0060] The wide-range resonant controller described in this paper significantly reduces the sensitivity of charge-type controllers to input and output voltage variations through pulse width compensation technology. Furthermore, a fine-tuning compensation circuit further compensates for the impact of errors generated by the resonant current during half-cycle integration in wide-input and wide-output applications on the control loop, optimizing the accuracy of the linear function relationship between the feedback voltage and the output load. First Embodiment

[0061] This embodiment provides a control method for a resonant DC-DC converter, used to generate a drive pulse width signal to control the high-side and low-side switching transistors of the resonant DC-DC converter, wherein the method includes:

[0062] The resonant sampling step obtains a first voltage signal characterizing the magnitude of the resonant cavity current of the resonant DC-DC converter;

[0063] The signal processing step converts the first voltage signal into a resonant charge signal that can be used for loop control;

[0064] The feedback acquisition step obtains the first feedback signal U, which characterizes the magnitude of the output voltage of the resonant DC-DC converter. FB ;

[0065] The first compensation step obtains the operating period T, which characterizes the resonant DC-DC converter. SW The second voltage signal is used to adjust the first feedback signal U. FB The second feedback signal U is obtained by performing compensation. FB2 And the second feedback signal U FB2 Satisfying relation U FB2 =k×U FB ×T SW where k is a constant;

[0066] The pulse width generation step involves comparing the resonant charge signal with the second feedback signal and generating a driving pulse width signal based on the comparison result.

[0067] This embodiment uses U FB2 Replace U in existing technology FB The difference between it and the resonant charge signal (hereinafter referred to as CS) rather than U FB The difference between the signal and CS is used to generate the drive pulse width signal. This is due to the second feedback signal U. FB2 It uses the second voltage signal to control the first feedback signal U. FB After compensation, the second feedback signal U is obtained. FB2 Satisfying relation U FB2 =k×U FB ×T SW Where k is a constant, the influence of the duty cycle on the control loop is eliminated, making the ratio of the average current output by the resonant DC-DC converter to the second feedback signal a constant value, thus ensuring the second feedback signal U... FB2 The size and changes can be used to more accurately reflect the size and changes of the output load.

[0068] In this embodiment, k is set according to the system design goals and is essentially used to calibrate load requirements (and U). FB2 The corresponding ratio between the relevant (related) and the total energy transfer (related to CS).

[0069] Furthermore, in this embodiment, the impact of input and output voltage changes on the loop is greatly reduced. The jump in input voltage will hardly cause changes in output voltage ripple. The first feedback signal is stable under different input and output voltages and only reflects the output load size. This makes the threshold for the resonant DC-DC converter to enter the light load control mode consistent, and achieves consistent output ripple size in the light no-load control mode when used in a wide range of applications. This effectively reduces power consumption, improves efficiency, limits ripple size, and reduces audio noise.

[0070] As a specific implementation, slope compensation is performed during the signal processing step to avoid the formation of resonant current amplitude fluctuations. Preferably, the compensation amplitude of the slope compensation is 30-60% of the total resonant charge signal at maximum power.

[0071] Furthermore, the signal processing step increases the resonant charge signal through a second compensation step when the driving pulse width signal decreases, or the first compensation step decreases the second voltage signal through a third compensation step when the driving pulse width signal decreases, further compensating for the problem of large resonant topology charge-type control error when the operating frequency deviates far from the resonant frequency point, and improving the stability and parameter consistency of load change control such as light load mode switching and overpower protection.

[0072] Furthermore, the second and third compensation steps are two-segment diagonal compensations. This is because the error in charge control of resonant topologies comes from the excitation and magnetizing current when the operating frequency is higher than the resonant frequency, and from the excitation current only when the operating frequency is lower than the resonant frequency. Therefore, segmented fine-tuning compensation can improve the consistency of compensation near the upper and lower frequency limits when used over a wide range of applications. Preferably, the inflection point of the two diagonal lines is located within a set range centered on the resonant frequency. Second Embodiment

[0073] This embodiment provides a resonant DC-DC converter control device for generating drive pulse width signals to control the high-side and low-side switching transistors of the resonant DC-DC converter, comprising:

[0074] A resonant sampling circuit is used to obtain a first voltage signal characterizing the magnitude of the resonant cavity current of the resonant DC-DC converter.

[0075] A signal processing circuit is used to convert the first voltage signal into a resonant charge signal that can be used for loop control;

[0076] The feedback acquisition circuit is used to acquire the first feedback signal U, which characterizes the magnitude of the output voltage of the resonant DC-DC converter. FB ;

[0077] The first compensation circuit is used to obtain the operating period T of the resonant DC-DC converter. SWThe second voltage signal is used to adjust the first feedback signal U. FB The second feedback signal U is obtained by performing compensation. FB2 And the second feedback signal U FB2 Satisfying relation U FB2 =k×U FB ×T SW where k is a constant;

[0078] The pulse width generation circuit is used to compare the resonant charge signal with the second feedback signal and generate the driving pulse width signal based on the comparison result.

[0079] Figure 1 is a first circuit block diagram of the resonant DC-DC converter control device according to the second embodiment of the present invention. A multiplier scheme is used to achieve pulse width compensation. The resonant converter includes a switching circuit and a control circuit. The switching circuit includes a voltage source 101, a high-side switching transistor 102, a low-side switching transistor 103, a resonant inductor 104, a resonant capacitor 106, a transformer 107, rectifier diodes 108 and 109, and an output capacitor 110. The control circuit includes a resonant sampling circuit 1001, a signal processing circuit 1002, a feedback acquisition circuit 1003, a pulse width acquisition circuit 1004, a multiplier 1005, and a pulse width generation circuit 1006. When the switching circuit is working, the resonant current signal can be acquired in the series resonant circuit composed of the resonant inductor 104, the primary winding 105 of the transformer 107, and the resonant capacitor 106. Since the resonant current will definitely pass through the resonant capacitor 106, the voltage across the resonant capacitor 106 can also be directly acquired as the integrated resonant current signal. The acquired resonant current or voltage signal serves as the input to the resonant sampling circuit 1001. After passing through the signal processing circuit 1002, it is converted into a resonant charge signal that can be used for loop control and fed into the pulse width generation circuit 1006. The output voltage is fed into the multiplier input through the feedback acquisition circuit 1003 and multiplied with the output of the pulse width acquisition circuit 1004 before being fed into the pulse width generation circuit 1006. The resonant charge signal and the feedback signal processed by the multiplier are compared in the pulse width generation circuit 1006, and together with other logic and enable signals, a high-side and low-side switching transistor drive pulse width signal is generated through a flip-flop.

[0080] Compared to conventional control circuits, the resonant DC-DC converter control device in Figure 1 replaces the input voltage acquisition and feedforward compensation circuits with the addition of a pulse width acquisition circuit 1004 and a multiplier 1005, which can simultaneously optimize the control loop's sensitivity to changes in input and output voltages.

[0081] Figure 2 shows a specific implementation circuit diagram of the resonant sampling circuit and signal processing circuit in Figure 1. Figure 3 shows a specific implementation circuit diagram of the feedback acquisition circuit in Figure 1. Figure 4 shows a specific implementation circuit diagram of the pulse width acquisition circuit in Figure 1. The resonant sampling circuit 1001 and signal processing circuit 1002 are half-cycle resonant current sampling and integration schemes, which are conventional circuits and can also be implemented using a transconductance circuit. Additionally, the voltage change across the resonant capacitor 106 is acquired, eliminating the integration stage; this is also a conventional circuit and is not shown in this diagram. The feedback acquisition circuit 1003 is a conventional current-type feedback acquisition circuit; it can also be implemented using a voltage-type feedback acquisition circuit, achieving the same effect, and is therefore not shown. The pulse width generation circuit 1006 consists of comparators 601 and 603, and triggers 602 and 604, and is a conventional circuit.

[0082] When the LLC resonant converter operates near the resonant point, the integral of the excitation current is approximately zero, and we have:

[0083] ∫I r =k1×U FB2 (1)

[0084] In the above formula, I r U is the resonant current, k1 is a coefficient related to the switching circuit parameters, and U FB2 It is the feedback signal output by the multiplier.

[0085] When the output load current Io of the switching converter changes, the primary resonant current Ir also changes. During integration, the excitation current Im is approximately zero, thus ∫Io=N PS ∫Ir, N PS Given the ratio of the number of turns in the primary and secondary windings of the transformer, and combining with equation (1), we have:

[0086] (2)

[0087] T SW is the operating period of the resonant converter, and k2 is a coefficient related to the switching circuit parameters.

[0088] The multiplier 1005 has inputs FB and VT2, and outputs FB2.

[0089] (3)

[0090] In the above formula, U FB It is the feedback acquisition signal voltage, UVT2 It is the voltage of the pulse width acquisition signal.

[0091] From the pulse width acquisition circuit 1004, we can see that:

[0092] (4)

[0093] Substituting equation (4) into equation (3), we get:

[0094] (5)

[0095] In the above formula, I 405 It is the current of the constant current source 405 in the pulse width acquisition circuit, C 407 K is the capacitance value of capacitor 407 in the pulse width acquisition circuit, and k3 is a coefficient related to the circuit parameters.

[0096] From (2) and (5), we can obtain:

[0097] (6)

[0098] As can be seen from equation (5) above, after introducing pulse width compensation through the multiplier, the working cycle T is eliminated. SW Regarding the impact on the control loop, the ratio of the average output current to the second feedback signal is a constant, while the magnitude and changes of the first feedback signal can be used to more accurately reflect the magnitude and changes of the output load.

[0099] When the operating frequency is significantly lower than the resonant frequency, the resonant current may be low before the end of the half-cycle. In this case, the integral change of the resonant current is very slow, resulting in a small angle between CS and FB2. This leads to a large difference in pulse width between the high-side and low-side drives, forming large and small waves in the resonant current. This increases system losses, reduces efficiency, and in severe cases, can cause magnetic components to saturate and fail. Therefore, a current source 208 needs to be added to the signal processing circuit to inject current into the integrating capacitor 211, forming a slope compensation. The compensation amplitude is recommended to be 30-60% of the total resonant charge signal at maximum power.

[0100] Figure 7 shows an example waveform of pulse width compensation implemented using a multiplier in Figure 1. The working principle of Figure 1 is analyzed below, combining the pulse width acquisition example circuit in Figure 4 and the multiplier example circuit in Figure 5:

[0101] The high-side and low-side pulse width signals GHS and GLS are input to the output of NOR gate 401 to drive the dead-time signal VG_DH, which is then fed to delay units 402, 404 and AND gate 403, respectively. The delay time of 402 is t1, and the delay time of 404 is t2. t2 should be slightly greater than t1, and as small as possible while ensuring that the reset and sample-and-hold functions are normal. It should be less than the dead time. It is recommended that t1 = 20~30 ns and t2 = 30~50 ns. If the VTG2 time exceeds the VG_DH time, an AND gate can be added to the output of 404. The input is the positive output of 404 and VG_DH, and the positive output is the original VTG2 signal. The VTG1 pulse waveform is shown in Figure 3. It goes high on the falling edge of GLS or GHS to control switch 409 in the sample-and-hold circuit, allowing sample-and-hold capacitor 410 to obtain the pulse width amplitude signal converted from half a cycle time. That is, VT is the product of the operating period of the resonant DC-DC converter and a constant. After a delay of t1, VTG1 goes low and remains low for about half a cycle, keeping switch 409 closed and ensuring that the voltage on capacitor 410 remains stable during the half cycle. The VTG2 signal goes high after a further delay of t2 on the falling edge of GLS or GHS, and goes low on the rising edge of GLS or GHS. It is used to control switch 406 to conduct, discharging capacitor 407 and resetting its voltage. Voltage source 412, comparator 411, switches 413 and 414 form a minimum value limiting circuit, converting VT to VT2 to prevent the loop from failing due to excessively small VT during startup or protection restart. VT2 is also the product of the operating period of the resonant DC-DC converter and a constant. As can be seen from Figure 3, the range of VT2 is generally 0.3~1.5V, and it varies when VTG1 is high, and remains constant at other times. VT2 is multiplied by FB to obtain FB2. FB remains basically unchanged during half a cycle. Therefore, the waveform of the multiplier output FB2 after multiplication is similar to that of VT2, but the voltage amplitude is different, and it satisfies the relationship of the above formula (5), which can achieve the purpose of the invention.

[0102] Figure 8 shows a specific implementation circuit diagram of adding a second compensation circuit 1011 to Figure 2. Referring to Figure 8, the second compensation circuit can increase the resonant charge signal when the driving pulse width signal decreases. Specifically, VF is an adjustable constant voltage source, and VT2 is the output of the pulse width acquisition circuit. As the pulse width changes, the higher the operating frequency of the resonant circuit, the smaller CS will be. The smaller the pulse width, the smaller VT2 will be. The difference between VF and VT2 will be greater, and the positive compensation current for capacitor 211 will be greater, thereby compensating CS and reducing its error influence. By fine-tuning the voltage of VF, more accurate compensation can be achieved. When the operating frequency deviates far from the resonant frequency point, the addition of the second compensation circuit 1011 in the signal processing circuit further compensates for the large error of the resonant topology charge-type control, improving the stability and parameter consistency of load change control such as light load mode switching and overpower protection.

[0103] Figure 9 shows a specific implementation circuit diagram of adding a third compensation circuit 1012 to Figure 4. The third compensation circuit 1012 can reduce the second voltage signal when the driving pulse width signal decreases. Specifically, VF is an adjustable constant voltage source, and VT2 is the output of the pulse width acquisition circuit. As the pulse width changes, the higher the operating frequency of the resonant circuit, the smaller CS will be. The smaller the pulse width, the smaller VT2 will be. The difference between VF and VT2 will be greater, and the negative compensation current of capacitor 407 will be greater, thereby reducing the size of VT2. This keeps the ratio of CS to VT2 relatively small. As we know from the multiplier, FB multiplied by VT2 equals FB2. Therefore, the compensation reduces the influence of CS on FB. By fine-tuning the voltage of VF, more accurate compensation can be achieved.

[0104] Furthermore, since the magnitude of the error caused by the same pulse width time variation differs between frequencies above and below the resonant frequency, the circuits shown in Figures 8 and 9 cannot perfectly compensate for the error. The compensation circuit function can be transformed from a single slant line to two slant lines, with the inflection point located near the resonant frequency. Besides the example circuits 1011 and 1012 shown in Figures 8 and 9, similar functionality can be achieved through combinations of transistors and other components.

[0105] Figure 10 is a second circuit block diagram of the resonant DC-DC converter control device according to the second embodiment of the present invention. It employs a feedback integration scheme to achieve pulse width compensation. The difference from Figure 1 is that the pulse width acquisition circuit 1004 is replaced by a pulse width compensation circuit 1007, and the multiplier 1005 is replaced by a feedback integration circuit 1008. Compared to conventional control circuits, the resonant DC-DC converter control device in Figure 1 replaces the input voltage acquisition and feedforward compensation circuits by adding a pulse width compensation circuit 1007 and a feedback integration circuit 1008, thus simultaneously optimizing the control loop's sensitivity to changes in input and output voltages.

[0106] The feedback integrator circuit 1008 has input FB and output FB2, and has the following properties:

[0107] (7)

[0108] In the above formula, U FB It is the feedback acquisition signal voltage, T SW It is the working switching cycle, U FB2 It is the output signal voltage of the feedback integrator circuit, k f These are coefficients related to the circuit parameters.

[0109] From equations (2) and (7), we can obtain:

[0110] (8)

[0111] As can be seen from equation (8) above, after introducing pulse width compensation through the feedback integrator circuit, the working period T is eliminated. SW Regarding the impact on the control loop, the ratio of the average output current to the second feedback signal is a constant, while the magnitude and changes of the first feedback signal can be used to more accurately reflect the magnitude and changes of the output load.

[0112] Figure 11 is a specific implementation circuit diagram of the pulse width compensation circuit in Figure 10, Figure 12 is a specific implementation circuit diagram of the feedback integration circuit in Figure 10, and Figure 13 is an example waveform of pulse width compensation implemented by the feedback integration scheme in Figure 10.

[0113] The working principle of the pulse width compensation circuit in Figure 11 and the feedback integration circuit in Figure 12 to achieve the purpose of the invention is the same as that of the pulse width acquisition circuit in Figure 4 and the multiplier circuit in Figure 5, so it will not be described in detail.

[0114] As shown in Figure 1, a second compensation circuit can be added to the signal processing circuit or a third compensation circuit can be added to the pulse width acquisition circuit to further compensate for the large error of the resonant topology charge-type control when the operating frequency deviates far from the resonant frequency point. Similarly, Figure 10 can achieve the same invention purpose by adding a second compensation circuit to the signal processing circuit or a third compensation circuit to the feedback integration circuit. The addition of a second compensation circuit to the signal processing circuit in Figure 10 is the same as that in Figure 1, so it will not be described in detail. Figure 14 provides a specific implementation circuit diagram of adding a third compensation circuit 1014 in Figure 12. Please refer to Figure 14. 1014 is located in the feedback integration circuit. VF is an adjustable constant voltage source, and VT2 is the output of the pulse width compensation circuit. As the pulse width changes, the higher the operating frequency of the resonant circuit, the smaller CS will be. The smaller the pulse width, the smaller VT2 will be. The difference between VF and VT2 will be larger, and the negative compensation current of capacitor 803 will be larger, thereby reducing the size of FB2. This keeps the ratio of CS to FB2 relatively small. By fine-tuning the voltage of VF, more accurate compensation can be achieved. Third Embodiment

[0115] This embodiment provides a switching power supply, including a resonant DC-DC converter, wherein: it further includes the resonant DC-DC converter control device described in any one of the second embodiments.

[0116] The switching power supply in this embodiment includes the resonant DC-DC converter control device described in any of the second embodiments. While retaining the advantage of fast loop response speed of charge-type resonant topology control, it not only solves the adverse effects of input and output voltage changes on the control loop and greatly reduces output voltage ripple, but also optimizes the changes in feedback signal when used in ultra-wide input and output range applications through pulse width compensation. This makes the feedback signal more accurately reflect the output load size, thereby reducing the changes in overpower protection point and light load mode switching point under wide input and output voltages, and reducing ripple and audio noise.

[0117] The above are merely preferred embodiments of the present invention. It should be noted that the above preferred embodiments should not be considered as limitations on the present invention. For those skilled in the art, several equivalent substitutions, improvements, and modifications can be made without departing from the spirit and scope of the present invention. These equivalent substitutions, improvements, and modifications should also be considered within the protection scope of the present invention. Further details will not be provided here, and the protection scope of the present invention should be determined by the scope defined in the claims.

Claims

1. A control method for a resonant DC-DC converter, used to generate a drive pulse width signal to control the high-side and low-side switching transistors of the resonant DC-DC converter, characterized in that, include: The resonant sampling step obtains a first voltage signal characterizing the magnitude of the resonant cavity current of the resonant DC-DC converter; The signal processing step converts the first voltage signal into a resonant charge signal that can be used for loop control; a feedback collection step of acquiring a first feedback signal U representing the magnitude of the output voltage of the resonant DC-DC converter FB ; a first compensation step, obtaining a second voltage signal representative of a duty cycle T of said resonant DC-DC converter SW a second compensation step, compensating said first feedback signal U FB with said second voltage signal obtaining a second feedback signal U FB2 , and said second feedback signal U FB2 satisfies the relation U FB2 =k x U FB x T SW where k is a constant; The pulse width generation step involves comparing the resonant charge signal with the second feedback signal and generating the driving pulse width signal based on the comparison result.

2. The control method according to claim 1, characterized in that: The signal processing steps require slope compensation to avoid the formation of resonant current waves of varying magnitudes.

3. The control method according to claim 2, characterized in that: The compensation range of the slope compensation is 30-60% of the total resonant charge signal at maximum power.

4. The control method according to claim 1, characterized in that: The signal processing step increases the resonant charge signal through a second compensation step when the driving pulse width signal decreases; Alternatively, the first compensation step may reduce the second voltage signal by a third compensation step when the driving pulse width signal decreases.

5. The control method according to claim 4, characterized in that: The second compensation step and the third compensation step are two-segment oblique line compensation.

6. The control method according to claim 4, characterized in that: The turning points of the two diagonal lines are located within a set range centered on the resonant frequency.

7. A resonant DC-DC converter control device, used to generate a drive pulse width signal to control the high-side switch and the low-side switch of the resonant DC-DC converter, characterized in that, include: A resonant sampling circuit is used to obtain a first voltage signal characterizing the magnitude of the resonant cavity current of the resonant DC-DC converter; A signal processing circuit is used to convert the first voltage signal into a resonant charge signal that can be used for loop control; The feedback acquisition circuit is used to acquire a first feedback signal U, which characterizes the magnitude of the output voltage of the resonant DC-DC converter. FB ; The first compensation circuit is used to obtain the operating period T of the resonant DC-DC converter. SW The second voltage signal is used to adjust the first feedback signal U. FB The second feedback signal U is obtained by performing compensation. FB2 And the second feedback signal U FB2 Satisfying relation U FB2 =k×U FB ×T SW where k is a constant; A pulse width generation circuit is used to compare the resonant charge signal with the second feedback signal and generate the driving pulse width signal based on the comparison result.

8. The resonant DC-DC converter control device according to claim 7, characterized in that, The first compensation circuit includes: The pulse width acquisition circuit is used to obtain the operating period T, which characterizes the resonant DC-DC converter. SW The second voltage signal; A multiplier is used to convert the first feedback signal U FB The second voltage signal is used to perform calculations, thereby enabling the first feedback signal U to be processed using the second voltage signal. FB The second feedback signal U is obtained by performing compensation. FB2 .

9. The resonant DC-DC converter control device according to claim 8, characterized in that: The signal processing circuit further includes a second compensation circuit, used to increase the resonant charge signal when the driving pulse width signal decreases; Alternatively, the pulse width acquisition circuit may further include a third compensation circuit, which reduces the second voltage signal when the driving pulse width signal decreases.

10. The resonant DC-DC converter control device according to claim 7, characterized in that, The first compensation circuit includes: The pulse width acquisition circuit is used to obtain the operating period T, which characterizes the resonant DC-DC converter. SW The second voltage signal; The feedback integrator circuit is used to integrate the first feedback signal U. FB The second voltage signal is used to perform calculations, thereby enabling the first feedback signal U to be processed using the second voltage signal. FB The second feedback signal U is obtained by performing compensation. FB2 .

11. The resonant DC-DC converter control device according to claim 10, characterized in that: The signal processing circuit further includes a second compensation circuit, used to increase the resonant charge signal when the driving pulse width signal decreases; Alternatively, the feedback integration circuit may further include a third compensation circuit that reduces the second voltage signal when the driving pulse width signal decreases.

12. A switching power supply, comprising a resonant DC-DC converter, characterized in that: It also includes the resonant DC-DC converter control device according to any one of claims 7 to 11.

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