Power conversion apparatus and method for estimating resonant frequency of power conversion apparatus

Abstract

[Problem] To provide a power conversion apparatus capable of accurately estimating a resonant frequency of an input filter online in real time. [Solution] A power conversion apparatus 1 generates an AC output from a DC power source 29 by means of an inverter 28, and applies the AC output to a motor 8. The power conversion apparatus comprises: a filter 40 that is connected to an input; a current sensor 45 that detects a filter resonant current flowing through a smoothing capacitor 32 constituting the filter 40; and a control device 21. The control device 21 has a resonant frequency estimation unit 50 for estimating a resonant frequency, which is the frequency of the filter resonant current, on the basis of the filter resonant current detected by the current sensor 45.

Description

Power converter and method for estimating the resonant frequency of a power converter

[0001] The present invention relates to a power conversion device that generates AC output from a DC power source using an inverter and applies it to a motor, and a method for estimating the resonant frequency of the power conversion device.

[0002] Conventionally, power conversion devices for driving motors built into electric compressors of vehicle air conditioning systems have configured a three-phase inverter using upper and lower arm switching elements for each phase (UVW), and controlled the switching elements of each phase using PWM (Pulse Width Modulation) to generate a voltage waveform close to a sine wave (three-phase AC output) from a DC power source (battery), which is then applied to the motor for driving.

[0003] In such power conversion devices, a smoothing capacitor is connected in parallel with the DC power supply. This smoothing capacitor is used to smooth the current, which has a large distortion component due to the inverter's PWM, and return it to the DC power supply. For this purpose, a capacitor with a relatively large capacity is required.

[0004] However, increasing the capacitance of a capacitor also increases its volume, leading to a larger electric compressor itself. Therefore, a small (low-capacity) smoothing capacitor is desirable. However, small capacitors can be damaged if the ripple current is large, as losses can cause a temperature rise.

[0005] Furthermore, there is inductance in the wiring between the DC power supply and the smoothing capacitor, and in addition, a normal mode choke coil is connected separately. These inductances and the smoothing capacitor mentioned above constitute an input filter to reduce the influence of conducted noise from other devices connected to the DC power supply and the influence of conducted noise returning to the DC power supply from the power converter (see, for example, Patent Document 1).

[0006] In such a smoothing capacitor, the high-frequency components of the inverter's generated high-frequency current that are higher than the resonant frequency of the input filter flow into the smoothing capacitor, while the components that are lower than the resonant frequency flow out to the DC power supply side.

[0007] Furthermore, the high-frequency current generated by the inverter should not include the resonant frequency of the input filter. This is because including the resonant frequency of the filter can cause resonance, leading to an increase in the current flowing through the smoothing capacitor (capacitor current), which can cause overvoltage, or the ripple current can cause the smoothing capacitor to overheat.

[0008] Therefore, for example, if the carrier frequency (carrier wave frequency) of the inverter is set to 20 kHz, the resonant frequency of the input filter will be designed to be 14 kHz or similar (design value), where the difference from the carrier frequency is sufficiently large.

[0009] On the other hand, the current flowing between the smoothing capacitor and the inverter, i.e., the DC link current I, dc The frequency components of the signal include multiple sidebands, centered around the carrier frequency, whose frequencies change depending on the motor's rotation speed (rotation frequency). When the motor's rotation speed increases, these sidebands reach the resonant frequency of the input filter. When the sidebands reach the resonant frequency, the ripple current of the smoothing capacitor increases sharply, leading to a problem of the smoothing capacitor's temperature rising.

[0010] Therefore, in the past, when the resonant frequency was high, such as by miniaturizing the smoothing capacitor that constitutes the filter, a resonance suppression operation was sometimes performed to suppress the temperature rise of the smoothing capacitor, for example by adjusting the switching timing of the inverter so that the sidebands do not reach the resonant frequency, thereby reducing the ripple of the current flowing through the smoothing capacitor (capacitor current).

[0011] Japanese Patent Publication No. 7157412, Japanese Unexamined Patent Publication No. 2024-104509

[0012] However, the resonant frequency of the input filter described above will deviate from the design value (for example, 14 kHz) due to parasitic components in the circuit board, parasitic components in the harness connecting the circuit board to the DC power supply, and aging degradation of passive elements such as smoothing capacitors and normal-mode choke coils. In other words, the difference between the pre-designed resonant frequency and the actual resonant frequency will degrade the effectiveness of the resonance suppression operation.

[0013] Therefore, conventional methods have been proposed that include means for detecting the resonant component contained in the line voltage of the power grid, but these have the problem of low accuracy (see, for example, Patent Document 2).

[0014] The present invention was made to solve the aforementioned conventional technical problems, and aims to provide a power converter that can accurately estimate the resonant frequency of an input filter online in real time, and a method for estimating the resonant frequency of a power converter.

[0015] The power conversion device of the first invention generates an AC output from a DC power source using an inverter and applies it to a motor, and comprises a filter connected to the input, a current detection unit that detects the filter resonant current flowing through a smoothing capacitor constituting the filter, and a control device, wherein the control device has a resonant frequency estimation unit that estimates the resonant frequency, which is the frequency of the filter resonant current, from the filter resonant current detected by the current detection unit.

[0016] The power conversion device of the second invention is characterized in that, in the above invention, the resonant frequency estimation unit estimates the resonant frequency from the time change of the filter resonant current detected by the current detection unit.

[0017] The power conversion device of the third invention is characterized in that, in the above invention, the resonant frequency estimation unit derives a phase change from the time change of the filter resonant current detected by the current detection unit and estimates the resonant frequency.

[0018] The power conversion device of the fourth invention is such that the resonant frequency estimation unit in the above invention determines the t-th sampling value i of the filter resonant current sampled at a predetermined sampling period Δt. f [t] and the (t+1)th sampling value i f [t+1] and the peak value i of the filter resonant current pp Based on this, the t-th sampling value i is obtained using the inverse trigonometric function. f The phase θ[t] of [t] and the (t+1)th sampling value i f The method is characterized by determining the phase θ[t+1] at [t+1], and then calculating the resonant frequency from the phase difference Δθ (the difference between these phases) and the sampling period Δt.

[0019] The power conversion device of the fifth invention is characterized in that, in the above invention, the resonance frequency estimation unit corrects the phase difference Δθ based on the domain of definition of the inverse trigonometric function.

[0020] The power conversion device of the sixth invention is characterized in that, in the above invention, the resonance frequency estimation unit uses the symmetry of the waveform to determine the symmetric phase of the phase θ[t] of the t-th sampling value i f [t] outside the domain of definition of the inverse trigonometric function, and the symmetric phase of the phase θ[t + 1] of the (t + 1)-th sampling value i f [t + 1], and determines the phase difference Δθ in the range of 0 to 2π including each derived symmetric phase.

[0021] The power conversion device of the seventh invention is characterized in that, in the above invention, the resonance frequency estimation unit adopts the phase difference Δθ at which the calculated resonance frequency becomes a value close to the design value, and determines the resonance frequency calculated from the adopted phase difference Δθ as the estimated value.

[0022] The power conversion device of the eighth invention is characterized in that, in the above invention, the resonance frequency estimation unit updates the resonance frequency determined as the estimated value as the design value.

[0023] The power conversion device of the ninth invention is characterized in that, in the fourth invention, the resonance frequency estimation unit determines the peak-to-peak value i pp from the sampling values of the filter resonance current in a section sufficiently longer than the resonance frequency.

[0024] The power conversion device of the tenth invention is characterized in that, in the first invention, the resonance frequency estimation unit estimates the resonance frequency in the operating state of the inverter in which the current flowing through the smoothing capacitor constituting the filter is only the filter resonance current.

[0025] The power conversion device of the eleventh invention is characterized in that, in the above invention, the control device flows a current through the inverter in a state where the motor is stopped, and then stops the operation of the inverter, so that only the filter resonance current flows through the smoothing capacitor to form a free oscillation state, and in that state, the resonance frequency estimation unit estimates the resonance frequency.

[0026] The power conversion device of the twelfth invention is characterized in that, in the first invention, the control device further includes a resonance suppression unit that performs a predetermined resonance suppression operation based on the resonance frequency estimated by the resonance frequency estimation unit.

[0027] The power conversion device of the 13th invention is characterized in that, in the above invention, the resonance suppression unit adjusts the switching timing of the inverter so that sidebands present around the carrier frequency do not reach the resonance frequency.

[0028] The fourteenth invention provides a method for estimating the resonant frequency of a power converter, characterized in that, when estimating the resonant frequency of a filter connected to the input of the power converter, the method involves detecting the filter resonant current flowing through a smoothing capacitor, deriving the phase change from the time change of the detected filter resonant current, and estimating the resonant frequency, which is the frequency of the filter resonant current.

[0029] The 15th invention: Method for estimating the resonant frequency of a power converter. In the above invention, the t-th sampling value of the filter resonant current i is obtained at a predetermined sampling period Δt. f [t] and the (t+1)th sampling value i f [t+1] is sampled, and the peak value i of the filter resonant current is also sampled. pp Determine this peak-peak value i pp And the t-th sampling value i f [t], and the (t+1)th sampling value i f Based on [t+1], the t-th sampling value i is obtained using the inverse trigonometric function. f The phase θ[t] of [t] and the (t+1)th sampling value i f The method is characterized by determining the phase θ[t+1] at [t+1], and then calculating the resonant frequency from the phase difference Δθ (the difference between these phases) and the sampling period Δt.

[0030] According to the present invention, in a power conversion device that generates AC output from a DC power source using an inverter and applies it to a motor, the device includes a filter connected to the input, a current detection unit that detects the filter resonant current flowing through a smoothing capacitor constituting the filter, and a control device. The control device has a resonant frequency estimation unit that estimates the resonant frequency, which is the frequency of the filter resonant current, from the filter resonant current detected by the current detection unit. For example, as in the second, third, and fourteenth inventions, the resonant frequency can be estimated by deriving the phase change from the time change of the filter resonant current detected by the current detection unit and estimating the resonant frequency, thereby enabling accurate, online, and real-time estimation of the resonant frequency of the input filter.

[0031] This makes it possible to accurately perform resonance suppression operations by the resonance suppression unit, as in the 12th and 13th inventions, and to always maintain a high level of resonance suppression effect.

[0032] In this case, as in the fourth and fifteenth inventions, the resonance frequency estimation unit actually determines the t-th sampling value i of the filter resonance current sampled with a predetermined sampling period Δt. f [t] and the (t+1)th sampling value i f [t+1] and the peak value i of the filter resonant current pp Based on this, the t-th sampling value i is obtained using the inverse trigonometric function. f The phase θ[t] of [t] and the (t+1)th sampling value i f By determining the phase θ[t+1] at [t+1], and calculating the resonant frequency from the phase difference Δθ and the sampling period Δt, it becomes possible to estimate the resonant frequency with high accuracy.

[0033] Furthermore, as in the fifth invention, the resonance frequency estimation unit can estimate the resonance frequency with greater accuracy by correcting the phase difference Δθ based on the domain of the inverse trigonometric function.

[0034] In that case, for example, as in the sixth invention, the resonance frequency estimation unit uses the waveform symmetry to determine the t-th sampling value i that lies outside the domain of the inverse trigonometric function. f The phase θ of [t] and the symmetric phase of [t], and the (t+1)th sampling value if By deriving the phase θ[t+1] at [t+1], determining the phase difference Δθ in the range of 0 to 2π, including each derived symmetric phase, and further adopting the phase difference Δθ such that the calculated resonant frequency, as in the seventh invention, is close to the design value, determining the resonant frequency calculated from the adopted phase difference Δθ as the estimated value, and updating the resonant frequency determined as the estimated value as in the eighth invention, it becomes possible to continue to estimate the resonant frequency with high accuracy.

[0035] Furthermore, as in the ninth invention, the resonance frequency estimation unit obtains the peak-to-peak value i from the sampling value of the filter resonance current over a sufficiently long interval beyond the resonance frequency. pp By determining this, we can obtain a more accurate peak-to-peak value i pp This allows for the determination of the resonant frequency and enables accurate estimation of the resonant frequency.

[0036] Furthermore, as in the tenth invention, by having the resonance frequency estimation unit estimate the resonance frequency in the operating state of the inverter where the current flowing through the smoothing capacitor constituting the filter is only the filter resonance current, it becomes possible to estimate the resonance frequency from the pure filter resonance current, which has been removed from non-resonant components such as carrier components.

[0037] In that case, for example, as in the 11th invention, if the control device supplies current to the inverter with the motor stopped, and then stops the operation of the inverter, thereby creating a free-vibration state where only the filter resonant current flows through the smoothing capacitor, and the resonant frequency estimation unit estimates the resonant frequency in that state, then it becomes possible to estimate the resonant frequency from the pure filter resonant current from which components other than resonance have been removed.

[0038] This is an electrical circuit diagram of a power converter according to one embodiment of the present invention. This diagram illustrates the filter resonant current of the power converter shown in Figure 1. This diagram illustrates the operating state of the inverter when the resonant frequency estimation unit of Figure 1 estimates the resonant frequency. This diagram illustrates the detection operation of the peak-to-peak value of the resonant current by the resonant frequency estimation unit of Figure 1. This diagram illustrates the estimation operation of the resonant frequency by the resonant frequency estimation unit of Figure 1. This diagram illustrates the phase difference correction operation by the resonant frequency estimation unit of Figure 1. This diagram also illustrates the phase difference correction operation by the resonant frequency estimation unit of Figure 1. This diagram illustrates the method of determining the phase difference by the resonant frequency estimation unit of Figure 1. This diagram illustrates the estimation result of the resonant frequency by the resonant frequency estimation unit of Figure 1.

[0039] Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. One embodiment of the present invention, the power conversion device 1, drives the motor 8 of a so-called inverter-integrated electric compressor that constitutes the refrigerant circuit of a vehicle air conditioning system mounted on a vehicle such as an electric vehicle.

[0040] (1) Power Conversion Device 1 In Figure 1, the power conversion device 1 of this embodiment includes a three-phase inverter 28 and a control device 21. The inverter 28 is a circuit that converts the DC voltage of a DC power source (vehicle battery: for example, 350V) 29 into a three-phase AC voltage (three-phase AC output) and applies it to the motor 8. In this case, the motor 8 of the embodiment is an IPMSM (Internal Permanent Magnet Synchronous Motor).

[0041] The inverter 28 of this embodiment has a U-phase half-bridge circuit 19U, a V-phase half-bridge circuit 19V, and a W-phase half-bridge circuit 19W, and each phase half-bridge circuit 19U to 19W has upper arm switching elements 18A to 18C and lower arm switching elements 18D to 18F individually.

[0042] Furthermore, each switching element 18A to 18F has a flywheel diode 31 connected in antiparallel. In this embodiment, each upper and lower arm switching element 18A to 18F is composed of an insulated gate bipolar transistor (IGBT) with a MOS structure incorporated into its gate portion.

[0043] In this embodiment, a normal mode choke coil 30 is connected to the upper arm power line (positive busbar) 10 of the DC power supply 29, and a smoothing capacitor 32 is connected between the upper arm power line 10 and the lower arm power line (negative busbar) 15 of the DC power supply 29 downstream of the normal mode choke coil 30.

[0044] The normal mode choke coil 30 (including the inductance of the wiring) and the smoothing capacitor 32 then constitute the input filter 40. In this case, when the carrier frequency is set to the aforementioned 20 kHz, the resonant frequency F of the input filter 40 composed of the normal mode choke coil 30 and the smoothing capacitor 32 is c It is designed to be 14 kHz (resonant frequency F) as mentioned above. c The design value is 14 kHz.

[0045] Furthermore, the normal mode choke coil 30 does not need to be provided. In that case, the inductance of the wiring of the upper arm power line 10 and the smoothing capacitor 32 will constitute the filter 40. And in that case as well, the resonant frequency F of the filter 40 c It shall be designed for 14 kHz.

[0046] Furthermore, a current sensor 45, consisting of a shunt resistor as a current detection unit, is connected to the upper arm power line 10 between the DC power supply 29 and the normal mode choke coil 30. This current sensor 45 is provided to detect the total current flowing from the DC power supply 29 into the circuit, but since it is located in a closed circuit consisting of the DC power supply 29, the normal mode choke coil 30 and the smoothing capacitor 32, it detects the filter resonant current i flowing through the smoothing capacitor 32 that constitutes the filter 40. f It is also possible to detect this.

[0047] The collectors of the upper arm switching elements 18A to 18C of the inverter 28 are connected to the upper arm power supply line 10 downstream of the smoothing capacitor 32. On the other hand, the emitters of the lower arm switching elements 18D to 18F of the inverter 28 are connected to the lower arm power supply line 15 downstream of the smoothing capacitor 32.

[0048] In this case, the emitter of the upper arm switching element 18A of the U-phase half-bridge circuit 19U and the collector of the lower arm switching element 18D are connected in series, the emitter of the upper arm switching element 18B of the V-phase half-bridge circuit 19V and the collector of the lower arm switching element 18E are connected in series, and the emitter of the upper arm switching element 18C of the W-phase half-bridge circuit 19W and the collector of the lower arm switching element 18F are connected in series.

[0049] Then, at the connection point of the upper arm switching element 18A and the lower arm switching element 18D of the U-phase half-bridge circuit 19U (U-phase voltage V u ) is connected to the U-phase armature coil of motor 8, and is the connection point (V-phase voltage V) between the upper arm switching element 18B and the lower arm switching element 18E of the V-phase half-bridge circuit 19V. v ) is connected to the V-phase armature coil of motor 8, and is the connection point (W-phase voltage V) between the upper arm switching element 18C and the lower arm switching element 18F of the W-phase half-bridge circuit 19W. w ) is connected to the W-phase armature coil of motor 8.

[0050] (2) Control device 21 Next, the control device 21 is composed of a microcomputer having a processor, and in this embodiment, the built-in vector control unit 25 controls the d-axis voltage command value V d and q-axis voltage command value V q The following is derived, and based on these, the ON / OFF state (switching) of each switching element 18A to 18F of the inverter 28 is controlled. Specifically, the gate voltage applied to the gates of each switching element 18A to 18F is controlled.

[0051] The control device 21 of this embodiment includes the vector control unit 25, the phase voltage command calculation unit 33, the line modulation calculation unit 34, the PWM signal generation unit 36, the gate driver 37, and the U-phase current I, which is the motor current (phase current) of each phase flowing through the motor 8. u , V phase current I v , W phase current I w It has current sensors 26A, 26B, and 26C, which consist of shunt resistors for measuring current, and the aforementioned current sensor 45.

[0052] In this embodiment, each current sensor 26A, 26B, and 26C is connected to the line modulation calculation unit 34. Furthermore, current sensor 26A is connected to the U-phase current I u The current sensor 26B measures the V-phase current I v The current sensor 26C measures the W-phase current I w The current sensor 26A measures the U-phase current I u The current sensor 26B measures the V-phase current I v Measure the W-phase current I w This can also be calculated from these values.

[0053] Furthermore, regarding the method for detecting the motor current of each phase, in addition to measuring it with current sensors 26A to 26C as in the embodiment, there are other methods such as detecting the current value of the lower arm power line 15 with a shunt resistor and having the phase voltage command calculation unit 33 estimate it from that current value and the operating state of the motor 8. Therefore, the method for detecting and estimating the current of each phase is not particularly limited.

[0054] (2-1) Vector control unit 25 The vector control unit 25 controls the speed command value ω rm ref and the mechanical angular velocity ω of motor 8 rm The difference is used in the PI calculation to obtain the q-axis current command value I q ref The d-axis current command value I is calculated. d ref and d-axis current I d q-axis current command value I q ref and q-axis current I q From the d-axis voltage command value V d and q-axis voltage command value V qThe result is calculated and output to the phase voltage command calculation unit 33.

[0055] Here, in order to rotate the motor 8 (IPMSM), it is necessary to generate a torque τ. This torque τ can be calculated using the following formula (I). Note that I d and I q These are the d-axis current and the q-axis current, L d and L q These are the d-axis inductance and the q-axis inductance. E is the power generation constant of motor 8, and P is the number of pole pairs.

[0056]

[0057] The d-axis and q-axis are two-dimensional axes, with the magnetic pole position of the motor 8 (IPMSM) being the d-axis and the coordinate system perpendicular to it being the q-axis. In this case, the output torque τ is proportional to the q-axis current Iq, so in order to easily control the torque τ, the q-axis current I q This will control the d-axis current I of the motor 8. d and q-axis current I q The d-axis voltage command value V for flowing the current d and q-axis voltage command value V q This can be calculated using the following formula (II).

[0058]

[0059] Here, ω re is the electrical angular rotation speed of motor 8, p is the derivative term, and R is the phase resistance. From this equation (II) and the aforementioned equation (I), the d-axis current I to be supplied to motor 8 is d and q-axis current I q And the electrical angular rotation speed ω of motor 8 re Once determined, the d-axis voltage command value V applied to motor 8 is determined. d and q-axis voltage command value V q It is determined.

[0060] As mentioned above, the motor 8 (IPMSM) is driven by the half-bridge three-phase inverter 28. The voltage that the inverter 28 applies to the three-phase motor 8 is the U-phase voltage V mentioned above. u V-phase voltage V v W-phase voltage V wThen, the d-axis voltage command value V d and the q-axis voltage command value V q and each phase voltage V of UVW u 、V v 、V w have the relationship of the following formula (III).

[0061]

[0062] (2-2) Phase voltage command calculation unit 33 The phase voltage command calculation unit 33 uses the d-axis voltage command value V d and the q-axis voltage command value V q obtained from the vector control unit 25 to calculate the U-phase voltage command value V u * 、the V-phase voltage command value V v * 、the W-phase voltage command value V w * using the following formula (IV). That is, dq-axis - three-phase conversion is performed. Note that V m and θ m in formula (IV) are obtained from formula (V) respectively. Also, θ is the magnetic pole position with respect to the U phase, and θ m is the voltage phase difference with respect to the magnetic pole position.

[0063]

[0064] (2-3) Line-to-line modulation calculation unit 34 The line-to-line modulation calculation unit 34 uses the phase voltage command values V u * 、V v * 、V w * calculated and output by the phase voltage command calculation unit 33 to calculate the PWM count values N dc of each phase normalized (corrected to 0 to 1) with the DC voltage V[[ID=*]] u1 (U-phase PWM count value), N v1 (V-phase PWM count value), N w1 (W-phase PWM count value) using the following formula (VI). These PWM count values N u1 [[ID=*]] v1 、N w1 are the three-phase modulation command values before line-to-line modulation.

[0065] [[ID=*]]

[0066] Furthermore, the line-to-line modulation calculation unit 34 calculates the PWM count value N of each phase. u1 , N v1 , N w1 Therefore, inter-line modulation is performed using equation (VII). In equation (VII), N u , N v , N w This is the PWM count value of each phase after line modulation (U-phase PWM count value N u V-phase PWM count value N v W-phase PWM count value N w )

[0067]

[0068] Note that N in formula (VII) mod This is the line-to-line modulation value for performing line-to-line modulation (two-phase modulation). This line-to-line modulation value N mod is zero ( mod If = 0, then the PWM count value N u , N v , N w This does not perform inter-line modulation, i.e., the PWM count value N before inter-line modulation. u1 , N v1 , N w1 This is identical to N in this embodiment. mod Set the value to 0 and assume that no inter-line modulation is performed.

[0069] Furthermore, in this embodiment, the line-to-line modulation calculation unit 34 includes a resonance frequency estimation unit 50 and a resonance suppression unit 35 in the present invention. The resonance frequency estimation operation by the resonance frequency estimation unit 50 and the resonance suppression operation by the resonance suppression unit 35 will be described in detail later.

[0070] (2-4) PWM signal generation unit 36 ​​The PWM signal generation unit 36 ​​generates the PWM count value N calculated by the line modulation calculation unit 34. u , N v , N wBy inputting the signal and comparing its magnitude with the carrier signal (carrier wave) cnt, the inverter 28 generates and outputs PWM pulses (PWM signals) that serve as drive command signals for the U-phase half-bridge circuit 19U, V-phase half-bridge circuit 19V, and W-phase half-bridge circuit 19W. That is, one cycle of the carrier signal cnt (carrier period) equals one switching period T. s And this is one switching period T s Each switching element 18A to 18F is controlled based on the PWM pulse of each phase.

[0071] Furthermore, the PWM signal generation unit 36 ​​generates the above PWM count value N u , N v , N w Based on this, PWM pulse S is used to switch the switching elements 18A and 18D of the U-phase half-bridge circuit 19U. u And, PWM pulse S for switching the switching elements 18B and 18E of the V-phase half-bridge circuit 19V v And, PWM pulse S for switching the switching elements 18C and 18F of the W-phase half-bridge circuit 19W. w This will cause it to happen.

[0072] Each PWM pulse S u S v S w When the signal rises to "1", the upper arm switching elements 18A to 18C of each phase turn ON, and when it falls to "0", the lower arm switching elements 18D to 18F turn ON (there is a dead time to avoid simultaneous ON of upper and lower). Here, each PWM pulse S u S v S w The period during which the signal strength is "1" is defined as the pulse width of the PWM pulse.

[0073] (2-5) Gate driver 37 The gate driver 37 receives the PWM pulse S output from the PWM signal generation unit 36. u Based on this, the gate voltage S of the switching element 18A of the U-phase half-bridge circuit 19U up And the gate voltage S of the switching element 18D unThis generates a PWM pulse S output from the PWM signal generation unit 36. v Based on this, the gate voltage S of the switching element 18B of the V-phase half-bridge circuit 19V vp And the gate voltage S of the switching element 18E. vn This generates a PWM pulse S output from the PWM signal generation unit 36. w Based on this, the gate voltage S of the switching element 18C of the W-phase half-bridge circuit 19W wp And the gate voltage S of the switching element 18F wn This will cause it to happen.

[0074] Here, the gate voltage S of the U-phase half-bridge circuit 19U up S un This indicates the switching timing that provides a dead time to avoid simultaneous ON of the upper and lower gates, and by providing a constant dead time with the gate driver 37, simultaneous ON of the upper and lower gates is reliably avoided. Gate voltage S of the V-phase half-bridge circuit 19V vp S vn , gate voltage S of W-phase half-bridge circuit 19W wp S wn Similarly, it also provides dead time.

[0075] Each switching element 18A to 18F of the inverter 28 receives the gate voltage S output from the gate driver 37. up S un S vp S vn S wp S wn Based on this, it is driven ON / OFF. That is, when the gate voltage is ON (a predetermined voltage value), the switching element operates ON, and when the gate voltage is OFF (zero), the switching element operates OFF. This gate driver 37 is a circuit for applying the gate voltage to the IGBTs based on the PWM pulse when the switching elements 18A to 18F are the aforementioned IGBTs, and is composed of a photocoupler, logic IC, transistor, etc.

[0076] Then, the voltage at the connection point between the upper arm switching element 18A and the lower arm switching element 18D of the U-phase half-bridge circuit 19U is the U-phase voltage V u The (phase voltage) is applied (output) to the U-phase armature coil of the motor 8, and the voltage at the connection point of the upper arm switching element 18B and lower arm switching element 18E of the V-phase half-bridge circuit 19V is the V-phase voltage V v The (phase voltage) is applied (output) to the V-phase armature coil of the motor 8, and the voltage at the connection point between the upper arm switching element 18C and the lower arm switching element 18F of the W-phase half-bridge circuit 19W is the W-phase voltage V w This (phase voltage) is applied (output) to the W-phase armature coil of motor 8.

[0077] During PWM switching, switching losses occur in the switching elements 18A to 18F. These losses in the switching elements 18A to 18F affect the efficiency of the inverter 28 and also cause a temperature rise in the switching elements 18A to 18F. Both efficiency and temperature rise are important factors in the power converter 1, and it is preferable to minimize switching losses. However, since electric compressors in vehicle air conditioning systems are required to be quiet, the PWM switching frequency, i.e., the frequency of the carrier signal cnt (carrier frequency), is often set to a high value such as 20 kHz. As a result, the number of switching cycles increases, and switching losses increase.

[0078] (3) High-frequency current generated by inverter 28 Next, the high-frequency current generated by inverter 28 will be explained using Figure 2. Note that the motors indicated by the same reference numerals in Figure 2 as in Figure 1 are the same. (3-1) Resonant frequency F of filter 40 c Design value in Figure 2, part I L The current flowing through the normal mode choke coil 30 is i c This is the current flowing through the smoothing capacitor 32 (capacitor current). In Figure 2, i f This is the filter resonant current flowing through the smoothing capacitor 32 that constitutes the filter 40, and the resonant frequency F of the filter 40 is determined by the inductance L of the normal mode choke coil 30 and the capacitance C of the smoothing capacitor 32.c This is the frequency of the filter resonant current if.

[0079] Therefore, the current generated by the inverter 28 is affected by the resonant frequency F of the filter 40, which is formed by the normal mode choke coil 30 and the smoothing capacitor 32. c If the frequency component is present, excessive noise will flow into the DC power supply 29, and the smoothing capacitor 32 will be destroyed by overcurrent. Also, the resonant frequency F c The current has the characteristic that currents with lower frequency components leak to the DC power supply 29 side, while higher frequency components flow into the smoothing capacitor 32 and do not leak to the power supply side.

[0080] Based on the above, the DC link current I flowing between the smoothing capacitor 32 and the inverter 28 dc The resonant frequency is F c Since it is desirable that the frequency components are not included, in this embodiment the carrier frequency (frequency of the carrier wave) is set to 20 kHz, and the resonant frequency F of the input filter 40, which is composed of a normal mode choke coil 30 and a smoothing capacitor 32, is set. c It is designed to be 14 kHz (resonant frequency F c The design value is 14 kHz.

[0081] (3-2) DC link current I dc The high-frequency current flowing through the smoothing capacitor 32 is the DC link current I dc It is determined by the characteristics of the DC link current I dc This changes depending on the magnitude of the current flowing through the motor 8 and the state of the switching elements 18A to 18F.

[0082] (3-3) DC link current I dc The high-frequency component of the current flowing through motor 8 is approximately a sine wave and therefore does not have a high-frequency component, but the DC link current I dc Because it is affected by PWM, it has sidebands, which are high-frequency components. That is, DC link current I dc This will result in sidebands that vary with rotation frequency, centered around the carrier frequency (20 kHz). These sidebands will be at the resonant frequency F of filter 40. cWhen it reaches this point, an excessive current amplitude flows into the smoothing capacitor 32.

[0083] (4) Resonance suppression operation by the resonance suppression unit 35 The resonance suppression unit 35 of the line modulation calculation unit 34 performs the aforementioned resonance frequency F c Based on the design value (14 kHz), a predetermined resonance suppression operation is performed. In this embodiment, the aforementioned sidebands that exist around the carrier frequency are at the resonance frequency F c The switching timing of each switching element 18A to 18F of the inverter 28 is adjusted so that it does not reach a certain level. This prevents excessive amplitude (ripple) current from flowing into the smoothing capacitor 32.

[0084] However, as mentioned above, the resonant frequency F of filter 40 c The resonant frequency deviates from its design value (14 kHz) due to parasitic components in the circuit board, parasitic components in the harness connecting the circuit board to the DC power supply 29, and aging degradation of the smoothing capacitor 32 and normal mode choke coil 30 (passive elements). In other words, the resonant frequency designed in advance differs from the actual resonant frequency. As a result, the adjustment of the switching timing by the resonance suppression unit 35, which is performed based on the resonant frequency, deviates from the appropriate setting, leading to a problem where the effectiveness of the resonance suppression operation deteriorates.

[0085] (5) Resonance frequency F calculated by the resonance frequency estimation unit 50 c In this embodiment, the estimated operation is the filter resonant current i flowing through the smoothing capacitor 32. f The current sensor 45 detects the current, and the detected filter resonance current i f The resonance frequency estimation unit 50 then determines the filter resonance current i f The resonant frequency F of filter 40 is the frequency of F. c This is estimated online in real time, and based on this estimation result, the resonance suppression unit 35 is able to accurately perform resonance suppression operations.

[0086] The following shows the resonance frequency F calculated by the resonance frequency estimation unit 50 using Figures 3 to 9. cThe estimated operation will be explained. (5-1) Operating state of inverter 28 Here, when the motor 8 is rotating, components other than resonance, such as carrier components, flow into the current flowing through the smoothing capacitor 32 that constitutes the filter 40. Therefore, the line modulation calculation unit 34 of the control device 21 estimates the resonance frequency F of the filter 40 using the resonance frequency estimation unit 50. c When estimating the filter resonant current i, first, the current flowing through the smoothing capacitor 32 that constitutes the filter 40 is considered to be the filter resonant current i. f This creates an operating state for inverter 28 where only this is the case.

[0087] In this embodiment, for example, before starting the motor 8 or after stopping the motor 8, the line modulation calculation unit 34 sequentially executes the current control mode in the motor-stopped state shown in Figure 3 and the resonant frequency estimation mode due to inverter stoppage. In Figure 3, the vertical axis represents the current flowing through the smoothing capacitor 32 constituting the filter 40, the horizontal axis represents time, and the solid line in Figure 3 represents the filter resonant current i f The dashed line represents the sampled value i of the acquired filter resonant current. fz This is shown. First, in the current control mode when the motor is stopped, the line modulation calculation unit 34 outputs a command value to supply current to the inverter 28 using the switching elements 18A to 18F while the motor 8 is stopped.

[0088] Next, in the resonant frequency estimation mode by stopping the inverter, the line modulation calculation unit 34 outputs a command value to turn off all switching elements 18A to 18F and stop the operation of the inverter 28. As a result, the filter resonant current i f Only the current flows through the smoothing capacitor 32 in a free-vibration state. In this free-vibration state, the resonance frequency estimation unit 50 determines the resonance frequency F c We estimate this.

[0089] This removes non-resonant components such as carrier components, resulting in a pure filter resonant current i. f From the resonant frequency F c It becomes possible to estimate the actual resonant frequency F. c This explains the estimated behavior.

[0090] (5-2) Peak-to-peak value i pp The resonant frequency determination unit 50 determines the filter resonant current i detected by the current sensor 45. f This is acquired with a predetermined sampling period Δt. In Figure 4, the solid line represents the filter resonant current i. f The dashed line represents the sampled value i of the acquired filter resonant current. fz This is shown. In Figure 4, the vertical axis represents the current flowing through the smoothing capacitor 32 that constitutes the filter 40, and the horizontal axis represents time.

[0091] Here, the resonant frequency F c If the design value is the aforementioned 14 kHz, the resonant frequency F is expected to fluctuate. c It is thought to be around 14 kHz. Therefore, this assumed resonant frequency F c Sampling value i over a sufficiently longer interval (shown as a square in Figure 4) fz Of these, the maximum value i max and minimum value i min Determine the maximum value i. max and minimum value i min From there, the filter resonant current i is calculated using the following formula (VIII). f Peak peak value i pp To decide.

[0092]

[0093] (5-3) Filter resonant current i f Resonant frequency F based on time evolution c Next, the t-th sampling value i of the filter resonant current sampled with a sampling period Δt, as shown in Figure 5. f [t] and the next t+1 sampling value i after a time interval of one sampling period Δt has elapsed. f Focusing on [t+1] (i.e., time variation), the filter resonant current i f Peak peak value i pp Using the inverse trigonometric function arcsin, the t-th sampling value i is obtained as shown in the following formula (IX): f The phase θ[t] of [t] and the (t+1)th sampling value i f Find the phase θ[t+1] at [t+1].

[0094]

[0095] Similarly, the solid line in Figure 5 represents the filter resonant current i. f The dashed line represents the sampled value i of the acquired filter resonant current. fz This is shown in Figure 5. In this Figure 5 as well, the vertical axis represents the current flowing through the smoothing capacitor 32 that constitutes the filter 40, and the horizontal axis represents time.

[0096] Next, the t-th sampling value i f The phase θ[t] of [t] and the (t+1)th sampling value i f The phase difference Δθ (i.e., phase change), which is the difference in phase θ[t+1] at [t+1], is derived, and the resonant frequency F is obtained using the following formula (X) from this phase difference Δθ and the sampling period Δt. c Calculate.

[0097]

[0098] (5-4) Correction of Phase Difference Δθ Here, since the domain of the inverse trigonometric function arcsin is in the range of plus or minus π / 2, the phase difference Δθ used in the calculation of the above formula (X) is corrected as follows to obtain a value of 0 to 2π. Next, an example of this correction of the phase difference Δθ will be explained using Figures 6 to 8.

[0099] As mentioned above, the inverse trigonometric function arcsin is defined as plus or minus π / 2, but there is another phase that can have the same amplitude. That is, there is a symmetric phase θ2 outside the domain of the definition of the output θ1 of the arcsin function. This symmetric phase θ2 is derived by the resonance frequency estimation unit 50 from the waveform symmetry of the output θ1 of the function, as shown in Figure 6.

[0100] Therefore, the t-th sampling value i can be obtained using the formula (IX). f If the phase θ[t] of [t] is θ1[t], then there exists a symmetric phase θ2[t], and the (t+1)th sampling value i f If the phase θ[t+1] at [t+1] is θ1[t+1], then there exists a symmetric phase θ2[t+1].

[0101] Furthermore, for negative values ​​of arcsin (-π / 2 to 0), the values ​​are corrected to 3π / 2 to 2π, as shown by the arrows in Figure 7, due to the symmetry of the waveform, so that the calculated phase difference Δθ is always positive.

[0102] And the sampling value i f Phase θ1[t] of [t] and its symmetric phase θ2[t], sampling value i f For the phase θ1[t+1] and its symmetric phase θ2[t+1] at [t+1], the resonance frequency estimation unit 50 calculates the phase difference Δθ from formula (X) by exhaustively trying all possibilities as shown in Figure 8.

[0103] The example in Figure 8 shows, for example, the tth sampling value i f [t] = 0.9 (A), phase θ1[t] = 64.16°, its symmetric phase θ2[t] = 115.84°, t+1th sampling value i f Assume that [t+1] = -0.18 (A), phase θ1[t+1] = 349.63°, and its symmetric phase θ2[t+1] = 190.37°. Resonant frequency F c The initial design value will be the aforementioned 14 kHz.

[0104] In this case, there are four possible combinations to substitute into equation (X). If the first combination is θ1[t+1] and θ1[t], then the phase difference Δθ1 is 285.47 degrees, and the resonant frequency F can be found from equation (X). c This results in 31.72 kHz. If the following combinations are θ²[t+1] and θ²[t], then the phase difference Δθ² is 74.53 degrees, and the resonant frequency F can be calculated from equation (X). c This corresponds to 8.28 kHz.

[0105] Furthermore, if we consider the following combinations as θ1[t+1] and θ2[t], the phase difference Δθ3 at that time is 233.79 degrees, and the resonant frequency F can be calculated from equation (X). c This becomes 25.98 kHz. Furthermore, if the following combination is θ2[t+1] and θ1[t], the phase difference Δθ4 at that time is 126.21 degrees, and the resonant frequency F can be obtained from equation (X). c This corresponds to 14.02 kHz.

[0106] Assuming that this is the initial estimation of the resonant frequency, the resonant frequency estimation unit 50 determines that 14.02 kHz (the value enclosed in a thick rectangle in Figure 8) is the closest value to the design value of 14 kHz, and therefore adopts the phase difference Δθ4 at that time as the phase difference Δθ in equation (X). This is the correction of the phase difference Δθ.

[0107] Furthermore, the resonance frequency estimation unit 50 estimates the resonance frequency of 14.02 kHz, calculated from the phase difference Δθ4 thus adopted, as the estimated value F c est The line-to-line modulation calculation unit 34 then determines this estimated value F. c est The resonant frequency of 14.02 kHz, which was determined as the optimal value, will be updated as the design value. In other words, 14.02 kHz will be the design value for the next estimation.

[0108] Then, the resonance suppression unit 35 of the line modulation calculation unit 34 uses the updated design value, which is the resonance frequency F. c Based on =14.02 kHz, a predetermined resonance suppression operation will be performed. The resonance frequency estimation unit 50 will perform the above estimation operation each time it estimates the resonance frequency F c We will update this as the design value.

[0109] (5-5) Resonant frequency F c Figure 9 shows the estimated resonant frequency F, which was estimated by the resonant frequency estimation unit 50 in this manner. c Estimated value F c est and the measured value F c real The above shows a comparison of the filter resonant current i shown in Figure 4. f (Solid line) and sampling value i fz (Dashed line). The middle section of Figure 9 shows the resonant frequency F. c Estimated value F c est (Dashed line) and measured value F c real The dashed lines are superimposed, and the lower part of Figure 9 shows the resonant frequency F. c Estimated value F c est and the measured value F c realThe difference is shown by the solid line. Therefore, in this invention, the resonant frequency F c This shows that it can be estimated with high accuracy.

[0110] As described in detail above, according to the present invention, the control device 21 receives the filter resonance current i detected by the current sensor 45. f The resonant frequency F is the frequency of the filter's resonant current. c Since a resonance frequency estimation unit 50 is provided to estimate the filter resonance current i as in the embodiment, the current sensor 45 (current detection unit) detects the filter resonance current i f The phase change is derived from the time change of and the resonant frequency F c By estimating this, the resonant frequency F of the input filter 40 can be determined. c This will enable accurate, real-time online estimation.

[0111] This makes it possible to accurately perform the resonance suppression operation by the resonance suppression unit 35 and to always maintain a high level of resonance suppression effect.

[0112] In particular, in the embodiment, the resonance frequency estimation unit 50 estimates the filter resonance current i sampled at a predetermined sampling period Δt. f The t-th sampling value i f [t] and the (t+1)th sampling value i f [t+1] and the filter resonant current i f Peak peak value i pp Based on this, the t-th sampling value i is obtained using the inverse trigonometric function arcsin. f The phase θ[t] of [t] and the (t+1)th sampling value i f The phase θ[t+1] at [t+1] is determined, and the resonant frequency F is obtained from the phase difference Δθ (the difference between these two phases) and the sampling period Δt. c Since we have made it so that the resonant frequency F can be calculated with high accuracy c It becomes possible to estimate this.

[0113] Furthermore, in this embodiment, the resonance frequency estimation unit 50 corrects the phase difference Δθ based on the domain of the inverse trigonometric function arcsin, thereby enabling more accurate estimation of the resonance frequency F c This will allow us to estimate the value.

[0114] That is, in this embodiment, the resonance frequency estimation unit 50 uses the waveform symmetry to determine the t-th sampling value i that is outside the definition domain of the inverse trigonometric function arcsin. f The phase θ of [t] and the symmetric phase of [t], and the (t+1)th sampling value i f The phase θ at [t+1] is derived, the symmetric phase of [t+1] is determined within the range of 0 to 2π, including each derived symmetric phase, and then the phase difference Δθ that results in a resonant frequency close to the design value is adopted, and the resonant frequency F is calculated from the adopted phase difference Δθ. c The resonant frequency F was determined as an estimated value. c We have updated the design value to reflect the resonant frequency F, so we can continue to maintain high accuracy. c This will enable us to achieve estimation.

[0115] In this embodiment, the resonance frequency estimation unit 50 determines the filter resonance current i over a sufficiently long interval compared to the resonance frequency. f From the sampling values, the peak peak value i pp By determining the peak peak value i, a more accurate peak peak value can be determined. pp This allows for the determination of the resonant frequency and enables accurate estimation of the resonant frequency.

[0116] In this embodiment, the resonance frequency estimation unit 50 determines that the current flowing through the smoothing capacitor 32 is the filter resonance current i f In the operating state of inverter 28, the resonant frequency F is the only one that occurs. c Since we have made it possible to estimate the pure filter resonant current i, which is obtained by removing non-resonant components such as carrier components. f From the resonant frequency F c It becomes possible to estimate this.

[0117] In this embodiment, the control device 21 supplies current to the inverter 28 with the motor 8 stopped, and then stops the operation of the inverter 28, thereby reducing the filter resonant current i f Only the current flowing through the smoothing capacitor 32 is in a free-vibration state, and in that state the resonance frequency estimation unit 50 determines the resonance frequency F c Since we have made it so that we can estimate the pure filter resonant current i, which is obtained by removing components other than resonance.f From the resonant frequency F c This will allow us to make estimations.

[0118] In this embodiment, the present invention was applied to a power conversion device 1 that drives and controls the motor 8 of an electric compressor. However, the present invention is not limited to this, and is effective when various devices are driven and controlled by an inverter with a filter at the input.

[0119] Furthermore, the resonance suppression operation and the operating state of the inverter when detecting the filter resonance current shown in the examples are not limited to those described, and it goes without saying that they can be modified without departing from the spirit of the present invention.

[0120] 1 Power conversion device 8 Motor 18A-18F Upper and lower arm switching element 19U U-phase half-bridge circuit 19V V-phase half-bridge circuit 19W W-phase half-bridge circuit 21 Control device 25 Vector control unit 28 Inverter 29 DC power supply 30 Normal mode choke coil 32 Smoothing capacitor 33 Phase voltage command calculation unit 34 Line-to-line modulation calculation unit 35 Resonance suppression unit 36 ​​PWM signal generation unit 37 Gate driver 40 Filter 45 Current sensor (current detection unit) 50 Resonance frequency estimation unit

Claims

1. A power conversion device that generates an AC output from a DC power source using an inverter and applies it to a motor, comprising: a filter connected to the input; a current detection unit that detects the filter resonant current flowing through a smoothing capacitor constituting the filter; and a control device, wherein the control device has a resonant frequency estimation unit that estimates the resonant frequency, which is the frequency of the filter resonant current, from the filter resonant current detected by the current detection unit.

2. The power conversion device according to claim 1, characterized in that the resonance frequency estimation unit estimates the resonance frequency from the time change of the filter resonance current detected by the current detection unit.

3. The power conversion device according to claim 2, characterized in that the resonance frequency estimation unit derives a phase change from the time change of the filter resonance current detected by the current detection unit and estimates the resonance frequency.

4. The resonance frequency estimation unit determines the t-th sampling value i of the filter resonance current sampled at a predetermined sampling period Δt. f [t] and the (t+1)th sampling value i f [t+1] and the peak value i of the filter resonant current. pp Based on this, the t-th sampling value i is obtained using the inverse trigonometric function. f The phase θ[t] of [t] and the t+1th sampling value i f The power conversion device according to claim 3, characterized in that the phase θ[t+1] at [t+1] is determined, and the resonant frequency is calculated from the phase difference Δθ, which is the difference between them, and the sampling period Δt.

5. The power conversion device according to claim 4, characterized in that the resonant frequency estimation unit applies a correction to the phase difference Δθ based on the domain of the inverse trigonometric function.

6. The resonance frequency estimation unit uses the waveform symmetry to determine the t-th sampling value i that is outside the definition domain of the inverse trigonometric function. f The phase θ[t] of [t] and the symmetric phase of the t+1th sampling value i f The power conversion device according to claim 5, characterized in that it derives the symmetric phase of the phase θ[t+1] at [t+1], and determines the phase difference Δθ in the range of 0 to 2π, including each of the derived symmetric phases.

7. The power conversion device according to claim 6, characterized in that the resonance frequency estimation unit adopts the phase difference Δθ such that the calculated resonance frequency is close to the design value, and determines the resonance frequency calculated from the adopted phase difference Δθ as an estimated value.

8. The power conversion device according to claim 7, characterized in that the resonance frequency estimation unit updates the resonance frequency determined as the estimated value as the design value.

9. The resonance frequency estimation unit calculates the peak-to-peak value i from the sampled values ​​of the filter resonance current over a sufficiently long interval compared to the resonance frequency. pp The power conversion device according to claim 4, characterized in that it determines the following.

10. The power conversion device according to claim 1, characterized in that the resonant frequency estimation unit estimates the resonant frequency in the operating state of the inverter where the current flowing through the smoothing capacitor constituting the filter is only the filter resonant current.

11. The power conversion device according to claim 10, characterized in that the control device supplies current to the inverter with the motor stopped, and then stops the operation of the inverter, thereby causing the filter resonant current to flow only through the smoothing capacitor in a free-vibration state, and the resonant frequency estimation unit estimates the resonant frequency in that state.

12. The power conversion device according to claim 1, further comprising a resonance suppression unit that performs a predetermined resonance suppression operation based on the resonance frequency estimated by the resonance frequency estimation unit.

13. The power conversion device according to claim 12, characterized in that the resonance suppression unit adjusts the switching timing of the inverter so that sidebands present around the carrier frequency do not reach the resonance frequency.

14. A method for estimating the resonant frequency of a filter connected to the input of a power converter, characterized by detecting the filter resonant current flowing through the smoothing capacitor, deriving a phase change from the time change of the detected filter resonant current, and estimating the resonant frequency, which is the frequency of the filter resonant current.

15. Sample the t-th sampling value i f [t] of the filter resonance current and the (t + 1)-th sampling value i f [t + 1], and determine the peak-to-peak value i pp of the filter resonance current. Based on the peak-to-peak value i pp , the t-th sampling value i f [t], and the (t + 1)-th sampling value i f [t + 1], use the inverse trigonometric function to obtain the phase θ[t] of the t-th sampling value i f [t] and the phase θ[t + 1] of the (t + 1)-th sampling value i f [t + 1]. Calculate the resonance frequency from the phase difference Δθ, which is the difference between them, and the sampling period Δt. The method for estimating the resonance frequency of the power conversion device according to claim 14, characterized in that.

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