Bidirectional switching unit and power converter

Abstract

A bidirectional switching unit, comprising an inductor, a first switch, a second switch, a first back-to-back switch pair, a first capacitor, a second back-to-back switch pair, and a second capacitor, wherein the first back-to-back switch pair and the first capacitor form a first switching branch, and the second back-to-back switch pair and the second capacitor form a second switching branch. The inductor has a first terminal and a second terminal. The first switch has a first terminal and a second terminal. The first terminal of the first switch is connected to the first terminal of the inductor to form a first common node. The second switch has a first terminal and a second terminal. The first terminal of the second switch is connected to the second terminal of the inductor to form a second common node. A first terminal of the first switching branch is connected to the second terminal of the first switch, and a second terminal of the first switching branch is connected to the second common node. A first terminal of the second switching branch is connected to the first common node, and a second terminal of the second switching branch is connected to the second terminal of the second switch. The present disclosure also relates to a power converter.

Description

Bidirectional switching unit and power converter Technical Field

[0001] This invention relates to a bidirectional switching unit and a power converter, and particularly to a flexible bidirectional switching unit and a power converter with zero-voltage switching. Background Technology

[0002] Single-phase two-wire to single-phase three-wire AC-AC converters can be used in applications such as on-board charging modules (OBCMs) or backup circuits for home energy storage systems providing household loads (V2H). Current technologies typically employ center-tapped power frequency transformers (as shown in Figure 1) or switching AC-AC converters (as shown in Figures 2 and 3).

[0003] Center-tapped power frequency transformers have disadvantages such as heavy weight and large size, which are easily limited by equipment installation space constraints. Single-phase two-wire to single-phase three-wire switching AC-to-AC converters achieve load-side voltage control through high-frequency switching of the switching unit, thereby reducing the size of magnetic components. However, the existing technology uses hard switching technology, which generates large switching losses and results in poor conversion efficiency.

[0004] Therefore, how to design a bidirectional switching unit and power converter to solve the problems and technical bottlenecks of the existing technology is an important research topic for the inventors of this disclosure. Summary of the Invention

[0005] One object of the present invention is to provide a bidirectional switching unit. The bidirectional switching unit includes an inductor, a first switch, a second switch, a first back-to-back switch pair and a first capacitor, and a second back-to-back switch pair and a second capacitor. The inductor has a first terminal and a second terminal. The first switch has a first terminal and a second terminal, and the first terminal of the first switch is connected to the first terminal of the inductor to form a first common contact. The second switch has a first terminal and a second terminal, and the first terminal of the second switch is connected to the second terminal of the inductor to form a second common contact. The first back-to-back switch pair is series-coupled with the first capacitor to form a first switch branch. The first switch branch has a first terminal and a second terminal, the first terminal of the first switch branch is connected to the second terminal of the first switch, and the second terminal of the first switch branch is connected to the second common contact. The second back-to-back switch pair is series-coupled with the second capacitor to form a second switch branch. The second switch branch has a first terminal and a second terminal, the first terminal of the second switch branch is connected to the first common contact, and the second terminal of the second switch branch is connected to the second terminal of the second switch.

[0006] Another object of the present invention is to provide a power converter. The power converter includes at least one switching bridge arm. The switching bridge arm includes a plurality of bidirectional switching units. Each bidirectional switching unit includes an inductor, a first switch, a second switch, a first back-to-back switch pair and a first capacitor, and a second back-to-back switch pair and a second capacitor. The inductor has a first terminal and a second terminal. The first switch has a first terminal and a second terminal, with the first terminal of the first switch connected to the first terminal of the inductor to form a first common contact. The second switch has a first terminal and a second terminal, with the first terminal of the second switch connected to the second terminal of the inductor to form a second common contact. The first back-to-back switch pair is series-coupled with the first capacitor to form a first switching branch, the first switching branch having a first terminal and a second terminal, the first terminal of the first switching branch connected to the second terminal of the first switch, and the second terminal of the first switching branch connected to the second common contact. The second back-to-back switch pair is series-coupled with the second capacitor to form a second switching branch, the second switching branch having a first terminal and a second terminal, the first terminal of the second switching branch connected to the first common contact, and the second terminal of the second switching branch connected to the second terminal of the second switch. At least one switching bridge arm receives the input power and converts the input power into the output power through a bidirectional switching unit, which can be used to provide multiple outputs.

[0007] Therefore, the bidirectional switching unit and power converter proposed in this invention have the following features and advantages: 1. The overall system architecture achieves a smooth switching effect through architectural adjustments within the switching unit, and the output voltage is distributed through the alternating operation of the two switching units. By adding auxiliary switches, inductors, and capacitors within the switching unit to provide an additional resonant circuit, the voltage across the two terminals is zero before the switch is turned on, achieving a smooth switching effect. Compared to traditional hard-switching schemes, this reduces losses during switching, thereby improving system efficiency and heat dissipation. Furthermore, compared to center-tapped power frequency transformers, it offers improvements in weight and size; 2. Balanced and unbalanced loads, combined with the positive and negative half-cycles of the input power supply, allow each high-frequency switch to achieve a smooth switching behavior with zero voltage switching.

[0008] To gain a deeper understanding of the techniques, means, and effects employed by this invention to achieve its intended purpose, please refer to the following detailed description and accompanying drawings. It is believed that the purpose, features, and characteristics of this invention can be understood in a thorough and specific manner from these drawings. However, the drawings are provided for reference and illustration only and are not intended to limit the scope of this invention. Attached Figure Description

[0009] Figure 1: A block diagram of a single-phase two-wire to single-phase three-wire AC-to-AC converter for an existing intermediate-tap power frequency transformer.

[0010] Figure 2: A block diagram of an existing single-phase two-wire to single-phase three-wire switching AC-to-AC converter.

[0011] Figure 3: Circuit diagram of an existing single-phase two-wire to single-phase three-wire switching AC-to-AC converter.

[0012] Figure 4: Circuit diagram of the bidirectional switching unit of the present invention.

[0013] Figure 5A: The equivalent circuit diagram of Figure 4 when the input power supply is positive.

[0014] Figure 5B: The equivalent circuit diagram of Figure 4 when the input power supply is negative.

[0015] Figure 6: A schematic circuit diagram of the load current flowing into Figure 4 when the input power supply is positive voltage.

[0016] Figure 7: A schematic circuit diagram of the load current flowing out when the input power supply is positive voltage as shown in Figure 4.

[0017] Figure 8: Circuit diagram of the power converter of the present invention.

[0018] Figure 9A: This is a control waveform diagram of the positive half-cycle bidirectional switching unit of the power converter of the present invention.

[0019] Figure 9B: This is a control waveform diagram of the negative half-cycle bidirectional switching unit of the power converter of the present invention.

[0020] Figure 10: A block diagram of the power converter control of the present invention.

[0021] Figure 11: A waveform diagram of the control signal of the power converter of the present invention.

[0022] Figure 12: A schematic circuit diagram of the power converter of the present invention operating under unbalanced load conditions where the input voltage is positive and the single-sided load is open.

[0023] Figure 13: A schematic circuit diagram of Figure 12 operating in the first time interval.

[0024] Figure 14: A schematic circuit diagram of Figure 12 operating in the second time interval.

[0025] Figure 15: A schematic circuit diagram of Figure 12 operating in the third time interval.

[0026] Figure 16: A schematic circuit diagram of Figure 12 operating in the fourth time interval.

[0027] Figure 17: Circuit diagram of the three-phase DC to AC converter of the present invention.

[0028] Figure Label Explanation: Vin: Input Power Supply; Vs: AC Power Supply; L1, L2, L2a, L2b, L2c, L3, L3a, L3b, L3c: Inductors; S1, S1a, S1b, S1c, S3, S3a, S3b, S3c: First Switch; S2, S2a, S2b, S2c, S4, S4a, S4b, S4c: Second Switch; SA, SAa, SAb, SAc, SC, SCa, SCb, SCc: Third Switch; SA2, SA2a, SA2b, SA2c, SC2, SC2a, SC2b, SC2c: Fourth Switch; SB, SBa, SBb, SBc, SD, SDa, SDb, SDc: Fifth Switch; SB2, ... SB2a, SB2b, SB2c, SD2, SD2a, SD2b, SD2c: Sixth switch SA, SA2: First back-to-back switch pair SC, SC2: First back-to-back switch pair SB, SB2: Second back-to-back switch pair SD, SD2: Second back-to-back switch pair P1: First common contact P2: Second common contact CA, CAa, CAb, CAc, CC, CCa, CCb, CCc: First capacitor CB, CBa, CBb, CBc, CD, CDa, CDb, CDc: Second capacitor L, La, Lb, Lc: Filter inductor C1: First filter capacitor C2: Second filter capacitor I: Load current VDC: DC power supply Detailed Implementation

[0029] The technical content and detailed description of the present invention are explained below with reference to the accompanying drawings.

[0030] This disclosure proposes a soft switching scheme for use in single-phase two-wire to single-phase three-wire AC-AC converters. The overall system architecture is shown in Figure 2. The soft switching effect is achieved through architectural adjustments within the switching unit, and the output voltage is distributed through the alternating operation of the two switching units. An auxiliary switch, inductor, and capacitor are added within the switching unit to provide an additional resonant circuit, ensuring that the voltage across the switch is zero before conduction, thus achieving the soft switching effect. Compared to traditional hard switching schemes, this reduces switching losses, thereby improving system efficiency and heat dissipation. Furthermore, compared to center-tapped power frequency transformers, it offers improvements in weight and size.

[0031] Please refer to Figure 4, which is a circuit diagram of the bidirectional switching unit of the present invention. This bidirectional switching unit has a flexible switching characteristic and can therefore be called a flexible bidirectional switching unit. It includes an inductor L1, a first switch S1, a second switch S2, a first back-to-back switch pair SA, SA2 and a first capacitor CA, and a second back-to-back switch pair SB, SB2 and a second capacitor CB.

[0032] Inductor L1 has a first terminal and a second terminal. First switch S1 has a first terminal and a second terminal that are not control terminals. In the embodiment shown in Figure 4, the first switch S1 is a metal-oxide-semiconductor field-effect transistor (MOSFET), but this is not intended to limit the invention. The first terminal of the first switch S1 (the source in this embodiment) is connected to the first terminal of inductor L1 to form a first common contact P1. Second switch S2 has a first terminal and a second terminal that are not control terminals. In the embodiment shown in Figure 4, the second switch S2 is a metal-oxide-semiconductor field-effect transistor (MOSFET), but this is not intended to limit the invention. The first terminal of the second switch S2 (the source in this embodiment) is connected to the second terminal of inductor L1 to form a second common contact P2.

[0033] The first back-to-back switch pair SA,SA2 has a first terminal and a second terminal that are not control terminals. The first back-to-back switch pair SA,SA2 comprises a third switch SA and a fourth switch SA2 connected back-to-back. As shown in the embodiment in Figure 4, the third switch SA and the fourth switch SA2 are metal-oxide-semiconductor field-effect transistors (MOSFETs), and the third switch SA and the fourth switch SA2 are common-drain connected. However, this is not a limitation of the invention; that is, the third switch SA and the fourth switch SA2 can also be common-source connected. The invention is illustrated using a common-drain connection as an example. The first terminal of the first back-to-back switch pair SA,SA2 (in this embodiment, the source of the third switch SA) is connected to the second terminal of the first switch S1 (in this embodiment, the drain), and the second terminal of the first back-to-back switch pair SA,SA2 (in this embodiment, the source of the fourth switch SA2) is connected to the second common contact point P2.

[0034] The second back-to-back switch pair SB, SB2 has a first terminal and a second terminal that are not control terminals. The second back-to-back switch pair SB, SB2 comprises a fifth switch SB and a sixth switch SB2 connected back-to-back. As shown in the embodiment in Figure 4, the fifth switch SB and the sixth switch SB2 are metal-oxide-semiconductor field-effect transistors (MOSFETs), and the fifth switch SB and the sixth switch SB2 are connected in a common-drain configuration. However, this is not a limitation of the invention; that is, the fifth switch SB and the sixth switch SB2 can also be connected in a common-source configuration. This invention uses a common-drain configuration as an example. The first terminal of the second back-to-back switch pair SB, SB2 (in this embodiment, the source of the sixth switch SB2) is connected to the first common connection point P1, and the second terminal of the second back-to-back switch pair SB, SB2 (in this embodiment, the source of the fifth switch SB) is connected to the second terminal of the second switch S2 (in this embodiment, the drain).

[0035] As shown in Figure 4, the first back-to-back switch pair SA, SA2 and the first capacitor CA form a first switch branch, and the two ends of the first switch branch are respectively connected to the second common contact P2 and the second end of the first switch S1. Therefore, the first back-to-back switch pair SA, SA2 is further connected between the second common contact P2 and the second end of the first switch S1 through the first capacitor CA. Alternatively, as shown in Figure 4, the first capacitor CA is connected between the third switch SA and the first switch S1, or the first capacitor CA (not shown in Figure 4) is connected between the fourth switch SA2 and the second common contact P2. Furthermore, the second back-to-back switch pair SB, SB2 and the second capacitor CB form a second switch branch, and the two ends of the second switch branch are respectively connected to the first common contact P1 and the second end of the second switch S2. Therefore, the second back-to-back switch pair SB, SB2 is further connected between the first common contact P1 and the second end of the second switch S2 through the second capacitor CB. That is, as shown in Figure 4, the second capacitor CB is connected between the fifth switch SB and the second switch S2, or the second capacitor CB, which is not shown in Figure 4, is connected between the sixth switch SB2 and the first common contact point P1.

[0036] The bidirectional switching unit shown in Figure 4 is connected to the AC power supply Vs. Therefore, depending on the voltage polarity caused by the positive and negative half-cycle changes of the AC power supply Vs, some switches will remain normally on and some switches will remain normally off, as explained below.

[0037] When the AC power supply Vs is in the positive half-cycle (i.e., Vs>0), the second switch S2 and the sixth switch SB2 of the second back-to-back switch pair SB,SB2 are normally on, and the third switch SA and the fourth switch SA2 of the first back-to-back switch pair SA,SA2 are normally off. Therefore, the bidirectional switching unit circuit in Figure 4 can be equivalent to the equivalent circuit in Figure 5A. When the AC power supply Vs is in the negative half-cycle (i.e., Vs<0), the first switch S1 and the fourth switch SA2 of the first back-to-back switch pair SA,SA2 are normally on, and the fifth switch SB and the sixth switch SB2 of the second back-to-back switch pair SB,SB2 are normally off. Therefore, the bidirectional switching unit circuit in Figure 4 can be equivalent to the equivalent circuit in Figure 5B.

[0038] As can be seen from Figure 5A (equivalent circuit under the positive half-cycle of AC power supply Vs) and Figure 5B (equivalent circuit under the negative half-cycle of AC power supply Vs), these two equivalent circuits are symmetrical. The equivalent circuit under the positive half-cycle of AC power supply Vs will be used as an example. As shown in Figure 6, the explanation focuses on the inflow of load current I, where current I1 > I is a necessary condition for the smooth switching. As shown in Figure 7. Incidentally, the load of this invention is not limited to resistive loads; in other words, it is applicable to both inductive and capacitive loads. Furthermore, not only can the smooth switching operation from single-phase two-wire to single-phase three-wire be achieved for balanced resistive, inductive, or capacitive loads, but it can also achieve the smooth switching operation from single-phase two-wire to single-phase three-wire for unbalanced resistive, inductive, or capacitive loads.

[0039] Please refer to Figure 8, which is a circuit diagram of the power converter of the present invention. Incidentally, the power converter shown in Figure 8 utilizes two sets of front-mounted bidirectional switching units as bridge arm switches. One set serves as a bidirectional upper bridge arm switch with a softening function, and the other set serves as a bidirectional lower bridge arm switch with a softening function. In other words, the embodiment in Figure 8 is merely one embodiment of a power converter and is not intended to limit the present invention. The application scope of the bidirectional switching unit of the present invention includes any switching application requiring softening and bidirectional control. In other words, in other applications, multiple sets of bidirectional switching units can be used as the required bridge arm switches according to the needs of the bridge arm topology.

[0040] Taking the power converter in Figure 8 as an example, it includes an upper bridge arm and a lower bridge arm. The upper bridge arm includes inductor L2, first switch S1, second switch S2, first back-to-back switch pairs SA and SA2, second back-to-back switch pairs SB and SB2, first capacitor CA, and second capacitor CB. The lower bridge arm correspondingly includes inductor L3, first switch S3, second switch S4, first back-to-back switch pairs SC and SC2, second back-to-back switch pairs SD and SD2, first capacitor CC, and second capacitor CD. For specific circuit connections, please refer to the preceding description; further details are omitted here. Furthermore, as shown in Figure 8, it also includes an output filter composed of filter inductor L and filter capacitors C1 and C2. Taking two bidirectional switching units as an example, it has a set of output filters, including filter inductor L, first filter capacitor C1, and second filter capacitor C2. This set of output filters is coupled between the two bidirectional switching units. As shown in Figure 8, filter inductor L, first filter capacitor C1, and second filter capacitor C2 are connected to a neutral point N. In addition, the other end of the filter inductor L is coupled between the upper and lower bridge arms. The other ends of the first filter capacitor C1 and the second filter capacitor C2 are respectively connected to the two ends of the input power supply Vin.

[0041] The switching unit and output filter proposed in Figure 8 establish a flexible switching scheme for a single-phase two-wire to single-phase three-wire AC-AC converter. This single-phase two-wire to single-phase three-wire AC-AC converter can be applied to vehicle charging modules or home energy storage systems, but this is not a limitation of the present invention. The control methods are divided into three types: positive half-cycle control, negative half-cycle control, and zero-crossing interval control (transition control from positive half-cycle to negative half-cycle or vice versa). In different situations, some switches remain always turned on, and some switches remain always turned off. Please refer to Figures 9A and 9B, which are the control waveform diagrams of the bidirectional switching unit for the positive and negative half-cycles of the power converter of the present invention, respectively, and refer to Figure 8 for further details.

[0042] As shown in Figure 9A, when the input power supply Vin is in the positive half-cycle (i.e., Vin>0), the second switch S2, the second switch S4, the sixth switch SB2 of the second back-to-back switch pair SB,SB2, and the sixth switch SD2 of the second back-to-back switch pair SD,SD2 are normally on, and the third switch SA and the fourth switch SA2 of the first back-to-back switch pair SA,SA2, and the third switch SC and the fourth switch SC2 of the first back-to-back switch pair SC,SC2 are normally off. The remaining switches S1,SB, and S3,SD are switched alternately at high frequency. Similarly, as shown in Figure 9B, when the input power supply Vin is in the negative half-cycle (i.e., Vin < 0), the first switch S1, the first switch S3, the fourth switch SA2 of the first back-to-back switch pair SA,SA2, and the fourth switch SC2 of the first back-to-back switch pair SC,SC2 are normally on, while the fifth switch SB and the sixth switch SB2 of the second back-to-back switch pair SB,SB2, and the fifth switch SD and the sixth switch SD2 of the second back-to-back switch pair SD,SD2 are normally off. The remaining switches S2, SA, S4, and SC are switched alternately at high frequency. Incidentally, the duty cycle of the switch control signal for alternating on and off is controlled at 50%, and no additional information on the output voltage is required.

[0043] Please refer to Figure 10, which is a block diagram of the power converter control of the present invention, and Figure 11, which is a waveform diagram of the control signal of the power converter of the present invention. During the zero-crossing interval, to avoid the simultaneous high-level conduction of the first switches S1, S3, the second switches S2, S4, and the back-to-back switch pairs of the third switches SA, SC, and the fourth switches SB, SD, the zero-crossing detection unit 101, or zero-crossing judgment circuit, firstly, the polarity of the input power supply Vin is determined using the zero-crossing detection unit 101. Then, the triangular wave and the DC level are compared via the comparison unit 102 to generate a pulse width modulation (PWM) signal. This PWM signal is processed by the dead-time unit 104 to generate complementary first PWM signal PWM1 and second PWM signal PWM2, with a dead-time interval between them. Therefore, the dead time unit 104 deals with the dead time (or short circuit prevention time) between the corresponding switches. The dead time generated by the delay can prevent the corresponding switches, such as the upper and lower switches of the same bridge arm, from short-circuiting.

[0044] Next, the positive edge (or rising edge) trigger control delay unit 103 determines the timing of the positive and negative half-cycle mode transition. Therefore, the delay provided by delay unit 103 is for the zero-crossing point, such as in an AC power supply, the zero-crossing point of a voltage waveform with a phase angle of 0 degrees or 180 degrees—that is, the instant the voltage changes from positive to negative or vice versa—extending the on / off time of a specific switch to improve triggering accuracy. Thus, delay unit 103, based on the polarity information obtained by zero-crossing detection unit 101, as shown in Figure 11, considers the sloping time interval before 0V as positive and the sloping time interval after 0V as negative, ultimately achieving the aforementioned purpose. Therefore, after the zero-crossing point where the positive voltage turns into a negative voltage (i.e., "0V" marked in Figure 11), and after a delay of PL_delay, the control signals of the fourth switch SA2 and the fourth switch SC2 change from low level (normally closed control) to high level (normally on control). Correspondingly, the control signals of the sixth switch SB2 and the sixth switch SD2 can be changed from high level (normally on control) to low level (normally closed control) through the inverting unit 106, thereby controlling the corresponding switches. Finally, the switch signals are modulated by the control unit 105 with modulation function to make them operate according to a specific mode. The control concept is shown in Figure 10, and the generated switch signals are shown in Figure 11.

[0045] The following section further explains the circuit behavior under the condition that the input power supply Vin is in the positive half-cycle and the load is unbalanced. As mentioned earlier, during the positive half-cycle, some switches remain normally on and some remain normally off. Therefore, the circuit is simplified based on this principle of normally on and normally off. Furthermore, considering the case of unbalanced load (taking the extreme unbalanced load 2 as an example with the load 2 considered as an open circuit), the final simplified circuit is shown in Figure 12.

[0046] As shown in Figure 9A, the positive half-cycle control signal is described by marking the states from time t10 to time t14. Time t14 is equivalent to the state at time t10, and the period from time t10 to time t14 is one pulse width modulation (PWM) signal cycle. The first switch S1 and the fifth switch SB of the second back-to-back switch pair are simultaneously turned on and off (i.e., synchronously switching on and off); the first switch S3 and the fifth switch SD of the second back-to-back switch pair are simultaneously turned on and off. Furthermore, the first switch S1 (the fifth switch SB of the second back-to-back switch pair) and the first switch S3 (the fifth switch SD of the second back-to-back switch pair) alternately turn on and off.

[0047] The operation from time t10 to time t11 is shown in Figure 13, which is a schematic circuit diagram of the operation in the first time interval of Figure 12. The first switches S1 and S3 and the fifth switches SB and SD of the second back-to-back switch pair are all off (indicated by dashed lines in Figure 13). This is the dead time of the circuit. The inductor current IL flowing through the inductor L freewheels and discharges the parasitic capacitance of the first switch S1 through the current IS1. The current ICB discharges the parasitic capacitance of the fifth switch SB of the second back-to-back switch pair. As a result, the voltage across the first switch S1 and the fifth switch SB of the second back-to-back switch pair drops to zero, that is, the source-drain voltage of the first switch S1 drops to zero and the source-drain voltage of the fifth switch SB drops to zero. The schematic circuit is shown in Figure 13.

[0048] The operation from time t11 to time t12 is illustrated in Figure 14, which is a schematic circuit diagram of the operation in the second time interval of Figure 12. The first switch S1 and the fifth switch SB of the second back-to-back switch pair are turned on, achieving zero-voltage switching (ZVS) smoothing to reduce the switching loss between the first switch S1 and the fifth switch SB. Inductor L2 and the second capacitor CB resonate, causing the inductor current IL2 flowing through inductor L2 to rise, and the direction of current ICB to change. The schematic circuit is shown in Figure 14. Therefore, from time t10 to time t12, the voltage across the first switch S1 and the fifth switch SB of the second back-to-back switch pair can be reduced to zero by discharging the parasitic capacitance of the switches, and then the fifth switch SB of the first switch S1 and the second back-to-back switch pair can be turned on again, achieving a smooth zero-voltage switching operation.

[0049] The operation from time t12 to time t13 is illustrated in Figure 15, which is a schematic circuit diagram of Figure 12 operating in the third time interval. The first switches S1 and S3, and the fifth switches SB and SD of the second back-to-back switch pair are all normally off (represented by dashed lines in Figure 15). This is the dead time of the circuit. The inductor current IL2 flowing through inductor L2 continues, and the inductor current IL2 is greater than the inductor current IL. Furthermore, the magnitude of current IS3 is equal to the inductor current IL2 minus the inductor current IL. Simultaneously, current ICD is generated. Current IS3 discharges the parasitic capacitance of the first switch S3, and current ICD discharges the parasitic capacitance of the fifth switch SD of the second back-to-back switch pair. Therefore, the voltage across the first switch S3 and the fifth switch SD of the second back-to-back switch pair drops to zero, i.e., the source-drain voltage of the first switch S3 drops to zero, and the source-drain voltage of the fifth switch SD drops to zero. The schematic circuit is shown in Figure 15.

[0050] The operation from time t13 to time t14 is illustrated in Figure 16, which is a schematic circuit diagram of the operation in the fourth time interval of Figure 12. The first switch S3 and the fifth switch SD of the second back-to-back switch pair are turned on, achieving zero-voltage switching (ZVS) smoothing to reduce the switching loss between the first switch S3 and the fifth switch SD. Inductor L3 resonates with the second capacitor CD, and the second capacitor CD is charged, as shown in Figure 16. Therefore, from time t12 to time t14, the voltage across the first switch S3 and the fifth switch SD of the second back-to-back switch pair can be reduced to zero by discharging the parasitic capacitance of the switches, and then the fifth switch SD of the first switch S3 and the second back-to-back switch pair can be turned on again, achieving a smooth zero-voltage switching operation.

[0051] Incidentally, the operation details described above for the positive half-cycle can achieve the smooth switching operation of the first switches S1, S3 and the fifth switches SB, SD of the second back-to-back switch pair. As for the operation of the negative half-cycle, since it is symmetrically related to the operation of the positive half-cycle, the operation from time t10 to time t14 described above can be performed in the same way, as shown in Figure 9B. The operation from time t20 to time t24 can also achieve the smooth switching operation of the second switches S2, S4 and the third switches SA, SC of the second back-to-back switch pair. These details will not be elaborated upon here.

[0052] For the flexible switching scheme of the single-phase two-wire to single-phase three-wire AC-AC (AC / AC) converter proposed in this invention, as shown in Figure 8, an unbalanced load simulation test was conducted. The simulation conditions were set with the maximum output currents IA and IB both at 32 Arms (RMS) and the rated output voltages VAN and VBN both at 120 Vrms. Load1 was set to 3.75Ω, corresponding to full-load output; load2 was set to 100kΩ to simulate a near-no-load condition. According to the simulation results, under extreme unbalanced load conditions, the output voltage deviates by approximately 9%. Considering the allowable output voltage range of ±10%, an open-loop control scheme remains feasible.

[0053] Furthermore, a balanced load simulation was conducted for the flexible switching scheme of the single-phase two-wire to single-phase three-wire AC-AC (AC / AC) converter proposed in this invention. The simulation conditions were set with the maximum output currents IA and IB both at 32 Arms, and the rated output voltages VAN and VBN both at 120 Vrms. Loads load1 and load2 were both set to 3.75Ω to simulate full-load output conditions. According to the simulation results, under balanced load conditions, the output voltage deviation is less than 1%, mainly affected by the on-resistance voltage drop of the switching elements.

[0054] Please refer to Figure 17, which is a circuit diagram of the three-phase DC-to-AC converter of the present invention. By utilizing the bidirectional switching units proposed in this disclosure, a flexible switching scheme for the three-phase DC-to-AC converter is established, which can be applied to drive three-phase AC loads. The flexible switching reduces switching losses, thereby increasing the switching frequency and improving system efficiency. As shown in Figure 17, the three-phase DC-to-AC converter includes six bidirectional switching units, and every two bidirectional switching units are connected in series to form a set of switching arms, thus forming a three-set parallel connection architecture. The three-phase DC-to-AC converter receives DC power VDC and converts the DC power VDC to output three-phase AC power, which can be applied to drive three-phase AC loads.

[0055] In summary, the present invention has the following features and advantages:

[0056] 1. The overall system architecture achieves a smooth switching effect through structural adjustments within the switching unit, and the output voltage is distributed through the alternating operation of the two switching units. Additional auxiliary switches, inductors, and capacitors are added within the switching unit to provide an extra resonant circuit, ensuring the voltage across the switches is zero before conduction, thus achieving a smooth switching effect. Compared to traditional hard-switching solutions, this reduces switching losses, thereby improving system efficiency and heat dissipation. Furthermore, compared to center-tapped power frequency transformers, it offers improvements in weight and size.

[0057] 2. Balanced and unbalanced loads, combined with the positive and negative half-cycle operation of the input power supply, enable each high-frequency switch to achieve the desired smooth switching behavior with zero voltage.

[0058] The above description is merely a detailed description and accompanying drawings of preferred embodiments of the present invention, and the features of the present invention are not limited thereto, nor are they intended to limit the present invention. The scope of the present invention should be determined by the following patent application scope. All embodiments that conform to the concept of the present invention and similar variations thereof should be included in the scope of the present invention. Any variations or modifications that can be easily conceived by those skilled in the art within the field of the present invention can be covered by the following patent disclosure scope.

Claims

1. A bidirectional switching unit, comprising: An inductor having a first terminal and a second terminal; A first switch has a first terminal and a second terminal, wherein the first terminal of the first switch is connected to the first terminal of the inductor to form a first common contact. A second switch having a first terminal and a second terminal, wherein the first terminal of the second switch is connected to the second terminal of the inductor to form a second common contact. A first back-to-back switch pair and a first capacitor, the first back-to-back switch pair being series-coupled to the first capacitor to form a first switch branch, the first switch branch having a first terminal and a second terminal, the first terminal of the first switch branch being connected to the second terminal of the first switch, and the second terminal of the first switch branch being connected to the second common contact; and A second back-to-back switch pair and a second capacitor are connected in series with the second capacitor to form a second switch branch. The second switch branch has a first terminal and a second terminal. The first terminal of the second switch branch is connected to the first common contact point, and the second terminal of the second switch branch is connected to the second terminal of the second switch.

2. The bidirectional switch unit as claimed in claim 1, wherein the first back-to-back switch pair includes a third switch and a fourth switch, the third switch and the fourth switch being connected back-to-back; wherein the second back-to-back switch pair includes a fifth switch and a sixth switch, the fifth switch and the sixth switch being connected back-to-back.

3. The bidirectional switching unit as claimed in claim 2, wherein the third switch and the fourth switch are metal-oxide-semiconductor field-effect transistors, and the third switch and the fourth switch are connected in a common-drain configuration; wherein the fifth switch and the sixth switch are metal-oxide-semiconductor field-effect transistors, and the fifth switch and the sixth switch are connected in a common-drain configuration.

4. The bidirectional switching unit as claimed in claim 2, wherein the third switch and the fourth switch are metal-oxide-semiconductor field-effect transistors, and the third switch and the fourth switch are connected to a common source; wherein the fifth switch and the sixth switch are metal-oxide-semiconductor field-effect transistors, and the fifth switch and the sixth switch are connected to a common source.

5. The bidirectional switching unit as claimed in claim 2, wherein the bidirectional switching unit receives an input power supply; when the input power supply is a positive voltage, the second switch and the sixth switch are normally on, and the third switch and the fourth switch are normally off; when the input power supply is a negative voltage, the first switch and the fourth switch are normally on, and the fifth switch and the sixth switch are normally off.

6. The bidirectional switching unit as described in claim 5, wherein when the input power supply is a positive voltage, the first switch and the fifth switch are controlled to switch, and when the voltage across the first switch and the fifth switch is zero during switching, the first switch and the fifth switch are controlled to be turned on; when the input power supply is a negative voltage, the second switch and the third switch are controlled to switch, and when the voltage across the second switch and the third switch is zero during switching, the second switch and the third switch are controlled to be turned on.

7. A power converter, comprising: At least one switching bridge arm includes a plurality of bidirectional switching units, each of the bidirectional switching units comprising: An inductor having a first terminal and a second terminal; A first switch has a first terminal and a second terminal, wherein the first terminal of the first switch is connected to the first terminal of the inductor to form a first common contact. A second switch having a first terminal and a second terminal, wherein the first terminal of the second switch is connected to the second terminal of the inductor to form a second common contact. A first back-to-back switch pair and a first capacitor, the first back-to-back switch pair being series-coupled to the first capacitor to form a first switch branch, the first switch branch having a first terminal and a second terminal, the first terminal of the first switch branch being connected to the second terminal of the first switch, and the second terminal of the first switch branch being connected to the second common contact; and A second back-to-back switch pair and a second capacitor are connected in series with the second capacitor to form a second switch branch. The second switch branch has a first terminal and a second terminal. The first terminal of the second switch branch is connected to the first common terminal, and the second terminal of the second back-to-back switch pair is connected to the second terminal of the second switch. The at least one switch arm receives an input power supply, which is converted into an output power supply by the bidirectional switch units, and can be used to provide multiple outputs.

8. The power converter of claim 7, wherein the first back-to-back switch pair includes a third switch and a fourth switch, the third switch and the fourth switch being connected back-to-back; wherein the second back-to-back switch pair includes a fifth switch and a sixth switch, the fifth switch and the sixth switch being connected back-to-back.

9. The power converter of claim 8, wherein when the input power supply is a positive voltage, the second switch and the sixth switch are normally on, and the third switch and the fourth switch are normally off; when the input power supply is a negative voltage, the first switch and the fourth switch are normally on, and the fifth switch and the sixth switch are normally off.

10. The power converter as claimed in claim 9, wherein when the input power supply is a positive voltage, the first switch and the fifth switch are controlled to switch at high frequency, and when the voltage across the first switch and the fifth switch is zero during the switching, the first switch and the fifth switch are controlled to be turned on; when the input power supply is a negative voltage, the second switch and the third switch are controlled to switch at high frequency, and when the voltage across the second switch and the third switch is zero during the switching, the second switch and the third switch are controlled to be turned on.

11. The power converter of claim 7, further comprising: At least one output filter, including a filter inductor, a first filter capacitor and a second filter capacitor; Each of the at least one output filter is coupled between two bidirectional switching units.

12. The power converter of claim 7, wherein the number of the bidirectional switching units is two, and the two bidirectional switching units are connected in series to form a switching bridge arm, so that the power converter forms a single-phase two-wire to single-phase three-wire AC-to-AC converter architecture.

13. The power converter of claim 7, wherein the number of bidirectional switch cells is six, and every two bidirectional switch cells are connected in series to form a switch leg, so as to form a structure of three switch legs connected in parallel.

14. The power converter of claim 13, wherein the power converter receives a direct current power source, and the three switch legs convert the direct current power source to output a three-phase alternating current power source.

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